JP2634959B2 - Speed sensorless speed control method - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a technique for controlling the rotation speed of an induction motor without using a speed sensor by using a PWM inverter.

[0002]

2. Description of the Related Art In a speed control system using a PWM inverter for a three-phase induction motor, a speed sensorless vector control method for controlling speed at high speed and high accuracy without using a speed sensor in the induction motor from the viewpoint of safety and economy. Are in the limelight, for example, the 54th of the materials of the Industrial Electric Power Application Research Group issued by the Institute of Electrical Engineers of Japan in October 1990.
It has also been published in a paper "Speed sensorless instantaneous space vector control" on page 66.

FIG. 2 is a block diagram of an example of a speed control system of a three-phase induction motor without a speed sensor using the instantaneous space vector method. The speed sensorless speed control according to the prior art will be described with reference to FIG.

A voltage type PWM inverter 1 supplies power to a three-phase induction motor 3. The primary state variable detector 2 detects a primary current and a primary voltage of the three-phase induction motor 3 and outputs the primary current and the primary voltage to the state variable calculator 4. In state variable processing unit 4 as follows, the primary interlinkage magnetic flux vector [psi ₁ and torque T, speed omega _m
And so on. In this calculation, the primary current and the primary voltage are respectively converted into d-axis components i _1d and v _1d and q-axis components i _1q and v ₁ of the instantaneous space vectors i ₁ and v ₁ with respect to the stator coordinates.
Decomposed into _1q, it calculates a d-axis component of the primary interlinkage magnetic flux vector [psi ₁ [psi _1d and [psi _1q like.

That is, first, the equation (1) ψ ₂ = ∫ ｛(L ₂ / M) (v ₁ −R ₁ i ₁ )｝ dt − ｛(L ₁ L ₂ / M) −M｝ i ₁ (1 ) To calculate the secondary linkage flux vector ψ ₂ . Here, L ₁ is the primary self-inductance, L ₂ is a secondary self-inductance, M is the mutual inductance, R ₁ is primary resistance. Subsequently, the primary interlinkage magnetic flux vector ψ ₁ is calculated by the equation (2) ψ ₁ = (M / L ₂ ) ψ ₂ + ｛L ₁ − (M ² / L ₂ )｝ i ₁ (2). Then the formula
(3) T = ψ _1d i _1q + ψ _1q i _1d Calculate the torque T according to (3) and _obtain the equation (4) ω ₂ = (d / dt) tan ^-1 (ψ _2q / ψ _2d ) (4) Calculate the rotational angular velocity ω ₂ of the secondary linkage magnetic flux vector ψ ₂ , and obtain the equation (5) ω _s = ｛R ₂ M (ψ _2d i _1q −ψ _2q i _1d )｝ / ｛L ₂ (ψ _2d ² + ψ _2q ²⁾ ) The slip angular velocity ω _s is calculated by｝ (5). Here, R ₂ is a secondary resistance. Thereafter, calculate the velocity omega _m by the equation _{(6) ω m = ω 2} -ω s (6).

[0006] The speed controller 5 as an input the speed omega _m of the output of the speed command omega _m ^* and the state variable processing unit 4, the speed omega _m is in the proportional-plus-integral calculator, etc. so as to follow the speed command omega _m ^* Outputs torque command T ^* . In the torque flux controller 6, the primary interlinkage flux | ψ ₁ | output from the state variable calculator 4 and the torque T follow the flux command φ ₁ ^* and the torque command T ^* output from the speed controller 5, Switching signal S is applied to voltage source P
Output to the WM inverter 1.

[0007]

As described in the description of the prior art, the velocity is calculated by the equation (6), and the slip angular velocity ω _s used therein is calculated by the equation (5) as shown in the equation (5). and using the following resistor R _2. The value of the secondary resistance R ₂ varies according to the temperature variation of the three-phase induction motor. However, since it can detect fluctuations of the state variable processing unit in 4 secondary resistance, the would contain errors, speed omega _m also to include an error in the slip angular velocity omega _s that is calculated by using the same. Therefore, the control of the prior art since the error in the speed omega _m by a secondary resistance variation occurs, there is a problem that can not be highly accurate speed control.

[0008]

In order to solve the above problems, a speed sensorless control system according to the present invention uses a speed sensor or a position sensor to measure the speed of a rotor of a three-phase induction motor fed by a voltage-type PWM inverter. In the speed sensorless speed control method obtained by calculation without using, the frequency of harmonics included in the state variable of the motor generated due to the slot ripple of the rotor of the motor is measured, and thereby the rotor of the motor is measured. The method is characterized in that an actual speed is estimated, and a secondary resistance of the electric motor is identified from a difference signal between the actual speed and the speed obtained by the calculation.

[0009]

When the rotor of a three-phase induction motor has a slot ripple, the magnetic flux distribution of the rotor is distorted by the influence of the slot, and the impedance seen from the primary side of the motor fluctuates at a frequency proportional to the rotation speed. Harmonics of a frequency proportional to the speed of the child occur in the state variables of the motor. For example, if a sinusoidal voltage power supply is symmetrical three-phase, the primary current of the induction motor includes f _r = harmonic of the frequency of the _{(S n / P n π)} ω mr ± Af i (7) is . Here, S _n is the number of slots rotor electric motor, P _n is the number of poles, [pi is pi, omega _mr
The actual rate, f _i is the frequency of the power supply. A is a proportional coefficient, and A = 1 in the case of a primary current.

[0010] Therefore, by measuring the harmonic frequency f _r of the primary current at harmonic frequency measuring means, by the equation (7) at the actual speed estimating means, independent of the actual speed omega _mr the secondary resistance R ₂ Can be estimated. The error between the speed ω _m calculated using the secondary resistance R ₂ and the actual speed ω _mr of the output of the actual speed estimating means is expressed by the following equation:
When used in the calculation of _m, it is considered to be caused by the secondary resistance variation, velocity omega _m in the secondary resistance identification means
And by adjusting the secondary resistance R ₂ to be used in the calculation as the actual speed omega _mr is equal, the secondary resistance can be identified.
As a result no longer error calculated speed omega _m by secondary resistance variation, it becomes possible to speed control with high accuracy. In addition, if the speed is controlled by the actual speed ω _mr of the output of the actual speed estimating means, the speed can be controlled with high accuracy without identifying the secondary resistance, but the estimation of the actual speed ω _mr takes time, Transient characteristics may be degraded.

[0011]

FIG. 1 shows a state variable of a three-phase induction motor for measuring a harmonic frequency caused by a slot ripple of a rotor. An example using ω _m is shown. Less than,
The description of the same parts as in FIG. 2 of this figure is omitted, and only different parts, that is, the state variable calculator 4 ′, the harmonic frequency measuring device 7, the actual speed estimator 8, and the secondary resistance identifier 9 will be described.

The harmonic frequency measuring device 7 calculates the calculation speed ω _m of the output of the state variable calculator 4 ′ and the power supply frequency f _i calculated from the equation (8) f _i = ω ₂ / (2π) (8). Enter the calculation speed ω
measuring the harmonic frequency f _r due to the slot ripple of the rotor are superimposed and outputs to _m. Here, the harmonic frequency _fr is represented by A =
It is represented by 2.

[0013] actual speed estimator 8 inputs an harmonic frequency f _r and the power supply frequency f _i, and outputs the calculated actual speed omega _mr by formula (7) (where A = 2). In the secondary resistance identification 9, performs identification seeking secondary resistance R ₂ as calculation speed omega _m by the equation (9) becomes equal to the actual speed omega _{mr. R 2 = {1 + ∫k r} (ω m -ω mr) dt} R 2n (9) where, k _r is the identifying coefficient takes the same polarity as that of the slip frequency omega _s of formula (5). R _2n is the initial value of the secondary resistance. The state variable calculator 4 'is the identified secondary resistance
Calculating a speed omega _m with R _2.

[0014] the rotor of the motor has slots ripple, but the harmonic frequency f _r of the waveform of the operation speed omega _m in A = 2 in equation (7) is superimposed, the waveform of the operation speed omega _m Includes not only harmonics of the frequency expressed by equation (7), but also a large number of harmonics such as harmonics of other frequencies caused by rotor slot ripple and harmonics caused by switching by a voltage-type PWM inverter. Have been. In the harmonic frequency detector 7, from the waveform of the operation speed omega _m containing harmonics of such a large number of frequencies to extract only the harmonic wave of the frequency component represented by the formula (7), the frequency f _r Is measured and output. FIG. 3 shows an example of a measuring method of the harmonic frequency measuring device 7. Hereinafter, this figure will be described. The harmonic frequency estimator 71 calculates the operation speed ω _m and the power supply frequency f _i
Enter the door, get an approximate value f _r 'harmonic frequency using the operation speed omega _m instead of the actual speed omega _mr of formula (7). The band-pass filter 72 receives the approximate value _fr 'and forms a band-pass filter using the approximate value _fr ' as a pass frequency. Therefore, the band-pass filter 72 can extract and output only the frequency components near _fr ′ in the signal input to the band-pass filter 72. Accordingly, when the calculation speed ω _m is input to the band-pass filter 72, the waveform of the harmonic component of the frequency component near f _r ′ from the harmonic frequency components included in the calculation speed ω _m is output from the band-pass filter 72. You. Since the difference between the calculation speed ω _m and the actual speed ω _mr is due to the setting error of the secondary resistance, the calculation speed ω _m and the actual speed ω _mr are close to each other, and therefore _fr ′ and f _r obtained from them are obtained. _r is considered to be a similar value, and the waveform of the output of the band-pass filter 72 is f _r ′.
Harmonics of the frequency components of f _r which is a frequency in the vicinity is also assumed that includes. Moreover, the largest component among the frequency components near f _r ′ is the frequency component of f _r , and the other frequency components are considered to be very small. Therefore, the output waveform of the band-pass filter 72 has almost no frequency component. f _r
Can be considered as the waveform of the higher harmonics. That by measuring the frequency of the largest components in the output waveform of the band-pass filter 72 by the frequency-voltage converter 73, it is possible to determine the harmonic frequency f _r.

[0015]

According to the speed sensorless speed control method of calculating the rotor speed of a three-phase induction motor without using a speed sensor, the speed sensorless speed control method according to the present invention does not use a fluctuating motor constant. The actual speed can be estimated by measuring the harmonic frequency of the motor state variable generated by the rotor slot ripple, and the secondary resistance can be identified sufficiently faster than the speed of the secondary resistance fluctuation. It is possible to reduce the speed calculation error due to the fluctuation of the secondary resistance due to the fluctuation, and to realize the speed sensorless speed control with high accuracy without deteriorating the transient characteristics.

[Brief description of the drawings]

FIG. 1 is a block diagram showing one embodiment of a speed sensorless speed control system according to the present invention.

FIG. 2 is a block diagram showing an example of a conventional speed sensorless speed control method.

FIG. 3 is a block diagram showing an example of a harmonic frequency measuring means used in the speed sensorless speed control method according to the present invention.

[Explanation of symbols]

 DESCRIPTION OF SYMBOLS 1 Voltage-type PWM inverter 2 Primary state variable detector 3 Three-phase induction motor 4, 4 'state variable calculator 5 Speed controller 6 Torque flux controller 7 Harmonic frequency measuring instrument 71 Harmonic frequency estimator 72 Band pass filter 73 Frequency-voltage converter 8 Actual speed estimator 9 Secondary resistance identifier

Claims (1)

(57) [Claims] In a speed sensorless speed control system for calculating the speed of a rotor of a three-phase induction motor fed by a voltage-source PWM inverter without using a speed sensor or a position sensor, the slot ripple of the rotor of the motor is reduced. The frequency of harmonics included in the state variable of the motor generated due to the cause is measured, the actual speed of the rotor of the motor is estimated therefrom, and the speed of the motor is calculated from the difference signal between the actual speed of the rotor and the speed obtained by the calculation. A speed sensorless speed control method characterized by identifying a secondary resistance.