JP7491735B2 - Energy Vibration Transmission System - Google Patents

Description

The present invention relates to a contactless power transmission system (wireless power transmission system), and in particular to an energy vibration transmission system that periodically excites wavelet-like damped vibrations by utilizing the resonance phenomenon that occurs when the movement of electromagnetic energy packets (lumps) vibrates.

Since the Massachusetts Institute of Technology (MIT) proposed it in 2005, research into wireless power transmission using sinusoidal electromagnetic waves has been active (see Patent Documents 1 and 2). For example, as described in Patent Document 2, conventional wireless power transmission methods are based on AC theory, which matches the resonant frequency 2π√LC of the power supply resonant circuit (LC circuit) with the resonant frequency 2π√LC of the power receiving resonant circuit (LC circuit). In conventional wireless power transmission methods, the technical common sense was that it was better not to have resonance (double resonance) between the power supply resonant circuit and the power receiving resonant circuit, with regard to new resonance that occurs as a result of the interaction between the power supply resonant circuit and the power receiving resonant circuit.

The power supply circuits (zero circuit) for the 10 kHz to 50 GHz frequency band described in Patent Documents 1 and 2 have a wasteful configuration in which commercial power is converted to DC by a switching power supply, and then equivalently converted to AC by switching a large number of power semiconductor elements such as PWM. This wasteful configuration generates power losses such as resistance losses in the power semiconductor elements and switching losses that increase rapidly with increasing frequency. In addition, switching elements are prone to destruction due to induced back electromotive force generated in the coil, and switching elements are prone to destruction due to excessive voltage rise caused by resonance, and the higher the frequency and the greater the power, the more difficult the circuit design becomes.

In view of the problems of the conventional technology described in Patent Documents 1 and 2, the present inventors have proposed a contactless transmission device based on non-AC theory that takes into account multiple resonance and focuses on transient response (see Patent Document 3). However, the invention described in Patent Document 3 has the problem that effective power transmission is not possible when the distance (transmission distance) between the power transmitting coil and the power receiving coil is 40 mm or more (see FIG. 17). Furthermore, when the position of the power receiving coil moves, the equivalent coupling coefficient k is unknown, so multiple resonance cannot be set according to the equivalent coupling coefficient k. For these reasons, the conventional technology has the problem of being unable to realize wireless power transmission that efficiently transmits the power that will be required in the future over long distances.

US Patent Application Publication No. 2008/0278264 Patent No. 5549745 World Intellectual Property Organization International Bureau International Publication No. 2020/039594 Pamphlet

In consideration of the above problems, the present invention aims to provide an energy vibration transmission system that takes into account the multiple resonance that occurs when packet-shaped electromagnetic energy moves, can effectively transmit power even when the distance between the power transmitting coil and the power receiving coil is 40 mm or more, can select the timing of circuit drive to enable optimal power transmission even in situations where the equivalent coupling coefficient k is unknown, and has a simplified circuit configuration and prevents damage to circuit elements.

The gist of the present invention is that it is an energy oscillation type transmission system that includes: (a) a primary circuit having a power transmission capacitor, a power transmission coil connected in parallel to the power transmission capacitor, and a detector that detects the terminal-to-terminal voltage of the power transmission capacitor; (b) a drive element that constitutes a circuit that inputs an intermittent DC voltage in steps between one terminal and the other terminal of the power transmission capacitor; (c) a secondary circuit having a power reception coil facing the power transmission coil and a power reception capacitor connected in parallel to the power reception coil; (d) a load that receives electrostatic energy from the power reception capacitor in a circuit that connects the terminals of the power reception capacitor; (e) a primary switching element drive circuit that sends a control signal to the control terminal of the drive element; and (f) an arithmetic logic circuit that detects the timing when the power transmission capacitor has completed at least one charge and discharge by the electromagnetic energy that flows back from the power transmission coil to the power transmission capacitor from the change in the output voltage of the detector, and causes the primary switching element drive circuit to periodically output a control signal to the control terminal in a drive cycle that sets this timing as the drive time.

According to the present invention, the driving period of the power transmission side can be selected to excite the system so that packet-shaped electromagnetic energy oscillates through super-resonance. This allows for effective power transmission even when the distance between the power transmission side coil and the power receiving side coil is 40 mm or more. Even in situations where the equivalent coupling coefficient k is unknown, optimal power transmission is possible by selecting the timing of circuit drive, and the circuit configuration is simplified, providing an energy oscillation type transmission system that can prevent damage to circuit elements.

1 is a schematic diagram showing a schematic structure of an example of an energy vibration type transmission system according to a first embodiment of the present invention; FIG. 2 is a block diagram mainly illustrating a drive control circuit shown in FIG. 1 . 1 is a circuit diagram showing an outline of a drive control circuit and a power receiving circuit constituting an example of an energy vibration type transmission system according to a first embodiment. FIG. 3B is an equivalent circuit illustrating an example of a load of the circuit shown in FIG. 3A. FIG. 3B is a circuit diagram showing an example of the detector shown in FIG. 3A. 3B is a circuit diagram illustrating a primary side circuit included in the drive control circuit shown in FIG. 3A and a secondary side circuit included in the power receiving circuit. FIG. 3B is a waveform diagram of the terminal voltage of the power transmitting side capacitor of the circuit shown in FIG. 3A. 1 is a cross-sectional view illustrating a schematic structure of a MOSFET used in an energy vibration type transmission system according to a first embodiment. FIG. 5B is a diagram illustrating a large signal equivalent circuit of the MOSFET shown in FIG. 5A. FIG. 6A is a waveform diagram showing changes in the inter-terminal voltages of the power transmitting side capacitor and the power receiving side capacitor in the energy vibration transmission system according to the first embodiment. FIG. 6B is a waveform diagram for explaining changes in the drive voltage, the terminal voltage of the load, the charging current to the load, etc., corresponding to FIG. 6A. 11 is a diagram for explaining the change in detector output voltage when the transmission distance d=2.0 cm and the equivalent coupling coefficient k=0.6. 11 is a diagram for explaining the change in detector output voltage when the transmission distance d=9.0 cm and the equivalent coupling coefficient k=0.3. 10 is a diagram for explaining the change in detector output voltage when the transmission distance d=20 cm and the equivalent coupling coefficient k=0.1. 2 is a schematic diagram illustrating the transfer of electromagnetic energy between a primary side circuit and a secondary side circuit in the energy vibration type transmission system according to the first embodiment. FIG. 9 is a schematic diagram illustrating the transfer of electromagnetic energy between the primary side circuit and the secondary side circuit in the energy vibration type transmission system according to the first embodiment at a timing subsequent to that in FIG. 8 . 10 is a schematic diagram illustrating the transfer of electromagnetic energy between the primary side circuit and the secondary side circuit in the energy vibration transmission system according to the first embodiment at a timing subsequent to that in FIG. 9 . 7C is a diagram explaining the relationship between the distance (transmission distance) between coils and the drive period in the energy vibration transmission system according to the first embodiment, using the zero cross time shown in FIG. 7C as a parameter. 7C is a diagram illustrating the relationship between the coupling coefficient k and the driving period in the energy vibration type transmission system according to the first embodiment, using some of the zero crossing times shown in FIG. 7C as parameters. 7D is a diagram illustrating changes in current flowing through the power transmitting coil and the power receiving coil corresponding to the waveforms shown in FIG. 7C. 7D are diagrams for explaining changes in the terminal voltage of the power receiving capacitor and the current flowing through the load, corresponding to the waveforms shown in FIG. 7C. This figure explains the change in the detector output voltage and the change in the current flowing through the transmitting side coil and the receiving side coil when, in the waveform shown in Figure 13, the driving element is turned on and a new driving voltage is stepped into the primary side circuit at zero crossing time T7. 16 is a diagram for explaining, in an enlarged manner, the change in the detector output voltage and the change in the current flowing through the power transmitting coil and the power receiving coil in the vicinity of the zero crossing time T7 in FIG. 15 . 1 is a diagram illustrating the relationship between transmission distance and transmission power in the energy vibration type transmission system according to the first embodiment in comparison with the prior art; FIG. 11 is a bird's-eye view illustrating another example of a spacing control mechanism that adjusts the surface spacing (transmission spacing) between coils when the energy vibration transmission system of the first embodiment is applied to charging the battery of an electric vehicle (EV). FIG. 11 is a circuit diagram showing an outline of a drive control circuit and a power receiving circuit constituting an example of an energy vibration transmission system according to a second embodiment of the present invention. FIG. 11 is a circuit diagram showing an outline of a drive control circuit and a power receiving circuit constituting an example of an energy vibration type transmission system according to a third embodiment of the present invention. FIG. 21 is a diagram for explaining a specific example in which the drive control circuit and the power receiving circuit shown in FIG. 20 are applied to charging a battery of an electric vehicle. FIG. 13 is a circuit diagram showing an outline of a drive control circuit and a power receiving circuit constituting an example of an energy vibration type transmission system according to a fourth embodiment of the present invention.

Next, first to fourth embodiments of the present invention will be described with reference to the drawings. In the following description of the drawings, the same or similar parts are given the same or similar reference numerals. However, it should be noted that the drawings are schematic, and the relationship between thickness and planar dimensions, the thickness ratio of each component, etc., differ from the actual ones. Therefore, specific thicknesses and dimensions should be determined with reference to the following explanation. In addition, it goes without saying that the drawings include parts with different dimensional relationships and ratios.

The first to fourth embodiments shown below are examples of devices and methods for embodying the technical idea of the present invention, and the technical idea of the present invention does not specify the materials, shapes, structures, arrangements, etc. of the components as described below. The technical idea of the present invention can be modified in various ways within the technical scope defined by the claims. Furthermore, the directions of "left and right" and "up and down" in the following explanation are simply defined for the convenience of explanation and do not limit the technical idea of the present invention. Therefore, for example, if the page is rotated 90 degrees, "left and right" and "up and down" are read interchangeably, and if the page is rotated 180 degrees, "left" becomes "right" and "right" becomes "left". Similarly, the direction of the spiral of the spiral as shown in Figure 18 is merely selected for the convenience of explanation, and it is also possible to select right-handed winding to left-handed winding and left-handed winding to right-handed winding according to the actual design circumstances.

First Embodiment
The vibrational energy transmission system according to the first embodiment of the present invention is a transmission system that supplies wavelet-shaped electromagnetic energy from a power supply device 29a in a contactless manner to a vehicle 31a having a power receiving circuit 27a, as shown in FIG. 1. "Wavelet-shaped electromagnetic energy" means a packet of electromagnetic energy that exhibits the characteristics of damped vibration localized in time. The power receiving circuit 27a includes a load (storage battery) 6. The vibrational energy transmission system according to the first embodiment has a power supply device 29a that supplies wavelet-shaped electromagnetic energy to the power receiving circuit 27a in a contactless manner, and a primary side operation unit 33 that is connected to the power supply device 29a and sends commands to the power supply device 29a.

Various structures and mechanisms can be adopted for the primary side operation unit 33, and for example, the primary side operation unit 33 may be provided with an imaging device. In the embodiment with an imaging device, a mechanism can be provided in which the vehicle height of the vehicle 31a is automatically linked by an AI function from an image of the vehicle 31a captured by the imaging device. Fig. 1 illustrates a schematic diagram showing that the power transmitting coil _L1 on the power supply device _29a side and the power receiving coil _L2 on the vehicle 31a side face each other, and wavelet-shaped electromagnetic energy is transmitted from the power transmitting coil _L1 to the power receiving coil _L2 without contact.

As shown in Fig. 1, the power supply device 29a is mainly composed of a power supply panel 11 in which the power transmission coil _L1 is housed in a disk-shaped dielectric, a distance control mechanism ₃₂ on which the power supply panel 11 is mounted and which controls the distance between the power transmission coil _L1 and the power receiving coil _L2 , a drive control circuit 34a which controls the power supply current flowing through the power transmission coil L1 and the distance control mechanism 32, a transmission data storage device 342a and a program storage device 342b connected to the drive control circuit 34a, and the power supply panel 11 whose transmission current is controlled by the drive control circuit 34a. The drive control circuit 34a and the power receiving circuit 27a transmit and receive wavelet-shaped electromagnetic energy to each other via the power transmission coil _L1 and the power receiving coil _L2 , and cause super-resonance. In the embodiment shown in Fig. 1, the distance control mechanism 32 is a vertical movement mechanism, and various well-known mechanisms can be adopted, such as a hydraulic vertical movement mechanism, an electromagnet vertical movement mechanism, and a movement mechanism in which a ball spiral is rotated by a step motor. On the other hand, in the embodiment shown in FIG. 18, the distance control mechanism 32 is a horizontal movement mechanism, but it is also possible to adopt various mechanisms such as a hydraulic horizontal movement mechanism, an electromagnet horizontal movement mechanism, or a ball screw horizontal movement mechanism.

1 is an example, and it is also possible to omit the power supply panel 11 that houses the power transmission side coil _L1 and use the power transmission side coil _L1 in a bare state. The power reception side coil _L2 is housed in a power reception panel 12 made of a disk-shaped dielectric. However, it is also possible to omit the power reception panel 12 that houses the power reception side coil _L2 and use the power reception side coil _L2 in a bare state. The wavelet-shaped electromagnetic energy from the power transmission side coil _L1 is supplied to the power reception side coil _L2 by electromagnetic induction so that the movement of the electromagnetic energy of the power supply side resonant circuit and the power reception side resonant circuit are superimposed.

The power supply panel 11 is installed or buried on the ground so that the upper surface of the power supply panel 11 is arranged parallel to the lower surface of the power receiving panel 12. Before the power supply operation, the upper surface of the power supply panel 11 is arranged parallel to the flat surface 30 on the ground, and the vehicle 31a is set to be able to enter by driving on a uniform flat surface. The power supply device 29a is installed, for example, in a parking space, and while the vehicle 31a is parked, it faces the power receiving panel 12 to supply wavelet-shaped electromagnetic energy to the power receiving panel 12 mounted on the vehicle 31a. When data on the transmission distance d of the energy vibration type transmission system according to the first embodiment as shown in FIG. 17 is acquired, if there is a case where a limit distance that can be transmitted exists due to special circumstances, the transmission distance d may not fall within the limit distance due to the vehicle height of the vehicle 31a. In such a case, the power supply panel 11 may be installed so that its upper surface protrudes above the ground. If the upper surface of the power supply panel 11 protrudes above the ground, the vehicle 31a entering the power supply field can be guided so that both wheels of the vehicle 31a straddle the upper surface of the power supply panel 11.

The load 6 is a storage battery represented by an equivalent circuit as shown in FIG. 3B, and stores wavelet-shaped electromagnetic energy supplied from the power supply device 29a via the power receiving panel 12. The vehicle 31a is, for example, a hybrid electric vehicle (HEV), a plug-in electric vehicle (PEV), or an electric vehicle (EV), and runs on electromagnetic energy stored in the storage battery as the load 6. The primary side operation unit 33 outputs a power supply start signal indicating the start of power supply or a power supply stop signal indicating the stop of power supply to the power supply device 29a by operation from outside. When the primary side operation unit 33 determines the vehicle height of the vehicle 31a by the AI function, the data on the vehicle height of the vehicle 31a is also transmitted to the drive control circuit 34a of the power supply device 29a.

As shown in FIG. 2, the drive control circuit 34a controls the power supply panel 11 to perform various drive controls to cause the transfer of electromagnetic energy between the power supply side resonant circuit and the power receiving side resonant circuit to undergo multiple resonance. For example, when a power supply start signal is input from the primary side operation unit 33, the drive control circuit 34a controls the current of the power supply panel 11 so as to supply wavelet-shaped electromagnetic energy at a set drive period. The drive control circuit 34a also includes an arithmetic logic circuit (ALU) 341 that acquires the vibration characteristics of the electromagnetic energy returned from the power receiving circuit 27a and performs processing to calculate the optimal drive period that maximizes the transmission efficiency due to multiple resonance between the power supply panel 11 and the power receiving panel 12. The transmission data storage device 342a and the program storage device 342b shown in FIG. 1 are connected to the arithmetic logic circuit 341 as shown in FIG. 2.

The drive control circuit 34a selects a drive timing that maximizes the transmission efficiency by double resonance from the vibration characteristics of the electromagnetic energy returned from the power receiving circuit 27a and does not damage the drive element _Q1 shown in FIG. 3A, etc., and controls the primary side switching element drive circuit 340a to operate at the selected drive timing. The primary side switching element drive circuit 340a sends a control signal to the control terminal of the drive element _Q1 shown in FIG. 3A, etc., to control the drive element _Q1 on/off. If the drive element _Q1 is a thyristor such as a field effect transistor (FET), a static induction transistor (SIT), a gate turn-off thyristor (GTO), or a static induction thyristor (SI thyristor), the "control terminal of the drive element _Q1" corresponds to the gate electrode of these power semiconductor elements. If the drive element _Q1 is a bipolar transistor (BJT), the base electrode of the BJT becomes the control terminal of the drive element _Q1 .

In the present invention, the period for cutting off the double resonance state accompanied by the movement of packet-shaped electromagnetic energy in each of the power supply side resonant circuit and the power receiving side resonant circuit is defined as the "driving period T". At the driving timing determined by the driving period T, a driving voltage E is input in a stepwise manner to the power supply side resonant circuit, and the free oscillation of the double resonance is cut off. Here, the "driving voltage E" is the terminal voltage E of the DC power source 5 shown in Figs. 3A and 4A. A wavelet-shaped damped oscillation is periodically excited in the driving period T. In order to operate the drive control circuit 34a so that the driving period T includes a plurality of oscillation peaks due to the oscillation accompanied by the movement of packet-shaped electromagnetic energy (hereinafter referred to as "energy packet oscillation"), the arithmetic logic circuit 341 includes a reflux voltage measurement control circuit 345, an energy peak counting circuit 346, a coupling coefficient calculation circuit 347, and a transmission condition setting circuit 348. That is, the arithmetic and logic circuit 341 controls the drive control circuit 34a so that the drive control circuit 34a and the power receiving circuit 27a can achieve super-resonance of energy packet vibration via the power transmitting coil _L1 and the power receiving coil _L2 .

The return voltage measurement control circuit 345 is a logic circuit that controls the measurement of the return voltage due to the electromagnetic energy returned from the power receiving circuit 27a to the drive control circuit 34a by double resonance using the detector 28 shown in FIG. 3A. The energy peak counting circuit 346 is a logic circuit that counts the number of energy peaks due to the electromagnetic energy returned from the power receiving circuit 27a from the return voltage measured by the return voltage measurement control circuit 345, and determines the drive period T at which the free vibration of double resonance accompanied by the movement of electromagnetic energy is cut off. The energy peak counting circuit 346 can determine the drive period T from the zero cross time when the change in the output voltage of the detector 28 falls below 0V. The coupling coefficient calculation circuit 347 is a logic circuit that calculates the equivalent coupling coefficient k corresponding to the drive period T determined by the energy peak counting circuit 346 from the relationship between the drive period T and the equivalent coupling coefficient k stored in the transmission data storage device 342a.

The transmission data storage device 342a stores the zero crossing time measured by the return voltage measurement control circuit 345 using the detector 28. The energy peak counting circuit 346 can read out the zero crossing time stored in the transmission data storage device 342a from the transmission data storage device 342a to determine the drive period T. Furthermore, the transmission data storage device 342a stores data showing the relationship between the distance (transmission distance) d between the coils and the drive period T as shown in FIG. 11 and data showing the relationship between the equivalent coupling coefficient k and the drive period as shown in FIG. 12. The transmission condition setting circuit 348 is a logic circuit that reads out the data showing the relationship between the transmission distance d and the drive period T and the data showing the relationship between the equivalent coupling coefficient k and the drive period from the transmission data storage device 342a, and determines the optimal transmission distance d from the equivalent coupling coefficient k calculated by the coupling coefficient calculation circuit 347. Here, the "transmission distance d" means the physical distance between the power transmitting coil _L1 and the power receiving coil _L2 as shown in FIG. 3A, and a specific example of the transmission distance d is shown in FIG. 18.

In general, the transmission distance d of the energy vibration transmission system according to the first embodiment may be unknown at first. When data on the limit of the transmission distance d of the energy vibration transmission system according to the first embodiment, as shown in FIG. 17, has been acquired, the transmission condition setting circuit 348 sends a command to the transmission interval control circuit 340b so that the transmission distance d falls within the range of the limit data, taking into account the height of the vehicle 31a to be supplied with power. When the primary side operation unit 33 has determined the height of the vehicle 31a using an AI function, it commands the interval control mechanism 32 the required travel distance, taking into account the data on the height of the vehicle 31a.

In the conventional technology described in Patent Document 3, fine adjustment of the transmission distance d was required to make it conform to the optimal equivalent coupling coefficient k = 0.6. In the energy vibration type transmission system according to the first embodiment, as described below with reference to FIG. 17 and the like, it is not necessary to set the optimal equivalent coupling coefficient k = 0.6. In particular, as shown in FIG. 11 and FIG. 12, in the energy vibration type transmission in the region away from the optimal equivalent coupling coefficient k = 0.6, the dependency of the drive period T on the transmission distance d is small. Therefore, in the case of long-distance transmission, since the operation is in the region away from k = 0.6, fine adjustment of the transmission distance d is not required. However, the transmission condition setting circuit 348 can also read the data indicating the relationship between the transmission distance d and the drive period T and the data indicating the relationship between the equivalent coupling coefficient k and the drive period stored in the transmission data storage device 342a, and determine the optimal transmission distance d or fine-tune the drive period T.

The arithmetic logic circuit 341 further includes an arithmetic sequence control circuit 344 that controls the sequence of the arithmetic processing of the return voltage measurement control circuit 345, the energy peak counting circuit 346, the coupling coefficient calculation circuit 347, and the transmission condition setting circuit 348. The arithmetic logic circuit 341 further includes an A bus 349a and a B bus 349b that transmit information and commands to the return voltage measurement control circuit 345, the energy peak counting circuit 346, the coupling coefficient calculation circuit 347, the transmission condition setting circuit 348, and the arithmetic sequence control circuit 344, respectively. In addition, when a power supply stop signal is input from the primary side operation unit 33, the drive control circuit 34a controls the primary side switching element drive circuit 340a so as not to start power supply or to stop power supply. The drive control circuit 34a operates the primary side switching element drive circuit 340a so that the drive voltage E is input in steps to the power supply side resonant circuit at the drive timing determined by the drive period T calculated by the arithmetic logic circuit 341, and supplies wavelet-shaped electromagnetic energy from the power supply device 29a to the power receiving circuit 27a.

The arithmetic logic circuit 341 can be configured as a computer system using a microprocessor (MPU) implemented as a microchip. In addition, a digital signal processor (DSP) with enhanced arithmetic operation functions and specialized for signal processing, or a microcontroller (microcomputer) equipped with memory and peripheral circuits and intended for embedded device control, may be used as the arithmetic logic circuit 341 that configures the computer system. Alternatively, the main CPU of a current general-purpose computer may be used as the arithmetic logic circuit 341. Furthermore, a part or all of the configuration of the arithmetic logic circuit 341 may be configured with a programmable logic device (PLD) such as a field programmable gate array (FPGA).

In the computer system of the drive control circuit 34a including the arithmetic logic circuit 341 shown in FIG. 2, the transmission data storage device 342a can be any combination appropriately selected from a group including a plurality of registers, a plurality of cache memories, a main storage device, and an auxiliary storage device. The cache memory may also be a combination of a primary cache memory and a secondary cache memory, and may further have a hierarchy including a tertiary cache memory. When a part or all of the arithmetic logic circuit 341 is configured by a PLD, the transmission data storage device 342a can be configured as a memory element such as a memory block included in a part of the logic block that configures the PLD. Furthermore, the arithmetic logic circuit 341 may have a structure in which a CPU core-like array and a PLD-like programmable core are mounted on the same chip. This CPU core-like array includes a hard macro CPU previously mounted inside the PLD and a soft macro CPU configured using the logic block of the PLD. In other words, the PLD may have a configuration in which software processing and hardware processing are mixed inside the PLD.

The return voltage measurement control circuit 345, the energy peak counting circuit 346, the coupling coefficient calculation circuit 347, the transmission condition setting circuit 348, and the calculation sequence control circuit 344 are connected to each other via the A bus 349a and the B bus 349b. The calculation sequence control circuit 344 controls the processing procedures of the return voltage measurement control circuit 345, the energy peak counting circuit 346, the coupling coefficient calculation circuit 347, and the transmission condition setting circuit 348 according to a computer software program. In FIG. 2, a configuration in which the primary side switching element drive circuit 340a and the transmission interval control circuit 340b are connected to the A bus 349a is illustrated. On the other hand, a configuration in which the transmission data storage device 342a, the program storage device 342b, and the output device 343 are connected to the B bus 349b is illustrated, but is not limited to the configuration shown in FIG. 2.

The hardware resources constituting the arithmetic logic circuit 341 shown in FIG. 2, such as the return voltage measurement control circuit 345, the energy peak counting circuit 346, the coupling coefficient calculation circuit 347, the transmission condition setting circuit 348, etc., formally express the hardware resources focusing on logical functions, and do not necessarily mean functional blocks that exist independently as physical areas on a semiconductor chip, but do not deny the actual configuration of programmable logic components implemented on a semiconductor chip such as the "logic block" of a PLD. If a part or all of the configuration of the arithmetic logic circuit 341 is configured with a PLD such as an FPGA, the program counter of the operation sequence control circuit 344 shown in FIG. 2 and data buses such as the A bus 349a and the B bus 349b can be omitted.

As shown in Fig. 3A, the energy vibration transmission system according to the first embodiment of the present invention includes a primary circuit 2a and a secondary circuit 3a. The primary circuit 2a is an LC resonant circuit having a power transmission side capacitor _C1 for storing electrostatic energy, and a power transmission side coil _L1 connected in parallel to the power transmission side capacitor _C1 for storing electrostatic energy sent from the power transmission side capacitor _C1 as magnetic energy, returning the magnetic energy to the power transmission side capacitor _C1 , and at the same time, magnetically coupled to the power receiving side coil _L2 of the secondary circuit 3a to transmit and receive magnetic energy. A DC power source 5 and a driving element _Q1 connected in series to each other are connected in parallel to the power transmission side capacitor _C1 . The DC power source 5 supplies a DC voltage to the power transmission side capacitor _C1 .

As described later, the "drive element Q ₁ " is a circuit element that generates wavelet-shaped electromagnetic energy by limiting the free damping oscillation of the primary circuit 2a to a specific number of times by the drive period T, and causes double resonance. By limiting the free damping oscillation to a specific number of times by the drive period T, the drive element Q ₁ realizes the oscillation of wavelet-shaped electromagnetic energy due to the transient current-voltage change in the primary circuit 2a. The DC power supply 5 may be a pseudo constant voltage source, or may be a DC power supply with a simple structure that is simply rectified and includes a large ripple component, so that a DC power supply 5 with a simple control circuit and peripheral circuits, which is not easily broken, easy to design, and inexpensive can be adopted. The secondary circuit 3a is an LC resonant circuit having a power receiving coil L ₂ that faces and is spaced apart from the power transmitting coil L ₁ and receives magnetic energy from the power transmitting coil L ₁ , and a power receiving capacitor C ₂ that is connected in parallel to the power receiving coil L ₂ and stores the magnetic energy stored in the power receiving coil L ₂ as electrostatic energy.

The load-side diode _D2 and the load 6, which are connected in series with each other, are connected in parallel to the receiving-side capacitor _C2 . The load 6 can be, for example, a rechargeable battery such as an on-board lithium (Li) ion battery for the vehicle 31a. In FIG. 3B, an equivalent circuit of a lithium ion battery is shown as a series-parallel circuit of a resistor and a capacitor as an example. The lithium ion battery includes the resistance of the current collector and the electrolyte, and the capacitor and resistor of the electric double layer formed at the interface in the battery. As shown in FIGS. 3A and 4A, the load-side diode _D2 is connected so that the anode faces the receiving-side resonator 2 and the cathode faces the load 6, limiting the direction in which the charging current I _CS flows to one direction. In FIGS. 3A and 4A, the floating resistance including the on-resistance of the load 6 is shown as _r2 .

In Figure 3A, the terminal voltage of the DC power supply 5 and the equivalent internal resistance _r1 is E, the terminal voltage of the power transmitting capacitor _C1 is V _C1 , the terminal voltage of the power receiving capacitor _C2 is V _C2 , the charging voltage measured between the terminals of the load 6 is V _CS , and the current flowing through the load side equivalent floating resistance r _L = r ₂ + _rs1 + _rs2 + _rs3 is the charging current I _CS , which are shown in Figures 4B, 6A, 6B, etc. The equivalent internal resistance r1 approximately represents the internal impedance of the DC power supply 5 as a resistance value. Figure _4B shows the transient response waveform of the terminal voltage V _C1 obtained by actual measurement when the driving element _Q1 is driven on and off.

In the first embodiment, it is assumed that the initial value of the charging voltage _VCS of the load 6 is a high value close to the charging completion voltage (close to full charge). At time t=0, the power transmitting capacitor _C1 is not charged and the terminal voltage _VC1 =0. When the driving element _Q1 is turned on at t=0, the driving voltage E from the DC power supply 5 is input in a step. Due to the step input at t=0, a charging current initially flows to the capacitor _C1 , and its value is E/ _r1 . At this time, the power transmitting coil _L1 generates a back electromotive force to block the inflowing current, so the current to the power transmitting coil _L1 is zero. 4B , the transmitting capacitor _C1 is charged with a rise time constant τ1 determined by the equivalent internal resistance _r1 , the capacitance _C1 of the transmitting capacitor _C1 , the parasitic resistance r _p1 of _the transmitting _capacitor C1, the on-resistance r _on1 of the driving element _Q1 , the inductance _L1 of the transmitting coil, and the dynamic mutual inductance M = M(t), and the terminal voltage V _C1 increases. The rise time constant _τ1 is a value that depends mainly on parameters related to the product _C1 · _r stray of the capacitance C1 of the transmitting capacitor _C1 and the equivalent _stray resistance r _stray = r ₁ + r _on1 + r _p1 .

As the voltage of _C1 gradually increases and the current decreases, the back electromotive force of the power transmission coil _L1 decreases and current begins to flow to the power transmission coil _L1 . This causes the voltage across the power transmission capacitor _C1 to drop slightly. At this point, the driving element _Q1 is turned off (t= _t1 ). The time _t1 at which the driving element _Q1 is turned off is set to the time when the current begins to flow through the coil and when the driving element _Q1 is turned off so that the driving element _Q1 is not destroyed by the voltage applied to the driving element Q1 caused by the back electromotive force generated by turning it off. When the driving element _Q1 is turned off at t= _t1 , current begins to flow from the power transmission capacitor _C1 to the power transmission coil _L1 and full-scale discharge begins. The electromagnetic energy stored in the power transmission capacitor _C1 tries to move to the power transmission coil _L1 .

At this time, the magnetic field generated around the power transmission coil _L1 by the current flowing through the power transmission coil _L1 generates an electromotive force in the power reception coil _L2 coupled with the dynamic mutual inductance M = M (t), and a current flows. If the characteristics of the primary circuit 2a and the secondary circuit 3a are in harmony at this time, the power reception capacitor _C2 is most efficiently charged by this transmitted power. That is, power is most efficiently transmitted from the primary circuit 2a to the secondary circuit 3a. The magnetic field generated around the power reception coil _L2 by the current flowing through the power reception coil _L2 generates an electromotive force in the power transmission coil _L1 . Due to the original voltage and this electromotive force, the voltage of the power transmission coil _L1 becomes a sawtooth wave that deviates from a sine wave, which is not a normal AC waveform. The sawtooth wave voltage has a nodule-like sawtooth wave characteristic that rises like a nodule just after passing the lowest voltage part.

The fall time constant _τ2 until the occurrence of this bump is a value that depends mainly on the capacitance _C1 of the power transmitting capacitor _C1 , the inductance _L1 of the power transmitting coil _L1 , and the parasitic resistance _Rstr ( _L1 ) of the power transmitting coil _L1 , and parameters that have a relationship similar to that of Equation (4) described below. However, for the inductance of the power transmitting coil _L1 , it is necessary to take into account the dynamic mutual inductance M=M(t) with the power receiving coil _L2 , which is a value that depends on time t.

At t= _t2 , the voltage of the primary capacitor again reaches a maximum peak, and at this time the voltage of the secondary capacitor reaches a value close to a minimum. In the invention described in Patent Document 3, the driving element _Q1 is turned on every time in accordance with all of these peaks, and the driving voltage is input in steps every time. However, in the energy vibration transmission system according to the first embodiment, after multiple vibration peaks are generated by free damping vibration as shown in Figures 7A, 7B, and 7C, the driving element _Q1 is turned on at a predetermined timing determined by the driving period T, and the driving voltage is input in steps intermittently and periodically.

The horizontal axis of FIG. 7A is the time axis, and the vertical axis is the output of the detector 28 shown in FIG. 3A. The waveform of the damped oscillation is shown when the physical distance (transmission distance) d between the power transmitting coil _L1 and the power receiving coil _L2 is 2.0 cm, and the equivalent coupling coefficient k is 0.6. The waveform of the free damped oscillation is shown, in which the driving voltage is not periodically input in steps. In a steady state where the AC theory is valid, as is well known, it is possible to approximate the power transmitting coil _L1 and the power receiving coil _L2 with a circuit magnetically coupled with a coupling coefficient K _AC and a mutual inductance M. The energy vibration transmission system according to the first embodiment is based on the non-AC theory, and the pseudo-coupling coefficient in a non-steady state defined at the time of a transient response, which is equivalent to the coupling coefficient K _AC derived from the AC theory, is defined as the "equivalent coupling coefficient k". The equivalent coupling coefficient k is a parameter that depends on time, strictly speaking. In the non-AC theory, a transient phenomenon (transient response mutual induction) between the primary circuit 2a and the secondary circuit 3a, which is caused by a time constant inherent in the circuit characteristics of the primary circuit 2a and a time constant inherent in the circuit characteristics of the secondary circuit 3a, can also be evaluated by an equivalent coupling coefficient k, which is the "degree of magnetic coupling" similar to the coupling coefficient _KAC in the AC theory.

7A, when the transmission distance d is 2.0 cm and the equivalent coupling coefficient k is 0.6, the output of the detector 28 has a wavelet-like free damped oscillation waveform that falls below 0 V at zero crossing times T1, T2, T3, T4, T5, T6, T7, and T8, but W-shaped secondary oscillations are sandwiched in the negative voltage region between the zero crossing times T1 and T2. Similarly, W-shaped secondary oscillations are also sandwiched in the negative voltage regions between the zero crossing times T3 and T4, between the zero crossing times T5 and T6, and between the zero crossing times T7 and T8. The detector 28 measures approximately the terminal voltage V _C1 of the transmitting side capacitor _C1 , but in the circuit shown in Figure 3A, as electromagnetic energy resonates back and forth between the primary circuit 2a and the secondary circuit 3a through double resonance, it is consumed as Joule heat in the parasitic resistance r _p1 of the transmitting side capacitor _C1 , the on resistance r _on1 of the driving element _Q1 , the parasitic resistance R _str (L ₁ ) of the transmitting side coil _L1 (not shown), the parasitic resistance r _p2 of the receiving side capacitor _C2, and the parasitic resistance R _str (L ₂ ) of the receiving side coil _L2 (not shown), so it can be seen that the electromagnetic energy gradually decays to a smaller value.

The zero crossing times T1, T2, and T3 in Fig. 7A correspond to the timings _T1 , _T4 , and _T5 indicated by subscripts in Fig. 8 to Fig. 10, respectively. The timings T2 and _T3 indicated by subscripts in Fig. 8 correspond to the rising and falling parts of the central peak of the W-shape between the zero crossing times _T1 and T2 in Fig. 7A. The timings _T6 and _T7 indicated by subscripts in Fig. 9 to Fig. 10 correspond to the rising and falling parts of the central peak of the W-shape between the zero crossing times T3 and T4 in Fig. 7A. In the case of the equivalent coupling coefficient k = 0.6 shown in Fig. _7A , the transmission efficiency due to the transient response mutual induction between the primary circuit 2a and the secondary circuit 3a is high, and at the timings _T2 and _T3 in Fig. 8, a sufficient return current is not returned from the power transmitting coil _L1 to the power transmitting capacitor C1, so it is considered that the output of the detector 28 is a negative value and zero crossing cannot occur. The same is true for timings _T6 and _T7 in Figures 9 and 10. When the equivalent coupling coefficient k = 0.6, the transmission efficiency due to the transient response mutual induction between the primary circuit 2a and the secondary circuit 3a is high, and a sufficient return current does not return from the power transmitting coil _L1 to the power transmitting capacitor _C1 , which is thought to be the reason for the formation of a small W-shaped peak in the negative voltage region between the zero-cross times T3 and T4.

The horizontal axis of FIG. 7B is the time axis as in FIG. 7A, and the vertical axis is the output of the detector 28 as in FIG. 7A, but the waveform of the damped oscillation when the transmission distance d=9.0 cm and the equivalent coupling coefficient k=0.3 are shown. As in FIG. 7A, the waveform of the damped oscillation is below 0V at zero cross times T1, T2, T3, T4, T5, T6, T7, and T8, but it is different from the damped oscillation of FIG. 7A in that a W-shaped secondary oscillation is sandwiched in the negative voltage region only between the zero cross times T3 and T4, and there is no W-shaped secondary oscillation in other time periods. The zero cross times T1, T2, T3, T4, and T5 of FIG. 7B correspond to the timings _T1 , _T2 , _T3 , _T6 , and _T7 indicated by subscripts in FIG. 8 to FIG. 10, respectively. It is considered that the timings _T4 and _T5 indicated by subscripts in FIG. 9 to FIG. 10 correspond to the W-shaped secondary oscillation that appears between the zero cross times T3 and T4. When the equivalent coupling coefficient k=0.3, the degree of magnetic coupling is lower than when k=0.6, and the transmission efficiency due to the transient response mutual induction between the primary circuit 2a and the secondary circuit 3a is low. As a result, at times _T4 and _T5 in Figures 9 and 10, a sufficient return current does not return from the transmitting coil _L1 to the transmitting capacitor _C1 , which is thought to be the reason for the small peak of negative voltage between the zero-cross times T3 and T4.

The horizontal axis of FIG. 7C is the time axis, as in FIG. 7A and FIG. 7B. Similarly to FIG. 7A and FIG. 7B, the vertical axis of FIG. 7C is the output of the detector 28, and shows the waveform of the damped oscillation when the transmission distance d=20 cm and the equivalent coupling coefficient k=0.1. Similarly to FIG. 7A and FIG. 7B, FIG. 7C shows the waveform of a wavelet-shaped free damped oscillation that falls below 0 V at zero crossing times T1, T2, T3, T4, T5, T6, T7, and T8, but differs from the damped oscillation of FIG. 7A and FIG. 7B in that there is no W-shaped secondary oscillation. The zero crossing times T1, T2, T3, T4, T5, T6, and T7 of FIG. 7C correspond to the timings T ₁ , T ₂ , T ₃ , T ₄ , T ₅ , T ₆ , and T ₇ indicated by subscripts in FIG. 8 to FIG. 10, respectively. The peak between zero crossing times T2 and T3 in FIG. 7C is a waveform indicating that the transmitting-side capacitor _C1 has been charged and discharged for the first time to a positive value.

7C are waveforms indicating that the second and third charging and discharging have been performed before the transmitting capacitor _C1 reaches a positive value, respectively. When the equivalent coupling coefficient k=0.1, the degree of magnetic coupling is lower than when k=0.3, and the transmission efficiency due to the transient response mutual induction between the primary circuit 2a and the secondary circuit 3a is even lower. Therefore, at all times _T1 to _T7 , a sufficient return current is returned from the transmitting coil _L1 to charge the transmitting capacitor _C1 to a positive value, which indicates that the charging and discharging of the transmitting capacitor _C1 can reach a zero crossing each time.

The peak between zero crossing times T2 and T3 in Fig. 7C corresponds to the phenomenon in which the drive voltage E is first stepped and input to the power supply resonant circuit to charge and discharge the power transmission capacitor _C1 . In contrast, the peak between zero crossing times T4 and T5 and the peak between zero crossing times T6 and T7 indicate the phenomenon in which the power transmission capacitor _C1 is charged and discharged by the electromagnetic energy returned in the heavy resonance. If we focus on the phenomenon in which the return energy due to the heavy resonance charges and discharges the power transmission capacitor _C1 to a positive value, the peak between zero crossing times T4 and T5 and the peak between zero crossing times T6 and T7 are the peaks of charging and discharging by the first and second return energy, respectively.

The damped oscillation shown in Figures 7A and 7B also indicates that electromagnetic energy _is consumed as Joule heat in the parasitic resistance r _p1 of the power transmitting capacitor C ₁ , the on-resistance r _on1 of the driving element Q ₁ , the parasitic resistance R _str (L ₁ ) of the power transmitting coil L ₁ (not shown), the parasitic resistance r _p2 of the power receiving capacitor C ₂ , and the parasitic resistance R _str (L ₂ ) of the power receiving coil L 2 (not shown). In the energy vibration transmission system according to the first embodiment, damped oscillation is obtained by modulating a sine wave as shown in Figures 7A, 7B, and 7C. In the present specification, a wave modulated by a sine wave as shown in Figures 7A, 7B, and 7C is referred to as a "pseudo sine wave". Figure 7A shows a W-shaped pseudo sine wave oscillation, and Figure 7B also shows a pseudo sine wave oscillation that includes a W-shaped portion.

In both the invention described in Patent Document 3 and the vibration energy transmission system according to the first embodiment, when the driving element _Q1 is turned on, a charging current flows to the power transmitting capacitor _C1 at first, and the value of the charging current is (E-Vc1)/ _r1 , which is the driving voltage E minus the voltage _Vc1 of the power transmitting capacitor _C1 at t= _t2 shown in Figures 4B, _6A , 6B, etc., divided by _r1 . In the vibration energy transmission system according to the first embodiment, the transient change when the driving element _Q1 is turned on at the timing _T7 at the end of the driving period T is shown in Figures 15 and 16. The difference between the invention described in Patent Document 3 and the energy vibration transmission system of the first embodiment is that, in the invention described in Patent Document 3, the drive voltage E is stepped up each time for each vibration indicating charging/discharging of the power transmitting capacitor _C1 , whereas the energy vibration transmission system of the first embodiment performs free damping vibration as shown in Figures 7A, 7B, and 7C, and after charging/discharging of the power transmitting capacitor _C1 by return current, the terminal voltage E is stepped up at a predetermined timing determined by the drive period T to stop the free damping vibration.

When the driving element _Q1 is turned on again, the power transmission coil _L1 generates a back electromotive force to block the current flowing in, so the current to the power transmission coil _L1 is zero. The voltage of the power transmission capacitor _C1 gradually rises, and as the current decreases, the back electromotive force of the power transmission coil _L1 decreases and a current begins to flow to the power transmission coil _L1 . This causes the voltage across the power transmission capacitor _C1 to drop slightly. At this point, the driving element _Q1 is turned off (t= _t3 in FIG. 4B). The time _t3 for turning off the driving element _Q1 is set to the time when the current begins to flow to the coil and when the driving element Q1 is turned off, so that the driving element _Q1 is not destroyed by the voltage applied to the driving element _Q1 by the back electromotive force generated in the power transmission coil _L1 . After the driving element _Q1 is turned off at t= _t3 , a current begins to flow from the power transmission capacitor _C1 to the power transmission coil _L1, and full-scale discharge begins. The electromagnetic energy stored in the power transmitting capacitor _C1 attempts to move to the power transmitting coil _L1 .

At this time, the magnetic field generated around the power transmitting coil _L1 by the current flowing through the power transmitting coil _L1 generates an electromotive force in the power receiving coil _L2 coupled by the dynamic mutual inductance M, and a current flows. As described later, if the time constants and the like involved in the transient response of the primary circuit 2a and the secondary circuit 3a are harmonized at this time, the power receiving capacitor _C2 is charged most efficiently. Then, the energy packet vibration between the primary circuit 2a and the secondary circuit 3a resonates heavily, and power is transmitted most efficiently from the primary circuit 2a to the secondary circuit 3a. The magnetic field generated around the power receiving coil _L2 by the current flowing through the power receiving coil _L2 generates an electromotive force in the power transmitting coil L1. As can be seen from Figures 4B, 6A, and 6B, even in the energy vibration type transmission system according to the first embodiment, the voltage due to the electromotive force of the power transmitting coil _L1 becomes a sawtooth wave that deviates from a sine wave, which is not a normal AC waveform. A little after passing the lowest voltage portion of this sawtooth wave, the voltage rises like a hump, resulting in a sawtooth wave characteristic with a hump.

As a result, after t= _t3 , the terminal voltage V _C1 of the power transmitting capacitor C ₁ decreases and becomes negative again, as shown on the right side of Fig. 4B. When the terminal voltage V _C1 of the power transmitting capacitor C ₁ becomes negative, the electromagnetic energy stored in the power transmitting coil L ₁ starts to return to the power transmitting capacitor C ₁ , and as shown on the right side of Fig. 4B, the terminal voltage V _C1 of the power transmitting capacitor C ₁ starts to increase due to the return current, becomes a positive value, further increases, and then decreases. In the energy vibration transmission system according to the first embodiment, this is shown as a peak between the zero crossing times T2 and T3, a peak between the zero crossing times T4 and T5, and a peak between T6 and T7 in Fig. 7C. It should be noted that the period of the energy packet oscillation in the energy oscillation transmission system according to the first embodiment depends on the fall time constant τ 2 determined by the capacitance C ₁ of the power transmission capacitor, the inductance L ₁ of the power transmission coil, the dynamic mutual inductance M, the parasitic resistance r _p1 of the power transmission capacitor C ₁ , the on-resistance r _on1 of the driving element Q ₁ , and the parasitic resistance R _str (L ₁ ) of the power transmission coil L ₁ (not shown). As shown in FIG. 4B, the change in the terminal voltage V _C1 due to the step input and cutoff of the driving voltage by the driving element _{Q 1} is not a sine wave waveform in the normal AC theory, but a waveform of a transient response of oscillation having a dull rise and fall characteristic of a nodule sawtooth wave. In the energy oscillation transmission system according to the first embodiment, as shown in FIGS. 7A, 7B, and 7C, a wavelet-shaped packet is formed in which multiple oscillation peaks caused by the movement of the energy packet are repeated while attenuating.

When the time constant inherent in the circuit characteristics of the primary circuit 2a and the time constant inherent in the circuit characteristics of the secondary circuit 3a are in harmony, "transient response mutual induction" occurs, and further, by making the energy packet vibration overlap, the electromagnetic energy of the primary circuit 2a is most efficiently transmitted to the secondary circuit 3a. In the invention described in Patent Document 3, the transient response waveforms of the terminal voltage V _{C1 of the primary circuit 2a and the terminal voltage V C2} of the secondary circuit 3a when the transient response mutual induction occurs between the primary circuit 2a and the secondary circuit _3a are shown in FIG. 6A. In FIG. 6A, as in FIG. 4B, the terminal voltage V _C1 shows a repeated transient response waveform with a dull rise and fall characteristic of a nodular sawtooth wave due to the step input and cutoff of the drive voltage by the drive element _Q1 . The transient response waveform shown by the solid line in Fig. 6A shows that the terminal voltage V _C1 of the power transmitting capacitor C ₁ starts to increase from a negative value, becomes a positive value, and further increases until the driving element Q ₁ reaches the ON state at t = _ti . In the energy vibration transmission system according to the first embodiment, as shown in Figs. 7A, 7B, and 7C, a wavelet-shaped vibration waveform having damped vibration due to the movement of the energy packet is obtained.

As shown by the dashed line on the left side of Fig. 6A, the step input of the drive voltage charges the power receiving side capacitor C2 of the secondary circuit 3a due to a transient phenomenon (transient response mutual induction) between the primary circuit 2a and the secondary circuit 3a, and after the terminal voltage V _C2 of the power receiving side capacitor _C2 reaches a peak value, the power receiving side capacitor _C2 starts to discharge and the terminal voltage _{V C2 starts} to decrease. Fig. 6A shows a case where the drive element _Q1 is turned on at any i-th time t = _ti , and the drive voltage E from the DC power source 5 is input in a step. The terminal voltage V _C2 of the power receiving side capacitor _C2 of the secondary circuit 3a decreases to a negative value at t = _ti .

In the invention described in Patent Document 3, by step-inputting the driving voltage E at t=t _i , as shown near the left center of Fig. 6A, the power transmission side capacitor C ₁ of the primary circuit 2a is charged with a rise time constant τ ₁ determined by the equivalent internal resistance r ₁ , the capacitance C ₁ of the power transmission side capacitor, the inductance L ₁ of the power transmission side coil, the dynamic mutual inductance M, the parasitic resistance r _p2 of the power reception side capacitor C ₂ , the load side equivalent floating resistance r _L , and the parasitic resistance of the power reception side coil L ₂ (not shown) R _str (L ₂ ), and the terminal voltage V _C1 increases. As shown in Fig. 6A, the terminal voltage V _C1 of the primary circuit 2a starts to decrease after reaching a peak value. When the driving element Q ₁ is turned off at t=t _i+1 , the power transmission side capacitor C ₁ of the primary circuit 2a starts to fully discharge, and the electromagnetic energy stored in the power transmission side capacitor C ₁ moves to the power transmission side coil L ₁ . The terminal voltage V _C2 of the power receiving capacitor _C2 in the secondary circuit 3a is a negative value between t=t _i and t=t _i+1, as indicated by the dashed line.

In the invention described in Patent Document 3, after t=t _i+1 , as shown by a solid line near the center of Fig. 6A, the terminal voltage V _C1 of the power transmitting capacitor C ₁ decreases and becomes negative with a fall time constant τ ₂ determined by the capacitance C ₁ of the power transmitting capacitor, the inductance L ₁ of the power transmitting coil, and the dynamic mutual inductance M, and the electromagnetic energy stored in the power transmitting capacitor C ₁ is transferred to the power transmitting coil L _1. In the energy vibration transmission system according to the first embodiment, as shown in Figs. 7A, 7B, and 7C, a damped oscillation having a plurality of oscillation peaks determined by the fall time constant τ ₂ determined by the capacitance C ₁ of the power transmitting capacitor, the inductance L ₁ of the power transmitting coil, and the dynamic mutual inductance M is formed. The electromagnetic energy stored in the power transmitting coil L ₁ is wirelessly transmitted to the power receiving coil L ₂ of the secondary circuit 3a by a transient phenomenon (transient response mutual induction) between the primary circuit 2a and the secondary circuit 3a. The electromagnetic energy wirelessly transmitted to the power receiving coil _L2 of the secondary circuit 3a is stored in the power receiving capacitor _C2 of the secondary circuit 3a.

As a result, after t=t _i+1 , the terminal voltage V _C2 of the receiving capacitor C ₂ of the secondary circuit 3a starts to increase as shown by the dashed line near the center of Fig. 6A. After the terminal voltage V _C2 of the receiving capacitor C ₂ of the secondary circuit 3a reaches a peak value, it starts to decrease and becomes a negative value as shown by the dashed line on the right side of Fig. 6A. In the energy vibration transmission system according to the first embodiment, the value of the charging voltage V _CS of the load 6 in the initial state is assumed to be a high value close to the charging completion voltage (close to full charge), and since the terminal voltage V _C2 of the receiving capacitor C ₂ exceeds the charging voltage V _CS of the load 6 near the peak value, a current flows through the load 6, and the electromagnetic energy stored in the receiving capacitor C2 moves to the load 6, and the rechargeable battery, which is the load 6, is charged.

In the invention described in Patent Document 3, after t=t _i+1 , when the terminal voltage V _C1 of the power transmitting capacitor C ₁ reaches the minimum negative value and then crosses 0V, as shown by the solid line in the center of Fig. 6A, the electromagnetic energy stored in the power transmitting coil L ₁ starts to return to the power transmitting capacitor C ₁ , and the terminal voltage V _C1 of the power transmitting capacitor C ₁ further increases due to the return current. As shown by the dashed line on the right side of Fig. 6A, when the terminal voltage V _C1 becomes positive, the terminal voltage V _C2 of the power receiving capacitor C ₂ becomes negative. In the energy vibration transmission system according to the first embodiment, the detector 28 shown in Fig. 3A is used to measure the vibration of the terminal voltage V _C1 of the power transmitting capacitor C ₁ due to the return current, as shown in Figs. 7A, 7B, and 7C.

In the invention described in Patent Document 3, when the terminal voltage V _C1 of the power transmission side capacitor C ₁ increases with a positive value and the driving element Q ₁ is turned on again at t=t _i+2 , the driving voltage E from the DC power supply 5 is input in a stepwise manner. In the energy vibration type transmission system according to the first embodiment, after the wavelet-shaped damped oscillation as shown in FIGS. 7A, 7B, and 7C is performed, the driving element Q ₁ is turned on at a predetermined timing determined by the driving period T, and the driving voltage E is input in a stepwise manner. By the stepwise input at t=t _i+2 , the power transmission side capacitor C ₁ is charged as shown by the solid line on the right side of FIG. 6A, and the terminal voltage V _C1 increases. As shown by the solid line in FIG. 6A, the terminal voltage V _C1 reaches a peak value E ₀ and then starts to decrease again. When the driving element Q ₁ is turned off at t=t _i+3 , the power transmission side capacitor C ₁ starts to discharge again, and the electromagnetic energy stored in the power transmission side capacitor C ₁ moves to the power transmission side coil L ₁ . The voltage V _C2 across the terminals of the power receiving capacitor C ₂ in the secondary circuit 3 a is a negative value between t=t _i+2 and t=t _i+3 , as indicated by the dashed line.

The electromagnetic energy stored in the power transmission coil _L1 is wirelessly transmitted to the power receiving coil L2 of the secondary circuit 3a by a transient phenomenon (transient response mutual induction) between the primary circuit 2a and the secondary circuit 3a. The electromagnetic energy wirelessly transmitted to the power receiving coil _L2 of the secondary circuit 3a is stored in the power receiving capacitor _C2 of the secondary circuit 3a. As shown in FIG. 6A, the change in the terminal voltage V _C1 of the primary circuit 2a due to the step input and cutoff of the drive voltage _E by the drive element _Q1 is not a sine wave waveform in the normal AC theory, but shows a transient response of a repetitive waveform with a dull rise and fall characteristic of a nodular sawtooth wave. On the other hand, the change in the terminal voltage V _C2 of the secondary circuit 3a shows a transient response of a repetitive waveform like a thinned out triangular wave, but is not a sine wave waveform in the normal AC theory. As can be seen from Fig. 6A, the "thinned out triangular wave" can also be interpreted as a waveform with the polarity of a trapezoidal wave reversed. In any case, the vibration waveform of the primary circuit 2a and the vibration waveform of the secondary circuit 3a are not symmetrical to each other.

Fig. 6B shows the measured waveforms of the transient response of the terminal voltage V _C1 and terminal voltage V _C2 shown in Fig. 6A in the invention described in Patent Document 3, which are further added with the driving voltage, the terminal voltage V _CS of the load 6, and the charging current I _CS to the rechargeable battery which is the load 6. When the driving element Q ₁ is turned on at t=t _i to store charge in the power transmission side capacitor C ₁ , and then the driving element Q ₁ is turned off at t=t _i+1 , a transient response mutual induction occurs between the primary circuit 2a and the secondary circuit 3a from the primary circuit 2a to the secondary circuit 3a. When the driving element Q ₁ is turned on, the equivalent internal resistance r ₁ of the DC power supply 5 is small, so the driving voltage shown by the thick solid line in Fig. 6B shows a change in which it is superimposed on the terminal voltage V _C1 of the primary circuit 2a.

The following description focuses on the transient response after t=t _i+1 in FIG. 6B. As shown by a solid line near the center of FIG. 6B, the terminal voltage V _C1 of the power transmitting capacitor C ₁ decreases and becomes a negative value after t=t _i+1 . The electromagnetic energy stored in the power transmitting capacitor C ₁ is transferred to the power transmitting coil L ₁ and stored in the power transmitting coil L _1. The electromagnetic energy stored in the power transmitting coil L ₁ is transmitted to the power receiving coil L ₂ of the secondary circuit 3a by the transient response mutual induction between the primary circuit 2a and the secondary circuit 3a. The electromagnetic energy transmitted to the power receiving coil L ₂ of the secondary circuit 3a is transmitted to the power receiving capacitor C ₂ of the secondary circuit 3a. Therefore, the terminal voltage V _C2 of the power receiving capacitor C ₂ starts to increase from a negative value after t=t _i+1 , as shown by a dashed line near the center of FIG. 6B.

The terminal voltage V _C2 of the receiving capacitor C ₂ of the secondary circuit 3a is a negative value between t=t _i and t=t _i+1 , as shown by the dashed line on the left side of the center. The terminal voltage V _C2 of the receiving capacitor C ₂ increases from the negative value at t=t _i+1 , becomes a positive value, increases further, reaches a peak value, and then starts to decrease as shown by the dashed line on the right side of FIG. 6B. When the terminal voltage V _C2 decreases, the electromagnetic energy stored in the receiving capacitor C ₂ flows to the load 6 as the charging current I _CS shown by the dashed line on the right side of FIG. 6B, and the load 6 is charged. Almost in synchronization with the increase in the charging current I _CS shown by the dashed line, the terminal voltage V _CS of the load 6 shown by the dotted line on the right side of FIG. 6B also increases slightly, and after passing through a peak value, shows a transient response of decreasing in synchronization with the decrease in the charging current I _CS . When the charging current I _CS decreases to zero, the decrease in the terminal voltage V _CS of the load 6 stops and starts to increase, and the terminal voltage of the load 6 reaches a steady value. The change in the terminal voltage V _CS according to the charging current I _CS occurs due to the presence of a series-parallel circuit of a resistor and a capacitor, as shown as an exemplary equivalent circuit in FIG.

In the invention described in Patent Document 3, after t=t _i+1 , when the terminal voltage V _C1 of the power transmitting capacitor C ₁ reaches the minimum negative value as shown by the solid line in the center of Fig. 6B, the electromagnetic energy stored in the power transmitting coil L ₁ starts to flow back to the power transmitting capacitor C ₁ , and the terminal voltage V _C1 of the power transmitting capacitor C ₁ starts to increase due to the return current, becomes a positive value, and continues to increase. As shown by the dashed line on the right side of Fig. 6B, when the terminal voltage V _C1 becomes a positive value, the terminal voltage V _C2 of the power receiving capacitor C ₂ becomes a negative value.

In the invention described in Patent Document 3, when the terminal voltage V _C1 of the power transmission side capacitor C ₁ increases with a positive value and the driving element Q ₁ is turned on again at t=t _i+2 , the driving voltage E from the DC power source 5 is input in a stepwise manner. In the energy vibration type transmission system according to the first embodiment, after wavelet-shaped damped oscillation as shown in FIGS. 7A, 7B, and 7C is performed, the driving element Q ₁ is turned on at a predetermined timing determined by the driving period T, and the driving voltage E is input in a stepwise manner. By periodically inputting the driving voltage E in a stepwise manner, the wavelet-shaped damped oscillation is periodically excited with the driving period T. By the stepwise input at t=t _i+2 , the power transmission side capacitor C ₁ is charged as shown by the solid line on the right side of FIG. 6B, and the terminal voltage V _C1 increases. As described above, since the equivalent internal resistance r ₁ of the DC power source 5 is small, when the driving element Q ₁ is turned on, the driving voltage E shown by the thick solid line in FIG. 6B changes so as to be superimposed on the terminal voltage V _C1 . As shown by the solid line superimposed on the terminal voltage E at the right end of Fig. 6B, the terminal voltage V _C1 reaches a peak value and then starts decreasing again. When the driving element _Q1 is turned off at t = t _{i + 3} , the power transmitting capacitor C ₁ starts discharging again, and the electromagnetic energy stored in the power transmitting capacitor C ₁ moves to the power transmitting coil L _1. The terminal voltage V _C2 of the power receiving capacitor C ₂ of the secondary circuit 3a is a negative value between t = t _{i + 2} and t = t _{i + 3,} as shown by the dashed line. In the energy vibration transmission system according to the first embodiment, when the driving element _Q1 is turned off, a wavelet-shaped damped oscillation is generated as shown in Figs. 7A, 7B, and 7C.

The behavior before an arbitrary time t=t _i is similar, and the terminal voltage V _C2 of the receiving capacitor C ₂ reaches a peak value and then starts to decrease, as shown by the dashed line on the left side of FIG. 6B. When the terminal voltage V _C2 increases and exceeds the terminal voltage V _CS , the electromagnetic energy stored in the receiving capacitor C ₂ flows to the load 6 as the charging current I _CS shown by the dashed line on the left side of FIG. 6B, and the load 6 is charged. Almost in sync with the increase in the charging current I _CS shown by the dashed line, the terminal voltage V _CS of the load 6 shown by the dotted line on the left side of FIG. 6B also increases slightly, and after passing a peak value, shows a transient response of decreasing in sync with the decrease in the charging current I _CS . When the charging current I _CS decreases to zero, the decrease in the terminal voltage V _CS of the load 6 stops and starts to increase, and the terminal voltage of the load 6 reaches a steady value. The change in the inter-terminal voltage _VCS according to the charging current _ICS occurs due to the presence of a series-parallel circuit of a resistor and a capacitor, the equivalent circuit of which is exemplarily shown in FIG. 3A.

Then, when the driving element _Q1 is turned off at t=t _i+1 as described above, the terminal voltage V _C1 of the power transmitting capacitor _C1 starts to decrease toward a negative value as shown by the solid line in the center of FIG. 6B. In this way, the electromagnetic energy stored in the power transmitting coil _L1 is wirelessly transmitted to the power receiving coil _L2 of the secondary circuit 3a by the transient response mutual induction between the primary circuit 2a and the secondary circuit 3a. The electromagnetic energy wirelessly transmitted to the power receiving coil _L2 of the secondary circuit 3a is stored in the power receiving capacitor _C2 of the secondary circuit 3a. As shown in FIG. 6B, the change in the terminal voltage V _C1 of the primary circuit 2a is not a sine wave waveform in the normal AC theory, but a repetitive transient response waveform with a dull rise and fall characteristic _of a nodular sawtooth wave, and the terminal voltage V _C2 of the secondary circuit 3a shows a transient response of a repetitive waveform like a thinned out triangular wave. In the energy vibration type transmission system according to the first embodiment, the vibration is a damped pseudo-sine wave as shown in FIGS. 7A, 7B and 7C, and is not a sine wave to which the AC theory with a constant amplitude can be applied.

As the driving element _Q1 shown in Fig. 3A, in addition to mechanical switching elements such as electromagnetic contactors, more preferably, power semiconductor elements capable of higher speed switching are used. As the power semiconductor elements, thyristors such as GTOs and SI thyristors are suitable as well as FETs, SITs, and BJTs. In particular, voltage-driven switching elements such as insulated gate field effect transistors (MISFETs), insulated gate static induction transistors (MISITs), insulated gate bipolar transistors (IGBTs), and MOS-controlled SI thyristors are suitable because they consume less power. From the viewpoint of market availability and evaluation of the internal resistance of power semiconductor elements, it is currently possible to adopt MOS field effect transistors (MOSFETs), which are a type of MISFET, as shown in the circuit shown in Fig. 3A.

In an energy vibration transmission system in which a rechargeable battery mounted on a vehicle 31a is used as the load 6, Joule heat is generated due to the flow of a large current, and heat of several hundred watts or more is generated, so that the energy vibration transmission system becomes a heating device (heater). In the energy vibration transmission system according to the first embodiment, only one power semiconductor element is required as the driving element _Q1 , so a structure can be easily designed to cover the driving element Q1 with a heat sink such as a copper block to increase thermal conductivity and prevent the element from being destroyed by heat generation, and the generation of floating resistance, floating capacitance, and floating inductance can also be minimized. Heat generation due to floating resistance (parasitic resistance) of the power transmitting coil _L1 and the power receiving coil _L2 is also large, so measures such as air cooling or water cooling of the power transmitting coil _L1 and the power receiving coil _L2 are preferable.

One method for suppressing the generation of Joule heat in a high-power energy vibration transmission system for in-vehicle use such as that of a vehicle 31a is to increase the voltage of the primary circuit 2a and set the voltage of the secondary circuit 3a to the optimum voltage of the load 6 by using the winding ratio of the power transmitting coil _L1 and the power receiving coil _L2 . When a power semiconductor element is used as the driving element _Q1 , a simple control for controlling the power semiconductor element on and off is all that is required, so that it is easy to design a circuit for increasing the voltage of the primary circuit 2a.

As described above, the vibration energy transmission system according to the first embodiment has a simple design with only one driving element _Q1 , so that it is easy to design the system to increase the voltage of the primary circuit 2a and minimize the power loss due to Joule heat generation on the primary circuit 2a side. Since the energy loss due to Joule heat generation can be reduced, the vibration energy transmission system according to the first embodiment improves the overall power transmission efficiency including the loss of the power supply circuit (zero circuit) in the case of high-power power transmission such as for on-board use in the vehicle 31a, and can contribute to solving the energy problems of mankind.

In the implementation circuit shown in Fig. 3A, a _first freewheel diode (freewheel diode) _FWD1 is connected in parallel as a protective element between the source and drain of the MOSFET as the driving element _Q1 in consideration of the freewheel current from the power transmitting coil L1. Also, in order to prevent the freewheel current from the power transmitting coil _L1 from flowing back to the DC power supply 5, a power supply side diode _D1 is connected in series between the DC power supply 5 and the driving element _Q1 . In the implementation circuit shown in Fig. 3A, the equivalent impedance _XLeq of the load 6 is approximated and expressed by the charging capacitance _Cs .

In the usual AC theory based on a steady-state sine wave, the mutual inductance M between the power transmitting coil _L1 and the power receiving coil _L2 is expressed as follows, using the coupling coefficient _KAC :

M = _KAC ( _{L1 L2} ) ^1/2 … (1)

It can be shown that:

The mutual induction between the series circuit of the power transmitting side capacitor _C1 and the power transmitting side coil _L1 and the series circuit of the power receiving side capacitor _C2 and the power receiving side coil _L2 is expressed by the following coupling equation using the mutual inductance M:

_L1di1 /dt+(1 _{/ C1} ) _∫i1dt + _Mdi2 /dt=0 ... (2)
_L2di2 / _dt +(1/ _C2 ) _∫i2dt + _Mdi2 /dt=0 ... (3)

In equations (2) and (3), ∫ is an integral symbol. That is, according to AC theory based on normal steady-state sine waves, the implementation circuit shown in FIG. 3A can be expressed as a T-type equivalent circuit using a coil with mutual inductance M between the power transmitting coil _L1 and the power receiving coil _L2 . In the T-type equivalent circuit, inductances _L1 -M and _L2 -M are connected in series to the upper horizontal bar, and inductance M is connected to the vertical center bar connected to the node of the series connection.

However, since the energy vibration type transmission system according to the first embodiment is a transient response transmission technology that does not rely on the AC theory of sinusoidal waves, the representation of the equivalent circuit used in the AC theory is merely a schematic diagram for considering an approximate physical model. Maxwell's equations in the case of time change can be analytically solved if the time change is based on a sine wave. However, when there is a sawtooth-wave time change as shown in Figures 4B, 6A, and 6B, it is extremely difficult to analytically solve Maxwell's equations. Therefore, in the present invention, the mutual inductance M used in the AC theory is a time-dependent parameter expressed by a function M(t) where t is time, and care must be taken.

Fig. 5A shows a large signal equivalent circuit of an nMOSFET used as an example of the driving element _Q1 in the mounting circuit shown in Fig. 3A. As shown in Fig. 5A, a typical nMOSFET has an n ⁺ type source region 72 and an n ⁺ type drain region 73 facing each other on a p-type substrate 71 with the surface of the p-type substrate 71, which serves as a channel region, sandwiched therebetween. A control terminal (gate electrode) 84 is provided on the channel region of the source region 72 and the drain region 73 via a gate oxide film 81 having a thickness of T _OX . A source electrode 82 is in ohmic contact with the source region 72, and a drain electrode 83 is in ohmic contact with the drain region 73.

As shown in Fig. 5A, in a typical nMOSFET, a gate-source capacitance C _GS exists between a control terminal (gate electrode) 84 and a source region 72, a gate-drain capacitance C _GD exists between the gate electrode 84 and a drain region 73, and a gate-substrate capacitance C _GB exists between the gate electrode 84 and a substrate 71. Furthermore, a source-substrate capacitance C _BS exists between the source region 72 and a substrate 71, and a drain-substrate capacitance C _BD exists between the drain region 73 and a substrate 71. The equivalent circuit shown in Fig. 5B shows a configuration in which a drain resistance R _D connected in series between the drain electrode and a channel region, and a source resistance R _S connected in series between the source electrode and a channel region are connected in series to a constant current source of a current I _DS provided in the channel region.

In the energy vibration type transmission system according to the first embodiment shown in Fig. 3A, the drain resistance R _D and source resistance R _S of the MOSFET shown in Fig. 5, which are the on-resistance of the driving element _Q1, are important, and it is necessary to select an element with a small on-resistance for the driving element _Q1 . Therefore, in the mounting circuit shown in Fig. 3A, it is necessary to determine the time constant inherent in the circuit characteristics of the primary side circuit 2a by including the on-resistance of the driving element _Q1 in the equivalent internal resistance _r1 of the DC power supply 5.

When the driving element _Q1 is turned off at t= _t1 in Fig. 4B, the primary circuit 2a becomes an RLC series circuit. According to AC theory, if the parasitic resistance of the power transmitting coil _L1 is _Rstr ( _L1 ), the damping coefficient _ζ1 of the RLC series circuit is

ζ ₁ = (R _str (L ₁ ) / 2) (C ₁ / L ₁ ) ^1/2 ... (4)

However, in practice, the inductance of the power transmitting coil _L1 needs to take into account the dynamic mutual inductance M=M(t) with the power receiving coil _L2 , but it is difficult to analytically explain the dynamic mutual inductance M=M(t).

In this case, if the parallel circuit consisting of the load side diode _D2 of the secondary circuit 3a and the load 6 is ignored, it can be regarded as an RLC series circuit. If the load side diode _D2 and the load 6 are ignored and AC theory is adopted, the damping coefficient _ζ2 of the secondary circuit 3a is expressed as follows, with the parasitic resistance of the power receiving coil _L2 being _Rstr ( _L2 ):

_ζ2 = ( _Rstr ( _L2 )/2) ( _C2 / _L2 ) ^1/2 ... (5)

In equation (5), the inductance of the power receiving coil _L2 must also take into account the time change in dynamic mutual inductance M=M(t).

According to AC theory, the resonant frequency of the RLC series circuit configured with the driving element _Q1 in the off state at t= _t1 in FIG. 4B is

f _o1 = (1/2π) (C ₁ · L ₁ ) - ^1/2 ... (6)

As described above, in equation (6), it is necessary to take into account the transient change of the time-dependent dynamic mutual inductance M=M(t) as the inductance of the power receiving coil _L1 . Similarly, the resonant frequency of the RLC series circuit when the load side diode _D2 and the load 6 of the secondary circuit 3a are ignored is given by the AC theory as follows:

f _o2 = (1/2π) ( _{C2 L2} ) - ^1/2 ... (7)

In the non-AC theory, even in equation (7), the inductance of the power receiving coil _L2 needs to take into account the dynamic mutual inductance M=M(t).

The power transmission coil _L1 and the power receiving coil _L2 of the energy vibration transmission system according to the first embodiment can be, for example, a spiral planar coil as shown in FIG. 18. FIG. 18 is a schematic diagram showing the physical structure of the power transmission coil _L1 and the power receiving coil _L2 of FIG. 3A in a concrete manner. In the energy vibration transmission system according to the first embodiment, for example, a wiring cable with a conductor cross-sectional area of 16 ^mm2 is wound nine times to form a spiral planar coil with a diameter of about 30 cm. These two spiral planar coils with a diameter of about 30 cm are arranged in parallel facing each other without contact with a gap of distance d. The transmission efficiency by the transient response mutual induction between the primary circuit 2a and the secondary circuit 3a from the primary circuit 2a to the secondary circuit 3a depends on the value of the magnetic coupling degree, which is similar to the coupling coefficient K _AC defined by AC theory. The magnetic coupling degree varies depending on the distance d between the two spiral planar coils, and it is necessary to control the distance d between the two spiral planar coils.

The degree of magnetic coupling can be adjusted by mechanically adjusting the relative positions of the two spiral planar coils, by inserting a magnetic body between the two spiral planar coils or by placing a magnetic body around the two spiral planar coils, or by attaching the two spiral planar coils to a pre-formed coupling using the attractive or repulsive forces acting between them.

A mutual relationship between the power transmitting coil _L1 and the power receiving coil _L2 that results in an equivalent coupling coefficient k that is approximately approximating the coupling coefficient K _AC = ^0.6 in AC theory is suitable for transient response mutual induction between the primary circuit 2a and the secondary circuit 3a. In the case of a spiral planar coil formed by winding a wiring cable with a conductor cross-sectional area of 16 mm2 nine times, the distance d needs to be about 0 cm to 2.0 cm to achieve the equivalent coupling coefficient k =0.6. On the other hand, the distance d needs to be about 10 cm to achieve a mutual relationship between the power transmitting coil _L1 and the power receiving coil _L2 that results in an equivalent coupling coefficient k _that is approximately approximating the coupling coefficient K AC =0.1 in AC theory. As shown in FIG. 14 with an exaggerated (enlarged) schematic view of the power transmitting coil _L1 and the power receiving coil _L2 , in order to charge a load 6, which is an on-board rechargeable battery of a vehicle 31b, using the energy vibration transmission system according to the first embodiment, the rear wheel stopper 33 is used as a distance control mechanism 32 to control the distance d between the power transmitting coil _L1 and the power receiving coil _L2 to about 10 cm, thereby enabling efficient contactless power transfer.

If a spacer having a thickness d ₀ is sandwiched between the power transmitting coil _L1 and the power receiving coil _L2 as the distance control mechanism 32 for controlling the distance d between the power transmitting coil _L1 and the power receiving coil _L2 , the distance d between the power transmitting coil _L1 and the power receiving coil _L2 can be controlled to d _0. Note that the mutual relationship between the power transmitting coil _L1 and the power receiving coil L2 corresponding to the value of the degree of magnetic coupling important for the transmission efficiency due to the transient response mutual induction between the primary circuit 2a and the secondary circuit _3a from the primary circuit 2a to the secondary circuit 3a can be configured by the distance control mechanism 32 using parameters other than the distance d between the power transmitting coil _L1 and the power receiving coil _L2 .

For example, a magnetic plate such as ferrite having a magnetic permeability _μr may be inserted between the power transmitting coil _L1 and the power receiving coil _L2 to serve as a coupling adjustment control mechanism. The value of the magnetic coupling degree between the power transmitting coil _L1 and the power receiving coil _L2 can be controlled by the insertion position of the magnetic plate in the vertical direction or the insertion area of the magnetic plate. The magnetic plate may be inserted behind the power transmitting coil _L1 instead of between the power transmitting coil _L1 and the power receiving coil _L2 . The value of the magnetic coupling degree between the power transmitting coil _L1 and the power receiving coil _L2 can be controlled by the insertion position of the magnetic plate in the vertical direction or the ratio of the insertion area of the magnetic plate to the area of the power transmitting coil _L1 . Although not shown in the figure, it goes without saying that the value of the magnetic coupling degree between the power transmitting coil _L1 and the power receiving coil _L2 can be similarly controlled by inserting the magnetic plate behind the power receiving coil _L2 .

Specifically, an optical distance measuring unit may be provided on the power supply device side where the power transmitting coil _L1 is provided. The distance measuring unit may be a distance control mechanism 32 having a light emitting section and a light receiving section. If the light receiving section is a time-of-flight (TOF) type distance measuring element d, pulsed light is emitted from the light emitting section. For example, a near-infrared LD (laser diode) or a near-infrared LED is used for the pulsed light emission. Pulsed light reflected from the power receiving coil _L2 or the bottom of the vehicle 31a is irradiated to the light receiving section through a lens or a BPF (band pass filter). The distance measuring unit may be configured as a laser interferometer or the like.

The light receiving unit of the distance measuring unit is connected to the transmission interval control circuit 340b of the drive control circuit 34a shown in FIG. 2. The transmission interval control circuit 340b is connected to the transmission condition setting circuit 348 of the arithmetic logic circuit 341. The output of the light receiving unit is input to the transmission condition setting circuit 348 that transmits the control condition to the interval control mechanism 32 via an output buffer or an interface not shown in the figure, and the transmission condition setting circuit 348 performs arithmetic processing necessary for measuring the distance between the power transmitting coil _L1 and the power receiving coil _L2 . The arithmetic logic circuit 341 is connected to a transmission data storage device 342a in which data necessary for logical operations such as calculation of the value of the magnetic coupling degree in the arithmetic logic circuit 341 and data on the insertion position in the vertical direction of the magnetic plate necessary to realize a desired equivalent coupling coefficient (pseudo coupling coefficient) are stored. Although not shown in the figure, the arithmetic logic circuit 341 may be connected to a program storage device or the like that stores a program that commands the operation of the arithmetic logic circuit 341.

When the value of the degree of magnetic coupling is controlled using a magnetic plate, data of the transmission distance d between the power transmitting coil _L1 and the power receiving coil _L2 calculated by the transmission condition setting circuit 348 is transmitted to a coupling coefficient calculation circuit 347 of the arithmetic logic circuit 341. The coupling coefficient calculation circuit 347 obtains the current value of the degree of magnetic coupling between the power transmitting coil _L1 and the power receiving coil _L2 from the data of the transmission distance d between the power transmitting coil _L1 and the power receiving coil _L2 calculated by the transmission condition setting circuit 348. The coupling coefficient calculation circuit 347 further calculates the movement distance of the magnetic plate from data of the insertion position in the vertical direction of the magnetic plate necessary to realize a desired equivalent coupling coefficient stored in the transmission data storage device 342a, and outputs the calculated distance to the transmission interval control circuit 340b. The transmission interval control circuit 340b of the interval control mechanism 32 shown in Fig. 2 controls the vertical insertion position of the magnetic plate and the vertical insertion position of the magnetic plate to a desired position based on the data on the moving distance of the magnetic plate sent from the coupling coefficient calculation circuit 347. A well-known position control mechanism such as a step motor can be used for the interval control mechanism 32. In this way, the vertical insertion position of the magnetic plate can be feedback-controlled to a desired position based on the output of the distance measurement unit.

8 to 10 are diagrams showing the operation of the implementation circuit shown in Fig. 3A in a time series, divided into eight timings _T0 to _T7 . In Fig. 8 to 10, a "differential connection" is assumed in which the magnetic fluxes of the power transmitting coil _L1 and the power receiving coil _L2 cancel each other out. According to AC theory, if the mutual inductance in AC theory is M, the combined inductance of the differential connection can be expressed as L = _L1 + _L2 - M, but it should be noted that the transient response mutual induction of the present invention does not necessarily follow AC theory.

As shown in Fig. 8(a), at the timing _T0 when the driving element _Q1 is turned on, a charge is first stored in the power transmitting capacitor _C1 as input electromagnetic energy E _IN1 . As shown in Fig. 8(a), the internal resistance r _on1 of the driving element _Q1 at this time is shown. When the terminal voltage V _C1 of the power transmitting capacitor _C1 starts to increase at the timing of Fig. 8(a), _a part of the input electromagnetic energy E _IN1 = (1/2) C ₁ V ² = Q ² / (2C ₁ ) stored in the power transmitting capacitor _C1 is transferred to the power transmitting coil L1 as the power transmitting coil current I _L1 and is stored in the power transmitting coil _L1 . Although the electromagnetic energy of the power transmitting coil _L1 is small, it is transmitted to the power receiving coil _L2 of the secondary circuit 3a. The electromagnetic energy transmitted to the power receiving coil _L2 of the secondary circuit 3a is consumed as the power receiving coil current _IL2 to charge the power receiving capacitor _C2 of the secondary circuit 3a. However, at the timing of FIG. 8(a), the terminal voltage _VC2 of the power receiving capacitor _C2 is a negative value.

Next, when the driving element _Q1 is switched to the cutoff state (off state) at the timing _T1 shown in FIG. 8B, the terminal voltage V _C1 of the power transmitting capacitor _C1 starts to decrease, and the transferred electromagnetic energy _E2 stored in the power transmitting capacitor _C1 is transferred to the power transmitting coil _L1 as the power transmitting coil current I _L1 and is stored in the power transmitting coil _L1 . The transferred electromagnetic energy _E2 is consumed as Joule heat in the equivalent internal resistance _r1 of the DC power source ₅ , the parasitic resistance r _p1 of the power transmitting capacitor C1, the on resistance r _on1 of the driving element _Q1 , and the parasitic resistance R _str (L ₁ ) of the power transmitting coil _L1 (not shown), and attenuates to a value smaller than the input electromagnetic energy E _IN1 . The first charging and discharging of the power transmitting capacitor _C1 is performed at the timings T ₀ to T ₁ shown in FIG. 8A and FIG. 8B. The transferred electromagnetic energy _E2 = ( ^1/2 ) _L1I2 stored in the transmitting coil _L1 is wirelessly transmitted to the receiving coil _L2 of the secondary circuit 3a by transient response mutual induction between the primary circuit 2a and the secondary circuit 3a.

Since a differential connection is assumed, according to normal AC theory, the current flows in the direction shown in FIG. 8B, and the transferred electromagnetic energy _E3 also flows. The transferred electromagnetic energy _E3 = (1/2 ⁾ _L2I2 transmitted to the power receiving coil _L2 of the secondary circuit 3a is stored in the power receiving capacitor _C2 of the secondary circuit 3a as the power receiving coil current I _L2 . The transferred electromagnetic energy _E3 is consumed as Joule heat in the parasitic resistance r _p2 of the power receiving capacitor _C2 and the parasitic resistance R _str ( _L2 ) of the power receiving coil _L2 (not shown), and is attenuated to a value smaller than the transferred electromagnetic energy _E2N1 . At the timing _T1 to _T2 in FIG. 8B, the terminal voltage V _C2 of the power receiving capacitor _C2 becomes a positive value. The terminal voltage V C2 of _the power receiving capacitor _C2 starts to decrease after reaching a peak value.

When the terminal voltage V _C2 decreases, the transfer electromagnetic energy E ₄ = (1/2) C ₂ V ² = Q ² / (2C ₂ ) stored in the power receiving capacitor C ₂ flows as a charging current I _CS to the load 6 at timing T ₂ as shown in FIG. 8(c), and the load 6 is charged. However, as the terminal voltage V _C2 decreases, a part of the electromagnetic energy stored in the power receiving capacitor C ₂ flows back to the power receiving coil L ₂ as the power receiving coil current I _L2 as shown in FIG. 8(c), and electrical energy is also stored in the power receiving coil L _2. The electromagnetic energy stored in the power receiving coil L ₂ is returned to the power transmitting coil L 1 of the primary circuit 2a by the transient response mutual induction between the primary circuit 2a and the secondary circuit 3a between the secondary circuit 3a and the primary circuit 2a as shown in FIG. ₈ (c). Since a differential connection is assumed, in accordance with normal AC theory, current flows in the direction shown in FIG. 8(c), and electromagnetic energy also flows.

As shown in Fig. 8C, the power transmitting coil current I _L1 returned to the power transmitting coil _L1 causes the transfer electromagnetic energy E stored in the power transmitting coil _L1 to start flowing back to the power transmitting capacitor _C1 , and the terminal voltage V _C1 of the power transmitting capacitor _C1 starts to increase due to the return current. Therefore, as shown in Fig. 8C, if a detector for measuring the power transmitting coil current I _L1 returned to the power transmitting coil _L1 is provided in the primary circuit 2a, the magnitude of the return voltage returned from the secondary circuit 3a can be measured. In other words, the detector provided in the primary circuit 2a can measure the efficiency of wireless transmission from the primary circuit 2a to the secondary circuit 3a due to the transient response mutual induction between the primary circuit 2a and the secondary circuit 3a.

When the driving element _Q1 is in the off state, at the timing _T3 shown in FIG. 9D, the terminal voltage V _C1 of the power transmitting capacitor _C1 starts to decrease, and the electromagnetic energy stored in the power transmitting capacitor _C1 is transferred to the power transmitting coil _L1 as the power transmitting coil current I _L1 and stored in the power transmitting coil _L1 . At the timings _T2 to _T3 in FIG. 8C and FIG. 9D, the power transmitting capacitor _C1 is charged and discharged for the second time. The electromagnetic energy stored in the power transmitting coil _L1 is wirelessly transmitted to the power receiving coil _L2 of the secondary circuit 3a by the transient response mutual induction between the primary circuit 2a and the secondary circuit 3a. The electromagnetic energy transmitted to the power receiving coil _L2 of the secondary circuit 3a is stored in the power receiving capacitor _C2 of the secondary circuit 3a as the power receiving coil current I _L2 . At this time, the remaining component of the electromagnetic energy that was intended to charge the load 6 flows back as transferred electromagnetic energy _E5 and is stored in the receiving-side capacitor _C2 . At timings _T3 to _T4 in Fig. 9(d), the terminal voltage V _C2 of the receiving-side capacitor _C2 becomes a positive value. After reaching a peak value, the terminal voltage V _C2 of the receiving-side capacitor _C2 starts to decrease.

When the inter-terminal voltage V _C2 decreases, the transferred electromagnetic energy E ₆ stored in the power receiving capacitor C ₂ flows back to the power receiving coil L ₂ as the power receiving coil current I _L2 as shown in Fig. 9(e), and electrical energy is also stored in the power receiving coil L _2. The electromagnetic energy stored in the power receiving coil L ₂ flows back to the power transmitting coil L ₁ of the primary circuit 2a as transferred electromagnetic energy E ₇ due to the transient response mutual induction between the secondary circuit 3a and the primary circuit 2a as shown in Fig. 9(e).

During the interval from _T4 to _T5 when the driving element _Q1 is in the off state, as shown in FIG. 9(e), the transferred electromagnetic energy _E7 stored in the power transmitting side coil _L1 begins to flow back to the power transmitting side capacitor _C1 due to the power transmitting side coil current I _L1 flowing back to the power transmitting side coil _L1 , and the inter-terminal voltage V _C1 of the power transmitting side capacitor _C1 begins to increase due to the return current.

At the timing _T5 shown in FIG. 9(f), the terminal voltage V _C1 of the power transmitting capacitor _C1 starts to decrease, and the transfer electromagnetic energy _E8 stored in the power transmitting capacitor _C1 is transferred to the power transmitting coil _L1 as the power transmitting coil current I _L1 and is stored in the power transmitting coil _L1 . At the timings _T4 to _T5 in FIG. 9(e) and FIG. 9(f), the third charge and discharge of the power transmitting capacitor _C1 is performed. The transfer electromagnetic energy _E8 stored in the power transmitting coil _L1 is wirelessly transmitted to the power receiving coil _L2 of the secondary circuit 3a by the transient response mutual induction between the primary circuit 2a and the secondary circuit 3a. The transfer electromagnetic energy _E9 transmitted to the power receiving coil _L2 of the secondary circuit 3a is stored in the power receiving capacitor _C2 of the secondary circuit 3a as the power receiving coil current I _L2 . 9F, the terminal voltage V _C2 of the power receiving capacitor C ₂ becomes a positive value from timing T ₅ to T _6. After the terminal voltage V _C2 of the power receiving capacitor C ₂ reaches a peak value, it starts to decrease.

When the inter-terminal voltage V _C2 decreases, the electromagnetic energy stored in the power receiving capacitor C ₂ flows back to the power receiving coil L ₂ as the power receiving coil current I _L2 at timing T ₆ to T ₇ -ΔT as shown in Fig. 10(g), and the transferred electromagnetic energy E ₁₁ is also stored in the power receiving coil L _2. The transferred electromagnetic energy E ₁₁ stored in the power receiving coil L ₂ flows back to the power transmitting coil L ₁ of the primary circuit 2a as transferred electromagnetic energy E ₁₂ due to the transient response mutual induction between the secondary circuit 3a and the primary circuit 2a as shown in Fig. 10(g). As shown in FIG. 10(g), from timing _T to T _- ΔT, the transferred electromagnetic energy E stored in the power transmitting side coil _L1 begins to flow back to the power transmitting side capacitor _C1 due to the power transmitting side coil current _{I L1} flowing back to the power transmitting side coil _L1 , and the inter-terminal voltage V _C1 of the power transmitting side capacitor _C1 begins to increase due to the return current.

When the driving element _Q1 is in the off state, at the timing _T7 -ΔT shown in FIG. 10(h), the terminal voltage V _C1 of the power transmitting capacitor _C1 starts to decrease, and the transferred electromagnetic energy _E13 stored in the power transmitting capacitor _C1 is transferred to the power transmitting coil _L1 as the power transmitting coil current I _L1 , and the transferred electromagnetic energy _E13 is stored in the power transmitting coil _L1 . In the case of differential connection, according to normal AC theory, a current should flow in the power receiving coil _L2 in the opposite direction to the power transmitting coil _L1 . However, as shown in FIG. 10(h), the direction in which the current flows in the power receiving coil _L2 is opposite to the prediction of normal AC theory. The transient response mutual induction of the present invention has time dependency, and the mutual coupling of the magnetic flux between the power transmitting coil _L1 and the power receiving coil _L2 is loose. As a result, a delay occurs in the movement of electrons that constitute the current in the receiving coil _L2 , and the remaining component of the transferred electromagnetic energy _E11 that was flowing in the receiving coil _L2 at timing _T6 continues to flow as transferred electromagnetic energy _E14 even at timing _T7 -ΔT.

That is, the fourth charging and discharging of the transmitting side capacitor _C1 is performed at timings _T6 to _T7 -ΔT in Figures 10(g) and 10(h), but at this point in time, the transferred electromagnetic energy _E13 stored in the transmitting side coil _L1 cannot cause transient response mutual induction between the primary side circuit 2a and the secondary side circuit 3a, and is not wirelessly transmitted to the receiving side coil _L2 of the secondary side circuit 3a. By repeating four charging and discharging operations, the transferred electromagnetic energy _E13 stored in the power transmitting coil _L1 is consumed as Joule heat in the parasitic resistance _rp1 of the power transmitting capacitor _C1 , the on-resistance _ron1 of the driving element _Q1 , the parasitic resistance _Rstr ( _L1 ) of the power transmitting coil _L1 ( _not shown), the parasitic resistance _rp2 of _the power receiving capacitor _C2 , and the parasitic resistance _Rstr ( _L2 ) of the power receiving coil _L2 (not shown) and gradually attenuates to a smaller value compared to the input electromagnetic energy EIN1 input at the initial timing T0. As shown in Figures 7A, 7B, and 7C, the energy loss due to Joule heat caused by four charging and discharging operations reduces the transferred electromagnetic energy _E13 , which can be considered to be the cause of the loosening of the mutual magnetic flux coupling between the power transmitting coil _L1 and the power receiving coil _L2 .

In the energy vibration transmission system according to the first embodiment, as shown in the output voltage of the detector 28 in Figs. 7A, 7B, and 7C, the state where the charge stored in the power transmission side capacitor _C1 becomes empty in the fourth charge/discharge is selected as the timing of expiration of the drive period T. Since the first (first) charge/discharge is a charge/discharge by the DC power source 5, counting the number of charges/discharges by the return current from the secondary circuit 3a, the state where the charge stored in the power transmission side capacitor _C1 becomes empty in the third charge/discharge by the return current is an example of the timing of expiration of the drive period T. When the drive period T expires and the power transmission side capacitor _C1 becomes empty, the drive element _Q1 is turned on again as shown in Fig. 10(i). At the timing _T7 when the drive element _Q1 is turned on again, the drive voltage E from the DC power source 5 is stepped and the charge is stored in the power transmission side capacitor _C1 as the input electromagnetic energy E _IN2 .

When a new drive voltage E is input to the primary circuit 2a, the double resonance state of the resonant circuit in the primary circuit 2a and the resonant circuit in the secondary circuit 3a is temporarily cut off, and the double resonance is immediately resumed by new excitation drive by the new input electromagnetic energy E _IN2 . When the new drive voltage E is input to the primary circuit 2a, the power transmitting capacitor C ₁ is empty, so the current constituting the input electromagnetic energy E _IN2 mainly flows into the power transmitting capacitor C _1. When the terminal voltage V _C1 of the power transmitting capacitor C ₁ starts to increase at the timing T ₇ to T ₇ +ΔT in FIG. 10(i), a part of the input electromagnetic energy E _IN2 = (1/2) C ₁ V ² = Q ² / (2C ₁ ) stored in the power transmitting capacitor C ₁ is transferred little by little to the power transmitting coil L ₁ as the power transmitting coil current I _L1 and is stored in the power transmitting coil L ₁ .

Figure 11 is a diagram for explaining the relationship between the transmission distance d and the drive period T in the energy vibration type transmission system according to the first embodiment, with the zero cross times T1, T2, T3, T4, T5, T6, and T7 shown in Figure 7C as parameters. Also, Figure 12 is a diagram for explaining the relationship between the equivalent coupling coefficient k and the drive period T in the energy vibration type transmission system according to the first embodiment, with the zero cross times T1, T4, and T7 shown in Figure 7C as parameters. The case of the transmission distance d = 2.0 cm in Figure 11 corresponds to the case of the equivalent coupling coefficient k = 0.6 shown in Figure 12. As can be seen from Figures 11 and 12, as the data for the transmission distance d = 2.0 cm and the equivalent coupling coefficient k = 0.6 is approached, a dip occurs in the curve showing the drive period T, and it can be seen that the drive period T is shortest when the transmission distance d = 2.0 cm and the equivalent coupling coefficient k = 0.6. Naturally, as time passes along the horizontal axis of FIG. 7C etc., from the zero crossing times T1, T2, T3, T4, T5, T6, T7, the drive period T shown on the vertical axis of FIG. 11 and FIG. 12 becomes longer.

If data showing the relationship between the transmission distance d and the drive cycle T shown in FIG. 11 and data showing the relationship between the equivalent coupling coefficient k and the drive cycle shown in FIG. 12 are measured in advance and stored in the transmission data storage device 342a shown in FIG. 2, the coupling coefficient calculation circuit 347 of the arithmetic logic circuit 341 can calculate the equivalent coupling coefficient k from the drive cycle T. In addition, the transmission condition setting circuit 348 can determine the optimal transmission distance d from the equivalent coupling coefficient k calculated by the coupling coefficient calculation circuit 347.

13 is a diagram showing the change in the current I _L1 flowing through the power transmitting coil L1 and the change in the current I _L2 flowing through the power receiving coil _L2 , corresponding to the waveform of the free damped oscillation shown in FIG. 7C, together with the terminal voltage (detector output voltage) V _C1 of the power transmitting capacitor _C1 measured by the detector 28. It can be seen that the pulsation of the current I _L1 flowing through the power transmitting coil _L1 , shown by the dashed line, is delayed from the _pulsation of the terminal voltage V _C1 of the power transmitting capacitor _C1 , shown by the dashed line. It can also be seen that the pulsation of the current I _L2 flowing through the power receiving coil _L2 , shown by the solid line, is in the opposite phase to the pulsation of the current I _L1 flowing through the power transmitting coil _L1 , shown by the dashed line. The drive period T measured up to the zero cross time T7 is 1.5 milliseconds.

14 is a diagram showing the change in the current I _L2 flowing through the power receiving coil _L2 , the terminal voltage V _C2 of the power receiving capacitor _C2 , and the load current I _{D2 flowing through the load diode D2, corresponding to the waveform of the free damped oscillation shown in FIG. 7C, together with the terminal voltage V C1 of the power transmitting capacitor C1 (detector output voltage) measured by the detector 28. The pulsation of the current I L2 flowing through the} power receiving coil _L2 , shown by the thin solid line, increases in opposite phase to the decaying pulsation of the _terminal voltage V _C1 of the power transmitting capacitor _C1 , shown by the dashed line. Also, the pulsation of the terminal voltage V _C2 of the power receiving capacitor _{C2, shown by the dashed line, is shown to gradually increase in amplitude with a phase delay from the pulsation of the current I L2 flowing} through the power receiving coil _L2 , shown by the thin solid line, and it can be seen that a large amount of electromagnetic energy is transmitted from the primary circuit 2a to the secondary circuit 3a over time. The load current I _D2 flowing through the load-side diode _D2 is shown intermittently as a peak just before the zero crossing time T6 and a peak just before the zero crossing time T7 of the waveform of the free damped oscillation shown in Fig. 7C. This is thought to correspond to the phenomenon in which the pulsation of the current I _L2 flowing through the power transmission coil _L2 shown by the thin solid line approaches a maximum value just before the zero crossing time T6. Fig. 14 also shows a drive period T of 1.5 milliseconds measured up to the zero crossing time T7.

13 and 14 show waveforms in the case of free vibration with multiple resonance, while FIG. 15 shows a transient response when the multiple resonance free vibration is temporarily turned off, the driving element _Q1 is turned on to generate a new multiple resonance, and the driving voltage E is input in a stepwise manner. The driving element Q1 is turned off when the power transmission side capacitor _C1 becomes full, and the driving voltage _E is input to the primary side circuit 2a in a Δ-function manner. It can be seen that the pulsation of the current I _L1 flowing through the power transmission side coil _L1 shown by the broken line is delayed with respect to the pulsation of the terminal voltage V _C1 of the power transmission side capacitor _C1 shown by the dashed line. It can also be seen that the pulsation of the current I _L2 flowing through the power receiving side coil _L2 shown by the thin solid line is in the opposite phase to the pulsation of the current I _L1 flowing through the power transmission side coil _L1 shown by the broken line. At the zero cross time T7 of the waveform of the free damping vibration shown in FIG. 7C, the driving _current I _Q1 flowing through the driving element _Q1 shown by the thick solid line is input in a Δ-function manner.

At the zero crossing time T7, when the drive current _IQ1 is input in a Δ-function manner, the terminal voltage V _C1 of the power transmitting capacitor _C1 , shown by the dashed line, tends to decrease until just before the zero crossing time T7, but increases suddenly, reaches a maximum value, and then decreases. With a delay from the rise of the terminal voltage V _C1 , the current I _L1 flowing through the power transmitting coil _L1 , shown by the dashed line, rises more slowly than the rise of the terminal voltage V _C1 , reaches a maximum value, and then decreases. The current I _L2 flowing through the power transmitting coil _L2 , shown by the thin solid line, was transferred to a maximum value in the negative region just before the zero crossing time T7, but when the drive current _IQ1 is input in a Δ-function manner, it increases suddenly in almost the same manner as the drive current _IQ1 , reaches a peak value, and then decreases. The current I _{L2 flowing through the power receiving coil L2 reaches} a peak value and then decreases to a large negative value in antiphase with the rise from the zero crossing time T7 of the current I _L1 flowing through the power transmitting coil _L1 shown by the dashed _line . Thereafter, the current I _L2 flowing through the power receiving coil _L2 performs free damping oscillation in antiphase with the current I _L1 flowing through the power transmitting coil _L1 .

FIG. 16 shows a transient response in the case where the driving element _Q1 is turned on so as to temporarily cut off the free vibration of the double resonance, and the driving voltage E is input in a stepwise manner, with the time scale enlarged from that of FIG. 15. The driving element _Q1 is turned off after the terminal voltage V _C1 of the power transmission side capacitor _C1 shown by the dashed line reaches its maximum value. That is, the driving current I _Q1 flowing through the driving element _Q1 shown by the thick solid line shows a change in which the terminal voltage V _C1 shown by the dashed line reaches its maximum value and then turns off. It can be seen that the rise and peak of the current I _L1 flowing through the power transmission side coil _L1 shown by the dashed line is delayed compared to the pulsation of the terminal voltage V _C1 shown by the dashed line. At the zero cross time T7, when the driving current I _Q1 is input in a Δ-functional manner, the terminal voltage V _C1 of the power transmission side capacitor _C1 shown by the dashed line shows a decreasing tendency until just before the zero cross time T7, but increases rapidly, reaches its maximum value, and then decreases. _The charge/discharge current _IC1 of the transmitting-side capacitor C1, indicated by a thin solid line, shows a rise similar to the change in the inter-terminal voltage V _C1 , indicated by the dashed dotted line, when the driving element _Q1 is in the on state, but after the driving element _Q1 is turned off, it changes with a phase that leads the change in the inter-terminal voltage V _C1 .

The current I _L1 shown by the dashed line rises slower than _{the terminal voltage V C1 ,} and after the drive element _Q1 is turned off, it reaches a maximum value and then decreases. The current I _L1 flowing through the power transmitting coil _L1 makes the total electromagnetic energy positive. When the power transmitting capacitor _C1 becomes full, the current I _Q1 flowing through the drive element Q1 becomes zero, and the drive element _Q1 is turned off, resulting in "zero current switching." The electromagnetic energy of the power transmitting coil _L1 is transmitted to the power receiving coil _L2 of the secondary circuit 3a _. The electromagnetic energy transmitted to the power receiving coil _L2 of the secondary circuit 3a is consumed as the power receiving coil current _IL2 to charge the power receiving capacitor _C2 of the secondary circuit 3a, and so a new double resonance mode proceeds. However, the behavior after the zero cross time T7 at the end of the drive period T, i.e., after the drive element _Q1 is turned off, is omitted because it overlaps with the items already described. According to the energy vibration type transmission system of the first embodiment of the present invention, the drive period T is set and the drive voltage E is periodically stepped, so that the wavelet-shaped damped oscillation can be periodically excited. That is, according to the energy vibration type transmission system of the first embodiment of the present invention, the drive period T is set in advance and the drive control circuit 34a can realize the periodic repetition of the double resonance state of energy, so that highly efficient contactless power transmission can be achieved even if the transmission distance d is long as shown in FIG. 17.

Figure 17 is a diagram showing the relationship between the transmission distance d and the transmission power in the energy vibration transmission system according to the first embodiment. In the case of the conventional technology described in Patent Document 3, shown by the thick dashed line, the transmission distance d = 40 mm was the limit. It can be seen that if the timing of the completion of the drive period T shown by the thick solid line is selected at the zero cross time T7, long-distance transmission is possible up to a transmission distance d = 160 mm, exceeding the transmission distance d = 40 mm. Note that although data exceeding the transmission distance d = 160 mm is not shown in Figure 17, this is merely for convenience's sake, and as shown in Figures 7A to 7C, contactless transmission is possible even when the transmission distance d = 200 mm is exceeded. It can be seen that if the timing of the completion of the drive period T shown by the thick dashed line is selected at the zero cross time T5, long-distance transmission is possible up to a transmission distance d = 120 mm, exceeding the transmission distance d = 40 mm. If the timing of the end of the drive cycle T is selected as the zero cross time T5, the power transmission efficiency is higher when the transmission distance d is between 60 and 100 mm than when the zero cross time is T7, which is indicated by the thick solid line.

The thin solid line, thin broken line, thin dashed line, and thin double-dashed line in FIG. 17 show data in the case of a comparative example in which a short drive period T that does not cross zero as in FIG. 7C is set. The drive period T was 1.5 milliseconds. In the comparative example, the drive period T was shortened to 490 μsec, 500 μsec, 510 μsec, and 515 μsec, respectively, so as to achieve a condition in which zero crossing does not occur, and the drive voltage E was step-input at this timing. In the case of the drive period T=490 μsec shown by the thin solid line, the transmission efficiency is higher than that of the conventional technology in the range of transmission distance d=30 to 40 mm, but the transmission distance d=50 mm is the limit for contactless transmission. In the case of the drive period T=500 μsec shown by the thin dashed line, the transmission efficiency is higher than that of the conventional technology in the range of transmission distance d=30 to 50 mm, but the transmission distance d=50 mm is the limit for contactless transmission. In the cases of the driving period T=510 μsec and 515 μsec shown by the thin dashed line and the thin double-dashed line, the transmission efficiency is higher than that of the conventional technology in the transmission distance d=30 to 60 mm range, but the transmission distance d=60 mm is the limit for contactless transmission. Therefore, it is preferable to set the driving period T in advance under the condition of zero crossing, and realize the repetition of the periodic energy super-resonance state by the driving control circuit 34a. In this way, according to the energy vibration type transmission system of the first embodiment, the system can be excited by selecting the driving period of the power transmission side so that the packet-shaped electromagnetic energy oscillates with super-resonance, so that effective power transmission can be achieved even when the distance between the power transmission side coil _L1 and the power receiving side coil _L2 is 40 mm or more. Moreover, according to the energy vibration type transmission system of the first embodiment, even in a situation where the equivalent coupling coefficient k is unknown, it is possible to provide an energy vibration type transmission system that can perform optimal power transmission by selecting the timing of turning on the driving element _Q1 , and can simplify the circuit configuration and prevent the destruction of the driving element _Q1, which is a circuit element.

Second Embodiment
As shown in Fig. 19, the vibration energy transmission system according to the second embodiment of the present invention is a transmission system that uses transient response mutual induction, supplies wavelet-shaped electromagnetic energy to the receiving circuit 27a without contact from the drive control circuit 34b, and causes super-resonance of energy packets between the drive control circuit 34b and the receiving circuit 27a. The vibration energy transmission system according to the second embodiment is configured by adding a transmission control element _Q2 to the circuit configuration of the vibration energy transmission system according to the first embodiment shown in Fig. 1. The transmission control element _Q2 is a circuit element that assists the drive element _Q1 in limiting the free damping oscillation of the primary circuit 2a to a specific number of times by the drive period T to realize the damping oscillation of the wavelet-shaped electromagnetic energy in the primary circuit 2a.

As the driving element _Q1 and the power transmission control element _Q2 shown in Fig. 19, power semiconductor elements including thyristors such as GTO thyristors, SI thyristors, etc., as well as FETs, SITs, and BJTs similar to those in the energy vibration type transmission system according to the first embodiment, are used. In particular, voltage-driven switching elements such as MISFETs, MISSITs, IGBTs, and MOS-controlled SI thyristors are suitable for the driving element _Q1 and the power transmission control element _Q2 because they reduce power consumption. From the viewpoint of market availability and evaluation of the internal resistance of power semiconductor elements, it is currently possible to employ MOSFETs as the driving element _Q1 and the power transmission control element _Q2 of the circuit shown in Fig. 10(b).

As already explained in the vibration energy transmission system according to the first embodiment, in a high-power vibration energy transmission system in which a rechargeable battery for an EV is used as the load 6, Joule heat is generated in a large amount. In the vibration energy transmission system according to the second embodiment, only two power semiconductor elements are required to be used as the driving element _Q1 and the power transmission control element _Q2 , so that a cooling structure that prevents the elements from being destroyed by heat generation can be designed easily, and the generation of floating resistance, floating capacitance, and floating inductance can be minimized. In addition, since only simple control of turning on/off the driving element _Q1 and the power transmission control element _Q2 is required, a design that suppresses the generation of Joule heat by increasing the voltage of the primary side circuit 2a can be easily performed.

In the implementation circuit shown in Fig. 19, in consideration of the return current from the power transmitting coil _L1 , a first return diode _FWD1 is connected in parallel between the source and drain of the MOSFET serving as the drive element _Q1 , and a second return diode _FWD2 is connected in parallel between the source and drain of the MOSFET serving as the power transmitting control element _Q2 , as protective elements. As in the circuit shown in Fig. 3A, in order to prevent the return current from the power transmitting coil _L1 from returning to the DC power supply 5, a power supply side diode _D1 is connected in series between the DC power supply 5 and the drive element _Q1 . In the implementation circuit shown in Fig. 19, the equivalent impedance _XLeq of the load 6 is also approximated and expressed by the charging capacitance _Cs .

As in the first embodiment, the wireless power transmission method according to the second embodiment will be described on the assumption that the coupling coefficient K is equivalent to the coupling coefficient K _AC =0.6 derived from AC theory. The value of the charging voltage V _CS in the initial state is assumed to be a sufficiently high voltage close to full charge. First, at timing _T0 similar to that shown in Fig. 8(a), the power transmission control element _Q2 is turned off and the drive element _Q1 is turned on, and a drive voltage E from the DC power supply 5 is input in a stepwise manner. When the drive voltage E from the DC power supply 5 is input in a stepwise manner, charge is stored in the power transmission side capacitor _C1 . In FIG. 8(a), the on state of driving element _Q1 is shown by the on resistance r _on1 of driving element _Q1 . However, when the power transmission control element _Q2 is in the off state, the primary side circuit 2b is not yet formed as a closed circuit. When driving element _Q1 is in the on state, a power supply side circuit is formed by the DC power supply 5, equivalent internal resistance r ₁ , a parallel circuit of driving element _Q1 and first free wheel diode (free wheel diode) _FWD1 , and a series circuit of the power transmission side capacitor _C1 .

The terminal voltage V _C1 of the power transmission side capacitor C ₁ is charged to a constant voltage while ringing. In the case of the energy vibration type transmission system according to the first embodiment shown in FIG. 3A, when the driving voltage E from the DC power source 5 is input in a step at timing T ₀ , a very small amount of current flows to the power transmission side coil L ₁ side, but in the energy vibration type transmission system according to the second embodiment, the power transmission control element Q ₂ is in an off state, so no current flows to the power transmission side coil L ₁ side. Therefore, in the energy vibration type transmission system according to the second embodiment, charge is stored in the power transmission side capacitor C ₁ more effectively. If the amount of charge flowing as electromagnetic energy when the driving voltage E from the DC power source 5 is input in a step is q ₀ , V _C1 =q ₀ /C ₁ , so the value of the terminal voltage V _C1 of the power transmission side capacitor C ₁ is larger than that in the case of the energy vibration type transmission system according to the first embodiment. At this timing, the terminal voltage V _C2 of the power receiving side capacitor C is a negative value.

Next, at timing _T1 similar to that shown in Fig. 8(b), when the driving element _Q1 is turned off and the power transmission control element _Q2 is turned on after a certain time, the electromagnetic energy stored in the power transmission side capacitor _C1 is accumulated in the power transmission side coil _L1 via the power transmission side coil current _IL1 , and further, a transient response mutual induction occurs between the primary side circuit 2b and the secondary side circuit 3a. At timing _T1 similar to that shown in Fig. 8(b), it is necessary to consider the on state of the power transmission control element _Q2 in terms of the on resistance r _on2 of the power transmission control element _Q2 . By turning on the power transmission control element _Q2 , the primary side circuit 2b is formed, and the power supply side circuit consisting of the DC power source 5, the equivalent internal resistance r ₁ , the parallel circuit of the driving element _Q1 and the first free wheel diode (free wheel diode) _FWD1 , and the power transmission side capacitor _C1 disappears.

When the electromagnetic energy stored in the power transmitting capacitor _C1 moves to the power transmitting coil _L1 , the terminal voltage V _C1 becomes 0V after reaching a negative maximum value. At timing T ₀ to T ₁ , the first charging and discharging of the power transmitting capacitor _C1 is performed. The electromagnetic energy transmitted to the power receiving coil _L2 by the transient response mutual induction from the primary circuit 2b to the secondary circuit 3a charges the power receiving capacitor _C2 by the power receiving coil current I _L2 . When charging of the power receiving capacitor _C2 starts, the terminal voltage V _C2 of the power receiving capacitor _C2 becomes a positive value after reaching a negative maximum value. It reaches a maximum value when the terminal voltage V _C1 reaches 0V. The terminal voltage V _C2 becomes a positive value after reaching a negative maximum value. It reaches a maximum value when _the terminal voltage V _C1 reaches 0V.

At a timing _T1 to _T2 similar to that shown in Fig. 8(b), as the terminal voltage V _C2 increases, a charging current I _CS is generated by a part of the electromagnetic energy stored in the power receiving capacitor _C2 , and charge is stored in the rechargeable battery as the load 6. Next, as shown in Fig. 8(c), a part of the electromagnetic energy stored in the power receiving capacitor _C2 flows back to _the power receiving coil _L2 as the power receiving coil current I L2, and electrical energy is also stored in the power receiving coil _L2 . As shown in Fig. 8(c), the electromagnetic energy stored in the power receiving coil _L2 flows back to the power transmitting coil _L1 of the primary circuit 2a due to the transient response mutual induction between the primary circuit 2a and the secondary circuit 3a. At timing _T3 similar to that shown in Fig. 9(d), the terminal voltage V _C1 of the power transmitting capacitor _C1 starts to decrease, and the electromagnetic energy stored in the power transmitting capacitor _C1 is transferred to the power transmitting coil _L1 as the power transmitting coil current I _L1 and is stored in the power transmitting coil _L1 . The second charging and discharging of the power transmitting capacitor _C1 is performed from timing _T2 to _T3 . At timing similar to that shown in Fig. 9(d), the charge that is excessive for charging the rechargeable battery flows back to the power receiving capacitor _C2 .

Another part of the electromagnetic energy stored in the power receiving capacitor _C2 flows back to the power receiving coil _L2 as the power receiving coil current I _L2 at the timing _T4 similar to that shown in FIG. 9E. At the timings _T4 to _T5 , when the charging current I _CS becomes 0, the terminal voltage V _C2 becomes the same value as the charging voltage V _CS . At the timings _T4 to _T5 , the electromagnetic energy stored in the power receiving capacitor _C2 flows back to the power receiving coil _L2 , causing a transient response mutual induction between the primary circuit 2b and the secondary circuit 3a, and a part of the electromagnetic energy of the secondary circuit 3a returns to the primary circuit 2b. At the timing _T5 similar to that shown in FIG. 9F, the terminal voltage V _C1 of the power transmitting capacitor _C1 starts to decrease, and the electromagnetic energy stored in the power transmitting capacitor _C1 moves to the power transmitting coil _L1 as the power transmitting coil current I _L1 and is stored in the power transmitting coil _L1 . At timings _T4 to _T5 , the third charging and discharging of the transmitting capacitor _C1 is performed. When the electromagnetic energy stored in the receiving capacitor _C2 is transferred to the load 6 and the receiving coil _L2 , the receiving capacitor _C2 is discharged. When the receiving capacitor _C2 is discharged, the terminal voltage V _C2 reaches a negative maximum value and then reaches 0 V. At timings _T3 to _T5 , the terminal voltage V _C1 reaches a negative maximum value and then increases to a positive value, reaching a maximum value of 0 V.

When the terminal voltage V _C2 of the secondary circuit 3a decreases, the electromagnetic energy stored in the power receiving capacitor _C2 flows back to the power receiving coil _L2 as the power receiving coil current I _L2 at the timing T ₆ to T ₇ -ΔT similar to that shown in Fig. 10(g). The electromagnetic energy stored in the power receiving coil L2 flows back to the power transmitting coil _L1 of the primary circuit 2a. As shown in Fig. 10(g), at the timing T ₆ to T ₇ -ΔT, the electromagnetic energy starts to flow back to the power transmitting capacitor _C1 due to the power transmitting coil current I _L1 flowing back to _the power transmitting coil _L1 , and the terminal voltage V _{C1 of the power transmitting capacitor C1 starts} to increase due to the return current. With the driving element _Q1 in the off state, at the timing _T7 -ΔT shown in Figure 10 (h), the terminal voltage _VC1 of the power transmitting capacitor _C1 begins to decrease, and the electromagnetic energy stored in the power transmitting capacitor _C1 is transferred to the power transmitting coil _L1 as the power transmitting coil current _IL1 and is stored in the power transmitting coil _L1 .

As described with reference to FIG. 10(h) in the energy vibration transmission system according to the first embodiment, the direction of current flowing in the power receiving coil _L2 at timings _T6 to _T7 -ΔT is opposite to the prediction in the case of differential connection in the normal AC theory. There is a delay in the movement of electrons constituting the current in the power receiving coil _L2 , and the remaining component of the transfer electromagnetic energy _E11 flowing in the power receiving coil _L2 at timing _T6 continues to flow as the transfer electromagnetic energy _E14 at timing _T7 -ΔT, and the mutual coupling of the magnetic flux between the power transmitting coil _L1 and the power receiving coil _L2 becomes loose. The fourth charge and discharge of the power transmitting capacitor _C1 is performed at timings _T6 to _T7 -ΔT, but at this point in time, the transfer electromagnetic energy _E13 stored in the power transmitting coil _L1 cannot generate a transient response mutual induction between the primary circuit 2a and the secondary circuit 3a, and is not wirelessly transmitted to the power receiving coil _L2 of the secondary circuit 3a.

In the vibration energy transmission system according to the second embodiment, the state where the charge stored in the power transmission side capacitor _C1 is empty in the fourth charge/discharge is selected as the timing of expiration of the drive period T, and the power transmission control element _Q2 is turned off in the state where the charge stored in the power transmission side capacitor _C1 is empty. As shown in FIG. 10(i), the drive element _Q1 is turned on again at timing _T7 . When the number of times of charging/discharging by the return current from the secondary circuit 3a is counted, the state where the charge stored in the power transmission side capacitor _C1 is empty in the third charge/discharge by the return current is an example of the timing of expiration of the drive period T defined in the vibration energy transmission system according to the second embodiment. At timing _T7 , the drive voltage E from the DC power source 5 is input in a stepwise manner, and the charge is stored in the power transmission side capacitor _C1 as electromagnetic energy.

When a new drive voltage E is input to the primary circuit 2b in a stepwise manner, the double resonance state of the resonant circuit in the primary circuit 2b and the resonant circuit in the secondary circuit 3a is temporarily cut off, and the double resonance is immediately resumed by a new excitation drive. When the new drive voltage E is input to the primary circuit 2b in a stepwise manner, the power transmitting capacitor _C1 is empty, so electromagnetic energy efficiently enters the power transmitting capacitor _C1 . In the new double resonance, the drive element _Q1 is then turned off, and after a certain time, the power transmitting control element _Q2 is turned on. The electromagnetic energy stored in the power transmitting capacitor _C1 is accumulated in the power transmitting coil _L1 through the power transmitting coil current _IL1 , and a transient response mutual induction is further performed from the primary circuit 2b to the secondary circuit 3a, and thus a new double resonance mode proceeds, but the following description will be omitted to avoid duplication. According to the energy vibration type transmission system of the second embodiment of the present invention, by periodically inputting a step-wise drive voltage E, a wavelet-shaped damped vibration is periodically excited with a drive period T, thereby realizing the repetition of a periodic energy super-resonance state, thereby enabling highly efficient contactless power transmission.

"Resonance" is a concept applied to a system that is freely vibrating. In contrast, in the energy vibration transmission system according to the second embodiment of the present invention, the free damped vibration of the primary circuit 2b is limited to a specific number of times by the drive period T, and the double resonance is temporarily cut off. In order to cut off the double resonance, a power transmission control element _Q2 and a drive element _Q1 are provided. In the energy vibration transmission system according to the second embodiment, in a state in which the double resonance of the energy packet vibration is cut off, it is possible to transmit it to the secondary circuit 3a by performing transient response mutual induction. Since the damped pseudo-sine wave is generated using the DC power supply 5, which has a simple and inexpensive control circuit configuration, and the transient response characteristics of the pseudo-sine wave can be used, an expensive AC power supply circuit that generates a sine wave vibration higher than the commercial frequency for the primary circuit 2b as in the conventional case is not required, and the circuit design is easy and less likely to break.

As described above, according to the vibration energy transmission system of the second embodiment, as in the vibration energy transmission system of the first embodiment, the control circuit and peripheral circuits can use a simple and inexpensive DC power supply 5, so that an expensive switching power supply is not required. The circuit configuration of the vibration energy transmission system of the second embodiment is simplified, the power loss on the control circuit side is minimized, and the circuit design is also easy. As a result, the overall configuration of the vibration energy transmission system is simplified, making it possible to reduce the weight and size and to increase the efficiency, and a wireless vibration energy transmission system with improved overall power transmission efficiency including the loss of the power supply circuit (zero circuit) can be manufactured at low cost. As described in the vibration energy transmission system of the first embodiment, the limit power of power transmission can be pushed up to a value that exceeds the limit power in the conventional AC theory, and in principle, pushed up to infinity, and the limit distance of power transmission can also be extended to infinity in principle. Furthermore, it is possible to increase the power transmission efficiency to a value close to 100% in principle. That is, according to the vibration energy transmission system of the second embodiment, the driving period of the power transmitting side can be selected to excite the system so that the packet-shaped electromagnetic energy oscillates by super-resonance, so that effective power transmission can be achieved even when the distance between the power transmitting coil _L1 and the power receiving coil _L2 is 40 mm or more. Also, according to the vibration energy transmission system of the second embodiment, even in a situation where the equivalent coupling coefficient k is unknown, the timing of turning on the driving element _Q1 can be selected to achieve optimal power transmission, and the circuit configuration can be simplified, making it possible to provide an vibration energy transmission system that can prevent damage to the driving element _Q1, which is a circuit element.

Third Embodiment
As shown in Fig. 20, the energy vibration transmission system according to the third embodiment of the present invention is a transmission system in which a wavelet-shaped electromagnetic energy is supplied to a power receiving circuit 27b from a drive control circuit 34b without contact using transient response mutual induction, and energy packets are caused to resonate multiple times between the drive control circuit 34b and the power receiving circuit 27b. The power receiving circuit 27b of the energy vibration transmission system according to the third embodiment is configured by adding a power receiving control element _Q3 to the energy vibration transmission system according to the second embodiment. As the drive element _Q1 , the power transmission control element _Q2 , and the power receiving control element _Q3 shown in Fig. 20, power semiconductor elements including thyristors such as GTO thyristors and SI thyristors as well as FETs, SITs, and BJTs similar to those in the energy vibration transmission systems according to the first and second embodiments are used. Due to the requirement for low internal resistance and availability in the market, it is considered industrially advantageous to adopt MOSFETs as the drive element _Q1 , the power transmission control element _Q2 , and the power receiving control element _Q3 of the mounting circuit shown in Fig. 20.

As explained in the first and second embodiments, the high power energy vibration transmission system generates a large amount of Joule heat. In the third embodiment, the energy vibration transmission system requires only three power semiconductor elements as the driving element _Q1 , the power transmission control element _Q2 , and the power reception control element _Q3 , so that a cooling structure that prevents the elements from being destroyed by heat can be designed easily, and the generation of floating resistance, floating capacitance, and floating inductance can be minimized. In addition, since the driving element _Q1 and the power transmission control element _Q2 only require simple on/off control, the voltage of the primary circuit 2b can be increased to easily suppress the generation of Joule heat.

In the implementation circuit shown in Fig. 20, the first freewheeling diode _FWD1 is connected in parallel between the source and drain of the MOSFET as the driving element _Q1 , the second freewheeling diode _FWD2 is connected in parallel between the source and drain of the MOSFET as the power transmission control element _Q2 , and the third freewheeling diode _FWD3 is connected in parallel between the source and drain of the MOSFET as the power reception control element _Q3 as protective elements. As shown in Fig. 20, the third freewheeling diode _FWD3 is provided in a direction to allow a freewheeling current from the power receiving coil _L2 to flow, and is therefore provided in the opposite direction to the second freewheeling diode _FWD2 . As in the circuits shown in Figs. 3A and 19, the power supply side diode _D1 is connected in series between the DC power supply 5 and the driving element _Q1 to prevent the freewheeling current from the power transmitting coil _L1 from returning to the DC power supply 5. In the implementation circuit shown in Fig. 20, the equivalent impedance _XLeq of the load 6 is also approximated and expressed by the charging capacitance _Cs .

As in the first and second embodiments, the energy vibration transmission system according to the third embodiment will be described assuming a transient response mutual induction under conditions corresponding to a coupling coefficient K _AC =0.6 according to AC theory, and assuming that the value of the charging voltage V _CS in the initial state is a sufficiently high voltage close to full charge.

First, at the same timing _T0 as shown in FIG. 8A, the power transmission control element _Q2 and the power reception control element _Q3 are turned off, and only the driving element _Q1 is turned on. The terminal voltage V _C1 of the power transmission side capacitor _C1 is charged to a constant voltage while ringing. At the timing _T0 , the terminal voltage V _C2 of the power reception side capacitor C is a negative value. As described in the energy vibration transmission system according to the second embodiment, in the case of the energy vibration transmission system according to the first embodiment, when the driving voltage E from the DC power source 5 is step-inputted at the timing _T0 , a very small current flows to the power transmission side coil _L1 side, but in the energy vibration transmission system according to the third embodiment, since the power transmission control element _Q2 is turned off, the charge is more effectively stored in the power transmission side capacitor _C1 . If the amount of charge flowing as electromagnetic energy when the drive voltage E from the DC power supply 5 is input in a stepwise manner is _q0 , then V _C1 = q ₀ /C ₁ , and therefore the value of the terminal voltage V _C1 of the power transmitting capacitor C ₁ is larger than in the case of the energy vibration type transmission system according to the first embodiment. After applying an initial voltage to the power transmitting capacitor C ₁ to store charge, the drive element Q ₁ is turned off. As described above, it is assumed that the charging voltage V _CS at this point is high.

At the same timing _T1 as shown in FIG. 8B, the driving element _Q1 is turned off, and then after a certain time, the power transmission control element _Q2 and the power reception control element _Q3 are simultaneously turned on. When the power transmission control element _Q2 is turned on, the electromagnetic energy stored in the power transmission side capacitor _C1 is stored in the power transmission side coil _L1 , and further, a transient response mutual induction occurs between the primary side circuit 2b and the secondary side circuit 3b. When the electromagnetic energy stored in the power transmission side capacitor _C1 moves to the power transmission side coil _L1 , the terminal voltage V _C1 becomes 0V after taking a negative maximum value. At the timing T ₀ to T ₁ , the first charging and discharging of the power transmission side capacitor _C1 is performed. The electromagnetic energy transmitted to the power reception side coil _L2 by the transient response mutual induction from the primary side circuit 2b to the secondary side circuit 3b charges the power reception side capacitor _C2 because the power reception control element _Q3 is on. When charging of the receiving capacitor _C2 begins, the terminal voltage V _C2 of the receiving capacitor C starts to increase from a negative maximum value and becomes a positive value. While the terminal voltage V _C2 is negative, the charging current I _CS does not flow, but when the terminal voltage V _C2 becomes positive at the timing _T2 similar to that shown in FIG. 8C, the charging current I _CS starts to rise.

At timing _T2 when the charging current I _CS starts to rise, the power transmission control element _Q2 and the power reception control element _Q3 are turned off. The power transmission control element _Q2 and the power reception control element _Q3 are turned off at the time when the terminal voltage V _C2 becomes maximum and the terminal voltage V _C1 becomes 0 V due to the transient response mutual induction between the primary circuit 2b and the secondary circuit 3b. In the case of the energy vibration type transmission system according to the second embodiment, at the stage when the terminal voltage V _C2 becomes maximum at timing _T2 , a very small current flows through the power reception coil _L2 , but in the energy vibration type transmission system according to the third embodiment, the power reception control element _Q3 is turned off, so that the charging current I _CS moves to the load 6 more effectively. The charging current I _CS increases even after the power transmission control element _Q2 and the power reception control element _Q3 are turned off, reaches a peak value, and then decreases to zero.

When the charging current I _CS starts to decrease, the maximum value of the terminal voltage V _C2 of the receiving capacitor C ₂ becomes a constant value that is slightly lower, forming a step-like (shoulder-shaped) waveform. Even after the charging current I _CS becomes zero, the value of the terminal voltage V _C2 remains lower than the maximum value when the power transmission control element Q ₂ is off. After a certain time has elapsed after the power transmission control element Q ₂ is turned off, the maximum value of the terminal voltage V _C2 decreases, but when the charging voltage V _CS is high, the amount of decrease in the maximum value of the terminal voltage V _C2 due to the charging current I _CS is small, and the effect on the transient response mutual induction between the primary circuit 2b and the secondary circuit 3b is small.

After the charging current I _CS becomes 0A, at the timing T ₃ similar to that shown in FIG. 9(d), when the power transmission control element Q ₂ and the power reception control element Q ₃ are simultaneously turned on again, charging of the power reception side capacitor C ₂ starts. The terminal voltage V _C1 of the power transmission side capacitor C ₁ starts decreasing from 0V to negative due to the electromagnetic coupling between the primary side circuit 2b and the secondary side circuit 3b. Furthermore, at the timing T ₄ similar to that shown in FIG. 9(e), discharging of the power reception side capacitor C ₂ starts, and a return current occurs due to the transient response mutual induction between the primary side circuit 2b and the secondary side circuit 3b. The power transmission side capacitor C ₁ is charged by the return current flowing through the power transmission side coil L ₁ , and the terminal voltage V _C1 becomes a positive value from a negative maximum value and starts to increase. At the timing T ₄ , the terminal voltage V _C2 starts decreasing, reaches a negative maximum value, and then becomes 0V.

At timing _T5 , similar to that shown in Figure 9(f), the terminal voltage V _C1 starts to decrease and becomes 0 V. From timing _T4 to _T5 , the second charging and discharging of the power transmitting capacitor _C1 is performed, during which electromagnetic energy is stored in the power receiving capacitor _C2 . From timing _T5 to _T6 , the electromagnetic energy stored in the power receiving capacitor _C2 is transferred to the load 6 and the power receiving coil _L2 , and the terminal voltage V _C2 of the power receiving capacitor _C2 decreases.

When the terminal voltage V _C2 of the secondary circuit 3a decreases, the electromagnetic energy stored in the receiving capacitor C2 flows back to the receiving coil _L2 as the receiving coil current I _L2 at timing _T6 to _T7 -ΔT similar to that shown in Figure 10(g _{). The electromagnetic energy stored in the receiving coil L2 flows back to the transmitting coil L1 of the} primary circuit 2a. At timing _T6 to _T7 -ΔT, the transmitting coil current I _L1 flowing back to the primary circuit 2a causes the electromagnetic energy to start flowing back to the transmitting capacitor _C1 , and the terminal voltage V _C1 of the transmitting capacitor _C1 starts to increase. With the driving element _Q1 in the off state, at the timing _T7 -ΔT shown in Figure 10 (h), the terminal voltage _VC1 of the power transmitting capacitor _C1 begins to decrease, and the electromagnetic energy stored in the power transmitting capacitor _C1 is transferred to the power transmitting coil _L1 as the power transmitting coil current _IL1 .

As described in the first embodiment of the vibration energy transmission system with reference to FIG. 10(h), the direction of current flow in the receiving coil _L2 at timings _T6 to _T7 -ΔT is opposite to the prediction in the case of differential connection in the normal AC theory. In the third embodiment of the vibration energy transmission system, the movement of electrons constituting the current in the receiving coil _L2 is delayed, and the mutual coupling of the magnetic flux between the transmitting coil _L1 and the receiving coil _L2 is loose. At timings _T6 to _T7 -ΔT, the transmitting capacitor _C1 is charged and discharged for the third time. At this time, the transfer electromagnetic energy _E13 stored in the transmitting coil _L1 cannot generate a transient response mutual induction between the primary circuit 2a and the secondary circuit 3a, and is not wirelessly transmitted to the receiving coil _L2 of the secondary circuit 3a. When the charge stored in the transmitting capacitor _C1 is depleted, the power transmission control element _Q2 is turned off.

In the vibration energy transmission system according to the third embodiment, the state in which the charge stored in the power transmission capacitor _C1 is emptied by the third charge and discharge is selected as the timing of expiration of the drive period T. By counting the number of charges and discharges by the return current from the secondary circuit 3b, the state in which the charge stored in the power transmission capacitor _C1 is emptied by the second charge and discharge by the return current is the appropriate timing of expiration of the drive period T in the vibration energy transmission system according to the third embodiment. At timing _T7 , the drive element _Q1 is turned on again, a new drive voltage E is stepped into the primary circuit 2b, and charge is stored in the power transmission capacitor _C1 as electromagnetic energy.

When a new drive voltage E is input to the primary circuit 2b in a stepwise manner, the double resonance state of the resonant circuit in the primary circuit 2b and the resonant circuit in the secondary circuit 3b is temporarily cut off, and the double resonance is immediately resumed by the new excitation drive. When the new drive voltage E is input to the primary circuit 2b in a stepwise manner, the power transmitting capacitor _C1 is empty, so electromagnetic energy efficiently enters the power transmitting capacitor _C1 from the DC power source 5. In the new double resonance, the drive element _Q1 is then turned off, and after a certain time, the power transmitting control element _Q2 and the power receiving control element _Q3 are simultaneously turned on. When the power transmitting control element _Q2 is turned on, the electromagnetic energy stored in the power transmitting capacitor _C1 is stored in the power transmitting coil _L1 , and further, a transient response mutual induction occurs between the primary circuit 2b and the secondary circuit 3b.

The electromagnetic energy transmitted to the power receiving coil _L2 by the transient response mutual induction from the primary circuit 2b to the secondary circuit 3b charges the power receiving capacitor _C2 because the power receiving control element _Q3 is in the on state. When the charging of the power receiving capacitor _C2 starts, the terminal voltage V _C2 of the power receiving capacitor C starts to increase from a negative maximum value and becomes a positive value. When the terminal voltage V _C2 becomes a positive value, the charging current I _CS starts to rise, and a new multiple resonance mode progresses, but the following description is omitted to avoid duplication. According to the energy vibration transmission system according to the third embodiment of the present invention, the drive voltage E is periodically stepped to periodically excite the wavelet-shaped damped oscillation with the drive period T, thereby realizing the repetition of the periodic energy multiple resonance state, and highly efficient contactless power transmission can be achieved.

In this way, even if the number of power receiving control elements _Q3 is increased by one, the waveforms showing the time changes (transient response) of the terminal voltages V _C1 and V _C2 in the transient response mutual induction between the primary circuit 2b and the secondary circuit 3b and the mode of the double resonance are almost the same as those in the energy vibration transmission systems according to the first and second embodiments. However, in the energy vibration transmission system according to the third embodiment, since there is a timing at which the power receiving control element _Q3 is turned off, the opportunity for return from the secondary circuit 3b to the primary circuit 2b is reduced by one, and the state in which the charge stored in the power transmitting capacitor _C1 becomes empty at the timing of the third charge/discharge is selected as the timing at which the drive period T expires.

"Resonance" is a concept used in an AC circuit that is freely vibrating, but in the energy vibration transmission system according to the third embodiment, the damped vibration of the pseudo-sine wave of the primary circuit 2b and the secondary circuit 3b is limited to a specific number of times, and the double resonance is temporarily cut off. In order to temporarily cut off the double resonance, the driving element _Q1 and the power transmission control element _Q2 are provided. Therefore, in the energy vibration transmission system according to the third embodiment, it is possible to transmit the electromagnetic energy of the primary circuit 2b to the secondary circuit 3b with high efficiency by performing transient response mutual induction in a state in which the double resonance of the energy packet vibration is cut off. Since the damped pseudo-sine wave is generated using the DC power supply 5 with a simple and inexpensive control circuit configuration and the electromagnetic energy can be transmitted using the transient response characteristics of the pseudo-sine wave, an expensive AC power supply circuit that generates a sine wave vibration higher than the commercial frequency is not required for the primary circuit 2b.

The vibration energy transmission system according to the third embodiment is a transmission system that supplies wavelet-shaped electromagnetic energy to a vehicle 31c having a power receiving circuit 27b with a built-in load (storage battery) 6 without contact. That is, the vibration energy transmission system according to the third embodiment has a power supply device 29b that supplies power to the power receiving circuit 27b, and a primary side operation unit 33 that is connected to the power supply device 29b and sends commands to the power supply device 29b. Figure 21 illustrates a schematic diagram showing that the primary side coil _L1 on the power supply device 29b side and the secondary side coil _L2 on the vehicle 31c side face each other, and the wavelet-shaped electromagnetic energy is transmitted from _the primary side coil _L1 to the secondary side coil _L2 without contact.

The power supply device 29b includes a power supply board 11 in which the primary coil _L1 is housed in a disk-shaped dielectric, a distance control mechanism (not shown) that is equipped with the power supply board 11 and controls the distance between the primary coil _L1 and the secondary coil _L2 , a drive control circuit 34b that controls the power supply current flowing through the primary coil _L1 and the distance control mechanism, a transmission data storage device (not shown) and a program storage device (not shown) connected to the drive control circuit 34b, the power supply board 11 whose transmission current is controlled by the drive control circuit 34b, and a primary communication unit 21. The secondary coil _L2 is housed in a power receiving board 12 made of a disk-shaped dielectric. FIG. 21 is an example, and it is also possible to omit the power supply board ₁₁ that houses the primary coil _L1 and the power receiving board 12 that houses the secondary coil L2, and use the primary coil _L1 and the secondary coil _L2 in a bare state. The secondary coil _L2 receives wavelet-shaped electromagnetic energy from the primary coil _L1 of the power supply panel 11 through electromagnetic induction and is fed with power.

The power supply panel 11 is installed or buried in the ground so that the upper surface of the power supply panel 11 is arranged parallel to the lower surface of the power receiving panel 12. The power supply device 29b is installed, for example, in a parking space, and while the vehicle 31c is parked, the power supply panel 11 faces the power receiving panel 12 to supply wavelet-shaped electromagnetic energy via the power receiving panel 12 to the power receiving circuit 27b mounted on the vehicle 31c.

The vehicle 31c is an automobile, such as an HEV, PEV, or EV, that runs on electromagnetic energy stored in a storage battery serving as a load 6. The load 6 is a storage battery that stores wavelet-shaped electromagnetic energy supplied from the power supply device 29b via the power receiving panel 12. The primary side operation unit 33 outputs a power supply start signal indicating the start of power supply or a power supply stop signal indicating the stop of power supply to the power supply device 29b, in response to an external operation.

As shown in Fig. 21, the vehicle 31c includes the secondary operation unit 23, the power receiving circuit 27b, the power receiving panel 12, the secondary communication unit 22, etc. The secondary operation unit 23 receives various operations by the driver and outputs a signal corresponding to the received operation to the power receiving circuit 27b. The power receiving circuit 27b controls the power receiving panel 12 and the secondary communication unit 22 to perform various processes associated with power supply or various processes associated with power supply stop based on the various signals input from the secondary operation unit 23. The power receiving panel 12 supplies the power received by the secondary coil _L2 to the load (storage battery) 6 according to the control of the power receiving circuit 27b.

The secondary communication unit 22 exchanges various information necessary for power supply with the primary communication unit 21. For example, the primary communication unit 21 transmits synchronization information related to the timing of the drive cycle T to the secondary communication unit 22. Conversely, the secondary communication unit 22 transmits received power information output from the power receiving circuit 27b to the primary communication unit 21 during power supply. In addition, the secondary communication unit 22 generates a power receiving possible signal that permits charging or a power receiving impossible signal that does not permit charging according to the control of the power receiving circuit 27b, and transmits the generated power receiving possible signal or power receiving impossible signal to the primary communication unit 21. Here, the power receiving impossible signal is transmitted when the load (storage battery) 6 is fully charged, etc.

The primary side communication unit 21 shown in FIG. 21 exchanges various information necessary for power supply with the secondary side communication unit 22. For example, the primary side communication unit 21 receives a power receiving possible signal or a power receiving impossible signal from the secondary side communication unit 22. The primary side communication unit 21 outputs the received power receiving possible signal or power receiving impossible signal to the drive control circuit 34b.

The drive control circuit 34b controls the power supply panel 11 to perform various controls related to power supply. For example, when a power supply start signal is input from the primary side operation unit 33, the drive control circuit 34b controls the power supply panel 11 to supply power at a set drive cycle T. The drive control circuit 34b also acquires the vibration characteristics of the energy returning from the power receiving circuit 27b and performs a process of calculating an optimal drive cycle T at which the transmission efficiency of power between the power supply panel 11 and the power receiving panel 12 is maximized. The drive control circuit 34b selects a drive timing at which the transmission efficiency is maximized from the acquired optimal drive cycle T, and controls the drive element _Q1 to operate the primary side circuit 2b at the selected drive timing. The drive control circuit 34b also controls the drive element Q1 not to start power supply or to stop power supply when a power supply stop signal is input from the primary side operation unit 33 or a power reception impossible signal is input from the primary side communication unit ₂₁ . The operation of the primary side circuit 2 b is driven and controlled by the drive timing of the drive control circuit 34 b , and wavelet-shaped electromagnetic energy is supplied from the power supply panel 11 to the power receiving panel 12 .

Therefore, according to the vibration energy transmission system according to the third embodiment of the present invention, as in the vibration energy transmission systems according to the first and second embodiments, the control circuit and peripheral circuits can use a simple and inexpensive DC power supply 5, so an expensive switching power supply is not required, the circuit configuration is simplified, and power loss on the control circuit side is also minimized. As a result, according to the vibration energy transmission system according to the third embodiment, the overall configuration of the vibration energy transmission system is simplified, making it possible to reduce the weight and size and increase the efficiency, and a wireless vibration energy transmission system with improved overall power transmission efficiency including the loss of the power supply circuit (zero circuit) can be manufactured at low cost. As described in the vibration energy transmission systems according to the first and second embodiments, according to the vibration energy transmission system according to the third embodiment, the circuit configuration is simplified, making it less likely to break and easier to design the circuit. In addition, it is possible to theoretically push up the limit power of power transmission to infinity, theoretically extend the limit distance of power transmission to infinity, and theoretically increase the power transmission efficiency to a value close to 100%. That is, according to the vibration energy transmission system of the third embodiment, the driving period of the power transmitting side can be selected to excite the system so that the packet-shaped electromagnetic energy oscillates by super-resonance, so that effective power transmission can be achieved even when the distance between the power transmitting coil _L1 and the power receiving coil _L2 is 40 mm or more. Also, according to the vibration energy transmission system of the third embodiment, even in a situation where the equivalent coupling coefficient k is unknown, the timing of turning on the driving element _Q1 can be selected to achieve optimal power transmission, and the circuit configuration can be simplified, making it possible to provide an vibration energy transmission system that can prevent damage to the driving element _Q1, which is a circuit element.

Fourth Embodiment
As shown in Fig. 22, the energy vibration transmission system according to the fourth embodiment of the present invention is a transmission system in which a wavelet-shaped electromagnetic energy is supplied to a power receiving circuit 27c from a drive control circuit 34b in a non-contact manner using transient response mutual induction, and energy packets are caused to resonate multiple times between the drive control circuit 34b and the power receiving circuit 27c. The power receiving circuit 27c of the energy vibration transmission system according to the fourth embodiment is configured by adding a load transfer control element _Q4 to the power receiving circuit 27b of the energy vibration transmission system according to the third embodiment. As the drive element _Q1 , power transmission control element _Q2 , power reception control element _Q3 , and load transfer control element _Q4 shown in Fig. 22, in addition to FET, SIT, and BJT, power semiconductor elements including thyristors such as GTO thyristors and SI thyristors can be used, as in the energy vibration transmission systems according to the first to third embodiments. Considering the requirement for low internal resistance, and due to current market availability, it is preferable to employ MOSFETs as the driver element _Q1 , the power transmission control element _Q2 , the power reception control element _Q3 , and the load transfer control element _Q4 in the implementation circuit shown in FIG. 22.

In the energy vibration type transmission system according to the fourth embodiment, only four power semiconductor elements are required as the driving element _Q1 , the power transmission control element _Q2 , the power reception control element _Q3 , and the load transfer control element _Q4 , so that a cooling structure for preventing the generation of Joule heat can be designed easily, and the generation of floating resistance, floating capacitance, and floating inductance can be minimized. In addition, since only simple control for controlling the driving element _Q1 and the power transmission control element _Q2 to be on/off is required, a design for suppressing the generation of Joule heat by increasing the voltage of the primary side circuit 2b can be easily designed.

In the mounting circuit shown in Fig. 22, the first freewheeling diode _FWD1 is connected in parallel between the source and drain of the MOSFET as the driving element _Q1 , the second freewheeling diode _FWD2 is connected in parallel between the source and drain of the MOSFET as the power transmission control element _Q2 , the third freewheeling diode _FWD3 is connected in parallel between the source and drain of the MOSFET as the power reception control element _Q3 , and the fourth freewheeling diode _FWD4 is connected in parallel between the source and drain of the MOSFET as the fourth semiconductor element _Q4 as protective elements. As shown in Fig. 22, the third freewheeling diode _FWD3 is provided in a direction to allow a freewheeling current from the power reception coil _L2 to flow, so that the second freewheeling diode _FWD2 is provided in the opposite direction to the freewheeling diode FWD3, as in Fig. 20. As in the circuits shown in Figs. 3A, 19, and 20, the power supply side diode _D1 is connected in series between the DC power supply 5 and the driving element _Q1 to prevent the freewheeling current from the power transmission coil L1 from returning to the DC power supply 5 _. In the implementation circuit shown in FIG. 22, the equivalent impedance X _Leq of the load 6 is also approximated and expressed by the charge capacitance C _s .

As shown in Fig. 22, even if the load transfer control element _Q4 is added to the energy vibration transmission system according to the third embodiment, the essence of the transient response mutual induction remains unchanged, and the basic operation and the waveforms showing the time changes (transient response) of the terminal voltages V _C1 and V _C2 in the transient response mutual induction between the primary circuit 2b and the secondary circuit 3b and the mode of the double resonance are almost the same. That is, at the same timing T ₀ as shown in Fig. 8(a), the power transmission control element Q ₂ , the power reception control element Q ₃ and the load transfer control element Q ₄ are turned off, and only the driving element Q ₁ is turned on. The terminal voltage V _C1 of the power transmission side capacitor C ₁ is charged to a constant voltage while ringing. At the timing T ₀ , the terminal voltage V _C2 of the power reception side capacitor C is a negative value. As described in the vibration energy transmission system according to the second embodiment, in the vibration energy transmission system according to the fourth embodiment, the power transmission control element _Q2 is in the off state, so that no current flows to the power transmission coil _L1 side, and charge is stored more effectively in the power transmission capacitor _C1 . After applying an initial voltage to the power transmission capacitor _C1 to store charge, the driving element _Q1 is turned off.

At timing _T1 similar to that shown in FIG. 8(b), the driving element _Q1 is turned off, and then after a certain time, the power transmission control element _Q2 and the power reception control element _Q3 are simultaneously turned on. The load transfer control element _Q4 is in the off state. When the power transmission control element _Q2 is turned on, the electromagnetic energy stored in the power transmission side capacitor _C1 is stored in the power transmission side coil _L1 , and further, a transient response mutual induction occurs between the primary side circuit 2b and the secondary side circuit 3b. When the electromagnetic energy stored in the power transmission side capacitor _C1 moves to the power transmission side coil _L1 , the inter-terminal voltage V _C1 becomes 0V after reaching a negative maximum value. The first charging and discharging of the power transmission side capacitor _C1 is performed from timing T ₀ to T ₁ . Electromagnetic energy transmitted to the power receiving coil _L2 by the transient response mutual induction from the primary circuit 2b to the secondary circuit 3b charges the power receiving capacitor _C2 because the power receiving control element _Q3 is in the ON state.

In the case of the vibration energy transmission system according to the third embodiment, at the timing _T1 , a very small amount of current flows to the load 6 side at the same time as the receiving capacitor _C2 is charged. However, in the vibration energy transmission system according to the fourth embodiment, since the load transfer control element _Q4 is in the OFF state, no current flows to the load 6 side, and charge is stored more effectively in the receiving capacitor _C2 . When charging of the receiving capacitor _C2 starts, the terminal voltage V _C2 of the receiving capacitor C starts to increase from a negative maximum value and becomes a positive value. When the terminal voltage V _C2 becomes a positive value, if the load transfer control element _Q4 is turned ON, the charging current I _CS starts to flow to the load 6.

At timing _T2 when the load transfer control element _Q4 is turned on, the power transmission control element _Q2 and the power reception control element _Q3 are turned off. The power transmission control element _Q2 and the power reception control element _Q3 are turned off at the time when the terminal voltage V _C2 becomes maximum and the terminal voltage V _C1 becomes 0 V due to the transient response mutual induction between the primary circuit 2b and the secondary circuit 3b. As in the energy vibration type transmission system according to the third embodiment, the power reception control element _Q3 is turned off, so that the charging current I _CS moves to the load 6 more effectively. The charging current I _CS increases even after the power transmission control element _Q2 and the power reception control element _Q3 are turned off, reaches a peak value, and then decreases to zero.

When the charging current I _CS starts to decrease, the maximum value of the inter-terminal voltage V _C2 becomes a slightly lower constant value, forming a step-like (shoulder-shaped) waveform. Even after the charging current I _CS becomes zero, the value of the inter-terminal voltage V _C2 remains lower than the maximum value when the power transmission control element _Q2 is off. After a certain time has passed since the power transmission control element _Q2 was turned off, the maximum value of the inter-terminal voltage V _C2 decreases, but when the charging voltage V _CS is high, the amount of decrease in the maximum value of the inter-terminal voltage V _C2 due to the charging current I _CS is small, and the effect on the transient response mutual induction between the primary circuit 2b and the secondary circuit 3b is small.

After the charging current I _CS becomes 0 A, at the same timing T ₃ as shown in FIG. 9(d), the power transmission control element Q ₂ and the power reception control element Q ₃ are simultaneously turned on again while maintaining the on state of the load transfer control element Q ₄ , and charging of the power reception side capacitor C ₂ starts. The terminal voltage V _C1 of the power transmission side capacitor C ₁ starts decreasing from 0 V to a negative value due to the electromagnetic coupling between the primary side circuit 2 b and the secondary side circuit 3 b. Furthermore, at the same timing T ₄ as shown in FIG. 9(e), when the load transfer control element Q ₄ is turned off and the power reception side capacitor C ₂ starts discharging, a reflux occurs due to the mutual induction of the transient response between the primary side circuit 2 b and the secondary side circuit 3 b. In the energy vibration transmission system according to the fourth embodiment, since the load transfer control element Q ₄ is in the off state, no current flows to the load 6, and the current flows more effectively to the power reception side coil L ₂ . The power transmitting capacitor _C1 is charged by the return current flowing through the power transmitting coil _L1 due to mutual induction between the power receiving coil _L2 and the power transmitting coil _L1 , and the terminal voltage V _C1 starts to increase from a negative maximum value to a positive value. At time _T4 , the terminal voltage V _C2 starts to decrease, reaches a negative maximum value, and then becomes 0 V.

At timing _T5 , similar to that shown in Figure 9(f), the terminal voltage V _C1 begins to decrease and becomes 0 V. From timing _T4 to _T5 , the power transmitting capacitor _C1 is charged and discharged for the second time, and electromagnetic energy is stored in the power receiving capacitor _C2 at this time. Because the load transfer control element _Q4 is in the off state, no current flows to the load 6, and charge is stored more effectively in the power receiving capacitor _C2 . When the electromagnetic energy stored in the power receiving capacitor _C2 moves to the power receiving coil _L2 from timing _T5 to _T6 , the terminal voltage V _C2 of the power receiving capacitor _C2 decreases.

When the terminal voltage V _C2 of the secondary circuit 3a decreases, the electromagnetic energy stored in the receiving capacitor C2 flows back to the receiving coil _L2 as the receiving coil current I _L2 at timing T to _T -ΔT similar to that shown in Figure 10(g). The electromagnetic energy stored in the receiving coil _L2 flows back to the transmitting coil _L1 of the primary circuit 2b. At timing T _to T _-ΔT , the transmitting coil current I _L1 flowing back to the primary circuit _2b causes the electromagnetic energy to start flowing back to _the transmitting capacitor _C1 , and the terminal voltage V _C1 of the transmitting capacitor _C1 starts to increase. With the driving element _Q1 in the off state, at the timing _T7 -ΔT shown in Figure 10 (h), the terminal voltage _VC1 of the power transmitting capacitor _C1 begins to decrease, and the electromagnetic energy stored in the power transmitting capacitor _C1 is transferred to the power transmitting coil _L1 as the power transmitting coil current _IL1 .

As described in the first embodiment of the vibration energy transmission system with reference to FIG. 10(h), the direction of current flow in the receiving coil _L2 at timings _T6 to _T7 -ΔT is opposite to the prediction in the case of differential connection in the normal AC theory. There is a delay in the movement of electrons constituting the current in the receiving coil _L2 , and in the fourth embodiment of the vibration energy transmission system, the mutual coupling of the magnetic flux between the transmitting coil _L1 and the receiving coil _L2 is loose. At timings _T6 to _T7 -ΔT, the transmitting capacitor _C1 is charged and discharged for the third time, but at this time, the transfer electromagnetic energy _E13 stored in the transmitting coil _L1 cannot cause a transient response mutual induction between the primary circuit 2a and the secondary circuit 3a, and is not wirelessly transmitted to the receiving coil _L2 of the secondary circuit 3a. When the charge stored in the transmitting capacitor _C1 is empty, the power transmission control element _Q2 is turned off. In the vibration energy transmission system according to the fourth embodiment, the state in which the charge stored in the power transmission side capacitor _C1 is depleted in the third charging and discharging is selected as the timing of expiration of the drive period T. When the number of charging and discharging by the return current from the secondary circuit 3c is counted, the state in which the charge stored in the power transmission side capacitor _C1 is depleted in the second charging and discharging by the return current becomes the appropriate timing of expiration of the drive period T in the vibration energy transmission system according to the fourth embodiment.

At timing _T7 , the driving element _Q1 is turned on again, a new driving voltage E is input from the DC power supply 5 to the primary circuit 2b, and an electric charge is stored as electromagnetic energy in the power transmission capacitor _C1 . By inputting the new driving voltage E to the primary circuit 2b, the double resonance state of the resonant circuit in the primary circuit 2b and the resonant circuit in the secondary circuit 3c is temporarily cut off, and the double resonance is immediately resumed by the new excitation drive. When the new driving voltage E is input to the primary circuit 2b, the power transmission capacitor _C1 is empty, so electromagnetic energy efficiently enters the power transmission capacitor _C1 from the DC power supply 5. Next, after the driving element _Q1 is turned off, after a certain time, the power transmission control element _Q2 and the power reception control element _Q3 are simultaneously turned on. At this time, the load transfer control element _Q4 is in the off state.

In the new double resonance, when the power transmission control element _Q2 is turned on, the electromagnetic energy stored in the power transmission capacitor _C1 is stored in the power transmission coil _L1 , and further, a transient response mutual induction occurs between the primary circuit 2b and the secondary circuit 3b. The electromagnetic energy transmitted to the power reception coil _L2 by the transient response mutual induction from the primary circuit 2b to the secondary circuit 3b charges the power reception capacitor _C2 because the power reception control element _Q3 is turned on. When charging of the power reception capacitor _C2 starts, the terminal voltage V _C2 of the power reception capacitor C starts to increase from a negative maximum value and becomes a positive value. When the terminal voltage V _C2 becomes a positive value, the charging current I _CS starts to rise, and the load transfer control element _Q4 is turned on to supply the charging current I _CS to the load 6, and so on, and thus a new double resonance mode proceeds, but the following description will be omitted to avoid duplication. According to the energy vibration type transmission system of the fourth embodiment of the present invention, by periodically inputting a step-wise drive voltage E, a wavelet-shaped damped vibration is periodically excited with a drive period T, thereby realizing the repetition of a periodic energy super-resonance state, thereby enabling highly efficient contactless power transmission.

In this way, even if the load transfer control element _Q4 is increased by one, the waveforms showing the time changes (transient response) of the terminal voltages V _C1 and V _C2 in the transient response mutual induction between the primary circuit 2b and the secondary circuit 3b and the mode of the double resonance are almost the same as those of the vibration energy transmission system according to the third embodiment. However, in the vibration energy transmission system according to the fourth embodiment, as in the vibration energy transmission system according to the third embodiment, there is a timing at which the power receiving control element _Q3 is turned off, so that the opportunity for return from the secondary circuit 3b to the primary circuit 2b is reduced by one, and the state in which the charge stored in the transmission side capacitor _C1 becomes empty at the timing of the third charge/discharge is selected as the timing of expiration of the drive period T.

22, in a configuration having four switching elements _Q1 , _Q2 , _Q3 , and _Q4 , namely, a driving element _Q1 , a power transmission control element _Q2 , a power reception control element _Q3 , and a load transfer control element _Q4 , when the driving element _Q1 and the load transfer control element _Q4 are turned off and the power transmission control element _Q2 and the power reception control element _Q3 are turned on, the circuit on the DC power source 5 side and the circuit on the load 6 side are isolated from the primary circuit 2b and the secondary circuit 3b, respectively, so that the primary circuit 2b and the secondary circuit 3b can vibrate freely. In other words, the LC resonant circuit of the primary circuit 2b and the LC resonant circuit of the secondary circuit 3b can be treated as circuits coupled by a mutual inductance M, so that the concept of multiple resonance in AC theory can be adopted. That is, at the timing when the driving element _Q1 and the load transfer control element _Q4 are in the cutoff state and the power transmission control element _Q2 and the power reception control element _Q3 are in the conduction state, the transmission efficiency due to the transient response mutual induction can be considered using the coupling equations (2) and (3) already described.

However, in the implemented circuit, the on-resistance of the power semiconductor elements used for each of the four switches, the driver element _Q1 , the power transmission control element _Q2 , the power reception control element _Q3 , and the load transfer control element _Q4 , must be taken into consideration, and therefore the coupling equations (2) and (3) cannot be used to describe it. Therefore, in the operation when the driver element _Q1 and the load transfer control element _Q4 are in the cutoff state and the power transmission control element _Q2 and the power reception control element _Q3 are in the conductive state, it is necessary to consider the LCR resonant circuit of the primary circuit 2b and the LCR resonant circuit of the secondary circuit 3b as a circuit coupled by a dynamic mutual inductance M.

In addition, since it is necessary to consider the energy transmission in the transient response such as the step response when the driving element _Q1 and the load transfer control element _Q4 are in the conductive state, not all of the vibrational energy transmission system according to the fourth embodiment of the present invention can _be interpreted by the conventional AC theory. That is, the conventional AC theory of the sine wave can be used in the free vibration region, but in _the operating environment of the vibrational energy transmission system according to the fourth embodiment in which the boundary conditions of the circuit are changed from moment to moment using _four switching elements _Q1 , _Q2 , _Q3 , and _Q4 , namely the driving element _Q1 , the power transmission control element Q2, the power reception control element Q3, and the load transfer control element Q4, it is necessary to analyze including the transient response such as the sawtooth wave rising characteristic. According to the vibrational energy transmission system according to the fourth embodiment, the system can be excited by selecting the driving period of the power transmission side so that the packet-shaped electromagnetic energy oscillates by multiple resonance, so that effective power transmission can be achieved even when the distance between the power transmission side coil _L1 and the power reception side coil _L2 is 40 mm or more. Moreover, according to the energy vibration transmission system of the fourth embodiment, even in a situation where the equivalent coupling coefficient k is unknown, by selecting the timing of turning on the driving element _Q1 , it is possible to provide an energy vibration transmission system in which optimal power transmission is possible, the circuit configuration is simplified, and damage to the driving element _Q1 , which is a circuit element, can be prevented.

Other Embodiments
As described above, the present invention has been described by the first to fourth embodiments, but the description and drawings forming part of this disclosure should not be understood as limiting the present invention. Various alternative embodiments, examples, and operation techniques will become apparent to those skilled in the art from this disclosure. As described above, the present invention includes various embodiments not described in this specification and drawings, and the technical scope of the present invention is defined only by the invention-specifying matters related to the scope of the claims that are appropriate from the above description.

2a, 2b...Primary side circuit, 3a, 3b, 3c...Secondary side circuit, 5...DC power supply, 6...Load, 21...Primary side communication unit, 22...Secondary side communication unit, 23...Secondary side operation unit, 27a, 27b, 27c...Power receiving circuit, 28...Detector, 29a, 29b...Power supply device, 30...Flat surface, 31a, 31b, 31c...Vehicle, 32...Gap control mechanism, 33...Primary side operation unit, 34a, 34b...Drive control circuit, 71...P-type substrate, 72...Source region, 73...Drain region, 81...Gate oxide film, 82...Source 3: primary side switching element drive circuit, 340a: primary side switching element drive circuit, 340b: transmission interval control circuit, 341: arithmetic logic circuit, 342a: transmission data storage device, 342b: program storage device, 343: output device, 344: calculation sequence control circuit, 345: return voltage measurement control circuit, 346: energy peak counting circuit, 347: coupling coefficient calculation circuit, 348: transmission condition setting circuit, 349a: A bus, 349b: B bus

Claims (4)

An energy vibration type transmission system using double resonance between a primary circuit and a secondary circuit,
a primary circuit including a power transmission side capacitor that accumulates electromagnetic energy that has returned from the secondary side circuit to the primary side circuit due to the double resonance, a power transmission side coil that is connected in parallel to the power transmission side capacitor, and a detector that detects a voltage between terminals of the power transmission side capacitor;
A driving element that configures a circuit that inputs an intermittent DC voltage stepwise between one terminal and the other terminal of the power transmission side capacitor;
the secondary circuit including a power receiving coil facing the power transmitting coil and a power receiving capacitor connected in parallel to the power receiving coil;
a load that receives electrostatic energy from the power-receiving-side capacitor in a circuit that connects terminals of the power-receiving-side capacitor;
a primary side switching element drive circuit for sending a control signal to a control terminal of the drive element;
an arithmetic and logic circuit that detects, from a change in an output voltage of the detector, a timing at which charging / discharging of the power transmitting side capacitor by the electromagnetic energy that has returned from the secondary side circuit to the power transmitting side capacitor via the power transmitting side coil has been completed at least once, and causes the primary side switching element drive circuit to periodically output the control signal to the control terminal in a drive cycle that has the detected timing as a drive time;
An energy vibration type transmission system comprising: The energy vibration transmission system according to claim 1, characterized in that the arithmetic logic circuit has an energy peak counting circuit that determines the drive period from the zero crossing time when the change in the output voltage of the detector falls below 0 V. The energy vibration type transmission system according to claim 2, characterized in that a transmission data storage device that stores the zero crossing time measured by the arithmetic logic circuit is connected to the arithmetic logic circuit. The energy vibration type transmission system according to claim 3, characterized in that the transmission data storage device further stores data indicating the relationship between the transmission distance defined between the power transmitting coil and the power receiving coil and the drive period T, and data indicating the relationship between the equivalent coupling coefficient k between the power transmitting coil and the power receiving coil and the drive period.

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