JP2015122903A - Switching power-supply device and power conversion device - Google Patents

Abstract

PROBLEM TO BE SOLVED: To improve the efficiency and characteristics of a switching power-supply device of a secondary-side phase shift system.SOLUTION: A bridge circuit converts a DC voltage into an AC voltage. A transformer includes a primary winding N1 connected to the bridge circuit and a secondary winding N2 electromagnetically coupled to the first winding N1. A rectifier circuit rectifies an AC voltage inputted from the secondary winding N2. A smoothing circuit smooths the voltage rectified by the rectifier circuit. The rectifier circuit includes a secondary-side switching element inserted into a current path. The on-period of the secondary-side switching element is set to be longer than the on-period of a primary-side switching element included in the bridge circuit.

Description

  The present invention relates to a switching power supply device and a power conversion device including the switching power supply device.

  In recent years, in order to reduce switching loss of a switching element and to suppress electromagnetic induction noise, a soft switching technique for switching the switching element to zero voltage switching (ZVS) or zero current switching (ZCS) has been studied. The characteristics of the DC-DC converter have been improved by applying the soft switching technique. There is a phase shift method as a control method of a DC-DC converter to which soft switching is applied.

  In the phase shift method, the primary side phase shift method that adjusts the phase of the control signal input to the switching element in the bridge circuit on the primary side, and the switching element in the secondary rectifier circuit, the secondary side There is a secondary side phase shift method in which the phase of the control signal input to the switching element is adjusted to control the stabilization of the output voltage (see, for example, Patent Document 1).

JP 2002-238257 A

  The present inventors have found a technique for improving efficiency and characteristics over the conventional secondary phase shift system. The present invention has been made in view of such circumstances, and an object thereof is to provide a technique for improving the efficiency and characteristics of a secondary-side phase shift switching power supply device.

  In order to solve the above problems, a switching power supply device according to an aspect of the present invention includes a bridge circuit that converts a DC voltage into an AC voltage, a primary winding connected to the bridge circuit, and an electromagnetic coupling with the primary winding. A transformer including a secondary winding, a rectifier circuit that rectifies an AC voltage input from the secondary winding, and a smoothing circuit that smoothes the voltage rectified by the rectifier circuit. The rectifier circuit includes a secondary side switching element inserted in a current path. The on period of the secondary side switching element is set longer than the on period of the primary side switching element included in the bridge circuit.

  Another aspect of the present invention is a power converter. This device includes a switching power supply device and a control device that controls the primary side switching element and the secondary side switching element.

  It should be noted that any combination of the above-described constituent elements and a conversion of the expression of the present invention among methods, circuits, devices, systems, etc. are also effective as an aspect of the present invention.

  ADVANTAGE OF THE INVENTION According to this invention, the efficiency and characteristic of a switching power supply device of a secondary side phase shift system can be improved.

It is a figure which shows the structure of the power converter device which concerns on embodiment of this invention. It is a figure which shows an example of the control method which concerns on the comparative example of the switching power supply device of FIG. It is a figure which shows an example of the control method which concerns on the Example of the switching power supply device of FIG. It is a figure which shows an example (another state) of the control method which concerns on the comparative example of the switching power supply device of FIG. It is a figure which shows an example (another state) of the control method which concerns on the Example of the switching power supply device of FIG. It is a figure for demonstrating the 1st structural example of the control apparatus of FIG. It is a figure for demonstrating the 2nd structural example of the control apparatus of FIG. It is a figure which shows an example (detail) of the control method which concerns on the comparative example of the switching power supply device of FIG. It is a figure which shows another example of the control method which concerns on the Example of the switching power supply device of FIG. It is a figure which shows the structure of the power converter device which concerns on the modification 1. As shown in FIG. It is a figure which shows the structure of the power converter device which concerns on the modification 2. It is a figure which shows the structure of the power converter device which concerns on the modification 3. It is a figure which shows the structure of the power converter device which concerns on the modification 4. It is a figure which shows the structure of the power converter device which concerns on the modification 5. FIG. It is a figure which shows the structure of the power converter device which concerns on the modification 6. It is a figure which shows the structure of the electrical storage system in which the power converter device which concerns on this Embodiment is used. It is a figure which shows the structure of the vehicle by which the power converter device which concerns on this Embodiment is used. It is a figure which shows the structure of the charger in which the power converter device which concerns on this Embodiment is used.

  FIG. 1 is a diagram showing a configuration of a power conversion device 100 according to an embodiment of the present invention. The power conversion device 100 includes a switching power supply device 10, an output voltage detection circuit 11, and a control device 20. The switching power supply apparatus 10 is a secondary side phase shift type insulation type DC-DC converter, and is a full bridge circuit, a first coil L1, a second coil L2, a transformer T, a third coil L3, a fourth coil L4, two Includes a secondary rectifier circuit and a smoothing circuit.

  The full bridge circuit converts the DC voltage supplied from the first DC power supply E1 to the fourth DC power supply E4 into an AC voltage. The full bridge circuit includes a first switching element S1, a second switching element S2, a third switching element S3, and a fourth switching element S4 that are connected in a full bridge. Specifically, a first arm including a first switching element S1 on the upper side and a third switching element S3 on the lower side, and a second arm including a second switching element S2 on the upper side and a fourth switching element S4 on the lower side. The first arm and the second arm are connected in parallel.

  A first diode D1 and a first capacitor C1 are connected in parallel with the first switching element S1. Similarly, the second switching element S2 to the fourth switching element S4 are connected in parallel with the second diode D2 to the fourth diode D4 and the second capacitor C2 to the fourth capacitor C4, respectively. The first diode D1 to the fourth diode D4 are reverse-biased to the first switching element S1 to the fourth switching element S4, respectively. The first capacitor C1 to the fourth capacitor C4 are snubber capacitors. For the first switching element S1 to the fourth switching element S4, for example, semiconductor switching elements such as MOSFETs and IGBTs are used. FIG. 1 shows an example using an n-channel MOSFET. A p-channel MOSFET may be used for the first switching element S1 and the second switching element S2.

  In the present embodiment, the primary side is constituted by a partial resonance type full bridge circuit. The partial resonance type full bridge circuit performs resonance commutation only at the time of switching, and operates in non-resonance in other modes. In FIG. 1, a resonance pole type configuration that can ensure zero voltage commutation even at light load is adopted. The first coil L1 is connected between a node between the first DC power supply E1 and the third DC power supply E3 and a node between the first switching element S1 and the third switching element S3. The second coil L2 is connected between a node between the second DC power supply E2 and the fourth DC power supply E4 and a node between the second switching element S2 and the fourth switching element S4.

  The first coil L1 and the second coil L2 are auxiliary inductors for resonance, and current is supplied from the first DC power supply E1 to the fourth DC power supply E4. Since the load current becomes small at light load, it may not be possible to secure a current necessary for zero voltage switching of the first switching element S1 to the fourth switching element S4. In order to perform zero voltage switching, the charge in the capacitor connected in parallel to the switching element needs to be zero at the time of switching. However, if the load current is small, the capacitor may not be completely discharged during the partial resonance period. On the other hand, as shown in FIG. 1, if the 1st coil L1 and the 2nd coil L2 are installed, the zero voltage switching of 1st switching element S1-4th switching element S4 can be ensured.

  The transformer T is a high-frequency transformer including a primary winding N1 and a secondary winding N2. The primary winding N1 and the secondary winding N2 are coupled by electromagnetic induction. The transformer T insulates the primary side and the secondary side, and transforms according to the winding ratio of the primary winding N1 and the secondary winding N2. Both ends of the primary winding N1 are connected to both output ends of the full bridge circuit. A third coil L3 and a fourth coil L4 are provided at both ends of the secondary winding N2, respectively. The third coil L3 and the fourth coil L4 may be a leakage inductance of the secondary winding N2, or may be an inductor element.

  The secondary side rectifier circuit rectifies the AC voltage input from the secondary winding N2. The secondary side rectifier circuit includes a fifth diode D5, a sixth diode D6, a seventh diode D7, an eighth diode D8, a fifth switching element S5, and a sixth switching element S6. The fifth diode D5 to the eighth diode D8 are connected by a full bridge to form a rectifier circuit. The fifth switching element S5 and the sixth switching element S6 are inserted into the current path of the rectifier circuit, and adjust the amount of power extracted from the transformer T. For the fifth switching element S5 and the sixth switching element S6, semiconductor switching elements such as MOSFET and IGBT are used, for example.

  The smoothing circuit smoothes the voltage rectified by the secondary side rectifier circuit. The smoothing circuit in FIG. 1 includes an LC filter including a fifth coil L5 and a fifth capacitor C5. The configuration of the smoothing circuit in FIG. 1 is an example, and other configurations may be used.

  The output voltage detection circuit 11 detects the output voltage of the switching power supply device 10 to be supplied to the load 30. In FIG. 1, the voltage across the fifth capacitor C5 is detected as the output voltage. The output voltage detection circuit 11 can be constituted by, for example, an error amplification circuit. The error amplifier circuit is composed of a combination of an operational amplifier and a passive element. The output voltage detection circuit 11 outputs the detected voltage to the control device 20.

  The control device 20 drives the switching power supply device 10 by controlling on / off of the first switching element S1 to the sixth switching element S6. The control device 20 adaptively changes the phases of the fifth switching element S5 and the sixth switching element S6 according to the output voltage of the switching power supply device 10 supplied from the output voltage detection circuit 11. Thereby, the output voltage of the switching power supply device 10 is stabilized. Details of the configuration and operation of the control device 20 will be described later.

  FIG. 2 is a diagram illustrating an example of a control method according to a comparative example of the switching power supply device 10 of FIG. The control device 20 supplies primary side drive signals to the control terminals (gate terminals in the case of FETs) of the first switching element S1 to the fourth switching element S4, and the control terminals of the fifth switching element S5 and the sixth switching element S6. To supply a secondary drive signal.

  A forward current flows through the transformer T when the first switching element S1 and the fourth switching element S4 are in the on state and the second switching element S2 and the third switching element S3 are in the off state. On the other hand, a reverse current flows through the transformer T when the second switching element S2 and the third switching element S3 are in the on state and the first switching element S1 and the fourth switching element S4 are in the off state. The first switching element S1 and the fourth switching element S4 function as a primary-side forward switching element that is inserted into a path that supplies a forward current to the primary winding N1. On the other hand, the second switching element S2 and the third switching element S3 function as a primary-side reverse switching element inserted in a path for supplying a reverse current to the primary winding N1.

  In addition, a dead time is provided when switching between a period in which a forward current flows and a period in which a reverse current flows. During the dead time, the first switching element S1 to the fourth switching element S4 are all turned off. The capacitance component and the inductance component resonate during the dead time, and the capacitor connected in parallel to the switching element that is turned on next is discharged.

  The control device 20 drives the primary-side forward switching element and the primary-side reverse switching element with a fixed duty ratio and a fixed phase. Specifically, the primary side forward switching element and the primary side reverse switching element are driven complementarily with a duty of 50%, excluding the dead time. In the present embodiment, since the output voltage is adjusted on the secondary side, the duty ratio and phase of the primary-side forward switching element and the primary-side reverse switching element are fixed. The full bridge circuit on the primary side always outputs power to the transformer T with full power.

  The forward current also flows through the secondary side rectifier circuit during at least a part of the period during which the forward current flows through the transformer T. Specifically, the forward current flows in the secondary rectifier circuit during the period in which the sixth switching element S6 is controlled to be in the ON state during the period in which the forward current flows in the transformer T. During the period in which the forward current flows, the sixth diode D6 and the seventh diode D7 are turned on, the fifth diode D5 and the eighth diode D8 are turned off, the sixth switching element S6 is turned on, and the fifth switching element S5 is turned off. It becomes a state.

  On the other hand, the reverse current also flows in the secondary rectifier circuit during at least a part of the period in which the reverse current flows in the transformer T. Specifically, the reverse current flows in the secondary rectifier circuit during the period in which the fifth switching element S5 is controlled to be in the ON state during the period in which the reverse current flows in the transformer T. During the period in which the reverse current flows, the fifth diode D5 and the eighth diode D8 are turned on, the sixth diode D6 and the seventh diode D7 are turned off, the fifth switching element S5 is turned on, and the sixth switching element S6 is turned off. It becomes a state.

  That is, the sixth diode D6 and the seventh diode D7 conduct the forward current from the secondary winding N2, and block the reverse current from the secondary winding N2. On the other hand, the fifth diode D5 and the eighth diode D8 block the forward current from the secondary winding N2 and conduct the reverse current from the secondary winding N2. Thereby, the AC voltage supplied from the secondary winding N2 is full-wave rectified.

  The fifth switching element S5 functions as a secondary-side forward switching element inserted in a forward current path including the fifth diode D5 and the eighth diode D8. On the other hand, the sixth switching element S6 functions as a secondary-side reverse switching element inserted in a reverse current path including the sixth diode D6 and the seventh diode D7.

  The control device 20 drives the secondary-side forward switching element and the secondary-side backward switching element with a fixed duty ratio and a varying phase. Specifically, the secondary-side forward switching element and the secondary-side reverse switching element are adaptively phase-shifted while being complementarily driven with a duty of 50%. In this embodiment, since the phase shift method is adopted instead of the PWM method, the control device 20 controls the phase rather than the duty ratio of the secondary forward switching element and the secondary reverse switching element. That is, the phases of the secondary forward switching element and the secondary reverse switching element are adaptively changed according to the output voltage of the switching power supply device 10.

  More specific description will be given below. The control device 20 determines a phase delay amount (hereinafter referred to as a phase difference) of the secondary forward switching element and the secondary reverse switching element with respect to the phase of the primary forward switching element and the primary reverse switching element. By changing, the output voltage of the switching power supply device 10 is stabilized. When the output voltage of the switching power supply device 10 increases, the control device 20 delays the phases of the secondary-side forward switching element and the secondary-side reverse switching element to increase the phase difference, and reduces the amount of power extracted from the transformer T. To do. On the other hand, when the output voltage of the switching power supply device 10 decreases, the control device 20 advances the phase of the secondary-side forward switching element and the secondary-side reverse switching element to reduce the phase difference, and takes out the power from the transformer T. Increase the amount. As shown in FIG. 2, the state in which the phase difference is zero is a state in which the most amount of power is extracted from the transformer T, and the amount of power to be extracted decreases as the phase difference increases.

  As shown in FIG. 2, in the comparative example, the unit on period of the primary side forward switching element and the primary side reverse switching element and the unit on period of the secondary side forward switching element and the secondary side reverse switching element are long. Are set substantially equal. That is, the pulse width of the control signal supplied from the control device 20 to the control terminals of the primary forward switching element and the primary reverse switching element, and the control terminals of the secondary forward switching element and the secondary reverse switching element The lengths of the pulse widths of the control signals supplied to are set to be substantially equal.

  In such a control method according to the comparative example, when the phase difference is small or zero as shown in FIG. 2, current is applied to the secondary forward switching element or secondary backward switching element that is an active switch. It will turn off (hard switching) while it is flowing. In that case, the stress of the switching element due to increased ringing increases. Also, switching loss and electromagnetic induction noise increase.

  More specific description will be given below. In the period from time t0 to time t1 in FIG. 2, on the primary side, the first DC power supply E1 to the fourth DC power supply E4 → the second switching element S2 → the primary winding N1 → the third switching element S3 → the first DC power supply E1. A reverse current flows through the fourth DC power supply E4. On the secondary side, the secondary winding N2 → the fourth coil L4 → the fifth switching element S5 → the fifth diode D5 → the fifth coil L5 → the load 30 → the eighth diode D8 → the third coil L3 → the secondary winding N2. Reverse current flows.

  Next, at time t1, the second switching element S2 and the third switching element S3 are turned off. Thereby, charging / discharging of the 1st capacitor | condenser C1-4th capacitor | condenser C4 respectively connected in parallel with 1st switching element S1-4th switching element S4 is started, and it will be in a partial resonance state. In this state, the primary-side transformer voltage V1, which is the output voltage of the primary-side full bridge circuit, rises and reverses positive and negative. Accordingly, when the secondary-side transformer voltage V2 having a polarity opposite to that of the primary-side transformer voltage V1 decreases and becomes less than zero, the fifth diode D5 enters a reverse bias state, and the fifth switching element S5 and the fifth diode D5 An attempt is made to cut off the flowing current is5 (id5).

  Since the diode is not instantaneously cut off due to its characteristics, the current is5 (id5) gradually tries to commutate to the seventh diode D7 that is in the forward bias state due to the polarity reversal of the transformer voltage V2 on the secondary side. At this time, in the comparative example, since the pulse widths of the drive signals on the primary side and the secondary side are set to be the same, the fifth switching element S5 is turned off in a state where the commutated current is5 (id5) flows. Will be. As a result, ringing increases (see the shadow portions t1 to t2 in FIG. 2), and the stress of the fifth switching element S5 increases. In addition, switching loss and electromagnetic induction noise also increase.

  FIG. 3 is a diagram illustrating an example of a control method according to the embodiment of the switching power supply device 10 of FIG. In the embodiment, the unit ON period of the secondary forward switching element and the secondary reverse switching element is set longer than the unit ON period of the primary forward switching element and the primary reverse switching element. That is, the pulse width of the control signal supplied from the control device 20 to the control terminals of the secondary forward switching element and the secondary reverse switching element is set to the control terminal of the primary forward switching element and the primary reverse switching element. It is set longer than the pulse width of the control signal supplied to. For example, the unit on period of the secondary side switching element may be set to a period obtained by adding a dead time to the unit on period of the primary side switching element.

  More specific description will be given below. The behavior of each element in the period from time t0 to time t1 in FIG. 3 is the same as the behavior in the period from time t0 to time t1 in FIG. Next, even if the second switching element S2 and the third switching element S3 are turned off at time t1, the fifth switching element S5 is not immediately turned off. As described above, when the second switching element S2 and the third switching element S3 are turned off and the polarity of the transformer voltage V2 on the secondary side is inverted, the fifth diode D5 enters a reverse bias state.

  In the present embodiment, the fifth switching element S5 maintains the on state for a certain period even when the fifth diode D5 is in the reverse bias state. During this period, the current is5 (id5) flowing through the fifth diode D5 and the fifth switching element S5 is commutated to the seventh diode D7 in the forward bias state, and at the time t2 after the commutation ends, the fifth switching element S5. Is turned off. Thereby, it is turned off (soft switching) in a state where no current flows through the fifth switching element S5, and zero current switching can be realized. Therefore, the stress of the fifth switching element S5 can be reduced, and the switching loss and electromagnetic induction noise can also be reduced.

  The above discussion is based on the fact that the first switching element S1 and the fourth switching element S4 are turned on and the second switching element S2 and the third switching element S3 are turned off. The same applies when the switching element S4 is turned off and the second switching element S2 and the third switching element S3 are turned on. In that case, the stress of the sixth switching element S6 can be reduced, and the switching loss and electromagnetic induction noise can be reduced.

  FIG. 4 is a diagram illustrating an example (another state) of a control method according to a comparative example of the switching power supply device 10 of FIG. FIG. 5 is a diagram illustrating an example (another state) of the control method according to the embodiment of the switching power supply device 10 of FIG. 4 and 5 show a state in which the phase difference between the primary side drive signal and the secondary side drive signal is enlarged as compared with FIGS. 2 and 3. The time at which the fifth switching element S5 in FIG. 4 is turned off is delayed from the time in FIG. 2, but the fifth switching element S5 is switched before the commutation of the current is5 (id5) flowing through the fifth diode D5 and the fifth switching element S5 is completed. Since the element S5 is turned off (hard switching), the above problem occurs.

  If the turn-off time of the fifth switching element S5 is delayed until commutation is completed, the above problem does not occur even if the pulse widths of the primary and secondary drive signals are set to be the same as in the comparative example. For example, the above problem does not occur if the low load state continues and the above phase difference can be maintained. However, normally, a high load state may occur due to load fluctuations. In that case, it becomes necessary to reduce the phase difference and increase the amount of power supplied to the load, and it becomes impossible to ensure turn-off (soft switching) after commutation is completed. On the other hand, in this embodiment, the turn-off (soft switching) after completion of commutation can be ensured regardless of the load state.

  FIG. 6 is a diagram for explaining a first configuration example of the control device 20 of FIG. The control device 20 according to the first configuration example includes a first drive circuit 21, a second drive circuit 22, a third drive circuit 23, a fourth drive circuit 24, and a microcomputer 25. The microcomputer 25 and the first drive circuit 21, the second drive circuit 22, the third drive circuit 23, and the fourth drive circuit 24 are connected by independent signal lines. The microcomputer 25 generates a control signal for generating drive signals for the first switching element S1 to the sixth switching element S6 in accordance with an external instruction signal (not shown) and a signal from the output voltage detection circuit 11. The microcomputer 25 supplies the generated control signals to the first drive circuit 21, the second drive circuit 22, the third drive circuit 23, and the fourth drive circuit 24, respectively.

  The first drive circuit 21 generates drive signals for the first switching element S1 and the fourth switching element S4 based on the control signals for the first switching element S1 and the fourth switching element S4 from the control device 20. . The first drive circuit 21 applies the generated drive signal to the control terminals of the first switching element S1 and the fourth switching element S4.

  Similarly, the second drive circuit 22 generates drive signals for the second switching element S2 and the third switching element S3 based on the control signals for the second switching element S2 and the third switching element S3 from the control device 20. Generate. The second drive circuit 22 applies the generated drive signal to the control terminals of the second switching element S2 and the third switching element S3.

  Similarly, the third drive circuit 23 generates a drive signal for the fifth switching element S5 based on the control signal for the fifth switching element S5 from the control device 20, and supplies the drive signal to the control terminal of the fifth switching element S5. Apply. Similarly, the fourth drive circuit 24 generates a drive signal for the sixth switching element S6 based on the control signal for the sixth switching element S6 from the control device 20, and supplies the drive signal to the control terminal of the sixth switching element S6. Apply.

  FIG. 7 is a diagram for explaining a second configuration example of the control device 20 of FIG. In the control device 20 according to the second configuration example, a NOT circuit 26 is added to the configuration of the control device 20 of FIG. The NOT circuit 26 inverts the logic of the control signal supplied from the microcomputer 25 to the third drive circuit 23 and supplies the inverted signal to the fourth drive circuit 24. For example, the NOT circuit 26 supplies a low-level control signal to the fourth drive circuit 24 when the control signal supplied from the microcomputer 25 to the third drive circuit 23 is at a high level.

  According to the second configuration example, the number of used channel ports of the microcomputer 25 can be reduced by one, and the number of signal lines connected to the microcomputer 25 can be reduced by one. In general, a microcomputer having a small number of channel ports is cheaper than a microcomputer having a large number of channel ports. Therefore, reducing the number of used channel ports leads to an increase in the probability that an inexpensive microcomputer can be adopted. Further, the reduction of signal lines leads to simplification of the circuit and cost reduction.

  In order to share a signal line as in the second configuration example, it is necessary that the two drive circuits connected to the signal line to be shared are drive circuits that generate completely opposite-phase drive signals. As described above, since the primary side employs a partial resonance type full bridge circuit, a dead time for the partial resonance period is essential. During the dead time, the drive signal supplied from the first drive circuit 21 to the first switching element S1 and the fourth switching element S4 and the second drive circuit 22 supplied to the second switching element S2 and the third switching element S3. The drive signals are all low level signals. That is, both drive signals during the dead time are in phase. Accordingly, the signal lines of the first drive circuit 21 and the second drive circuit 22 that generate drive signals for the first switching element S1 to the fourth switching element S4 on the primary side including the dead time cannot be shared. This is because a dead time cannot be provided if they are shared.

  On the other hand, the secondary-side fifth switching element S5 and sixth switching element S6 can theoretically operate without dead time. As described above, when the second switching element S2 and the third switching element S3 are turned off from the state where the reverse current flows in the primary winding N1, the fifth diode D5 enters the reverse bias state. The eighth diode D8 is in a forward bias state. As a result, current is commutated from the fifth diode D5 to the eighth diode D8. In the control method according to the embodiment shown in FIGS. 3 and 5, the fifth switching element S5 is turned off after the commutation is completed. When the fifth switching element S5 is turned off, no current flows through the fifth diode D5 in either the forward direction or the reverse direction. Therefore, there is no change in characteristics even if the turn-off time of the fifth switching element S5 is delayed until the turn-on time of the sixth switching element S6. The same applies to the case where the first switching element S1 and the fourth switching element S4 are turned off from the state in which the forward current flows in the primary winding N1 to enter the partial resonance period.

  When the turn-off time of the fifth switching element S5 is delayed until the turn-on time of the sixth switching element S6 and the turn-off time of the sixth switching element S6 is delayed until the turn-on time of the fifth switching element S5, the fifth switching element S5 and the sixth switching element are switched. The element S6 has a completely complementary operation that does not include dead time. In this setting, the signal lines of the third drive circuit 23 that generates the drive signal for the fifth switching element S5 and the fourth drive circuit 24 that generates the drive signal for the sixth switching element S6 are connected to the third drive circuit 23. Alternatively, it can be made common by providing a NOT circuit 26 in the previous stage of the fourth drive circuit 24.

  In FIG. 7, a NOT circuit 26 is provided in front of the fourth drive circuit 24, the microcomputer 25 generates a control signal for the fifth switching element S5, and the NOT circuit 26 inverts the logic of the control signal to perform the fourth drive. A configuration for outputting to the circuit 24 is shown. In this regard, a NOT circuit 26 is provided in the preceding stage of the third drive circuit 23, the microcomputer 25 generates a control signal for the sixth switching element S6, and the NOT circuit 26 inverts the logic of the control signal to perform the third drive. It may be configured to output to the circuit 23.

  As described above, in the comparative example, the pulse widths of the primary side and secondary side drive signals are set to be the same. Therefore, if the primary side drive signal includes a dead time, the secondary side drive signal also has a dead time. Will be included. Therefore, in the comparative example, the control device 20 according to the second configuration example cannot be employed. Hereinafter, description will be given with reference to the drawings.

  FIG. 8 is a diagram illustrating an example (details) of a control method according to a comparative example of the switching power supply apparatus 10 of FIG. In FIG. 8, the transitions of the voltages Vc1 and Vc4 of the first capacitor C1 and the fourth capacitor C4 are drawn by bold lines, and the transitions of the voltages Vc2 and Vc3 of the second capacitor C2 and the third capacitor C3 are drawn by thin lines. The transition of the current id7 flowing through the seventh diode D7 on the secondary side is drawn with a thick line, and the transition of the current is5 (id5) flowing through the fifth switching element S5 and the fifth diode D5 on the secondary side is drawn with a thin line. Yes. Similarly, the transition of the current id8 flowing through the secondary-side eighth diode D8 is drawn with a bold line, and the transition of the current is6 (id6) flowing through the secondary-side sixth switching element S6 and the sixth diode D6 is drawn with a thin line. ing.

  The phase difference α between the drive signals of the second switching element S2 and the third switching element S3 on the primary side and the drive signal of the fifth switching element S5 on the secondary side, and the first switching element S1 and the fourth switching element S4 on the primary side. The phase difference α between the drive signal and the drive signal of the secondary-side sixth switching element S6 is adaptively controlled, and the output voltage of the switching power supply device 10 is kept substantially constant.

  When the pulse widths of the primary and secondary drive signals are set to be the same as shown in FIG. 8, the drive signal pulse of the fifth switching element S5 and the drive signal pulse of the sixth switching element S6 A blank period occurs. Therefore, the drive signal for the fifth switching element S5 and the signal line for the drive signal for the sixth switching element S6 cannot be shared.

  FIG. 9 is a diagram illustrating another example of the control method according to the embodiment of the switching power supply device 10 of FIG. In FIG. 9, no blank period is provided between the pulse of the drive signal for the fifth switching element S5 and the pulse of the drive signal for the sixth switching element S6. That is, the drive signal of the fifth switching element S5 and the drive signal of the sixth switching element S6 are completely inverted signals. Therefore, the sixth switching element S6 can be driven by the inverted drive signal of the fifth switching element S5 during the period when the sixth switching element S6 is on. Therefore, the drive signal for the fifth switching element S5 and the signal line for the drive signal for the sixth switching element S6 can be shared.

  Comparing FIG. 8 and FIG. 9, the current id7 flowing through the seventh diode D7, the current is5 (id5) flowing through the fifth switching element S5 and the fifth diode D5, the current id8 flowing through the eighth diode D8, and the sixth switching element The waveforms of the current is6 (id6) flowing through S6 and the sixth diode D6 are the same. Therefore, even when the control device 20 according to the second configuration example is employed and the control method that does not provide the dead time for the secondary side drive shown in FIG. 9 is used, the same characteristics as the control method according to the comparative example shown in FIG. Can be realized. That is, the current waveform on the secondary side is the same between the two, and the amount of power extracted from the primary side to the secondary side is also the same.

  As described above, according to the present embodiment, secondary-side hard-switching can be avoided in secondary-side phase shift type power conversion device 100, so that efficiency and characteristics can be improved. In addition, by using the drive signals for the fifth switching element S5 and the sixth switching element S6 on the secondary side as inverted signals, the signal lines can be reduced, and the circuit scale and cost can be reduced.

  Hereinafter, modifications of the configuration of the power conversion device 100 according to the embodiment shown in FIG. 1 will be described. The power conversion apparatus 100 of FIG. 1 is an example provided with a full bridge type inverter on the primary side and a full bridge type rectifier circuit on the secondary side.

  FIG. 10 is a diagram illustrating a configuration of the power conversion device 100 according to the first modification. The power conversion device 100 according to the modification 1 is an example in which a full bridge type inverter is provided on the primary side and a center tap type rectifier circuit is provided on the secondary side. The center tap type rectifier circuit includes a fifth diode D5, a fifth switching element S5, a sixth diode D6, a sixth switching element S6, and a ninth diode D9.

  The output terminal of the rectifier circuit is connected to the plus terminal of the smoothing circuit composed of the fifth coil L5 and the fifth capacitor C5, and the midpoint of the secondary winding N2 is connected to the minus terminal of the smoothing circuit. The anode terminal of the fifth diode D5 is connected to one end of the secondary winding N2 via the fifth switching element S5 and the fourth coil L4, and the anode terminal of the sixth diode D6 is connected to the sixth switching element S6 and the third coil The other end of the secondary winding N2 is connected via the coil L3.

  The fifth diode D5 conducts the forward current from the secondary winding N2, and interrupts the reverse current from the secondary winding N2. On the other hand, the sixth diode D6 conducts the reverse current from the secondary winding N2, and cuts off the forward current from the secondary winding N2. The fifth switching element S5 acts as a secondary forward switching element that switches forward current, and the sixth switching element S6 acts as a secondary backward switching element that switches reverse current. The anode terminal of the ninth diode D9 is connected to the middle point of the secondary winding N2, and the cathode terminal of the ninth diode D9 is connected in common to the cathode terminals of the fifth diode D5 and the sixth diode D6.

  FIG. 11 is a diagram illustrating a configuration of the power conversion device 100 according to the second modification. The power conversion device 100 according to the modification 2 is an example including a half-bridge type inverter on the primary side and a full-bridge type rectifier circuit on the secondary side. The half-bridge type inverter shown in FIG. 11 includes a first switching element S1, a first diode D1, a first capacitor C1, a third switching element S3, a third diode D3, a first switching element, and the like from the full-bridge type inverter shown in FIGS. In this configuration, the three capacitors C3 and the first coil L1 are omitted. In FIG. 11, the exciting inductor of the primary winding N1 is depicted as the sixth coil L6.

  FIG. 12 is a diagram illustrating a configuration of the power conversion device 100 according to the third modification. The power conversion device 100 according to Modification 3 is an example in which a half-bridge inverter is provided on the primary side and a center tap rectifier circuit is provided on the secondary side.

  As described above, in the DC-DC converter in which the primary side and the secondary side are AC-linked by the high-frequency transformer in the secondary side phase shift method, zero voltage switching, zero current switching, or both soft switching is performed in all switching. The operation can be realized regardless of the load state. In addition, since the circulating current is suppressed, a significant reduction in conduction loss and an improvement in conversion efficiency can be realized. This operation principle can also be applied to a circuit topology in which the primary-side inverter circuit is a half-bridge type. The secondary side rectifier circuit can also be applied to a center tap type circuit topology. The power converters 100 in FIGS. 1 and 10 to 12 all have the same effect, so that the designer selects an optimum circuit topology in consideration of the voltage and current to be used and the number of power devices to be used. do it.

  Hereinafter, the modification of the structure of the primary side of the power converter device 100 which concerns on this Embodiment is further mentioned. In addition, although the example provided with a full bridge type rectifier circuit is drawn as a secondary side configuration in the following modification, a center tap type rectifier circuit may be provided.

  FIG. 13 is a diagram illustrating a configuration of the power conversion device 100 according to the fourth modification. The power conversion device 100 according to the modified example 4 replaces the first DC power supply E1 to the fourth DC power supply E4 of FIGS. 1 and 10 to 12 with the sixth capacitor C6 to the ninth capacitor C9, respectively, and uses the DC power supply E as one. I'm stuck. In Modification 4, the first coil L1, the first DC power supply E1, and the third DC power supply E3, not the first coil L1, the sixth capacitor C6, and the eighth capacitor C8 form a resonance pole circuit. The same applies to the right arm.

  FIG. 14 is a diagram illustrating a configuration of the power conversion device 100 according to the fifth modification. In the power conversion device 100 according to the modification 5, the seventh capacitor C7 and the eighth capacitor C8 are deleted from the power conversion device 100 according to the modification 4, and one end of the second coil L2 is connected to the sixth capacitor C6 and the eighth capacitor. The configuration is connected to a node between C8. That is, the sixth capacitor C6 and the eighth capacitor C8 are shared by the resonance pole circuits of the left and right arms.

  FIG. 15 is a diagram illustrating a configuration of the power conversion device 100 according to the sixth modification. The power conversion device 100 according to Modification 6 has a configuration in which the first coil L1 and the second coil L2 are realized by the leakage inductance of the primary winding N1. The power conversion device 100 according to Modification 4-6 described above can achieve the same effects as those of the power conversion device 100 of FIGS. 1 and 10 to 12.

  The power conversion device 100 according to the present embodiment described above can be used for various purposes. Hereinafter, examples of use in a power storage system and a vehicle will be given. In addition, it is also used for applications that require highly efficient power conversion and insulation, such as a data center power supply device.

  FIG. 16 is a diagram showing a configuration of power storage system 400 in which power conversion device 100 according to the present embodiment is used. A power storage system 400 illustrated in FIG. 16 includes a solar battery 200a, a storage battery 200b, a DC-DC converter 100a, a DC-DC converter 100b, and an inverter 300a. The storage battery 200b may be a stationary storage battery or a portable storage battery such as an in-vehicle battery. The DC power generated by the solar cell 200a is converted into DC power having a predetermined voltage by the DC-DC converter 100a. The DC power is converted into AC power by the inverter 300a and output to the system 500, or is converted into DC power of the storage voltage by the DC-DC converter 100b and stored in the storage battery 200b. The power conversion device 100 according to the present embodiment is used for at least one of the DC-DC converter 100a and the DC-DC converter 100b.

  Note that the storage battery 200b and the DC-DC converter 100b may be omitted. In this case, a solar power generation system without a power storage function is obtained. Further, the solar cell 200a and the DC-DC converter 100a may be omitted. In this case, the power storage system has no power generation function.

  FIG. 17 is a diagram showing a configuration of a vehicle 700 in which the power conversion device 100 according to the present embodiment is used. A vehicle 700 shown in FIG. 17 is a hybrid car (HV), a plug-in hybrid car (PHV), or an electric car (EV) on which a traveling motor 600 is mounted. The motor 600 is not limited to a high-power motor that can run on its own, but may be a travel assist motor mounted on a mild hybrid car. Usually, an AC synchronous motor is used as the motor 600.

  A vehicle 700 shown in FIG. 17 includes a traveling battery 200c, an auxiliary battery 200d, a DC-DC converter 100c, a bidirectional DC-DC converter 150, an inverter 300b, and a motor 600. A storage battery such as a lithium ion battery or a nickel metal hydride battery can be used as the traveling battery 200c. During power running, bidirectional DC-DC converter 150 converts the DC power supplied from battery for traveling 200c into DC power of a predetermined voltage and outputs it to inverter 300b. Inverter 300 b converts the DC power supplied from bidirectional DC-DC converter 150 into AC power and supplies the AC power to motor 600. During regeneration, inverter 300 b converts AC power generated based on the deceleration energy into DC power and outputs the DC power to bidirectional DC-DC converter 150. The bidirectional DC-DC converter 150 converts the direct current power supplied from the inverter 300b into the direct current power of the battery voltage and charges the traveling battery 200c.

  As the auxiliary battery 200d, a 12V output lead battery is usually used. In the mild hybrid car, the traveling battery 200c is designed to output 48V, for example. The 12V system to which the auxiliary battery 200d is connected and the 48V system to which the traveling battery 200c is connected are connected via the DC-DC converter 100c. The DC-DC converter 100c boosts the voltage of the auxiliary battery 200d to the voltage of the traveling battery 200c. Thus, when the capacity of the traveling battery 200c is insufficient, power can be supplied to the motor 600 from the auxiliary battery 200d. The DC-DC converter 100c uses the power conversion device 100 according to the present embodiment.

  FIG. 18 is a diagram illustrating a configuration of a charger 800 in which the power conversion device 100 according to the present embodiment is used. A vehicle 700 shown in FIG. 18 is a vehicle in which a plug-in charging function is added to the vehicle shown in FIG. The charger 800 includes a rectifier circuit 810, a PFC circuit 820, and a DC-DC converter 100d. The rectifier circuit 810 rectifies the AC voltage supplied from the system 500. The PFC circuit 820 improves the power factor of the rectified power. The DC-DC converter 100d converts the input voltage from the PFC circuit 820 into a charging voltage. The DC-DC converter 100d uses the power conversion device 100 according to the present embodiment. As shown in FIG. 18, the charger 800 may be a (rapid) charger installed outside the vehicle, or an in-vehicle charger mounted in the vehicle 700.

  As described above, by using the power conversion device 100 according to the present embodiment for the DC-DC converter used in the power storage system 400, the vehicle 700, or the charger 800, there is little loss and characteristics are improved. A good power supply system can be constructed.

  The present invention has been described based on the embodiments. The embodiments are exemplifications, and it will be understood by those skilled in the art that various modifications can be made to combinations of the respective constituent elements and processing processes, and such modifications are within the scope of the present invention. .

  For example, the first capacitor C1 to the fourth capacitor C1 to C4 may be configured by the respective parasitic capacitances of the first switching element S1 to the fourth switching element S4 instead of the snubber capacitor. Further, the first diode D1 to the fourth diode D4 may be constituted by the respective parasitic diodes of the first switching element S1 to the fourth switching element S4.

  The present invention can be used for DC-DC converters used in power storage systems, vehicles, and the like.

  DESCRIPTION OF SYMBOLS 100 Power converter device, 10 Switching power supply device, 20 Control apparatus, S1 1st switching element, S2 2nd switching element, S3 3rd switching element, S4 4th switching element, S5 5th switching element, S6 6th switching element, C1 first capacitor, C2 second capacitor, C3 third capacitor, C4 fourth capacitor, C5 fifth capacitor, C6 sixth capacitor, C7 seventh capacitor, C8 eighth capacitor, C9 ninth capacitor, D1 first diode, D2 second diode, D3 third diode, D4 fourth diode, D5 fifth diode, D6 sixth diode, D7 seventh diode, D8 eighth diode, D9 ninth diode, L1 first coil, L2 2nd coil, L3 3rd coil, L4 4th coil, L5 5th coil, L6 6th coil, T transformer, N1 primary winding, N2 secondary winding, E DC power supply, E1 1st DC power supply, E2 Second DC power supply, E3 Third DC power supply, E4 Fourth DC power supply, 11 Output voltage detection circuit, 21 First drive circuit, 22 Second drive circuit, 23 Third drive circuit, 24 Fourth drive circuit, 25 Microcomputer , 26 NOT circuit, 30 load, 100a, 100b, 100c, 100d DC-DC converter, 150 bidirectional DC-DC converter, 200a solar battery, 200b storage battery, 200c battery for traveling, 200d auxiliary battery, 300a, 300b inverter, 400 power storage system, 500 systems, 600 Over data, 700 vehicle, 800 charger, 810 rectifier circuit, 820 PFC circuit.

Claims (5)

A bridge circuit that converts DC voltage to AC voltage;
A transformer including a primary winding connected to the bridge circuit, and a secondary winding electromagnetically coupled to the primary winding;
A rectifier circuit for rectifying an AC voltage input from the secondary winding;
A smoothing circuit that smoothes the voltage rectified by the rectifier circuit,
The rectifier circuit includes a secondary side switching element inserted in a current path,
The switching power supply device according to claim 1, wherein an on period of the secondary side switching element is set longer than an on period of a primary side switching element included in the bridge circuit. The bridge circuit as the primary side switching element,
A primary-side forward switching element inserted in a path for supplying a forward current to the primary winding;
A primary-side reverse switching element inserted in a path for supplying a reverse current to the primary winding,
The rectifier circuit is
A first diode that conducts forward current from the secondary winding;
A second diode for conducting reverse current from the secondary winding;
As the secondary side switching element,
A secondary forward switching element inserted in a forward current path including the first diode;
A secondary reverse switching element inserted into a reverse current path including the second diode,
The primary-side forward switching element and the primary-side reverse switching element operate with a fixed duty ratio and a fixed phase,
The secondary-side forward switching element and the secondary-side reverse switching element operate with a fixed duty ratio and a phase that varies according to the output voltage of the switching power supply device,
Unit on periods of the secondary side forward switching element and the secondary side reverse switching element are set longer than unit on periods of the primary side forward switching element and the primary side reverse switching element. The switching power supply device according to claim 1.   The primary-side forward switching element and the primary-side reverse switching element include a dead time, and the secondary-side forward switching element and the secondary-side reverse switching element do not include a dead time. Item 3. The switching power supply device according to Item 2. A switching power supply device according to claim 1;
A control device for controlling the primary side switching element and the secondary side switching element;
A power conversion device comprising: The switching power supply device according to claim 2 or 3,
A controller for controlling the primary-side forward switching element, the primary-side forward switching element, the secondary-side forward switching element, and the secondary-side reverse switching element;
The controller is
A control circuit for generating a control signal for generating drive signals for the primary-side forward switching element, the primary-side forward switching element, the secondary-side forward switching element, and the secondary-side reverse switching element; ,
A secondary-side forward drive circuit that generates a drive signal for driving the secondary-side forward switching element based on a control signal from the control circuit;
Based on a control signal from the control circuit, a secondary reverse drive circuit for driving the secondary forward switching element,
The control signal supplied from the control circuit to the secondary forward drive circuit is inverted and supplied to the secondary reverse drive circuit, or from the control circuit to the secondary reverse drive circuit. A NOT circuit that inverts the logic of the supplied control signal and supplies it to the secondary forward drive circuit;
The power converter device characterized by including.

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