JP6487874B2 - Switching power supply - Google Patents

Description

The present invention relates to a flyback switching power supply that operates in a current discontinuous mode or a critical mode.

  Conventionally, for example, as disclosed in Patent Document 1, there has been a flyback switching power supply that operates in a critical mode. In this switching power supply, the on-time and off-time of the main switching element are controlled so that the output voltage approaches the target value, and the excitation energy stored in the transformer is released through the output rectifying element while the main switching element is on. Then, the main switching element is controlled so as to invert from OFF to ON immediately. In order to reduce the cross loss of the main switching element, the turn-on of the main switching element is controlled at a timing when the voltage across the main switching element resonates and falls below a certain level (so-called pseudo-resonance type control).

  In addition, a current detection circuit that converts the switching current into a voltage to generate a switching current signal is provided, and a configuration that uses the switching current signal as a carrier when modulating a signal obtained by amplifying the difference between the output voltage and the target value (so-called so-called carrier) Current mode control). Furthermore, when the switching current signal exceeds a predetermined threshold, an overcurrent protection means (so-called pulse-by-pulse overcurrent protection means) that forcibly turns off the main switching element is provided, and the output current becomes excessive. The output voltage is reduced and the increase in output current is suppressed.

  Patent Document 2 discloses a technique for improving the above-described pulse-by-pulse overcurrent protection means and reducing the shift due to the input voltage of the overcurrent protection characteristics. For example, a flyback switching power supply that operates in a current discontinuous mode or a critical mode is provided with a current detection circuit that converts the switching current into a voltage to generate a switching current signal and a unique input correction circuit. A configuration for correcting the switching current signal accordingly is described.

JP-A-10-191630 Japanese Patent Laying-Open No. 2015-186281

  The current detection circuit of the switching power supply device disclosed in Patent Literatures 1 and 2 is configured to insert a current detection resistor in a path through which a switching current flows and to output a voltage generated at both ends of the current detection resistor as a switching current signal. That is, since a large current flows through the current detection resistor, a large loss occurs and the efficiency is significantly reduced.

  In order to suppress the loss, a method of using a current transformer instead of the current detection resistor can be considered. However, since the current transformer is relatively expensive, it is difficult to use it particularly for a low-cost type switching power supply device.

The present invention has been made in view of the above-described background art, and an object of the present invention is to provide a switching power supply device that can generate a switching current signal that can be used for controlling a main switching element in a simple and low loss manner. And

The present invention is provided with a main switching element, a transformer having an input winding and an output winding, an output rectifying element, and an output smoothing capacitor. The main switching element is turned on and off to change the input voltage to a predetermined output voltage. A flyback power conversion circuit for converting, and a circuit for controlling on / off of the main switching element, wherein the excitation energy stored in the transformer is released through the output rectifying element while the main switching element is on. Then, a switching power supply device comprising a switching control circuit for performing current discontinuous mode control or critical mode control for inverting the main switching element from off to on,
The switching current waveform flowing through the main switching element to generate a switching current signal is a voltage signal of similar shape, the switching current signal generation circuit is provided for output to the switching control circuit, the switching current signal generation circuit A timer capacitor that has one end connected to the ground and outputs the switching current signal from the other end, generates an input proportional current that is substantially proportional to the input voltage, and charges the timer capacitor while the main switching element is on. And a reset circuit that discharges the timer capacitor while the main switching element is off, and a switching control circuit utilizes the switching current signal generated at both ends of the timer capacitor. Switch that controls on / off of the switching element A quenching power supply.

The switching control circuit generates a driving pulse for driving the main switching element, and the main switching element includes a transistor element that is turned on when the driving pulse is at a high level and turned off when the driving pulse is at a low level. The reset circuit can be configured by a reset diode having an anode connected to the other end of the timer capacitor and a cathode connected to a position where the drive pulse or a pulse having the same phase as that is generated. .

The switching control circuit is provided with overcurrent protection means for forcibly turning off the main switching element when the switching current signal exceeds a predetermined threshold, reducing the output voltage, and suppressing an increase in output current. In the switching current signal generation circuit, a correction resistor is inserted in a position in series with the timer capacitor, and the switching current signal is generated by a change in the input voltage by superimposing a voltage generated in the correction resistor. It is corrected in a direction in which fluctuation of the overcurrent protection characteristics is reduced.

Alternatively , the switching control circuit includes overcurrent protection means for forcibly turning off the main switching element when the switching current signal exceeds a predetermined threshold, reducing the output voltage, and suppressing an increase in output current. The charging circuit of the switching current signal generating circuit is provided with a voltage substantially proportional to the input voltage applied to both ends thereof to generate the input proportional current and output the current to the timer capacitor. The current generating resistor is formed of a series circuit of a plurality of resistance elements, a correction zener diode is connected in parallel to both ends of a specific resistance element of the plurality of resistance elements, and the switching current signal is The correction Zener diode performs an operation of limiting the voltage generated by the specific resistance element to a certain level or less. More variation overcurrent protection characteristics due to a change in the input voltage is configured to be corrected in the direction to become smaller.

The switching power supply device according to the present invention generates an input proportional current substantially proportional to the input voltage, charges a timer capacitor while the main switching element is on, and the timer capacitor when the main switching element is off. To generate a switching current signal that is a voltage signal having a waveform similar to that of the switching current flowing through the main switching element at both ends of the timer capacitor, and using the switching current signal, the main switching element It is driven by a control method for controlling on / off of the.

The switching power supply device of the present invention is an optimum technology for a power supply device having a sawtooth waveform in which the switching current rises from zero ampere to the right, that is, a flyback power supply circuit that operates in a current discontinuous mode or a critical mode The switching current signal, which is a voltage signal having a waveform similar to the switching current, can be generated simply and with low loss, and can be used for various controls.

It is a circuit diagram which shows one form of a switching power supply device . FIG. 2 is a circuit diagram illustrating a specific configuration example (first configuration example) of the switching current signal generation circuit illustrated in FIG. 1. FIG. 2 is a circuit diagram illustrating a specific configuration example (second configuration example) of the switching current signal generation circuit illustrated in FIG. 1. FIG. 3 is a circuit diagram illustrating a specific configuration example (third configuration example) of the switching current signal generation circuit illustrated in FIG. 1. It is a circuit diagram which shows the other form of a switching power supply device . Output current of the switching power supply device shown in FIG. 5 - is a graph showing the output voltage characteristics (overcurrent protection characteristics). 6 is an operation waveform at an operating point P1 (L) of the switching power supply device shown in FIG. Operation waveforms (a), (b), (c), and (d) at the operating points P1 (L), P1 (H), P2 (L), and P2 (H) of the switching power supply device shown in FIG. is there. It is a circuit diagram which shows the switching current signal generation circuit with which 1st embodiment of the switching power supply device of this invention is provided . Operation waveforms (a), (b), (c) at the operation points P1 (L), P1 (H), P2 (L), P2 (H) when the switching current signal generation circuit shown in FIG. 9 is used. ), (D). The circuit diagram (a) which shows the switching current signal generation circuit with which 2nd embodiment of the switching power supply device of this invention is provided , The relationship between the input proportional current and input voltage at the time of using this switching current signal generation circuit was represented. It is a graph (b). It is a circuit diagram which shows the further another form of a switching power supply device .

Hereinafter, an embodiment of the switching power supply device will be described with reference to FIGS. As shown in FIG. 1, the switching power supply 10 of this embodiment converts a DC input voltage Vi input from an input power supply 12 into a DC output voltage Vo, and outputs the output voltage Vo and output to an externally connected load 14. It is a DC-DC converter that supplies a current Io.

  In the power conversion circuit 16, a series circuit of the input winding 20 a of the transformer 20 and the main switching element 22 is connected to both ends of the input power supply 12. The main switching element 22 is, for example, an N-channel MOS FET, the drain is connected to one end of the input winding 20a, and the source is connected to a negative line (hereinafter referred to as ground) of the input power supply 12. Yes. A drive pulse Vg output from a switching control circuit 18 to be described later is input between the gate and source of the main switching element 22, and the main switching element 22 is turned on while the drive pulse Vg is at a high level and is at a low level. Turn off.

  The transformer 20 has an input winding 20a and an output winding 20b. A dot attached to each winding indicates polarity, and one end of the input winding 20 a to which the dot is attached is connected to the main switching element 22, and the other end is connected to the plus side of the input power supply 12. The transformer 20 performs an operation in which the input voltage Vi is applied to the input winding 20a while the main switching element 22 is on to accumulate excitation energy, and this is discharged from the output winding 20b during the off period.

  The anode of the output rectifying element 24 that is a diode is connected to one end of the output winding 20b that is not marked with a dot. The output rectifying element 24 becomes conductive during a period in which the main switching element 22 is off, rectifies the voltage generated in the output winding 20b, and serves as a current path through which the transformer 20 emits excitation energy. Further, an output smoothing capacitor 26 is connected between the cathode of the output rectifying element 24 and the other end of the output winding 20b to which the dots are attached. The output smoothing capacitor 26 outputs the rectified voltage output by the output rectifying element 24. Smoothes and generates a DC output voltage Vo.

  As described above, the power conversion circuit 16 is a flyback converter provided with the transformer 20, the main switching element 22, the output rectifying element 24, and the output smoothing capacitor 26.

  The switching control circuit 18 is a circuit that controls on / off of the main switching element 22, and after the excitation energy accumulated in the transformer 20 is released through the output rectifying element 24 while the main switching element 22 is on, the main switching element 22 is turned on. Current discontinuous mode control or critical mode control for inverting 22 from off to on is performed. Therefore, the switching current Id flowing through the main switching element 22 has a sawtooth waveform that rises from zero ampere to the right, and the slope of the right rise is substantially proportional to the input voltage Vi, and the higher the input voltage Vi is, the higher the input voltage Vi is. The slope becomes steep.

  The switching current signal generation circuit 28 is a circuit that generates a switching current signal Vid that is a voltage signal having a waveform similar to the switching current Id, and outputs the switching current signal Vid to the switching control circuit 18, and includes a timer capacitor 30, a charging circuit 32, And a reset circuit 34. The timer capacitor 30 is a capacitor element that has one end connected to the ground and outputs the switching current signal Vid from the other end. The charging circuit 32 is a circuit that generates an input proportional current Jin that is substantially proportional to the input voltage Vi and charges the timer capacitor 30 during a period in which the main switching element 22 is on. The reset circuit 34 is a circuit that discharges the timer capacitor 30 while the main switching element 22 is off.

  FIG. 2 shows a specific configuration example (first configuration example) of the switching current signal generation circuit 28. The switching current signal generation circuit 36 of this configuration example has a timer capacitor 30 having one end connected to the ground, and the charging circuit has a current connected between the line of the input voltage Vi and the other end of the timer capacitor 30. The generating resistor 38 is configured. The reset circuit is constituted by a reset diode 40. The reset diode 40 has an anode connected to the other end of the timer capacitor 30 and a cathode at a location where the drive pulse Vg is generated (the gate of the main switching element 22). It is connected. The position to which the cathode is connected may be a position where a pulse having the same phase as the drive pulse Vg is generated, and may be a specific position in the switching control circuit 18, for example.

  The current generation resistor 38 is substantially applied with the input voltage Vi at both ends thereof, generates an input proportional current Jin proportional to the input voltage Vi, and outputs it to the timer capacitor 30. During a period in which the drive pulse Vg is low (a period in which the main switching element 22 is off), the reset diode 40 is conducted, the input proportional current Jin is bypassed, the timer capacitor 30 is discharged, and both ends of the timer capacitor 30 are discharged. The voltage is almost zero volts. During the period in which the drive pulse Vg is high (the period in which the main switching element 22 is on), the reset diode 40 is non-conductive, the timer capacitor 30 is charged with the input proportional current Jin, and the voltage at both ends rises from zero volts to the right. To rise.

  The waveform of the voltage across the timer capacitor 30 is a sawtooth waveform that rises from zero volts to the right, and the slope of the right rise is approximately proportional to the input proportional current Jin, and the slope becomes steeper as the input voltage Vi is higher. become. That is, the switching current signal Vid that is a voltage signal having a waveform similar to the switching current Id is generated at both ends of the timer capacitor 30.

  FIG. 3 shows another configuration example (second configuration example) of the switching current signal generation circuit 28. The switching current signal generation circuit 42 of this configuration example includes a timer capacitor 30 having one end connected to the ground, and the reset circuit is configured by a reset diode 40 similar to the above. The charging circuit 44 includes an auxiliary winding 20c provided in the transformer 20, an auxiliary rectifying / smoothing circuit 46 that rectifies and smoothes the voltage of the auxiliary winding 20c and outputs a DC voltage Vi-1 that is substantially proportional to the input voltage Vi, and an auxiliary circuit. The current generating resistor 48 is connected between the output of the rectifying and smoothing circuit 46 and the other end of the timer capacitor 30.

  The current generating resistor 48 is applied with substantially the DC voltage Vi-1 at both ends, generates an input proportional current Jin proportional to the input voltage Vi, and outputs it to the timer capacitor 30. During a period in which the drive pulse Vg is low (a period in which the main switching element 22 is off), the reset diode 40 is conducted, the input proportional current Jin is bypassed, the timer capacitor 30 is discharged, and both ends of the timer capacitor 30 are discharged. The voltage is almost zero volts. During the period in which the drive pulse Vg is high (the period in which the main switching element 22 is on), the reset diode 40 is non-conductive, the timer capacitor 30 is charged with the input proportional current Jin, and the voltage at both ends rises from zero volts to the right. To rise. That is, similarly to the above, the switching current signal Vid, which is a voltage signal having a waveform similar to the switching current Id, is generated at both ends of the timer capacitor 30.

  Further, FIG. 4 shows another configuration example (third configuration example) of the switching current signal generation circuit 28. The switching current signal generation circuit 50 of this configuration example includes a timer capacitor 30 having one end connected to the ground, and the reset circuit is configured by a reset diode 40 similar to the above. The charging circuit 52 rectifies and smoothes the voltage of the auxiliary winding 20c provided in the transformer 20 and the auxiliary winding 20c, and outputs a constant DC voltage Vo-1 approximately proportional to the output voltage Vo. It has. Also, a current generating resistor 56 having one end connected to the line of the input voltage Vi, and a current I56 output from the other end of the current generating resistor 56 and the ground are input, and an output terminal And a first current mirror circuit 58 for drawing a current corresponding to the current I56. Further, it is provided between the auxiliary rectifying / smoothing circuit 54 and the other end of the timer capacitor 30 and operates using the DC voltage Vo-1 as a power source to generate a current corresponding to the current drawn by the first current mirror circuit 58. , A second current mirror circuit 60 is provided for output toward the other end of the timer capacitor 30.

  The current generating resistor 56 is substantially applied with the input voltage Vi at both ends, and generates a current I56 proportional to the input voltage Vi, that is, an input proportional current Jin, and the second current mirror circuit 60 corresponds to the input proportional current Jin. The current Jin-1 is output toward the timer capacitor 30. During the period in which the drive pulse Vg is low (the period in which the main switching element 22 is off), the reset diode 40 is conducted, the current Jin-1 is bypassed, the timer capacitor 30 is discharged, and both ends of the timer capacitor 30 are discharged. The voltage is almost zero volts. During the period when the drive pulse Vg is at a high level (the period when the main switching element 22 is on), the reset diode 40 becomes non-conductive, the timer capacitor 30 is charged with the current Jin-1, and the voltage at both ends rises from zero volts to the right. To rise. That is, similarly to the above, the switching current signal Vid, which is a voltage signal having a waveform similar to the switching current Id, is generated at both ends of the timer capacitor 30.

  Since the switching current signal generation circuits 36, 42, and 50 have their characteristics, it is preferable to use them according to the specifications of the apparatus. The switching current signal generation circuit 36 shown in FIG. 2 has a very simple configuration. For example, when the input voltage Vi is a high voltage (for example, several hundred volts), a high voltage is applied across the current generation resistor 38. As a result, a large loss occurs. On the other hand, in the case of the switching current signal generation circuit 42 shown in FIG. 3, the voltage applied to the current generation resistor 48 is set to a moderately low voltage by adjusting the turn ratio of the input winding 20a and the auxiliary winding 20c. Therefore, the loss of the current generating resistor 48 can be easily suppressed. In the case of the switching current signal generation circuit 50 shown in FIG. 4, the current mirror circuits 58 and 60 can generate a current Jin-1 (> Jin) obtained by amplifying the input proportional current Jin. Therefore, while increasing the resistance value of the current generating resistor 56 to suppress loss, the reduced input proportional current Jin is amplified by the current mirror circuits 58 and 60 to secure a current necessary for charging the timer capacitor 30. it can.

  In addition, the switching current signal generation circuits 36, 42, 50 accurately input the input voltage at both ends of each of the current generation resistors 38, 48, 56 in order to increase the similarity of the waveforms of the switching current Id and the switching current signal Vid. It is preferable to apply a voltage proportional to Vi. However, it is difficult for the switching current signal generation circuit 36 to realize this when the input voltage Vi is very low. For example, if the input voltage Vi is 5 volts and the switching current signal Vid to be generated is 1 to 2 volts, the voltage across the current generating resistor 38 cannot be made almost equal to the input voltage Vi. On the other hand, in the case of the switching current signal generation circuit 42 shown in FIG. 3, the voltage applied to the current generation resistor 48 is set to a moderately high voltage by adjusting the turns ratio of the input winding 20a and the auxiliary winding 20c. Therefore, a voltage Vi-1 that is substantially proportional to the input voltage Vi can be applied to both ends of the current generating resistor 48.

The control method of the switching power supply device of this embodiment generates an input proportional current Jin that is substantially proportional to the input voltage Vi, charges the timer capacitor 30 during the period when the main switching element 22 is on, and turns off the main switching element 22. By discharging the timer capacitor 30 during the period, a switching current signal Vid that is a voltage signal having a waveform similar to that of the switching current Id flowing through the main switching element 22 is generated at both ends of the timer capacitor 30. The configuration is such that the on / off of the main switching element 22 is controlled by using the switching current signal generation circuit 28 (36, 42, 50) and the switching control circuit 18 described above.

  As described above, the switching power supply device 10 and the control method thereof are a power supply device having a sawtooth waveform in which the switching current Id rises from zero ampere to the upper right, that is, a flyback system that operates in a current discontinuous mode or a critical mode. The switching current signal Vid, which is a voltage signal having a waveform similar to the switching current Id, can be generated simply and with low loss and used for various controls.

Next, another embodiment of the switching power supply device will be described with reference to FIGS. Here, the same components as those in the above embodiment are denoted by the same reference numerals and description thereof is omitted. As shown in FIG. 5, the switching power supply 62 of this embodiment converts a DC input voltage Vi input from the input power supply 12 into a DC output voltage Vo, and outputs the output voltage Vo and output to an externally connected load 14. This is a DC-DC converter that supplies a current Io, and includes a power conversion circuit 16 and a switching current signal generation circuit 36 similar to those described above, and a novel switching control circuit 64.

  The switching control circuit 64 controls the power conversion circuit 16 to operate in a quasi-resonant critical mode. Further, the switching current signal Vid is used to control the output voltage Vo to be close to the target value in the current mode and to perform the pulse-by-pulse overcurrent protection. Hereinafter, the internal configuration of the switching control circuit 64 will be described.

  The switching control circuit 64 includes an auxiliary winding 20 c of the transformer 20, an integration circuit 66, and a first comparator 68 as a circuit block for determining the turn-on timing of the main switching element 22. The auxiliary winding 20c has a dot-attached end connected to the ground and the other end outputs a voltage V20c that is a positive voltage during a period in which the main switching element 22 is off. The integrating circuit 66 is connected to both ends of the auxiliary winding 20c, and outputs a transformer voltage signal Vot in which the phase of a specific frequency component included in the input voltage V20c is delayed. The first comparator 68 has one inverting input and one non-inverting input, the transformer voltage signal Vot is inputted to the inverting input, and the non-inverting input is held at a predetermined transformer voltage threshold Vdr. . The transformer voltage threshold Vdr is a voltage slightly higher than zero volts, for example, about 0.1V.

  The first comparator 68 compares the transformer voltage signal Vot with the transformer voltage threshold value Vdr, and outputs a first pulse V1 corresponding to this. Although the detailed operation will be described later, the timing at which the first pulse V1 is inverted from the low level to the high level is the turn-on timing of the main switching element 22. The diode 68 a is an element for preventing a negative voltage from being input to the inverting input side of the first comparator 68, and is provided to protect the first comparator 68.

  The switching control circuit 64 includes a voltage control signal generation circuit 70 and a second comparator 72 as circuit blocks for determining the turn-off timing of the main switching element 22. The voltage control signal generation circuit 70 includes an inverting amplifier circuit 70a and an insulating element 70b. The inverting amplifier circuit 70a generates a voltage control signal Vfb by inverting and amplifying the difference between the output voltage Vo and the reference voltage Vor. The signal is output through an insulating element 70b such as a coupler. The reference voltage Vor is a voltage corresponding to the target value of the output voltage Vo.

  The second comparator 72 has two inverting inputs and one non-inverting input, the voltage control signal Vfb is input to one inverting input, and the other inverting input has a predetermined threshold Vth (for example, about 2V). ) And the switching current signal Vid generated by the switching current signal generation circuit 36 is input to the non-inverting input. When the voltage control signal Vfb is lower than the threshold value Vth, the second comparator 72 compares the switching current signal Vid with the voltage control signal Vfb, and outputs a second pulse V2 corresponding thereto. On the other hand, when the voltage control signal Vfb is higher than the threshold value Vth, the switching current signal Vid and the threshold value Vth are compared, and a second pulse V2 corresponding to this is output. Although the detailed operation will be described later, the timing at which the second pulse V2 is inverted from the low level to the high level is the timing at which the main switching element 22 is turned off.

Further, the switching control circuit 64 is a circuit block for generating a drive pulse Vg for turning on and off the main switching element 22, and is used for a set / reset flip-flop 74 (hereinafter referred to as RS-FF 74) and current amplification. A non-inverting buffer 76 is provided.
In the RS-FF 74, the first pulse V1 is input to the S terminal, the second pulse V2 is input to the R terminal, and the corresponding driving pulse Vg is output from the Q terminal. The non-inverting buffer 76 outputs a high-power drive pulse Vg between the gate and source of the main switching element 22.

  Next, the operation of the switching power supply device 62 will be described. Here, it is assumed that each transistor and diode included in the switching power supply device 62 are small enough to ignore a voltage drop when a current flows through the transistor and the diode. FIG. 6 shows the output current Io-output voltage Vo characteristic of the switching power supply 62. When the output current Io is small, the output voltage Vo is controlled to be equal to the target value Vo1 (normal operation), and the output current is shown. When Io exceeds a certain level, an increase in the output current Io is suppressed by the overcurrent protection operation, and the output voltage Vo starts to decrease (overcurrent protection operation).

  P1 (L), P1 (H), P2 (L), and P2 (H) indicate operating points of the switching power supply 62. At the operating point P1 (L), the input voltage Vi is held at a relatively low voltage within the specification range, and the operation is performed with the output current Io = Io1 and the output voltage Vo = Vo1. At the operating point P1 (H), the input voltage Vi is held at a relatively high voltage within the specification range, and the operation is performed with the output current Io = Io1 and the output voltage Vo = Vo1. At the operating point P2 (L), the input voltage Vi is held at a relatively low voltage within the specified range, and the output current Vo is slightly lowered from Vo1 with the output current Io = Io2 (L). Yes. At the operating point P2 (H), the input voltage Vi is held at a relatively high voltage within the specified range, and the output current Vo operates slightly with the output voltage Vo1 slightly lower than Vo1 at the output current Io = Io2 (H). Yes.

  First, the normal operation performed at the operating point P1 (L) will be described with reference to FIG. Each of Tsw (1) and Tsw (2) shown in FIG. 7 is one cycle of switching, and the operation of the switching power supply device 62 can be described by dividing one cycle of switching into periods T1 to T4.

  The period T1 starts when the main switching element 22 is turned on, and the drain-source voltage Vd of the main switching element 22 becomes approximately zero volts. Further, since the input voltage Vi is applied to the dot side of the input winding 20a and the output rectifying element 24 connected to the output winding 20b is reverse-biased, the output rectifying element 24 becomes non-conductive, and the output rectifying element 24 The current If of 24 becomes zero amperes. Then, a switching current Id that increases from zero ampere with a predetermined inclination starts to flow through the path of the input power supply 12, the input winding 20 a, and the main switching element 22, and excitation energy is accumulated in the transformer 20. Although the waveform of the switching current Id is omitted in FIG. 7, it is almost similar to the waveform of the switching current signal Vid.

  The voltage V20c of the auxiliary winding 20c is a negative voltage proportional to the input voltage Vi. The transformer voltage signal Vot is input to the inverting input of the first comparator 68 after the diode 68a becomes conductive and becomes approximately zero volts. On the other hand, the transformer voltage threshold value Vdr of the non-inverting input is about 0.1 V, and the period T1 is “Vot <Vdr”, so the first pulse V1 output from the first comparator 68 is at the high level.

  The voltage control signal Vfb generated by the voltage control signal generation circuit 70 is a substantially constant voltage and is input to the inverting input of the second comparator 72. In normal operation, the voltage control signal Vfb is lower than the threshold value Vth. The switching current signal Vid generated by the switching current signal generation circuit 36 has a waveform that rises from zero volts with a predetermined slope, and is input to the non-inverting input of the second comparator 72. Since the period T1 is “Vfb> Vid”, the second pulse V2 output from the second comparator 72 is at a low level. Accordingly, the drive pulse Vg output from the RS-FF 74 and the non-inverting buffer 76 becomes high level, and the main switching element 22 continues to be turned on during the period T1. The period T1 ends when the switching current signal Vid reaches the voltage control signal Vfb.

  Since “Vfb <Vid” is satisfied for a short period immediately after the start of the period T2, the second comparator 72 operates to invert the second pulse V2 to a high level. On the other hand, the first pulse V1 is held at a high level because “Vot <Vdr” continues. Therefore, the RS-FF 74 inverts the drive pulse Vg to a low level, and the main switching element 22 is turned off.

  When the drive pulse Vg becomes low level, the reset diode 40 is turned on, the input proportional current Jin is bypassed, the timer capacitor 30 is discharged, the switching current signal Vid is instantaneously reduced to almost zero volts, and again “Vfb > Vid ”. Then, the second comparator 72 operates and the second pulse V2 returns to the low level again during the period T2. Further, when the main switching element 22 is turned off, the voltage Vd of the main switching element 22 rises sharply, the voltage V20c of the auxiliary winding 20c rises in the positive direction exceeding zero volts, and accordingly, the transformer voltage signal Vot is increased. Since it rises to “Vot> Vdr”, the first pulse V1 returns to the low level during the period T2.

  Thus, the logic of the first and second pulses V1, V2 changes in the middle of the period T2, but the RS-FF 74 performs an operation of holding the drive pulse Vg at a low level, and during the period T2, The switching element 22 continues to be turned off.

  During the period T2, the output rectifying element 24 is non-conductive while being reverse-biased. The period T2 ends when the voltage V20c increases and the reverse bias is switched to the forward bias.

  During the period T3, the output rectifier element 24 is forward biased and becomes conductive, the voltage V20c of the auxiliary winding 20c becomes a constant voltage substantially proportional to the output voltage Vo, and the voltage Vd of the main switching element 22 also becomes a constant voltage. . In the output rectifying element 24, a current that releases the excitation energy accumulated in the transformer 20, that is, a current that decreases to the right with a predetermined inclination flows as shown in the current If waveform.

  Since “Vot> Vdr” during the period T3, the first pulse V1 is held at a low level. Further, since the switching current signal Vid is substantially zero volts and “Vfb> Vid”, the second pulse V2 is held at a low level. As a result, the drive pulse Vg is held at a low level, and the main switching element 22 continues to be turned off. The period T3 ends when the current Id of the output rectifying element 24 decreases to zero ampere, that is, when the operation of releasing the excitation energy of the transformer 20 ends.

  Even when the period T4 starts, the states of the first and second pulses V1, V2 and the drive pulse Vg are maintained at the state at the end of the period T3, and the main switching element 22 is off. During the period T4, the output rectifying element 24 is also turned off, so that the voltage of each winding of the transformer 20 is not fixed, and free resonance occurs between the inductance component of the transformer 20 and the capacitor component around the transformer 20. The voltage V20c of the auxiliary winding 20c is reduced to a sine wave and becomes a negative voltage. Along with this, the transformer voltage signal Vot also decreases in a sine wave form, but the state of “Vot> Vdr” is maintained during the period T4 due to the phase shift action of the integrating circuit 66. The period T4 ends when the transformer voltage signal Vot reaches the transformer voltage threshold Vdr.

  When the next period T1 starts, “Vot <Vdr” is established, so the first pulse V1 turns to the high level. On the other hand, since “Vfb> Vid” continues, the second pulse V2 continues to be at the low level. Therefore, the drive pulse Vg turns to high level, and the main switching element 22 is turned on. Thereafter, the operations in the above-described periods T1 to T4 are repeated.

  At the operating point P1 (L) where the normal operation is performed, the switching power supply device 62 operates the power conversion circuit 16 in the critical mode, and controls the output voltage Vo to be close to the target value using current mode control using the switching current signal Vid. It is done by. Further, by providing the integrating circuit 66, after the period T4 starts, the timing at which the transformer voltage signal Vot decreases and reaches the transformer voltage threshold Vdr (timing at which the period T4 ends) is delayed, and the main switching element 22 is turned on. The timing (timing at which the period T1 starts) is adjusted so that the voltage Vd of the main switching element 22 becomes close to zero volts. Therefore, quasi-resonant operation is realized, and the switching loss and switching noise of the main switching element 22 are reduced.

  The normal operation performed at the operating point P1 (H) is basically the same as the normal operation performed at the operating point P1 (L) described above. However, since the input voltage Vi is relatively high at the operating point P1 (H), as can be seen from FIGS. 8A and 8B, the slope at which the switching current signal Vid increases is higher than the operating point P1 (L). The period T1 during which the main switching element 22 is turned on is short.

  The overcurrent protection operation performed at the operating point P2 (L) is performed by the second comparator 72 that determines the turn-off timing of the main switching element 22 as compared with the normal operation performed at the operating point P1 (L). The operation is different. In the normal operation performed at the operating point P1 (L), as shown in FIG. 8A, the relationship between the voltage control signal Vfb and the threshold value Vth is “Vfb <Vth”. Therefore, the turn-off timing of the main switching element 22 is The timing when the switching current signal Vid reaches the voltage control signal Vfb. The voltage control signal Vfb is a voltage signal that is varied so that the output voltage Vo approaches the target value Vo1. On the other hand, in the overcurrent protection operation performed at the operating point P2 (L), the relationship between the voltage control signal Vfb and the threshold value Vth becomes “Vfb> Vth” as shown in FIG. The turn-off timing of the element 22 is the timing when the switching current signal Vid reaches the threshold value Vth. Since the threshold value Vth is a constant voltage and the peak value of the switching current signal Vid is limited by the threshold value Vth, as shown in FIG. 6, when the output current Io = Io2 (L), the output voltage Vo is slightly lower than Vo1. Operates in a degraded state.

  The overcurrent protection operation performed at the operating point P2 (H) is basically the same as the overcurrent protection operation performed at the operating point P2 (L) described above. However, since the input voltage Vi is relatively high at the operating point P2 (H), as can be seen from FIGS. 8C and 8D, the slope at which the switching current signal Vid increases is higher than the operating point P2 (L). The period T1 during which the main switching element 22 is turned on is short.

  As described above, the switching power supply device 62 generates the switching current signal Vid using the switching current signal generation circuit 36, and uses this switching current signal Vid to control the output voltage Vo to be close to the target value. This is done in current mode, and pulse-by-pulse overcurrent protection. In particular, the switching current signal generation circuit 36 has a simple configuration and can generate the switching current signal Vid with low loss, which contributes to downsizing and high efficiency of the device.

Incidentally, in the switching power supply device 62, the limit value of the output current Io due to overcurrent protection varies depending on the input voltage Vi, and as shown in FIG. 6, the input value Vio (L) when the input voltage Vi is low is input. The limit value Io2 (H) when the voltage Vi is high is relatively high. This property is caused by the delay time of the second comparator 72 as described in detail in the above-mentioned Patent Document 2. Therefore, when the input voltage range of the switching power supply 62 is wide, there arises a problem that the difference between the limit values Io2 (L) and Io2 (H) becomes large. This problem can be solved by using the switching power supply device of the present invention described below.

In a first embodiment of the present invention, a switching current signal generation circuit 78 instead of the switching current signal generation circuit 36. As shown in FIG. 9, the switching current signal generation circuit 78 includes a timer capacitor 30, a current generation resistor 38, and a reset diode 40 in the same manner as the switching current signal generation circuit 36. A correction resistor 80 is inserted.

  In the correction resistor 80, the input proportional current Jin flows during the period in which the main switching element 22 is on, and a voltage is generated at both ends. This generated voltage is superimposed on the switching current signal Vid. The generated voltage V80 (H) when the input voltage Vi is high is higher than the generated voltage V80 (L) when the input voltage Vi is low. That is, as can be seen from FIGS. 10C and 10D, when the input voltage Vi is high, the peak value of the switching current signal Vid is more likely to reach the threshold value Vth. Accordingly, the difference between the limit values Io2 (H) and Io2 (L) can be reduced by adjusting the resistance value of the correction resistor 80 to an appropriate value.

Further, in the second embodiment of the present invention, a switching current signal generation circuit 82 instead of the switching current signal generation circuit 36. As shown in FIG. 11A, the switching current signal generation circuit 82 includes a timer capacitor 30, a current generation resistor 38, and a reset diode 40 as in the switching current signal generation circuit 36. Here, the current generation resistor 38 is provided. A series circuit of two resistance elements 38 (1) and 38 (2) is provided, and a correction zener diode 84 is connected in parallel to both ends of one resistance element 38 (2).

  The Zener voltage of the correction Zener diode 84 is set to be conductive when the input voltage Vi becomes equal to or higher than a predetermined voltage Vi1. Therefore, as shown in FIG. 11B, when the input voltage Vi is lower than the voltage Vi1, the input proportional current Jin increases in proportion to the input voltage Vi, and when the input voltage Vi is higher than the voltage Vi1, The proportional current Jin increases by ΔJ and the slope becomes a little steep. That is, when the input voltage Vi is high, the peak value of the switching current signal Vid is more likely to reach the threshold value Vth. Therefore, the difference between the limit values Io2 (H) and Io2 (L) is reduced by adjusting the resistance values of the resistance elements 38 (1) and 38 (2) and the Zener voltage of the correction Zener diode 84 to appropriate values. be able to.

Next, still another embodiment of the switching power supply device will be described with reference to FIG. Here, the same components as those in the above embodiment are denoted by the same reference numerals and description thereof is omitted. The switching power supply 86 of this embodiment converts the DC input voltage Vi input from the input power supply 12 into a DC output voltage Vo, and supplies the output voltage Vo and the output current Io to the externally connected load 14. It is a DC converter, and includes a power conversion circuit 16 and a switching current signal generation circuit 36 similar to those described above, and a novel switching control circuit 88.

  The switching control circuit 88 controls the power conversion circuit 16 to operate in a current discontinuous mode with a fixed switching frequency. The switching current signal Vid is used to control the output voltage Vo to be close to the target value in the current mode and to perform pulse-by-pulse overcurrent protection. Hereinafter, the internal configuration of the switching control circuit 88 will be described focusing on differences from the switching control circuit 64.

  In the case of the switching control circuit 64 described above, the auxiliary winding 20c of the transformer 20 is integrated as a circuit block for determining the turn-on timing of the main switching element 22 in order to operate the power conversion circuit 16 in a quasi-resonant critical mode. A circuit 66 and a first comparator 68 were provided. On the other hand, the switching control circuit 88 sets a constant cycle as a circuit block for determining the turn-on timing of the main switching element 22 in order to operate the power conversion circuit 16 in the current discontinuous mode with a fixed switching cycle. A set signal generator 90 for outputting the signal Vset toward the S terminal of the RS-FF 74 is provided, and the auxiliary winding 20c, the integrating circuit 66, and the first comparator 68 are omitted. Other configurations are the same as those of the switching control circuit 64.

  The operation of the switching power supply 86 is different from the switching power supply 62 described above in that the switching cycle is constant, but the control is performed in the current mode in order to bring the output voltage Vo close to the target value, and the operation of the pulse-by-pulse method is performed. The operation for performing the current protection is basically the same as that of the switching power supply device 62, and the same effect can be obtained by the switching current signal generation circuit 36 having a unique configuration. Further, by replacing the switching current signal generation circuit 36 with the switching current signal generation circuits 78 and 82, it is possible to reduce the fluctuation of the overcurrent protection characteristic due to the change of the input voltage Vi.

The switching power supply device of the present invention is not limited to the above embodiment. For example, a specific configuration for generating an input proportional current substantially proportional to the input voltage and charging the timer capacitor while the main switching element is on, and discharging the timer capacitor when the main switching element is off The specific configuration for doing so is not limited to the charging circuit and the reset circuit of the above embodiment, and can be appropriately configured by combining known circuits.

  The switching power supply devices 62 and 86 use the switching current signal Vid for current mode control in order to control the on / off of the main switching element 22 in the current mode, but control the on / off of the main switching element in the voltage mode. In such a case, the switching current signal Vid may be used only for pulse-by-pulse overcurrent protection control. Further, control other than the current mode and overcurrent protection may be performed using the switching current signal Vid.

  Further, the switching current signal generation circuit 82 shown in FIG. 11 may be further provided with a correction resistor 80 in a position in series with the timer capacitor 30 so that the switching current signal can be corrected more effectively.

10, 62, 86 Switching power supply device 16 Power conversion circuit 18, 64, 88 Switching control circuit 20 Transformer 20a Input winding 20b Output winding 22 Main switching element 24 Output rectifier 26 Output smoothing capacitors 28, 36, 42, 50, 78, 82 Switching current signal generation circuit 30 Timer capacitors 32, 44, 52 Charging circuit 34 Reset circuits 38, 48, 56 Current generation resistors 38 (1), 38 (2) Resistance element 40 Reset diode 80 Correction resistor 84 Correction Zener diode
Id switching current
Io output current
Jin input proportional current
Vg drive pulse
Vi input voltage
Vid switching current signal
Vo output voltage
Vth threshold

Claims (3)

A flyback that includes a main switching element, a transformer having an input winding and an output winding, an output rectifying element, and an output smoothing capacitor, and converts the input voltage to a predetermined output voltage by turning on and off the main switching element. A power conversion circuit of a system and a circuit for controlling on / off of the main switching element, wherein the excitation energy accumulated in the transformer during the period in which the main switching element is on is discharged through the output rectifying element, In a switching power supply device comprising a switching control circuit for performing current discontinuous mode control or critical mode control for inverting the main switching element from off to on,
A switching current signal generating circuit that generates a switching current signal that is a voltage signal having a waveform similar to the switching current flowing in the main switching element and outputs the switching current signal to the switching control circuit;
The switching current signal generation circuit is configured such that one end is connected to the ground, the timer capacitor that outputs the switching current signal from the other end, an input proportional current that is substantially proportional to the input voltage, and the main switching element is turned on A charging circuit that charges the timer capacitor during a period, and a reset circuit that discharges the timer capacitor during a period when the main switching element is off,
The switching control circuit controls on / off of the main switching element using the switching current signal generated at both ends of the timer capacitor,
Furthermore, the switching control circuit is provided with overcurrent protection means for forcibly turning off the main switching element when the switching current signal exceeds a predetermined threshold, reducing the output voltage and suppressing an increase in output current. And
In the switching current signal generation circuit, a correction resistor is inserted in a position in series with the timer capacitor,
The switching power supply device is characterized in that the switching current signal is corrected in a direction in which a fluctuation in an overcurrent protection characteristic due to a change in the input voltage is reduced by superimposing a voltage generated in the correction resistor . A flyback that includes a main switching element, a transformer having an input winding and an output winding, an output rectifying element, and an output smoothing capacitor, and converts the input voltage to a predetermined output voltage by turning on and off the main switching element. A power conversion circuit of a system and a circuit for controlling on / off of the main switching element, wherein the excitation energy accumulated in the transformer during the period in which the main switching element is on is discharged through the output rectifying element, In a switching power supply device comprising a switching control circuit for performing current discontinuous mode control or critical mode control for inverting the main switching element from off to on,
A switching current signal generating circuit that generates a switching current signal that is a voltage signal having a waveform similar to the switching current flowing in the main switching element and outputs the switching current signal to the switching control circuit;
The switching current signal generation circuit is configured such that one end is connected to the ground, the timer capacitor that outputs the switching current signal from the other end, an input proportional current that is substantially proportional to the input voltage, and the main switching element is turned on A charging circuit that charges the timer capacitor during a period, and a reset circuit that discharges the timer capacitor during a period when the main switching element is off,
The switching control circuit controls on / off of the main switching element using the switching current signal generated at both ends of the timer capacitor,
Furthermore, the switching control circuit is provided with overcurrent protection means for forcibly turning off the main switching element when the switching current signal exceeds a predetermined threshold, reducing the output voltage and suppressing an increase in output current. And
The charging circuit of the switching current signal generation circuit generates an input proportional current when a voltage approximately proportional to the input voltage is applied to both ends, and a current generation resistor that outputs the generated current to the timer capacitor. The current generating resistor is composed of a series circuit of a plurality of resistance elements, and a correction zener diode is connected in parallel to both ends of a specific resistance element among the plurality of resistance elements,
The switching current signal is corrected so that the fluctuation of the overcurrent protection characteristic due to the change of the input voltage is reduced by the operation of the correction Zener diode limiting the generated voltage of the specific resistance element to a certain value or less. The switching power supply device characterized by the above-mentioned . The switching control circuit generates a driving pulse for driving the main switching element;
The main switching element is configured by a transistor element that is turned on when the drive pulse is at a high level and turned off when the drive pulse is at a low level.
2. The reset circuit includes a reset diode having an anode connected to the other end of the timer capacitor and a cathode connected to a position where the drive pulse or a pulse having the same phase as that is generated. Or the switching power supply device of 2 .

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