JP6379978B2 - Power converter control device - Google Patents

Description

  The present invention relates to a technique for controlling a power converter, and more particularly to a technique for controlling a power converter including a rectifier circuit and an inverter coupled through an LC filter.

  Conventionally, it has been required to reduce harmonic components of current flowing in a power converter. For example, Patent Document 1 listed below exemplifies a configuration in which an LC filter provided between a diode bridge and an inverter unit includes an inductance element. And based on the both-ends voltage of the said inductance element, the inverter part is controlled and the resonance by LC filter is suppressed by it.

  In Patent Document 2 below, the current flowing through the load is Fourier transformed to obtain the (6n-1) -order component and the (6n + 1) -order component for the fundamental frequency of the voltage output from the inverter, and the power source current is based on them. The harmonics are reduced. At this time, in order to reduce power supply harmonics, the modulation factor of the inverter and the voltage command are corrected.

  In Patent Document 3 listed below, a Fourier transform is performed on the command value of the motor current, a sixth-order component for the fundamental frequency of the motor voltage is extracted, and this is removed from the command value, whereby harmonics based on motor current distortion are obtained. The wave component is reduced.

  Patent Document 4 listed below also exemplifies a configuration in which an LC filter provided between a diode bridge and an inverter unit includes an inductance element. Based on the voltage across the inductance element, the voltage control rate (modulation rate) is corrected to reduce the harmonic component of the DC voltage.

Japanese Patent No. 4067021 Japanese Patent No. 5229419 Japanese Patent No. 528809 JP 2014-68498 A

  In Patent Document 2, although the (6n-1) -order component and the (6n + 1) -order component are obtained by performing Fourier transform on the current flowing through the load, there is no mention of the integration period employed in the Fourier transform.

  In Patent Document 3, although the above-described sixth-order component is extracted by performing Fourier transform on the command value of the motor current, an integral multiple of a half cycle of the AC power supply is employed for the integration period employed in the Fourier transform. Yes.

  However, since the Fourier transform is performed using a periodic function, it is desirable to appropriately select the integration period in order to reduce the error.

  Accordingly, an object of the present invention is to provide a technique for performing higher-precision harmonic component reduction in a power converter by performing Fourier transform using an appropriate integration period.

  The power converter control device according to the present invention is a device for controlling the power converter (9). The power converter includes a rectifying circuit (3) that rectifies the first AC voltage (Vr, Vs, Vt) and a rectified voltage (Vdc) obtained from the rectifying circuit via the LC filter (5). Inductivity connected between the inverter (4) that converts the voltage (Vu, Vv, Vw) and applies it to the load (2), and one output terminal of the rectifier circuit and one input terminal of the inverter. An element (L1) and a capacitive element (C1) that is connected between the one input terminal and the other input terminal of the inverter and constitutes the LC filter together with the inductive element.

The first aspect of the power converter control device according to the present invention is that the modulation rate (k) of the inverter or the command value (Vd ^* , Vq ^* ) for the inverter is changed to a correction amount (1-Vh · cosΘ; Vdh). , Vqh), a correction unit (713; 717, 718) for correction, a correction amount generation unit (716A; 716B) for generating the correction amount using a correction phase (Θ), the inductive element or the capacitive element A first phase (δ) is obtained by performing a Fourier transform on the voltage (VL, Vdc) at both ends of the element at a cycle (2π / ω _e ) of the second AC voltage to obtain a phase (θ) of the second AC voltage. Is multiplied by an integer to obtain a second phase (6θ), and a harmonic phase calculation unit (703) that generates the modified phase by adding the first phase and the second phase is provided.

The second mode of the power converter control device according to the present invention is the first mode, wherein the amplitude (Vh) of the correction amount is set to an angular velocity (Vu, Vv, Vw) of the second AC voltage (Vu, Vv, Vw). A harmonic amplitude calculation unit (702) determined based on ω _e ) is further provided.

  A third aspect of the power converter control device according to the present invention is the second aspect, wherein the modulation factor (k) is corrected by multiplying the correction amount (1-Vh · cosΘ). . The correction amount generation unit (716A) includes a cosine value acquisition unit (716g) that calculates a cosine value (cosΘ) of the correction phase, and a multiplier (716h) that multiplies the amplitude (Vh) of the correction amount by the cosine value. ) And a subtracter (716j) for subtracting the multiplication result of the multiplier from the value 1 to obtain the correction amount.

The 4th aspect of the power converter control device concerning this invention is the 2nd aspect, Comprising: The said load (2) is a rotary electric machine. The command values (Vd ^* , Vq ^* ) are corrected by subtracting the correction amounts (Vdh, Vqh). The command value includes a first command value (Vd ^* ) in phase with the field of the rotating electrical machine and a second command value (Vq ^* ) whose phase is advanced by 90 degrees with respect to the field. The correction amount generation unit (716B) includes a sine value acquisition unit (716a) that calculates a sine value (sinΘ) of the correction phase, a cosine value acquisition unit (716b) that calculates a cosine value (cosΘ) of the correction phase, A first multiplier (716c) that multiplies the amplitude (Vh) of the correction amount and the sine value; and a second multiplier (716d) that multiplies the amplitude of the correction amount and the cosine value. The multiplication result (Vdh) of the first multiplier is adopted as the correction amount for the first command value, and the multiplication result (Vqh) of the second multiplier is adopted as the correction amount for the second command value. The

  A fifth aspect of the power converter control device according to the present invention is any one of the first to fourth aspects, wherein the second AC voltage (Vu, Vv, Vw) is a three-phase voltage. The second phase (6θ) is obtained by multiplying the phase (θ) of the second AC voltage by an integer multiple of 6.

  By performing the Fourier transform that employs an appropriate integration period, the harmonic components in the power converter can be reduced more accurately.

It is a circuit diagram which illustrates the composition of the direct form power converter to which the art shown in an embodiment is applicable. It is a block diagram which illustrates the composition of the control device concerning a 1st embodiment. It is a block diagram which illustrates the composition of a harmonic phase operation part. It is a graph explaining the integration period used as a comparative example. It is a graph explaining the integration period in 1st Embodiment. It is a block diagram which illustrates the composition of the compensation term generating part in a 1st embodiment. It is a block diagram which illustrates a part of structure of the control apparatus concerning 2nd Embodiment. It is a block diagram which illustrates the composition of the compensation term generating part in a 2nd embodiment.

A. Configuration of power converter.
FIG. 1 is a circuit diagram illustrating the configuration of a direct power converter 9 that is a power converter to which the technique described in the following embodiment can be applied. The direct power converter 9 includes a converter 3 and an inverter 4 and a pair of DC power supply lines LH and LL that connect them.

  The converter 3 functions as a rectifier circuit and is a three-phase AC voltage Vr, Vs, Vt (hereinafter also referred to as “first AC voltage”) obtained from the AC power source 1 (here, R phase, S phase, and T phase). And rectified voltage Vdc is output to the pair of DC power supply lines LH and LL via the LC filter 5.

  The converter 3 is a current source rectifier, for example, and operates by pulse width modulation. Converter 3 has a plurality of current paths connected in parallel between DC power supply lines LH and LL. The current path of converter 3 corresponding to the R phase includes a pair of switching elements Srp and Srn connected in series between DC power supply lines LH and LL. An AC voltage Vr is applied to a connection point between the switching elements Srp and Srn. The current path of converter 3 corresponding to the S phase includes a pair of switching elements Ssp and Ssn connected in series between DC power supply lines LH and LL. A voltage Vs is applied to a connection point between the switching elements Ssp and Ssn. The current path of converter 3 corresponding to the T phase includes a pair of switching elements Stp and Stn connected in series between DC power supply lines LH and LL. A voltage Vt is applied to a connection point between the switching elements Stp and Stn.

  The switching elements Srp, Ssp, Stp are connected to the DC power supply line LH side, and the switching elements Srn, Ssn, Stn are connected to the DC power supply line LL side.

  The inverter 4 is, for example, a voltage source inverter, and operates by pulse width modulation according to, for example, instantaneous space vector control (hereinafter simply referred to as “vector control”). The inverter 4 outputs three-phase AC voltages Vu, Vv, Vw (hereinafter also referred to as “second AC voltage”) (here, U phase, V phase, and W phase).

  Inverter 4 has a plurality of current paths connected in parallel between DC power supply lines LH and LL.

  The current path of the inverter 4 corresponding to the U phase includes a pair of switching elements Sup and Sun connected in series between the DC power supply lines LH and LL. An AC voltage Vu is obtained from the connection point between the switching elements Sup and Sun. The current path of inverter 4 corresponding to the V phase includes a pair of switching elements Svp and Svn connected in series between DC power supply lines LH and LL. An AC voltage Vv is obtained from a connection point between the switching elements Svp and Svn. The current path of inverter 4 corresponding to the W phase includes a pair of switching elements Swp and Swn connected in series between DC power supply lines LH and LL. An AC voltage Vw is obtained from a connection point between the switching elements Swp and Swn.

  The switching elements Sup, Svp, Swp are connected to the DC power supply line LH side. Hereinafter, these switching elements are grasped as switching elements on the upper arm side. Switching elements Sun, Svn, and Swn are connected to the DC power supply line LL side. Hereinafter, these switching elements will be grasped as switching elements on the lower arm side. That is, the potential of the DC power supply line LH is higher than the potential of the DC power supply line LL.

  Since the configuration of the switching elements Srp, Ssp, Stp, Srn, Ssn, Stn, Sup, Svp, Swp, Sun, Svn, and Swn itself is known, detailed description is omitted.

  The load 2 is an inductive load and is connected to the inverter 4. Specifically, the load 2 is a rotating electrical machine, such as a motor, having a three-phase coil that is Y-connected and to which AC voltages Vu, Vv, and Vw are applied. On the circuit diagram, each resistance component of the three-phase coil is described as a resistor connected in series to the coil. Among the coils, load currents iu, iv, and iw flow through the coils corresponding to the U phase, the V phase, and the W phase, respectively. These currents are monitored by a current sensor (not shown).

  An inductor L1 that is an inductive element is inserted into the DC power supply line LH. Further, a capacitor C1, which is a capacitive element, is provided between the DC power supply lines LH and LL on the inverter 4 side of the inductor L1.

  More specifically, the inductor L1 is connected to one output terminal of the converter 3 (on the side connected to the DC power supply line LH in accordance with FIG. 1) and one of the inverter 4 (in accordance with FIG. 1). In other words, it is connected to the input end (on the side connected to the DC power supply line LH). The capacitor C1 is connected between the one input end of the inverter 4 and the other input end (on the side connected to the DC power supply line LL in accordance with FIG. 1). The capacitor C1 constitutes the LC filter 5 together with the inductor L1.

  For convenience of explanation to be described later, a voltage Vdc across the capacitor C1 (this is equated with the rectified voltage Vdc described above) and a voltage VL across the inductor L1 are respectively introduced. These can be measured by known techniques.

B. First embodiment.
FIG. 2 is a block diagram illustrating the configuration of the control device 7A according to the first embodiment of the invention. The control device 7A functions as a power converter control device that controls the operation of the direct power converter 9.

The control device 7A includes the rotational angular velocity ω _{m of the} motor as the load 2 and its command value ω _m ^* , the angular velocity ω _{e of} the second AC voltages Vu, Vv, and Vw, the induced voltage coefficient Ke, the both-end voltage VL, and the modulation rate before correction. k0, load currents iu, iv, iw, a phase control command value β ^* , and a phase θ which is a motor position angle (electrical angle) are inputted, and gate signals Sup ^* , Svp ^* , Swp ^* , Sun ^* , Svn ^* , Swn. ^{* Is} output.

  In the present embodiment, a technique for obtaining a modulation factor k that reduces harmonics by correcting the modulation factor k0 when the voltage Vdc at both ends is constant will be described.

  The resonance suppression control calculation unit 701 receives the both-end voltage VL, multiplies it by a predetermined gain, or further delays the time, and outputs a resonance suppression amount. The resonance suppression amount is subtracted from the modulation factor k0 in the subtractor 715, and a modulation factor that suppresses fluctuations in the voltage Vdc at both ends due to resonance of the LC filter 5 is obtained. Since this technique is a known technique in Patent Document 4, for example, detailed description thereof is omitted.

The harmonic amplitude calculation unit 702 determines a harmonic amplitude Vh that is proportional to the product of the angular velocity ω _e and the induced voltage coefficient Ke.

Since the technique for obtaining the harmonic amplitude Vh is a known technique such as “Gain adjustment section (54b)” disclosed in Patent Document 3, for example, detailed description thereof is omitted. However, the no-load induced voltage ωφ shown in Patent Document 3 is a quantity corresponding to the product of the angular velocity omega _e and the induced voltage coefficient Ke according to the present embodiment.

Harmonic phase calculating unit 703 inputs the voltage across VL, and phase theta, and the angular velocity omega _e, and outputs the modified phase theta. The configuration and function of the harmonic phase calculation unit 703 will be described in detail later with reference to FIG.

  The compensation term generation unit 716A generates a compensation term for the modulation rate based on the harmonic amplitude Vh and the modified phase Θ. The configuration and function of the compensation term generator 716A will be described in detail later with reference to FIG.

  Multiplier 713 multiplies the compensation term and the output of subtractor 715, and outputs modulation factor k.

The PWM modulation unit 714 uses the d-axis voltage command value Vd ^* , the q-axis voltage command value Vq ^*, and the modulation factor k to generate gate signals Sup ^* , Svp ^* , Swp ^* , Sun ^* , Svn ^* , Swn ^* . Generate. Here, the d-axis is in phase with the motor field (not shown), and the q-axis advances 90 degrees with respect to the field.

  Since the generation of such a gate signal is realized by adopting a known technique, the details of the calculation are omitted. In other words, it is possible to reduce the harmonic power based on the 6 ± 1st harmonic by using a known technique simply by replacing the modulation factor k0 with the modulation factor k.

  If there is no need to suppress the resonance of the LC filter 5, the resonance suppression control calculation unit 701 and the subtractor 715 may be omitted, and the multiplier 713 may multiply the modulation factor k0 and the compensation term.

Since the technique for obtaining the command values Vd ^* and Vq ^* is also a well-known technique, the details will be omitted, but will be briefly described below.

The subtractor 705 obtains the deviation between the rotational angular velocity ω _m and the command value ω _m ^* and inputs it to the PI control unit 706. The PI control unit 706 performs a known PI control (proportional / integral control) to generate a current command value Ia ^* . The current command value Ia ^* and the phase control command value β ^* are input to the d-axis current command value generation unit 707 and the q-axis current command value generation unit 708, and the sign of the phase control command value β ^* is set to be negative ( The sine component and cosine component of the current command value Ia ^* for -β ^* ) are obtained as the d-axis current command value Id ^* and the q-axis current command value Iq ^* .

  The coordinate system conversion unit 704 calculates the d-axis current Id and the q-axis current Iq based on the load currents iu, iv, iw and the phase θ, and outputs them to the subtracters 709 and 710, respectively.

The subtractor 709 outputs a deviation ΔId between the d-axis current Id and its command value Id ^* . The subtractor 710 outputs a deviation ΔIq between the q-axis current Iq and its command value Iq ^* .

The interference compensator 711 performs an operation for compensating for the mutual interference between the d-axis and the q-axis due to the impedances ω _e · Ld, ω _e · Lq based on the motor inductances Ld and Lq and the induced voltage coefficient Ke. d-axis current Id and q-axis current Iq and the difference .DELTA.Id, the ΔIq and the angular velocity ω instruction value serving as the d-axis voltage command Vd of type d-axis voltage _e ^* and the q-axis voltage command value serving q-axis voltage command Vq ^* Generate. Since this calculation is a well-known technique, its details are omitted.

  FIG. 3 is a block diagram illustrating the configuration of the harmonic phase calculation unit 703.

Multiplier 703a is multiplied six times this by entering the angular velocity omega _e, and outputs a sixth-order angular 6ω _e.

The sine value calculator 703c outputs the sine value sin (6ω _e · t) of the product of the input angular velocity 6ω _e and time t. The cosine value calculator 703d outputs a cosine value cos (6ω _e · t) of the product of the input angular velocity 6ω _e and time t. For example, the time t can be taken on the basis of the time when the load current iu becomes zero.

  The multiplier 703e outputs a product of the sine value output from the sine value calculator 703c and the both-end voltage VL. The multiplier 703f outputs the product of the cosine value output from the cosine value calculator 703d and the both-end voltage VL.

  The sum calculator 703g adds the outputs from the multiplier 703e over time Tf. The sum calculator 703i adds the output from the multiplier 703f over time Tf. This addition period Tf will be described in detail later as an integration period Tf.

  The Fourier transform calculation unit 703h obtains the sum of the square of the output of the sum calculator 703g and the square of the sum calculator 703i (hereinafter also referred to as “sum of squares”). Such a so-called product-sum operation is well-known, and the details thereof are omitted.

Phase adjusting unit 703j receives the square sum obtained from the Fourier transform operation unit 703h, and outputs the phase δ of angular velocity 6Omega _e across voltage VL. Multiplier 703k outputs a phase 6θ by multiplying the time t to the angular velocity 6Omega _e derived from multiplier 703a. The adder 703m adds the phase 6θ and the phase δ, and outputs a corrected phase Θ = 6θ + δ. Such functions of the phase adjustment unit 703j and the adder 703m are known (see, for example, Patent Document 3), and detailed description thereof is omitted.

The integration period Tf employed by the sum calculator 703g and the sum calculator 703i is output from the Fourier transform integration period calculator 703b. The Fourier transform integration period calculation unit 703b receives the angular velocity ω _e and outputs an integral multiple of π / ω _e as the integration period Tf. Hereinafter, the integration period Tf will be described with a comparative example.

4 and 5 are graphs for explaining the integration period. By introducing a sixth-order component VL ^ of the both-end voltage VL, the period is 2π / (6ω _e ) = π / (3ω _e ).

  FIG. 4 shows a waveform conforming to the technique disclosed in Patent Document 3. When the frequency fs of the AC voltage Vr obtained from the AC power supply 1 is introduced, an integral multiple of the period 1 / (2fs) is an integration period employed in the Fourier transform.

The second AC voltages Vu, Vv, Vw output from the inverter 4 are not necessarily in an integer multiple relationship with the first AC voltages Vr, Vs, Vt obtained from the AC power source 1. Therefore, the integral multiple of the period 1 / (2fs) is not necessarily an integral multiple of the period π / (3ω _e ) of the sixth-order component VL ^.

As shown in FIG. 4, consider a case where the period 1 / (2fs) itself is adopted as an integer multiple of the period 1 / (2fs). In the example of FIG. 4, a positive integer N1 is introduced in a half cycle in which the AC voltage Vr is positive, and 1 / (2fs) = N1 · π / (3ω _e ) + Δ1 is established, and the AC voltage Vr is negative. By introducing a positive integer N2 in the cycle, 1 / (2fs) = N2 · π / (3ω _e ) + Δ2 is established. However, Δ1 <π / (3ω _e ) and Δ2 <π / (3ω _e ).

If Δ1 = Δ2 = 0 is always established, the Fourier transform error due to the integration period can be ignored in principle. Sixth-order component VL ^ is a periodic function, N1 times the period π / (3ω _e), or the integration result over a N2 times is because matching the integration result over the period π / (3ω _e) .

  However, as described above, in general, it cannot be said that Δ1 = Δ2 = 0 is always established. Therefore, if an integral multiple of the period 1 / (2fs) is set as the integration period of the Fourier transform, an error caused by the periods Δ1 and Δ2 occurs in the result of the Fourier transform corresponding to the sixth-order component VL ^.

On the other hand, FIG. 5 shows a waveform conforming to the technique employed in the present embodiment. One cycle of the second AC voltage Vu or the load current iu is 2π / ω _e , and the half cycle π / ω _e is three times the cycle π / (3ω _e ) of the sixth-order component VL ^. Therefore, by adopting an integral multiple of π / ω _e as the integral period Tf, the integral period Tf is always an integral multiple of the period π / (3ω _e ). Thereby, the error of the Fourier transform can be ignored in principle.

  In this way, the harmonic component in the power converter can be reduced more accurately by performing the Fourier transform employing an appropriate integration period.

  FIG. 6 is a block diagram illustrating the configuration of the compensation term generator 716A.

  The cosine value calculator 716g obtains the cosine value cos (Θ) of the input corrected phase Θ. The multiplier 716h multiplies the cosine value cos (Θ) by the harmonic amplitude Vh. The subtractor 716j subtracts the multiplication result of the multiplier 716h from the value 1, and outputs a correction amount (1-Vh · cos (Θ)). This correction amount is used for the multiplication of the multiplier 713 (see FIG. 2) as the compensation term. That is, the compensation term generation unit 716A is a correction amount generation unit that generates the correction amount (1-Vh · cos Θ) using the correction phase Θ, and the multiplier 713 corrects the modulation factor with the correction amount (1-Vh · cos Θ). Each functions as a correction unit.

  A technique that can reduce the harmonics of the 6 ± 1st order components of the load currents iu, iv, iw by multiplying such a compensation term (correction amount) by the modulation factor is known, for example, in Patent Document 2 and the like. Therefore, the details are omitted.

C. Second embodiment.
FIG. 7 is a block diagram illustrating a part of the configuration of the control device 7B according to the second embodiment of this invention. The control device 7B functions as a power converter control device that controls the operation of the direct power converter 9.

  The control device 7B is similar to the control device 7A shown in FIG. 2 in the first embodiment, and includes a resonance suppression control calculation unit 701, a harmonic amplitude calculation unit 702, a harmonic phase calculation unit 703, and a coordinate system conversion unit. 704, a subtractor 705, a PI control unit 706, a d-axis current command value generation unit 707, a q-axis current command value generation unit 708, subtracters 709 and 710, an interference compensation unit 711, and a subtractor 715. Therefore, in FIG. 7, constituent elements that are the same as those in FIG. 2 are omitted or simplified.

The control device 7B includes a compensation term generation unit 716B instead of the compensation term generation unit 716A of the control device 7A. Subtractors 718 and 717 are also provided. The compensation terms output from the compensation term generator 716B modify the command values Vd ^* and Vq ^* in the subtracters 718 and 717, respectively.

  Also in this embodiment, a case where the modulation rate is corrected using the resonance suppression control calculation unit 701 and the subtractor 715 is illustrated. However, as described in the first embodiment, if it is not necessary to suppress the resonance of the LC filter 5, the resonance suppression control calculation unit 701 and the subtracter 715 are omitted, and the modulation factor k0 is input to the PWM modulation unit 714. May be.

  FIG. 8 is a block diagram illustrating the configuration of the compensation term generation unit 716B.

The sine value calculator 716a functions as a sine value acquisition unit that calculates the sine value sin (Θ) of the input corrected phase Θ. The cosine value calculator 716b functions as a cosine value acquisition unit that calculates the cosine value cos (Θ) of the input corrected phase Θ. The multiplier 716c multiplies the sine value sin (Θ) by the harmonic amplitude Vh. The multiplier 716d multiplies the cosine value cos (Θ) by the harmonic amplitude Vh. The multiplication result of the multiplier 716c, that is, Vh · sin (Θ)) is output as the correction amount Vdh. The multiplication result of the multiplier 716d, that is, Vh · cos (Θ) is output as the correction amount Vqh. The correction amounts Vdh and Vqh are compensation terms in the present embodiment. That is, the compensation term generation unit 716B is a correction amount generation unit that generates the correction amounts Vdh and Vqh using the correction phase Θ, and the subtractors 718 and 717 correct the command values Vd ^* and Vq ^* with the correction amounts Vdh and Vqh. Each functions as a correction unit.

A technique that can reduce the harmonics of the 6 ± 1st order components of the load currents iu, iv, iw by subtracting the correction amounts Vdh, Vqh from the command values Vd ^* , Vq ^* , respectively, is disclosed in, for example, Patent Document 2 The details are omitted here.

  In the present embodiment, similarly to the first embodiment, the harmonic component in the power converter can be more accurately reduced by performing the Fourier transform employing an appropriate integration period.

  In the present invention, the above-described various embodiments can be appropriately modified and omitted within the scope of the present invention as long as they do not contradict each other.

For example multiplier 703a This was multiplied to 6n times to input angular velocity omega _e, may output 6n following angular 6n · omega _e. In this case, the harmonics of the 6n ± 1st order components of the load currents iu, iv, iw can be reduced.

  Further, the sum of the both-end voltages VL and Vdc becomes the output voltage of the converter 3, which is determined by the first AC voltages Vr, Vs and Vt. Therefore, the 6n-order component of the both-end voltage Vdc is only opposite in sign to the 6n-order component of the both-end voltage VL. Therefore, it is possible to change the design such that the voltage Vdc at both ends is input to the harmonic amplitude calculation unit 702 and the harmonic phase calculation unit 703 to obtain the harmonic amplitude Vh and the corrected phase Θ, respectively, instead of the voltage VL at both ends. .

2 Load 3 Rectifier circuit 4 Inverter 702 Harmonic amplitude calculator 703 Harmonic phase calculator 713, 716c, 716d, 716h Multiplier 716a Sine value calculator 716b Cosine value calculator 716A, 716B Compensation term generator 716j, 717, 718 Subtractor 9 Power converter C1 Capacitor L1 Inductor

Claims (5)

A rectifier circuit (3) for rectifying the first AC voltage (Vr, Vs, Vt);
An inverter (4) for converting a rectified voltage (Vdc) obtained from the rectifier circuit through an LC filter (5) into a second AC voltage (Vu, Vv, Vw) and applying it to a load (2);
An inductive element (L1) connected between one output terminal of the rectifier circuit and one input terminal of the inverter, and connected between the one input terminal and the other input terminal of the inverter. A power converter (9) comprising a capacitive element (C1) that constitutes the LC filter together with the inductive element,
A correction unit (713; 717, 718) for correcting the modulation factor (k) of the inverter or a command value (Vd ^* , Vq ^* ) for the inverter with a correction amount (1-Vh · cosΘ; Vdh, Vqh);
A correction amount generation unit (716A; 716B) that generates the correction amount using a correction phase (Θ);
A first phase (δ) is obtained by performing a Fourier transform on the voltage (VL, Vdc) across the inductive element or the capacitive element at a period (2π / ω _e ) of the second AC voltage, (2) A harmonic phase calculation unit (703) for obtaining the second phase (6θ) by multiplying the phase (θ) of the AC voltage by an integer and adding the first phase and the second phase to generate the corrected phase A power converter control device. The harmonic amplitude calculation unit (702) according to claim 1, further comprising a harmonic amplitude calculation unit (702) that determines the amplitude (Vh) of the correction amount based on an angular velocity (ω _e ) of the second AC voltage (Vu, Vv, Vw). Power converter control device. The modulation factor (k) is corrected by multiplying the correction amount (1-Vh · cos Θ),
The correction amount generation unit (716A)
A cosine value obtaining unit (716 g) for obtaining a cosine value (cos Θ) of the corrected phase;
A multiplier (716h) for multiplying the amplitude (Vh) of the correction amount by the cosine value;
The power converter control device according to claim 2, further comprising: a subtractor (716 j) that subtracts the multiplication result of the multiplier from the value 1 to obtain the correction amount. The load (2) is a rotating electric machine,
The command value (Vd ^* , Vq ^* ) is corrected by subtracting the correction amount (Vdh, Vqh),
The command value includes a first command value (Vd ^* ) in phase with the field of the rotating electrical machine and a second command value (Vq ^* ) whose phase is advanced by 90 degrees with respect to the field.
The correction amount generation unit (716B)
A sine value acquisition unit (716a) for obtaining a sine value (sin Θ) of the correction phase;
A cosine value obtaining unit (716b) for obtaining a cosine value (cos Θ) of the corrected phase;
A first multiplier (716c) that multiplies the amplitude (Vh) of the correction amount by the sine value;
A second multiplier (716d) for multiplying the amplitude of the correction amount by the cosine value;
The multiplication result (Vdh) of the first multiplier is adopted as the correction amount for the first command value, and the multiplication result (Vqh) of the second multiplier is adopted as the correction amount for the second command value. The power converter control device according to claim 2. The second AC voltage (Vu, Vv, Vw) is a three-phase voltage,
The power converter control device according to any one of claims 1 to 4, wherein the second phase (6θ) is obtained by multiplying the phase (θ) of the second AC voltage by six.

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