JP5929492B2 - Induction machine control device - Google Patents

Description

  The present invention relates to a control device for an induction machine that operates an AC voltage application unit that applies an AC voltage to the induction machine by using a secondary frequency of the induction machine estimated based on an electrical state quantity of the induction machine as an input.

  In a hybrid vehicle in which a rotating machine is mounted as an in-vehicle main engine together with an internal combustion engine, it is well known to employ a synchronous machine as the rotating machine. However, since the synchronous machine generally includes a permanent magnet made of a rare metal material, there is a concern that a stable supply of a necessary material is not always guaranteed.

  On the other hand, since the induction machine does not include a permanent magnet, there is no such concern.

  On the other hand, in a vehicle, there is a concern that it is difficult to secure even a space for mounting a sensor for detecting the rotation state of the rotor of the rotating machine due to restrictions on the mounting space. On the other hand, Patent Document 1 below describes a sensorless control device that does not require a speed sensor in an induction machine.

Japanese Patent No. 2808709

  However, when performing sensorless control, the magnetic flux control becomes unstable during the period until the estimated value of the rotor rotation speed (secondary frequency) approximates the actual value, and an overcurrent flows through the induction machine. There are concerns that it will be impossible.

  The present invention has been made in the process of solving the above-described problems, and its purpose is to input an induction machine secondary frequency estimated based on the electrical state quantity of the induction machine, and to supply an AC voltage to the induction machine. It is an object of the present invention to provide a new induction machine control device that controls the control amount of an induction machine by operating an AC voltage application means to be applied.

  Hereinafter, means for solving the above-described problems and the operation and effect thereof will be described.

According to a first aspect of the present invention, in order to control a control amount of an induction machine (10) in which a rotor (10a) is mechanically connected to a rotation shaft (11a) of an internal combustion engine (11), an AC voltage is applied to the induction machine. An operation means (20) for generating an operation signal of an AC voltage application means (INV) to be applied and outputting the operation signal to the AC voltage application means is provided, and the operation means is estimated based on an electrical state quantity of the induction machine. The sensorless control means (48, 52) for generating the operation signal using the secondary frequency of the induction machine as an input and the detected value of the rotational speed of the rotating shaft of the internal combustion engine as an input for generating the operation signal. When control of the control amount of the induction machine is started by operating the output voltage of the engine speed using means (60, 62, 64) and the AC voltage applying means, the operation signal generated by the engine speed using means is Output and Switching means (66, 68) for switching to the output of the operation signal generated by the sensorless control means when the fixed period has elapsed, the sensorless control means receiving the primary voltage of the induction machine And a secondary magnetic flux calculating means (48) for calculating a secondary magnetic flux of the induction machine, and generating the operation signal with the secondary magnetic flux calculated by the secondary magnetic flux calculating means as an input, Prior to switching from the engine speed utilization unit to the sensorless control unit, the secondary magnetic flux calculation unit calculates the secondary magnetic flux by using a primary voltage corresponding to an operation signal generated by the engine speed utilization unit as an input. It is characterized by doing.

  When sensorless control means is used when starting control of the control amount of the induction machine, there is a concern that the controllability is lowered because the accuracy of the secondary frequency information is low. On the other hand, when using the engine speed utilization means, the controllability is higher at the beginning of control compared to using the sensorless control means, but compared with the sensorless control means as time passes from the start of control. Therefore, controllability is inferior. This is because the responsiveness of the internal combustion engine is usually lower than the responsiveness of the induction machine, so that the detection accuracy of the rotational speed of the rotary shaft of the internal combustion engine is unsatisfactory for the responsiveness required of the induction machine. It is.

  In the above invention, in view of this point, by providing the switching means, it is possible to selectively use the higher controllability of the sensorless control means and the engine speed control means.

  In addition, about the expansion of the concept regarding the following typical embodiment concerning this invention, it describes in the column of "other embodiment" after typical embodiment.

The system block diagram concerning one Embodiment. The flowchart which shows the process sequence at the time of starting of the induction machine concerning the embodiment. The time chart which shows the effect of the embodiment.

  Hereinafter, an embodiment in which an induction machine control device according to the present invention is applied to an induction machine control device in a parallel hybrid vehicle will be described with reference to the drawings.

  The motor generator 10 shown in FIG. 1 is an induction machine. Specifically, it is a three-phase cage induction machine. The motor generator 10 is an in-vehicle main machine, and a rotor 10a thereof is mechanically connected to drive wheels. The rotor 10 a of the motor generator 10 is further mechanically coupled to the crankshaft 11 a of the internal combustion engine 11.

  The motor generator 10 is connected to the high voltage battery 12 via the inverter INV. The inverter INV includes three sets of series connection bodies of switching elements S ¥ p, S ¥ n (¥ = u, v, w), and the connection points of these series connection bodies are U, V, Each is connected to the W phase. As these switching elements S ¥ # (¥ = u, v, w; # = p, n), an insulated gate bipolar transistor (IGBT) is used in the present embodiment. In addition, a diode D ¥ # is connected in antiparallel to each of these.

  In the present embodiment, the following is provided as detection means for detecting the state of the motor generator 10 and the inverter INV. First, a current sensor 14 for detecting currents iv and iw flowing through the V phase and the W phase of the motor generator 10 is provided. Moreover, the voltage sensor 16 which detects the input voltage (power supply voltage VDC) of the inverter INV is provided.

  The detection values of the various sensors are taken into the control device 20. The control device 20 generates and outputs an operation signal for operating the inverter INV based on the detection values of these various sensors. Here, the signal for operating the switching element S ¥ # of the inverter INV is the operation signal g ¥ #.

  The control device 20 operates the inverter INV so as to control the torque of the motor generator 10 to the torque command value Trq *. In the figure, processing relating to control to the torque command value Trq * is shown as a block diagram. This will be described below.

  The currents iv and iw detected by the current sensor 14 are converted into primary currents id1 and iq1, which are currents on the dq axis, in the dq converter 22. Here, the dq coordinate system is an orthogonal two-dimensional fixed coordinate system. Specifically, in the present embodiment, the d-axis is the positive direction of the U-phase of the stator of the motor generator 10, and the q-axis is the direction orthogonal to the direction.

  The primary currents id1 and iq1 are input to the γδ converter 24 and converted into primary currents iγ1 and iδ1 on the γδ axis. The γδ coordinate system is an orthogonal two-dimensional rotational coordinate system that rotates at a power supply angular frequency (hereinafter, primary frequency ω1) that is a rotational frequency of the output voltage vector of the inverter INV. Incidentally, the conversion process by the γδ conversion unit 24 is performed based on the power supply angle (hereinafter referred to as the primary phase θ1) which is the phase of the fundamental component of the output voltage vector of the inverter INV calculated by the process described later.

  In the present embodiment, the control system for controlling the torque command value Trq * is configured as a control system that performs feedback control of the primary currents iγ1 and iδ1 to the primary current command values iγ1 * and iδ1 *. At this time, the primary current command values iγ1 * and iδ1 * cannot be uniquely determined only by giving the torque command value Trq *. Therefore, in this embodiment, the secondary magnetic flux φδ2 of the δ axis is controlled to zero, and the primary current command value iγ1 * is based on the torque command value Trq * and the secondary magnetic flux command value φγ2 * of the γ-axis component. , Iδ1 * is generated.

  That is, the torque feedback control unit 26 calculates the primary current command value iδ1 * of the δ-axis component as an operation amount for feeding back the torque Trq estimated by the processing described later to the torque command value Trq *. More specifically, the primary current command value iδ1 * is calculated as the sum of the proportional element and the integral element of the difference between the torque command value Trq * and the torque Trq.

  On the other hand, the magnetic flux feedback control unit 28 uses the γ-axis component primary current command as an operation amount for feedback-controlling the secondary magnetic flux φγ2 of the γ-axis component estimated by the processing described later to the secondary magnetic flux command value φγ2 *. The value iγ1 * is calculated. Specifically, the primary current command value iγ1 * is calculated as the sum of the proportional elements of the difference between the secondary magnetic flux command value φγ2 * and the secondary magnetic flux φγ2 and the outputs of the integral elements.

  The δ-axis current feedback control unit 30 calculates a δ-axis primary voltage component as an operation amount for performing feedback control of the primary current iδ1 to the primary current command value iδ1 *. Specifically, the primary voltage component on the δ axis is calculated as the sum of the proportional elements and the integral elements of the difference between the primary current command value iδ1 * and the primary current iδ1.

  Further, the γ-axis current feedback control unit 32 calculates a γ-axis primary voltage component as an operation amount for performing feedback control of the primary current iγ1 to the primary current command value iγ1 *. Specifically, the primary voltage component of the γ-axis is calculated as the sum of the proportional element and the integral element of the difference between the primary current command value iγ1 * and the primary current iγ1.

  By adding the feedforward manipulated variable Δvγ1 output from the non-interference control unit 34 to the output value of the γ-axis current feedback control unit 32, the primary voltage command value vγ1 * of the γ-axis is calculated. On the other hand, the δ-axis primary voltage command value vδ1 * is calculated by adding the feedforward manipulated variable Δvδ1 output from the non-interference control unit 34 to the output value of the δ-axis current feedback control unit 30. Note that the feedforward manipulated variables Δvγ1 and Δvδ1 are well known and therefore will not be described here. The definitions of symbols used to express the feedforward manipulated variables Δvγ1 and Δvδ1 in the figure are as follows. It is written in the remarks column at the end of the book.

  The primary voltage command values vγ1 * and vδ1 * are converted into primary voltage command values vd * 1 and vq1 * on the dq axis by the dq converter 36. The primary voltage command values vd * 1 and vq1 * are converted by the uvw converter 38 into a U-phase voltage command value vu *, a V-phase voltage command value vv *, and a W-phase voltage command value vw *. Are input to the operation signal generator 40.

  The operation signal generation unit 40 performs a process of generating an operation signal g ¥ # for the inverter INV so that the output voltage of the inverter INV is a simulation of the voltage command values vu *, vv *, and vw *. In the present embodiment, in particular, based on a PWM signal generated based on a magnitude comparison between a signal in which the voltage command value v ¥ * (¥ = u, v, w) is normalized by the power supply voltage VDC and a triangular wave carrier, An operation signal g ¥ # is generated.

  Next, calculation (estimation) processing of parameters necessary for performing the above-described processing will be described.

  The primary currents iγ1 and iδ1 are input to the slip frequency estimation unit 42. The slip frequency estimating unit 42 receives the primary currents iγ1 and iδ1 and estimates the slip frequency ωs based on the following equation (c1).

ωs = iδ1 / (iγ1 · τ2) (c1)
The above equation (c1) is premised on the setting specific to the present embodiment. That is, since the secondary magnetic flux φδ2 of the δ-axis component is controlled to zero and the secondary magnetic flux command value φγ2 * of the γ-axis component is constant when the torque command value Trq * is constant, the torque command When the value Trq * is constant, the γ-axis primary current iγ1 is assumed to be constant and is calculated.

  On the other hand, the primary frequency calculation unit 44 calculates the primary frequency ω1a by differential calculation of the primary phase θ1a. The secondary frequency calculation unit 46 calculates the secondary frequency ω2a by subtracting the slip frequency ωs from the primary frequency ω1a.

  The secondary magnetic flux calculation means (magnetic flux observer 48) receives the primary currents id1, iq1, primary voltage command values vd1 *, vq1 *, and the secondary frequency ω2 as the output value of the selector 66 as a secondary. The magnetic fluxes φd2e and φq2e are estimated. Here, the subscript “e” indicates an estimated value by the observer. The magnetic flux observer 48 is a minimum dimension observer. The derivation of the minimum dimension observer according to this embodiment is given in the remarks column at the end of this specification.

  On the other hand, the primary phase calculation unit 52 calculates the primary phase θ1a using the following formula (c2) based on the secondary magnetic fluxes φd2e and φq2e.

θ1a = arctan (φq2e / φd2e) (c2)
The primary phase θ1a calculated in this way is used for the calculation process of the primary frequency ω1a by the primary frequency calculation unit 44, and is selected by the selector 68, whereby the primary phase θ1 is used as the γδ conversion unit 24, It is used for coordinate conversion processing by the dq conversion unit 36 and the like.

  The torque estimation unit 56 receives the secondary magnetic fluxes φd2e and φq2e and the primary currents id1 and iq1, and estimates the torque Trq using the following equation (c3).

Trq = Pn · M · (iq1 · φd2e−id1 · φq2e) / L2 (c3)
In the above formula (c3), the number of pole pairs Pn, the secondary inductance L2, and the mutual inductance M are used. This torque Trq is a control amount for torque feedback control.

  As can be seen from the above-described processing, in the present embodiment, sensorless control is performed that does not use a detection value by hardware means that detects rotation speed information of the motor generator 10. In this case, the estimation accuracy of the magnetic flux observer 48 is particularly low during the period after the motor generator 10 is started until the secondary frequency ω2 converges to the actual frequency. Therefore, in the present embodiment, the primary phase θ1b calculated based on the output value CR of the crank angle sensor 19 that detects the rotation angle of the crankshaft 11a of the internal combustion engine 11 is limited to a predetermined time after the motor generator 10 is started. It uses to control the motor generator 10.

  Specifically, the output value CR is input to the secondary frequency calculation unit 60, where the secondary frequency ω2b is calculated. The secondary frequency ω <b> 2 b can be input to the magnetic flux observer 48 via the selector 66. In addition, the adding unit 62 adds the slip frequency ωs to the secondary frequency ω2b, thereby calculating the primary frequency ω1b. The primary frequency ω1b is input to the integrator 64, where the primary phase θ1b is calculated. The primary phase θ1b is selected by the selector 68 immediately after the activation.

  Furthermore, an open-loop control unit 70 that gives a primary voltage command value vγ1 * of the γ component by open-loop control is provided, and by the selector 72, the sum of the output value of the γ-axis current feedback control unit 32 and the feedforward manipulated variable Δvγ1; One of the output values of the open loop control unit 70 can be selected. Here, the output value of the open loop control unit 70 gradually increases the absolute value of the primary voltage command value vγ1 * of the γ component, and is a function having the time t as an independent variable. Specifically, in the present embodiment, the primary voltage command value vγ1 * of the γ component is raised in a ramp shape by adopting the linear function “a · t; a is a proportionality constant”.

  FIG. 2 shows the procedure for switching the selectors 66, 68, 72. This process is repeatedly executed by the control device 20 at a predetermined cycle, for example.

  In this series of processing, first, in step S10, it is determined whether or not a start request for the motor generator 10 has occurred. If an affirmative determination is made in step S10, the output value CR of the crank angle sensor 19 is acquired in step S12. In the subsequent step S14, the secondary frequency ω2b is calculated based on the output value CR. In step S18, the primary frequency ω1b is calculated by adding the secondary frequency ω2b and the slip frequency ωs.

  In subsequent step S18, the primary phase θ1b is calculated as the time integral value of the primary frequency ω1b. The primary phase θ1 is the primary phase θ1b, and the secondary frequency ω2 is the secondary frequency ω2b. These are the selection processes of the selectors 66 and 68. Further, the torque command value Trq * is set to zero. This process constitutes zero setting means in the present embodiment.

  In addition, the primary voltage command value vγ1 * of the γ axis is set as the output value of the open loop control unit 70. This is a selection process of the selector 72.

  In the subsequent step S20, the secondary magnetic flux φd2, φq2 is calculated by the magnetic flux observer 48 based on the secondary frequency ω2b, the primary voltage command values vd1 *, vq1 *, and the primary currents id1, iq1. In step S22, the primary phase θ1a is calculated based on the secondary magnetic fluxes φd2e and φq2e, and the secondary frequency ω2a is calculated based on this.

  In subsequent step S24, it is determined whether or not a predetermined time has elapsed since the affirmative determination in step S10. Here, the predetermined time is set to a time when the absolute value of the difference between the secondary frequency ω2a and the secondary frequency ω2b is assumed to be equal to or less than a specified value.

  When an affirmative determination is made in step S24, the primary phase θ1 is set to the primary phase θ1a and the secondary frequency ω2 is set to the secondary frequency ω2a in step S26. Further, the γ-axis primary voltage command value vγ1 * is switched to the sum of the output value of the γ-axis current feedback control unit 32 and the feedforward manipulated variable Δvγ1. When the process of step S26 is completed or when a negative determination is made in step S10, this series of processes is temporarily terminated.

  FIG. 3 shows the effect of this embodiment. As shown in the figure, although the primary phase θ1b changes periodically immediately after the motor generator 10 is started, it takes time until the primary phase θ1a changes periodically. This means that it takes time until the estimation accuracy of the primary phase θ1a is improved. The primary phase θ1a changes periodically and approximates to the primary phase θ1b, so that the primary phase θ1 used for control is switched to the primary phase θ1a. As a result, the torque Trq can be made to follow the torque command value Trq * with high accuracy immediately after startup.

  Hereinafter, some of the effects of this embodiment will be described.

  (1) After startup of the motor generator 10, the primary phase θ1b calculated based on the secondary frequency ω2b based on the output value CR of the crank angle sensor 19 is used for a predetermined time (engine speed utilization means), and the predetermined time As time passed, the primary phase θ1a (sensorless control means) calculated based on the secondary magnetic fluxes φd2e and φq2e was used. As a result, when the accuracy of the primary phase θ1a and the secondary frequency ω2a is low immediately after startup, the primary phase θ1b and the secondary frequency ω2b with higher accuracy can be used. Further, the control of the motor generator 10 is performed by using the primary phase θ1a and the secondary frequency ω2a that have higher responsiveness than the primary phase θ1b and the secondary frequency ω2b after a predetermined time has elapsed after the start. Can be improved.

  Incidentally, the primary phase θ1a and the secondary frequency ω2a are more responsive than the primary phase θ1b and the secondary frequency ω2b. The response of the internal combustion engine 11 is lower than the response of the motor generator 10. Therefore, the resolution of the secondary frequency ω <b> 2 b calculated from the output value CR of the crank angle sensor 19 is insufficient for the control of the motor generator 10. Such a situation becomes more prominent as the number of poles of the motor generator 10 increases.

  (2) The secondary magnetic fluxes φd2e and φq2e were calculated by the magnetic flux observer 48 with the primary voltage command values id1 * and iq1 * and the secondary frequency ω2b as inputs before the elapse of a predetermined time after starting. Thereby, it is possible to switch to the sensorless control at the timing when the estimation accuracy of the secondary magnetic fluxes φd2e and φq2e by the magnetic flux observer 48 is improved.

  (3) After the motor generator 10 is started, the torque command value Trq * is set to zero over a predetermined time. As a result, it is possible to avoid a situation in which the torque of the motor generator 10 becomes excessively large, and thus it is possible to avoid a decrease in drivability.

(4) The above control is applied to the motor generator 10 as the in-vehicle main machine. In the case of the main engine, since the secondary frequency ω2 varies greatly, the response of the secondary frequency ω2b calculated from the output value CR of the crank angle sensor 19 is a response required to maintain the controllability of the motor generator 10. It tends to be lower than gender. For this reason, the utility value of the switching process to the secondary frequency ω2a is particularly high.
<Other embodiments>
The above embodiment may be modified as follows.

"Sensorless control means"
a) Secondary magnetic flux calculation means The observer is not limited to the minimum dimension observer. For example, “speed sensorless direct vector control of an induction motor using an adaptive secondary magnetic flux observer: Kubota et al., T. IEEE Japan Vol 111- As exemplified in “D, No. 11, 91”, it may be a same-dimensional observer.

  Further, the observer is not limited to the observer that estimates the secondary magnetic flux component in the fixed coordinate system, but may be an observer that estimates the secondary magnetic flux component in the rotating coordinate system, for example.

  However, it is not limited to an observer.

b) Primary Phase Calculation Unit The primary phase calculation unit is not limited to the primary phase calculation unit 52. For example, the output value of the primary phase calculation unit 52 is set as a temporary primary phase, and a value obtained by integrating the differentiated value and an operation for feedback control of the final primary phase to the temporary primary phase are performed. The amount may be added to make the added value the final primary phase.

"Zero setting method"
The torque command value Trq * is not limited to zero. For example, the torque feedback control unit 26 may be stopped and the primary current command value iδ1 * may be fixed to zero.

  Further, for example, when a control system for controlling the secondary magnetic flux φγ2 of the γ axis to zero instead of controlling the secondary magnetic flux φδ2 of the δ axis to zero, the torque is proportional to the primary current iγ1. Therefore, the primary current command value iγ1 * may be controlled to zero.

  However, the zero setting means itself is not essential.

About engine speed utilization means
The rotation shaft of the engine is not limited to the crankshaft 11a, but may be a camshaft (not shown) that rotates in conjunction with the crankshaft 11a, for example.

  For example, in the above-described embodiment, when the torque feedback control unit 26 is stopped as described in the column “about zero setting unit”, it is not necessary to include the secondary magnetic flux calculating unit.

  Moreover, it is not restricted to what is provided with an open loop control means. For example, in the above embodiment, the magnetic flux feedback control unit 28 or the γ-axis current feedback control unit 32 may be used. However, immediately after the start of estimation of the secondary magnetic fluxes φd2e and φq2e by the magnetic flux observer 48, it is desirable to reduce the gain of feedback control in view of the low estimation accuracy.

"About open loop control means"
The method of gradually increasing the absolute value of the primary voltage command value vγ1 * of the γ-axis component is not limited to the method of changing it into a ramp shape. For example, the absolute value may increase in proportion to the square root of time t.

  Further, for example, when a control system for controlling the secondary magnetic flux φγ2 of the γ-axis to zero instead of controlling the secondary magnetic flux φδ2 of the δ-axis to zero is absolute of the primary voltage command value vδ1 * of the d-axis component The value may be increased with time.

About switching means
In the above-described embodiment, the time when the absolute value of the difference between the primary frequency ω1a and the primary frequency ω1b is assumed to be equal to or less than the specified value has elapsed so that the primary phase θ1a changes from the primary phase θ1b. (S20) is not limited to this. For example, switching from the primary phase θ1b to the primary phase θ1a on condition that a state where the absolute value of the difference between the primary frequency ω1b and the primary frequency ω1a is not more than a specified value is continuously detected for a specified time. Also good. Further, for example, the absolute value of the difference between the primary phase θ1a and the primary phase θ1b is equal to or less than a specified value, or a state where the absolute value is equal to or less than the specified value is continuously detected for a specified time. It may be switched to the primary phase θ1a. Further, for example, the primary phase θ1b may be switched to the primary phase θ1a on condition that it is detected that the primary phase θ1a periodically changes with a period of 0 to 360 °.

  Moreover, it is not restricted to what switches from primary phase (theta) 1b to primary phase (theta) 1a discontinuously. For example, the weighted average processing value of the primary phase θ1b and the primary phase θ1a is set as the final primary phase θ1, the weighting factor of the primary phase θ1b is gradually decreased, and the weighting factor of the primary phase θ1a is gradually increased. It may be a thing.

  Further, the estimation of the secondary magnetic fluxes φd2e and φq2e by the magnetic flux observer 48 at the same time as the calculation of the primary phase θ1b is started. For example, after controlling the motor generator 10 over a specified time based on the secondary frequency ω2b and the primary phase θ1b, the secondary frequency ω2b, the primary voltage command values vd1 *, vq1 *, and the primary currents id1, iq1 are obtained. Input to the magnetic flux observer 48 may start estimation of the secondary magnetic fluxes φd2e and φq2e.

  Further, the secondary frequency ω2 as the input of the magnetic flux observer 48 is not limited to the secondary frequency ω2b, but may be the secondary frequency ω2a. In this case, although the time required until the estimation accuracy of the secondary magnetic fluxes φd2e and φq2e is improved, the effect according to the above embodiment can be obtained by appropriately adjusting the switching timing to the sensorless control. be able to.

"About operation means"
The control amount is not limited to torque, but may be rotation speed.

About AC voltage application means
The present invention is not limited to a DC / AC conversion circuit (inverter INV) including a switching element that connects a terminal of a rotating machine to each of a positive electrode and a negative electrode of a DC voltage source. For example, as described in Japanese Patent Application No. 2008-30825, a converter connected to each terminal of the rotating machine may be used.

"About the vehicle"
Not limited to parallel hybrid vehicles. For example, a series hybrid vehicle may be used. Even in this case, the application of the present invention is effective if an induction machine is employed as the generator connected to the internal combustion engine.
<Remarks>
The state equation of the cage induction machine is expressed by the following equations (ca) and (cb).

ただし、１次インダクタンスＬ１，２次インダクタンスＬ２、１次抵抗Ｒ１，２次抵抗Ｒ２，相互インダクタンスＭを用いている。

However, primary inductance L1, secondary inductance L2, primary resistance R1, secondary resistance R2, and mutual inductance M are used.

  Here, the primary current vector I1 = (id1, iq1) is an amount that can be directly observed. If the estimated value of the secondary magnetic flux vector Φ2 is the secondary magnetic flux vector Φ2e, the following equation (cc) is established.

ただし、電流の微分演算を回避する上では中間変数ｘ「＝Φ２−ＧＩ１」を導入することが望ましい。

However, it is desirable to introduce an intermediate variable x “= Φ2−GI1” in order to avoid current differentiation.

  The observer gain G is determined by the following equations (cd) and (ce).

ただし、上記においてオブザーバの極の実部−αおよび虚部βを用いている。ここで、「α＞０」とすることで、２次磁束ベクトルΦ２ｅの各成分を真の値に収束させることができる。

However, in the above, the real part -α and the imaginary part β of the pole of the observer are used. Here, by setting “α> 0”, each component of the secondary magnetic flux vector Φ2e can be converged to a true value.

  DESCRIPTION OF SYMBOLS 10 ... Motor generator (one Embodiment of induction machine), 20 ... Control apparatus, 48 ... Magnetic flux observer.

Claims (1)

AC voltage applying means for applying an AC voltage to the induction machine in order to control a control amount of the induction machine (10) in which the rotor (10a) is mechanically coupled to the rotation shaft (11a) of the internal combustion engine (11). An operation means (20) for generating an operation signal of (INV) and outputting the operation signal to the AC voltage application means;
The operation means includes
Sensorless control means (48, 52) for receiving the secondary frequency of the induction machine estimated based on the electrical state quantity of the induction machine and generating the operation signal;
The detected value of the rotational speed of the rotating shaft of the internal combustion engine is input, the operation signal is generated , and the induction machine is started so that the absolute value of the secondary magnetic flux of the induction machine is gradually increased when starting the induction machine. Engine speed utilization means (60, 62, 64) comprising open loop control means (70) for open loop control of the primary voltage of the machine ;
And outputs the operation signal generated by the engine speed utilizing unit in a predetermined period after the start of the induction machine, that the predetermined period has elapsed, the output of the operation signal generated by the sensor-less control unit Switching means (66, 68) for switching to
With
The sensorless control means includes a secondary magnetic flux calculation means (48) that receives the primary voltage of the induction machine and calculates a secondary magnetic flux of the induction machine, and the secondary flux calculated by the secondary magnetic flux calculation means. The operation signal is generated with magnetic flux as an input,
Prior to switching from the engine speed utilization means to the sensorless control means, the secondary magnetic flux calculation means inputs the primary voltage corresponding to the operation signal generated by the engine speed utilization means and inputs the secondary magnetic flux. To calculate ,
In the case where the operation signal is generated by the engine speed utilization means, there is provided zero setting means (S18) that uses the operation signal as an operation signal for controlling the torque of the induction machine as the control amount to zero. A control device for an induction machine.

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