JP2014050123A - Rotor position estimation apparatus, motor control system and rotor position estimation method - Google Patents

Abstract

PROBLEM TO BE SOLVED: To estimate the rotor position accurately without being affected by the variation or dead time of switching operation of an inverter, in a rotor position estimation apparatus for estimating the rotor position of a motor including a rotor having a salient pole.SOLUTION: A rotor position estimation unit 270 applies a d-axis voltage to an AC motor M1 during stoppage thereof, and detects a q-axis current flowing to the AC motor M1 when the d-axis voltage is applied. Based on the q-axis current thus detected, rotor position during stoppage of the AC motor M1 is estimated. Depending on the magnetic saturation characteristics of the AC motor M1, the rotor position estimation unit 270 sets a d-axis voltage command value so that the motor current has a maximum value within a current range capable of maintaining the salient pole ratio of the AC motor M1 at an acceptable value or more, and controls the system voltage VH to have a minimum value within a range of the system voltage VH where pulse width modulation control is applicable, depending on the d-axis voltage command value.

Description

  The present invention relates to a rotor position estimation device and a rotor position estimation method for estimating a rotor position of an electric motor including a rotor having saliency.

  In an electric vehicle (an electric vehicle, a hybrid vehicle, etc.) equipped with an AC motor as a power source of the vehicle, a synchronous motor such as a permanent magnet type synchronous motor using a permanent magnet as a rotor is applied to the AC motor, and the position of the rotor The energization control of the synchronous motor is performed based on the information. In this energization control of the synchronous motor, the current supplied to each phase coil of the synchronous motor is controlled based on the position information of the rotor. Therefore, a rotational position sensor that detects a rotor position such as a resolver is usually used.

  However, in the configuration in which the rotor position is detected using the rotational position sensor as described above, if there is an error in the mounting position of the rotational position sensor, this becomes an offset error, and the rotor position detected by the rotational position sensor is the actual rotor. It will shift from the position. As a result, the accuracy of energization control of the synchronous motor may be reduced. Therefore, it is necessary to estimate the actual rotor position of the synchronous motor and detect the offset error of the rotational position sensor based on the estimated actual rotor position.

  As a technique for estimating the rotor position of a synchronous motor, for example, Japanese Patent Laid-Open No. 7-245981 (Patent Document 1) detects a motor current when an alternating voltage is applied to a motor with a current detector, A magnetic pole position detection device for an electric motor that estimates the magnetic pole position (rotor position) of the electric motor based on the detection value is described. According to Patent Document 1, the detected motor current is separated into a parallel component and an orthogonal component with respect to the applied alternating voltage, and the magnetic pole position of the motor is detected based on at least one of the parallel component and the orthogonal component of the motor current. To do.

JP 7-245981 A JP 2010-35353 A JP 2010-35352 A JP 2010-35351 A JP 2011-78295 A JP 2007-124836 A

  In the above Patent Document 1, the command value of the alternating voltage applied to the electric motor is converted into a three-phase voltage command value and input to the inverter. Then, when the inverter performs a switching operation according to the three-phase voltage command value, an alternating voltage corresponding to the command value is applied to the motor.

  However, if the switching operation of the switching element varies due to variations in the characteristics of the switching elements constituting the inverter, the voltage (actual voltage) actually applied to the motor by the switching operation deviates from the command value. .

  In addition, in an inverter, in order to prevent two switching elements from being turned on (conductive) at the same time, a dead time for temporarily turning off both switching elements is provided. Due to the influence of this dead time, a deviation occurs between the actual voltage and the command value. As described above, when a deviation occurs between the actual voltage and the command value, it is difficult to accurately estimate the rotor position. Thereby, the precision of the energization control of a synchronous motor may fall.

  The present invention has been made to solve such problems, and the object of the present invention is to accurately estimate the rotor position of the motor without being affected by variations in switching operation and dead time in the inverter. That is.

  In one aspect of the present invention, a rotor position estimation device estimates a rotor position of an electric motor including a rotor having saliency. The rotor position estimation device is detected by a voltage application unit that applies a d-axis voltage while the motor is stopped, a current detection unit that detects a q-axis current that flows through the motor when the d-axis voltage is applied, and a current detection unit. And estimating means for estimating the rotor position when the electric motor is stopped based on the q-axis current. The voltage application means converts the DC voltage into an AC voltage by setting the setting means for setting the command value of the d-axis voltage to be applied to the motor, and performing pulse width modulation control of the inverter according to the d-axis voltage command value. Voltage converting means for applying to the inverter, and DC voltage generating means for variably controlling the input voltage to the inverter according to the d-axis voltage command value. The setting means sets the d-axis voltage command value in accordance with the magnetic saturation characteristics of the motor so that the current flowing through the motor becomes a maximum value within a current range in which the salient pole ratio of the motor can be maintained at an allowable value or more. . The DC voltage generating means controls the input voltage so as to be the minimum value within the range of the input voltage to which the pulse width modulation control can be applied, according to the d-axis voltage command value.

  In another aspect of the present invention, an electric motor control system for controlling an electric motor including a rotor having saliency has a rotational position detecting means for detecting the rotor position of the electric motor, and estimates the rotor position of the electric motor while the electric motor is stopped. A rotor position estimating means, an error detecting means for detecting an error of a detected value of the rotor position by the rotating position detecting means with respect to an estimated value of the rotor position by the rotor position estimating means, and an error detected by the error detecting means Correction means for correcting the rotor position detected by the position detection means, and control means for performing energization control of the electric motor based on the rotor position corrected by the correction means. The rotor position estimating means is detected by a voltage applying means for applying a d-axis voltage while the motor is stopped, a current detecting means for detecting a q-axis current flowing through the motor when the d-axis voltage is applied, and a current detecting means. And estimating means for estimating the rotor position when the electric motor is stopped based on the q-axis current. The voltage application means converts the DC voltage into an AC voltage by setting the setting means for setting the command value of the d-axis voltage to be applied to the motor, and performing pulse width modulation control of the inverter according to the d-axis voltage command value. Voltage converting means for applying to the inverter, and DC voltage generating means for variably controlling the input voltage to the inverter according to the d-axis voltage command value. The setting means sets the d-axis voltage command value in accordance with the magnetic saturation characteristics of the motor so that the current flowing through the motor becomes a maximum value within a current range in which the salient pole ratio of the motor can be maintained at an allowable value or more. . The DC voltage generating means controls the input voltage so as to be the minimum value within the range of the input voltage to which the pulse width modulation control can be applied, according to the d-axis voltage command value.

  In another aspect of the present invention, a rotor position estimation method for estimating a rotor position of an electric motor including a rotor having saliency is provided with a step of applying a d-axis voltage while the electric motor is stopped, and a d-axis voltage is applied. And a step of detecting a q-axis current flowing through the motor and a step of estimating a rotor position when the motor is stopped based on the q-axis current detected by the step of detecting the q-axis current. The step of applying the d-axis voltage includes a step of setting a command value of the d-axis voltage to be applied to the motor, and a pulse width modulation control of the inverter according to the d-axis voltage command value, thereby converting the DC voltage into an AC voltage. Applying to the motor, and variably controlling the input voltage to the inverter in accordance with the d-axis voltage command value. The step of setting the command value of the d-axis voltage is such that the current flowing through the motor becomes a maximum value within a current range in which the salient pole ratio of the motor can be maintained at an allowable value or more according to the magnetic saturation characteristics of the motor. Set the d-axis voltage command value. In the step of variably controlling the input voltage to the inverter, the input voltage is controlled so as to become a minimum value within the range of the input voltage to which the pulse width modulation control can be applied, according to the d-axis voltage command value.

  According to the present invention, it is possible to accurately estimate the rotor position of an electric motor including a rotor having saliency.

1 is a schematic configuration diagram of an electric motor control system to which a rotor position estimation device according to an embodiment of the present invention is applied. It is a functional block diagram of a pulse width modulation control system executed by the control device. It is a figure for demonstrating the offset error of the rotation coordinate system of an AC motor. It is a flowchart explaining the control processing procedure for implement | achieving rotor position estimation according to embodiment of this invention. It is an output waveform figure of d axis voltage applied to an AC motor, and q axis current which flows into an AC motor. It is a figure explaining the sine wave PWM control in the inverter by step S07 of FIG. It is a figure for demonstrating the influence of the dispersion | variation in the switching operation in an inverter. It is a figure explaining the magnetic saturation characteristic of an AC motor. It is a figure which shows the relationship between a carrier signal, a U-phase voltage command value, and a switching control signal. It is a figure which shows the relationship between the voltage (actual voltage) actually applied to an alternating current motor, and d-axis voltage command value. It is a figure which shows the relationship between a carrier signal and a U-phase voltage command value. It is a figure explaining the setting of a system voltage command value.

  Embodiments of the present invention will be described below in detail with reference to the drawings. In the following, the same or corresponding parts in the drawings are denoted by the same reference numerals, and the description thereof will not be repeated in principle.

(overall structure)
FIG. 1 is a schematic configuration diagram of an electric motor control system to which a rotor position estimation device according to an embodiment of the present invention is applied.

  Referring to FIG. 1, electric motor control system 100 includes a DC voltage generation unit 10 #, a smoothing capacitor C0, an inverter 14, a control device 30, and an AC electric motor M1.

  The AC motor M1 is a synchronous motor including a rotor having saliency, and for example, a permanent magnet type synchronous motor using a permanent magnet for the rotor is applied.

  In the present embodiment, AC electric motor M1 is a driving electric motor that generates torque for driving the driving wheels of an electric vehicle such as a hybrid vehicle or an electric vehicle. The electric vehicle includes all vehicles equipped with an electric motor for generating vehicle driving force, including an electric vehicle not equipped with an engine. Note that AC motor M1 is generally configured to have both functions of an electric motor and a generator. Further, this AC motor M1 may be configured to have a function of a generator driven by an engine in a hybrid vehicle. Further, AC electric motor M1 may operate as an electric motor for the engine, and may be incorporated in a hybrid vehicle as one that can start the engine, for example.

  DC voltage generation unit 10 # includes a DC power supply B, system relays SR1 and SR2, a smoothing capacitor C1, and a converter 12.

  The DC power source B is composed of a secondary battery such as nickel metal hydride or lithium ion, a fuel cell, an electric double layer capacitor, or a combination thereof. The voltage (Vb), current, and temperature of the DC power supply B are detected by the sensor 10 provided in the DC power supply B. A value detected by the sensor 10 is output to the control device 30.

  System relay SR 1 is connected between the positive terminal of DC power supply B and power line 6, and system relay SR 2 is connected between the negative terminal of DC power supply B and power line 5. System relays SR1 and SR2 are turned on / off by signal SE from control device 30. Smoothing capacitor C <b> 1 is connected between power line 6 and power line 5. The DC voltage VL between the power line 6 and the power line 5 is detected by the voltage sensor 11. The value detected by the voltage sensor 11 is sent to the control device 30.

  Converter 12 includes a reactor L1, power semiconductor switching elements Q1, Q2, and diodes D1, D2.

  Power semiconductor switching elements Q 1 and Q 2 are connected in series between power line 7 and power line 5. On / off of power semiconductor switching elements Q 1 and Q 2 is controlled by switching control signals S 1 and S 2 from control device 30.

  In the embodiment of the present invention, an IGBT (Insulated Gate Bipolar Transistor), a power MOS (Metal Oxide Semiconductor) transistor, or a power bipolar transistor is used as a power semiconductor switching element (hereinafter simply referred to as “switching element”). Etc. can be used. Anti-parallel diodes D1, D2 are arranged for switching elements Q1, Q2.

Reactor L1 is connected between a connection node of switching elements Q1 and Q2 and power line 6. Further, the smoothing capacitor C0 is connected between the power line 7 and the power line 5.
Inverter 14 includes a U-phase arm 15, a V-phase arm 16, and a W-phase arm 17 provided in parallel between power line 7 and power line 5. Each phase arm is composed of switching elements connected in series between power line 7 and power line 5. For example, U-phase arm 15 includes switching elements Q3 and Q4, V-phase arm 16 includes switching elements Q5 and Q6, and W-phase arm 17 includes switching elements Q7 and Q8. Further, antiparallel diodes D3 to D8 are connected to switching elements Q3 to Q8, respectively. Switching elements Q3 to Q8 are turned on / off by switching control signals S3 to S8 from control device 30.

  An intermediate point of each phase arm is connected to each phase end of each phase coil of AC electric motor M1. Typically, AC electric motor M1 is a three-phase permanent magnet motor, and is configured such that one end of three coils of U, V, and W phases are commonly connected to a midpoint. Furthermore, the other end of each phase coil is connected to the midpoint of the switching elements of each phase arm 15-17.

  Converter 12 is basically controlled such that switching elements Q1 and Q2 are turned on and off in a complementary manner in each switching period. During the boosting operation, converter 12 boosts DC voltage VL supplied from DC power supply B to DC voltage VH (this DC voltage corresponding to the input voltage to inverter 14 is hereinafter also referred to as “system voltage”). This step-up operation is performed by supplying the electromagnetic energy stored in reactor L1 during the ON period of switching element Q2 to power line 7 via switching element Q1 and antiparallel diode D1.

  Converter 12 steps down DC voltage VH to DC voltage VL during the step-down operation. This step-down operation is performed by supplying the electromagnetic energy accumulated in the reactor L1 during the ON period of the switching element Q1 to the power line 6 via the switching element Q2 and the antiparallel diode D2. The voltage conversion ratio (the ratio of VH and VL) in these step-up or step-down operations is controlled by the on-period ratio (duty ratio) of the switching elements Q1 and Q2 with respect to the switching period. If switching elements Q1 and Q2 are fixed to ON and OFF, respectively, VH = VL (voltage conversion ratio = 1.0) can be obtained.

  The smoothing capacitor C0 smoothes the DC voltage on the power line 7. The voltage sensor 13 detects the voltage across the smoothing capacitor C 0, that is, the system voltage VH, and outputs the detected value to the control device 30.

  When the torque command value of AC motor M1 is positive (Trqcom> 0), inverter 14 operates on power line 7 by the switching operation of switching elements Q3 to Q8 in response to switching control signals S3 to S8 from control device 30. The AC motor M1 is driven so as to convert the DC voltage into an AC voltage and output a positive torque. Further, when the torque command value of AC electric motor M1 is zero (Trqcom = 0), inverter 14 converts the DC voltage to the AC voltage by the switching operation in response to switching control signals S3 to S8, and the torque is zero. The AC motor M1 is driven so that Thus, AC electric motor M1 is driven to generate zero or positive torque designated by torque command value Trqcom.

  Furthermore, during regenerative braking of an electric vehicle equipped with electric motor control system 100, torque command value Trqcom of AC electric motor M1 is set to be negative (Trqcom <0). In this case, the inverter 14 converts the AC voltage generated by the AC motor M1 into a DC voltage by a switching operation in response to the switching control signals S3 to S8, and converts the converted DC voltage (system voltage) to the smoothing capacitor C0. To the converter 12 via The regenerative braking here refers to braking with regenerative power generation when the driver operating the electric vehicle performs a footbrake operation, or regenerative braking by turning off the accelerator pedal while driving, although the footbrake is not operated. This includes decelerating (or stopping acceleration) the vehicle while generating electricity.

  Current sensor 24 detects a current (phase current) flowing through AC electric motor M <b> 1 and outputs the detected value to control device 30. Since the sum of instantaneous values of the three-phase currents iu, iv, and iw is zero, the motor currents for two phases (for example, the V-phase current iv and the W-phase current iw) are detected as shown in FIG. You may arrange in.

  The rotational position sensor (resolver) 25 detects the rotor position (rotor magnetic pole position) θc of the AC motor M1, and sends the detected rotor position θc to the control device 30. The control device 30 can calculate the rotational speed and the rotational angular velocity ω of the AC motor M1 based on the rotor position θc.

  The control device 30 is configured by an electronic control unit (ECU), and performs an electric motor by software processing by executing a program stored in advance by a CPU (Central Processing Unit) (not shown) and / or hardware processing by a dedicated electronic circuit. The operation of the control system 100 is controlled.

  As a representative function, the control device 30 is detected by the detection value by the sensor 10, the torque command value Trqcom, the DC voltage VL detected by the voltage sensor 11, the system voltage VH detected by the voltage sensor 13, and the current sensor 24. The operation of the converter 12 and the inverter 14 is controlled based on the motor currents iv and iw, the rotor position θc from the rotational position sensor 25, and the like. That is, switching control signals S1 to S8 for controlling converter 12 and inverter 14 as described above are generated and output to converter 12 and inverter 14.

  Specifically, control device 30 feedback-controls system voltage VH and generates switching control signals S1 and S2 so that system voltage VH matches the voltage command value. In addition, control device 30 generates switching control signals S3 to S8 and outputs them to inverter 14 so that AC electric motor M1 outputs torque according to torque command value Trqcom by a control method described later. Further, control device 30 controls on / off of system relays SR1 and SR2 in response to the start / stop of electric motor control system 100.

(Control configuration)
Next, power conversion in the inverter 14 controlled by the control device 30 will be described.

  FIG. 2 is a functional block diagram of a pulse width modulation (PWM) control method executed by the control device 30. In the embodiment of the present invention, sine wave PWM control which is a general PWM control method is applied to power conversion in the inverter 14. It should be noted that the sine wave PWM control method, the overmodulation PWM control method, and the rectangular wave voltage control may be switched and applied in accordance with the operating conditions (typically torque / rotation speed) of the AC motor M1. Good.

  Referring to FIG. 2, control device 30 includes current command generation unit 210, coordinate conversion units 220 and 250, rotation speed calculation unit 230, PI calculation unit 240, PWM signal generation unit 260, and rotor position estimation. Part 270.

  Current command generation unit 210 generates a d-axis current command value Idcom and a q-axis current command value Iqcom corresponding to torque command value Trqcom according to a table or the like created in advance.

  The coordinate conversion unit 220 performs the V-phase current iv detected by the current sensor 24 by coordinate conversion (3 phase → 2 phase) using the rotor position θc # of the AC motor M1 output from the rotor position estimation unit 270 described later. The d-axis current Id and the q-axis current Iq are calculated based on the W-phase current iw. Rotational speed calculation unit 230 calculates rotational speed Nmt (or rotational angular velocity ω) of AC electric motor M1 based on rotor position θc # from rotor position estimation unit 270.

  A deviation ΔId (ΔId = Idcom−Id) with respect to the command value of the d-axis current and a deviation ΔIq (ΔIq = Iqcom−Iq) with respect to the command value of the q-axis current are input to the PI calculation unit 240. PI calculation unit 240 performs PI calculation with a predetermined gain on d-axis current deviation ΔId and q-axis current deviation ΔIq to obtain a control deviation, and d-axis voltage command value Vd # and q-axis voltage command corresponding to the control deviation. A value Vq # is generated.

  Coordinate conversion unit 250 performs d-axis voltage command value Vd # and q-axis voltage command value Vq # by coordinate conversion (2 phase → 3 phase) using rotor position θc # of AC electric motor M1 from rotor position estimation unit 270. Is converted into U-phase, V-phase, and W-phase voltage command values Vu, Vv, and Vw. The system voltage VH is also reflected in the conversion from the d-axis and q-axis voltage command values Vd # and Vq # to the phase voltage command values Vu, Vv and Vw.

  The PWM signal generation unit 260 generates the switching control signals S3 to S8 shown in FIG. 1 based on the comparison between the voltage command values Vu, Vv, Vw in each phase and a predetermined carrier wave (carrier signal). When the inverter 14 is subjected to switching control according to the switching control signals S3 to S8 generated by the control device 30, an AC voltage for outputting torque according to the torque command value Trqcom input to the current command generation unit 210 is generated. Applied.

  As described above, a closed loop for controlling the motor current to the current command value (Idcom, Iqcom) corresponding to the torque command value Trqcom is configured, whereby the output torque of the AC motor M1 is controlled according to the torque command value Trqcom.

(Rotor position estimation)
As described above, the energization control of the AC motor M1 is performed based on the rotor position θc detected by the rotational position sensor 25. However, if there is an error in the mounting position of the rotational position sensor 25, this becomes an offset error, and the output θc of the rotational position sensor 25 may deviate from the actual rotor position (hereinafter also referred to as “actual rotor position”). There is sex. As a result, the accuracy of energization control of AC electric motor M1 decreases, and it may be difficult to output torque according to torque command value Trqcom. Therefore, it is necessary to obtain an offset error of the rotational position sensor 25 and correct the output of the rotational position sensor 25 using the offset error.

  In the present embodiment, rotor position estimating unit 270 uses the inductance saliency to estimate the actual rotor position when AC motor M1 is stopped. Then, an offset error which is a deviation of the output θc of the rotational position sensor 25 with respect to the actual rotor position is obtained.

FIG. 3 is a diagram for explaining an offset error in the rotational coordinate system of AC electric motor M1.
In the control of the AC motor M1, the currents iu, iv, and iw that flow through the three-phase coils are used in a two-axis rotational coordinate system that has a magnetic flux direction (d-axis) of the permanent magnet of the rotor and a direction (q-axis) orthogonal thereto. It is done. FIG. 3 shows a motor model on a rotational coordinate (dq coordinate) system based on the actual rotor position θ. FIG. 3 further shows a rotation coordinate (dc-qc coordinate) system used for energization control of AC electric motor M1. The rotational coordinate system used for this control has a magnetic flux direction (dc axis) of the permanent magnet of the rotor determined based on the output θc of the rotational position sensor 25 and a direction (qc axis) perpendicular thereto. Hereinafter, the dc-qc axis of the rotational coordinate system (dc-qc coordinate) system used for this control is also referred to as a “control axis”.

  Here, if there is an error in the mounting position of the rotational position sensor 25, an offset error Δθ that is a deviation of the output θc of the rotational position sensor 25 from the actual rotor position θ occurs. As a result, the control axis (dc-qc axis) has an offset error Δθ with respect to the dq axis due to the actual rotor position θ.

  Rotor position estimation unit 270 estimates actual rotor position θ when AC electric motor M1 is stopped. Then, the offset error Δθ of the rotational position sensor 25 is calculated using the estimated actual rotor position θ. The rotor position estimation unit 270 corrects the rotor position θc detected by the rotation position sensor 25 using the calculated offset error Δθ, and the corrected rotor position θc # is converted into the coordinate conversion units 220 and 250 and the rotation speed calculation unit. 230.

  Hereinafter, the rotor position estimation in the rotor position estimation unit 270 according to the present embodiment will be described in detail.

  Returning to FIG. 2, the rotor position estimation unit 270 applies a predetermined pulse voltage to the dc axis that is the control axis when the AC motor M <b> 1 is stopped, and based on the output of the current sensor 24 when the pulse voltage is applied. To estimate the actual rotor position θ.

  Specifically, the d-axis voltage command value in the dc-qc coordinate system (control axis) used for the energization control of the AC motor M1 is set to a predetermined value Vd other than 0, and the q-axis voltage command value is set to 0. . That is, the voltage vector is set to be controlled on the dc axis of the dc-qc coordinate system. In the following description, in order to distinguish from d-axis voltage command value Vd # and q-axis voltage command value Vq # in energization control of AC electric motor M1, d-axis voltage command value and q-axis voltage command value for rotor position estimation are used. Are also expressed as “d-axis voltage command value Vdc” and “q-axis voltage command value Vqc”.

  The coordinate conversion unit 250 receives the d-axis voltage command value Vdc (= Vd) and the q-axis voltage command value Vqc (= 0) from the rotor position estimation unit 270, and the rotor position (AC motor M1) detected by the rotational position sensor 25. (Rotor position at the time of stopping) θc and d-axis voltage application time Td. The d-axis voltage application time Td corresponds to the pulse width of a predetermined pulse voltage applied to the dc axis. The d-axis voltage command value Vdc corresponds to the pulse height of the pulse voltage.

  The coordinate conversion unit 250 converts the d-axis voltage command value Vdc and the q-axis voltage command value Vqc into the U phase and the V phase by coordinate conversion using the rotor position θc detected by the rotational position sensor 25 (2 phase → 3 phase). , W-phase voltage command values Vu, Vv, Vw are converted. The PWM signal generation unit 260 generates the switching control signals S3 to S8 based on the comparison of the voltage command values Vu, Vv, Vw and the carrier signal in each phase. The inverter 14 is subjected to switching control according to the switching control signals S3 to S8 generated by the control device 30, whereby a predetermined pulse voltage is applied to the dc axis of the AC motor M1.

  Here, as is generally known, the voltage equation of each phase of the U phase, the V phase, and the W phase in the permanent magnet type synchronous motor is expressed by the following equation (1).

  In Equation (1), R represents the armature winding resistance, and φ represents the number of armature linkage fluxes of the permanent magnet.

  By treating the AC motor M1 as a motor model on the rotational coordinate (dq axis) system shown in FIG. 3, the voltage equation of the above equation (1) is on the dq axis shown in the following equation (2). Is converted to the voltage equation.

  In Expression (2), ω represents the electrical angular velocity of the AC motor M1, Ld represents the d-axis inductance, and Lq represents the q-axis inductance.

  Further, the voltage equation on the dq axis shown in the above equation (2) is converted into a voltage equation on the control axis (dc-qc axis) having an offset error Δθ with respect to the dq axis, thereby performing control. The voltage equation on the axis (dc-qc axis) is expressed by the following expression (3).

  In Equation (3), Vdc and Vqc indicate the d-axis voltage and the q-axis voltage on the control axis (dc-qc axis), and Idc and Iqc indicate the d-axis current and the control axis (dc-qc axis) and The q-axis current is shown. That is, Expression (3) is a mathematical expression of the motor model on the control axis (dc-qc axis) based on the output θc of the rotational position sensor 25.

  Since AC motor M1 is now stopped, the electrical angular velocity ω = 0 of AC motor M1 in equation (3) is set. Then, the following equations (4) and (5) are obtained by solving the equation (3) for the currents Idc and Iqc on the control axis (dc-qc axis).

  In the current equations (4) and (5), consider the case where the offset error Δθ = 0 of the rotational position sensor 25, that is, the case where the actual rotor position θ and the output θc of the rotational position sensor 25 coincide. In this case, since sin2Δθ = 0 and cos2Δθ = 1 in each equation, equations (4) and (5) are replaced by the following equations (6) and (7), respectively.

  When the above-described d-axis voltage command value Vdc (= Vd) and q-axis voltage command value Vqc (= 0) are applied to the d-axis voltage Vdc and q-axis voltage in the above formulas (6) and (7), control is performed. The current equation on the axis (dc-qc axis) is expressed by the following expressions (8) and (9).

  As shown in the above formulas (8) and (9), when a predetermined pulse voltage is applied to the dc axis as the control axis while the AC motor M1 is stopped, the offset error Δθ = 0 of the rotational position sensor 25 The d-axis current Idc is a predetermined value other than 0, while the q-axis current Iqc = 0. That is, when the offset error Δθ = 0, only the d-axis current Idc flows and the q-axis current Iqc does not flow.

  Therefore, by obtaining a control axis where the q-axis current Iqc = 0 when a predetermined pulse voltage is applied to the dc axis, the rotor position θc corresponding to the control axis is determined as the actual rotor position θ when the AC motor M1 is stopped. Can be estimated.

  In the present embodiment, the rotor position estimation unit 270 rotates the control axis (dc-qc axis) based on the output θc of the rotational position sensor 25 within a predetermined rotation angle range, and controls the control axis for each rotation angle. A predetermined pulse voltage is applied to the dc axis. Then, the rotor position θc corresponding to the control axis when the q-axis current Iqc when the predetermined pulse voltage is applied is closest to 0 is estimated as the actual rotor position θc.

  FIG. 4 is a flowchart illustrating a control processing procedure for realizing rotor position estimation according to the embodiment of the present invention. The control process shown in FIG. 4 is executed by the control device 30 while the AC motor M1 is stopped. It is assumed that the control process of each step in FIG. 4 is realized by a control calculation process by a predetermined program executed by the control device 30 and / or an electronic circuit (hardware) in the control device 30.

  Referring to FIG. 4, control device 30 obtains rotor position θc when AC motor M <b> 1 is stopped detected by rotational position sensor 25 in step S <b> 01.

  In step S02, the control device 30 sets a control axis (dc-qc axis) based on the output θc of the rotational position sensor 25 acquired in step S01. Specifically, the control device 30 changes the output θc of the rotational position sensor 25 by a predetermined angle θs for each execution cycle of steps S02 to S10 in FIG. Accordingly, the control axis is rotated by a predetermined angle θs for each execution cycle. The rotation angle range of the control shaft is set to have a width of ± θs × N (N is a natural number) with respect to the output θc of the rotation position sensor 25. As an example, the control device 30 changes the output θc of the rotational position sensor 25 so as to increase by a predetermined angle θs with (θc−θs × N) as an initial value.

  In step S03, control device 30 sets d-axis voltage command value Vdc and q-axis voltage command value Vqc so that a predetermined pulse voltage is applied to the dc axis of the control axis. As described above, the control device 30 sets the d-axis voltage command value Vdc corresponding to the pulse height of the predetermined pulse voltage to a predetermined value Vd other than 0, and the d-axis corresponding to the pulse width of the pulse voltage. The voltage application time Td is set. Control device 30 sets q-axis voltage command value Vqc = 0. Furthermore, control device 30 uses d-axis voltage command value Vdc and q-axis voltage command value Vqc to set a command value (hereinafter also referred to as “system voltage command value”) VH # of system voltage VH by a method described later. To do.

  FIG. 5 shows output waveforms of the d-axis voltage Vdc applied to the AC motor M1 and the q-axis current Iqc flowing through the AC motor M1.

  Referring to FIG. 5, when a constant voltage Vdc is applied to the dc axis, the q-axis current Iqc shows a different output waveform depending on the offset error Δθ of the rotational position sensor 25.

  Specifically, when the offset error Δθ = 0 of the rotational position sensor 25, the q-axis current Iqc = 0 when a constant voltage Vdc is applied to the dc axis (corresponding to the waveform k1 in the figure).

  On the other hand, the q-axis current Iqc when the offset error Δθ ≠ 0 of the rotational position sensor 25 is expressed by the above equation (7), and shows a transient response characteristic as shown by the waveform k2. That is, when the d-axis voltage Vdc is applied (time t1), the q-axis current Iqc increases with a delay of the inductances Ld and Lq with respect to the d-axis voltage, and after reaching a peak at time t3, gradually Decrease. Note that the q-axis current Iqc when the offset error Δθ ≠ 0 increases as the offset error Δθ increases.

  By rotating the control axis by a predetermined angle θs in step S02 of FIG. 4, the offset error Δθ changes by a predetermined angle θs. Then, as the offset error Δθ changes, the q-axis current Iqc changes when the d-axis voltage Vdc is applied. The rotor position estimation unit 270 obtains a control axis when the q-axis current Iqc is closest to 0 by detecting the change in the q-axis current Iqc.

  Specifically, returning to FIG. 4, in step S04, the d-axis voltage command values Vdc and q set in step S03 are converted by coordinate conversion using the rotor position θc from the rotational position sensor 25 (2 phase → 3 phase). The shaft voltage command value Vqc is converted into phase voltage command values Vu, Vv, Vw for the U phase, V phase, and W phase.

  In step S05, control device 30 generates switching control signals S3 to S8 based on comparison of voltage command values Vu, Vv, Vw and carrier signals in each phase. Control device 30 generates switching control signals S1 and S2 so that system voltage VH becomes system voltage command value VH #.

  In step S06, converter 12 is switching-controlled according to switching control signals S1 and S2, and inverter 14 is switching-controlled according to switching control signals S3 to S8, whereby a predetermined pulse voltage is applied to the dc axis of the control axis. The

  The current sensor 24 detects currents (V-phase current iv and W-phase current iw) that flow through the AC motor M1 when a predetermined pulse voltage is applied to the dc axis of the control axis in step S06. When controller 30 acquires the detected value of current sensor 24 in step S07, control device 30 based on V-phase current iv and W-phase current iw detected by current sensor 24 at the end of the d-axis voltage application time in step S08. Then, the d-axis current Idc and the q-axis current Iqc are calculated by coordinate conversion (3-phase → 2-phase) using the rotor position θc output from the rotational position sensor 25. The control device 30 holds the calculated q-axis current Iqc in the internal memory.

  Next, in step S09, the control device 30 determines whether or not the output θc of the rotational position sensor 25 has reached the final value (θc + θs × N) in a predetermined rotational angle range. When the output θc of the rotational position sensor 25 has not reached the final value (NO in step S09), the control device 30 proceeds to step S10 and increases the output θc of the rotational position sensor 25 by a predetermined angle θs. Then, the process returns to step S02.

  On the other hand, when the output θc of the rotational position sensor 25 reaches the final value (when YES is determined in step S09), the control device 30 performs a plurality of q-axes held in the internal memory in step S11. Among the currents Iqc, a q-axis current Iqc closest to 0 is selected, and a control axis corresponding to the selected q-axis current Iqc is obtained. Then, the control device 30 estimates the rotor position θc corresponding to this control axis as the actual rotor position θ.

  When the actual rotor position θ is estimated while AC electric motor M1 is stopped, control device 30 calculates offset error Δθ of rotational position sensor 25 using the estimated actual rotor position θ in step S12. The control device 30 holds the calculated offset error Δθ in the internal memory.

  When the offset error Δθ of the rotational position sensor 25 is calculated in this way, the control device 30 determines the rotor position θc detected by the rotational position sensor 25 using the offset error Δθ during the operation of the AC motor M1. to correct. Then, the corrected rotor position θc # is output to the coordinate converters 220 and 250 and the rotation speed calculator 230.

  FIG. 6 is a diagram for explaining the sine wave PWM control in the inverter 14 in step S07 of FIG.

  Referring to FIG. 6, the d-axis voltage command value Vdc and the q-axis voltage command value Vqc are converted into the U-phase, V-phase, and the like by the coordinate conversion using the rotor position θc from the rotational position sensor 25 (2 phase → 3 phase). It is converted into each phase voltage command value Vu, Vv, Vw of W phase. Then, switching control signals S3 to S8 are generated based on a comparison between the voltage command values Vu, Vv, Vw in each phase and a carrier signal CR composed of a triangular wave. For example, when the carrier signal CR is lower than the voltage command value Vu, the switching control signal S3 is turned on (and the switching control signal S4 is turned off). Otherwise, the switching control signal S4 is turned on (and the switching control signal S3). Off).

  The currents (V-phase current iv and W-phase current iw) that flow through AC motor M1 by the switching control of inverter 14 are converted by coordinate conversion (3-phase → 2-phase) using rotor position θc output from rotational position sensor 25. It is converted into a d-axis current Idc and a q-axis current Iqc. Then, the q-axis current Iqc at the end point (time t2) of the d-axis voltage application time Td is sampled.

  In this way, when the inverter 14 is subjected to switching control using the sine wave PWM control method, a voltage that matches the d-axis voltage command value Vdc and the q-axis voltage command value Vqc (= 0) is dc−. Applied to the qc axis. However, if the switching operation of the switching element varies due to variations in characteristics of the switching elements constituting the inverter 14, a voltage (hereinafter also referred to as “actual voltage”) that is actually applied to the dc-qc axis during switching control. It becomes a value deviated from the voltage command values Vdc and Vqc.

  FIG. 7 is a diagram for explaining the influence of variation in switching operation in the inverter 14.

  Referring to FIG. 7, a voltage deviation dVsw is generated on the dc-qc axis in a direction in which the phase is delayed with respect to the d-axis voltage command value Vdc due to variations in the switching operation. The actual voltage Vdr is delayed in phase from the d-axis voltage command value Vdc by the voltage deviation dVsw. Here, since the voltage deviation dVsw is a value unique to the inverter 14, as shown in FIG. 7, the phase shift of the actual voltage Vdr with respect to the dc-axis voltage command value Vdc increases as the d-axis voltage command value Vdc decreases. Become. In other words, by setting the d-axis voltage command value Vdc to a large value, it is possible to reduce the influence of variations in switching operations.

  On the other hand, when the d-axis voltage command value Vdc is increased, the currents (d-axis current Idc and q-axis current Iqc) flowing through the AC motor M1 are increased according to the above formulas (6) and (7). The AC motor M1 normally has a magnetic saturation characteristic in which the d and q axis inductances Ld and Lq are reduced due to magnetic saturation of the armature winding in a region where the current is large. Therefore, for example, when AC electric motor M1 has the magnetic saturation characteristics shown in FIG. 8, the salient pole ratio (Ld / Lq) decreases as the current increases. In this embodiment, since the rotor position is estimated using the saliency of the inductance, the estimation accuracy of the rotor position may be reduced in a region where the current is large. Therefore, it is necessary to limit the current flowing through AC electric motor M1 so that the salient pole ratio is maintained within a range suitable for estimation of the rotor position.

  Therefore, in the present embodiment, according to the magnetic saturation characteristic of AC motor M1 (FIG. 8), the current flowing through AC motor M1 is set to the maximum value within the current range in which the salient pole ratio of AC motor M1 can be maintained above the allowable value. The d-axis voltage command value Vdc is set so that The allowable value is set to a lower limit value within a range in which the estimation accuracy of the rotor position is guaranteed based on the magnetic saturation characteristics of AC electric motor M1. Thereby, the influence of the dispersion | variation in the switching operation of the inverter 14 can be reduced, without reducing the salient pole ratio of AC electric motor M1.

  Further, in the switching control of the inverter 14, in order to prevent the two switching elements from being turned on at the same time, a dead time DT in which both the switching elements are temporarily turned off with respect to the switching control signals S3 to S8. Is set. Also due to the influence of the dead time DT, the actual voltage deviates from the voltage command values Vdc and Vqc.

  FIG. 9 shows the relationship among the carrier signal CR, the voltage command value Vu, and the switching control signals S3 and S4. As described above, when the carrier signal CR is smaller than the voltage command value Vu, the switching control signal S3 is turned on (and the switching control signal S4 is turned off). Otherwise, the switching control signal S4 is turned on (and the switching control signal is turned on). S3 is off). Therefore, in the state where the dead time DT is not set, the state of the switching control signals S3 and S4 also changes when the magnitude relationship between the carrier signal CR and the voltage command value Vu changes (see the one-dot chain line in FIG. 9). .

  On the other hand, after setting the dead time DT, as shown by the solid line in FIG. 9, the magnitude relationship between the carrier signal CR and the voltage command value Vu changes when the switching control signals S3 and S4 change from on to off. Although the time remains unchanged, the time when the switching control signals S3 and S4 change from OFF to ON is the time when the dead time DT has elapsed from the time when the magnitude relationship between the carrier signal CR and the voltage command value Vu changes.

In the dead time DT, a state in which the U-phase current iu flows in the diode D3 and a state in which the U-phase current iu flows in the diode D4 occur depending on the polarity of the U-phase current iu. Specifically, when the U-phase current iu is positive, the current iu flows through the diode D1, while when the U-phase current iu is negative, the current iu flows through the diode D2. Thus, the U-phase current iu flows through the diodes D3 and D4 not only during the period when the switching control signals S3 and S4 are ON but also during the dead time period DT. If the voltage deviation caused by the current flowing through the diodes D1 and D2 during the dead time period DT is dV _DT , the voltage deviation dV _DT can be expressed by Expression (10).

  In Expression (10), Tc represents the period of the carrier signal CR (carrier period), and VH represents the system voltage (input voltage to the inverter 14).

Due to this voltage deviation dV _DT , as shown in FIG. 10, the actual voltage Vdr is in a phase shifted from the d-axis voltage command value Vdc. Here, as is clear from the equation (10), the voltage deviation dV _DT increases as the system voltage VH increases. Moreover, it becomes larger as the carrier period Tc becomes shorter (that is, the carrier frequency becomes higher). Therefore, the voltage deviation dV _DT can be reduced by reducing the system voltage VH or reducing the frequency (carrier frequency) of the carrier signal CR.

  Therefore, in the present embodiment, system voltage VH is set to the minimum value within the range of system voltage VH to which sine wave PWM control can be applied.

  Specifically, whether to apply the sine wave PWM control method is determined according to the modulation rate range of the voltage command value according to the vector control. The “modulation rate” is represented by the ratio of the fundamental component (effective value) of the motor applied voltage (line voltage) to the input voltage to the inverter 14 (that is, the system voltage VH). When the modulation factor is equal to or lower than a predetermined upper limit UL, sine wave PWM control is applied. When the modulation rate exceeds the upper limit value UL, overmodulation PWM control is applied.

  In the present specification, this modulation factor can be expressed by the following equation (11) using the d-axis voltage command value Vdc and the q-axis voltage command value Vqc.

  In the sine wave PWM control, typically, the amplitude of the voltage command value is set to be in a range equal to or smaller than the amplitude of the carrier signal CR, and in this case, the upper limit value UL is set to about 0.61. In the above equation (11), the q-axis voltage command value Vqc = 0 is set, so that the modulation rate = Vdc / VH. In the present embodiment, system voltage command value VH # is set (that is, VH # = Vdc / UL) so that this modulation factor (= Vdc / VH) becomes upper limit value UL (about 0.61). In converter 12, feedback control of output voltage VH is performed so that output voltage VH matches system voltage command value VH #.

FIG. 11 shows the relationship between the carrier signal CR and the U-phase voltage command value Vu when the modulation rate is smaller than the upper limit value UL (see the solid line in the figure), the carrier signal CR when the modulation rate becomes the upper limit value UL, and The relationship of U-phase voltage command value Vu is shown (refer the dashed-dotted line in FIG. 11). Since the voltage deviation dV _DT is reduced by setting the system voltage VH to be the minimum value within the range of the system voltage VH to which the sine wave PWM control can be applied, the deviation of the actual voltage Vdr from the voltage command values Vdc and Vqc. Can be reduced. Furthermore, this shift can be further reduced by setting the carrier frequency low.

  As described above, in the rotor position estimation device according to the embodiment of the present invention, the d-axis voltage command value Vdc is a current that allows the current flowing through AC motor M1 to maintain the salient pole ratio of AC motor M1 at or above the allowable value. The system voltage command value is set so as to be the maximum value within the range, and the system voltage command value is set to the minimum value within the range of the system voltage VH to which the sine wave PWM control can be applied according to the set d-axis voltage command value Vdc. VH # is set. Thereby, since the shift of the switching operation in the inverter 14 and the influence of the dead time can be reduced, the shift between the voltage command values Vdc, Vqc and the actual voltage Vdr can be reduced. As a result, the estimation accuracy of the rotor position can be improved.

  Here, when system voltage command value VH # is set, converter 12 performs voltage feedback control for matching detected value of system voltage VH by voltage sensor 13 with system voltage command value VH #.

  In such a voltage feedback control, if a deviation occurs between the detected value of the system voltage VH and the actual system voltage VH due to a detection error of the voltage sensor 13, in the worst case, the actual system voltage VH is changed to the system voltage command. There is a possibility that the sine wave PWM control in the inverter 14 cannot be executed because the value becomes lower than the value VH #.

  As a countermeasure, as shown in FIG. 12, the detection error of the voltage sensor 13 is added to the system voltage command value VH # set by the method described above. Specifically, assuming that the minimum value within the range of system voltage VH to which sine wave PWM control can be applied is “minimum voltage”, a value obtained by adding the detection error of voltage sensor 13 to this minimum voltage is set as system voltage command value VH #. Set to.

  Thereby, when voltage feedback control is performed so that the detected value of voltage sensor 13 matches system voltage command value VH #, it is possible to avoid a situation in which actual system voltage VH falls below the minimum voltage.

  In the above embodiment, the motor control system to which the rotor position estimation device according to the present invention is applied is an electric motor configured to perform energization control of the AC motor based on the rotor position detected by the rotation position sensor. The control system has been described. However, the application of the present invention is not limited to such an electric motor control system. Specifically, the present invention can also be applied to an electric motor control system that employs a so-called position sensorless control system that performs energization control by estimating a rotor position without using a rotational position sensor.

  The embodiment disclosed this time should be considered as illustrative in all points and not restrictive. The scope of the present invention is defined by the terms of the claims, rather than the description above, and is intended to include any modifications within the scope and meaning equivalent to the terms of the claims.

  10 # DC voltage generation unit, 12 converter, 13 voltage sensor, 14 inverter, 24 current sensor, 25 rotational position sensor, 30 control device, 100 motor control system, 210 current command generation unit, 220, 250 coordinate conversion unit, 230 rotation Number calculation unit, 240 PI calculation unit, 260 PWM signal generation unit, 270 rotor position estimation unit, M1 AC motor.

Claims (11)

A rotor position estimation device for estimating a rotor position of an electric motor including a rotor having saliency,
Voltage application means for applying the d-axis voltage while the motor is stopped;
Current detection means for detecting a q-axis current flowing in the electric motor when the d-axis voltage is applied;
Estimating means for estimating the rotor position when the electric motor is stopped based on the q-axis current detected by the current detecting means;
The voltage applying means includes
Setting means for setting a command value of the d-axis voltage applied to the electric motor;
Voltage conversion means for converting a DC voltage into an AC voltage and applying it to the motor by performing pulse width modulation control of the inverter according to the d-axis voltage command value;
DC voltage generating means for variably controlling the input voltage to the inverter according to the d-axis voltage command value,
In accordance with the magnetic saturation characteristics of the motor, the setting means is configured so that the current flowing through the motor becomes a maximum value within a current range in which the salient pole ratio of the motor can be maintained at an allowable value or more. Set the command value,
The DC voltage generating means controls the input voltage so as to be a minimum value within the range of the input voltage to which the pulse width modulation control can be applied, according to the d-axis voltage command value. The DC voltage generating means includes
Command value setting means for setting a command value of the input voltage;
Voltage detecting means for detecting the input voltage;
Feedback control means for feedback-controlling the input voltage so that the detected value of the input voltage matches the input voltage command value;
The rotor position estimating device according to claim 1, wherein the command value setting means sets a minimum value within the range of the input voltage to which the pulse width modulation control can be applied, to the input voltage command value.   The command value setting means is configured such that a modulation rate of voltage conversion by the inverter calculated based on the d-axis voltage command value and an input voltage to the inverter is an upper limit value of a modulation rate to which the pulse width modulation control can be applied. The rotor position estimation device according to claim 2, wherein the input voltage command value is set so as to be.   The feedback control means is configured so that the detected value of the input voltage coincides with a value obtained by adding a detection error in the voltage detecting means to the input voltage command value set by the command value setting means. The rotor position estimation device according to claim 2 or 3, wherein the voltage is feedback-controlled. The voltage application means applies the d-axis voltage for each rotation angle while rotating the dq axes set as a rotation coordinate system of the electric motor by a predetermined angle,
The estimation means estimates the rotor position corresponding to the dq axis when the detected value of the q-axis current is closest to zero as the rotor position when the electric motor is stopped. The rotor position estimation apparatus of any one of Claims. An electric motor control system for controlling an electric motor including a rotor having saliency,
Rotational position detecting means for detecting a rotor position of the electric motor;
Rotor position estimating means for estimating a rotor position of the electric motor when the electric motor is stopped;
An error detecting means for detecting an error in a detected value of the rotor position by the rotational position detecting means with respect to an estimated value of the rotor position by the rotor position estimating means;
Correction means for correcting the rotor position detected by the rotational position detection means using the error detected by the error detection means;
Control means for performing energization control of the electric motor based on the rotor position corrected by the correction means,
The rotor position estimating means includes
Voltage application means for applying the d-axis voltage while the motor is stopped;
Current detection means for detecting a q-axis current flowing in the electric motor when the d-axis voltage is applied;
Estimating means for estimating the rotor position when the electric motor is stopped based on the q-axis current detected by the current detecting means,
The voltage applying means includes
Setting means for setting a command value of the d-axis voltage applied to the electric motor;
Voltage conversion means for converting a DC voltage into an AC voltage and applying it to the motor by performing pulse width modulation control of the inverter according to the d-axis voltage command value;
DC voltage generating means for variably controlling the input voltage to the inverter according to the d-axis voltage command value,
In accordance with the magnetic saturation characteristics of the motor, the setting means is configured so that the current flowing through the motor becomes a maximum value within a current range in which the salient pole ratio of the motor can be maintained at an allowable value or more. Set the command value,
The motor control system, wherein the DC voltage generation means controls the input voltage so as to be a minimum value within the range of the input voltage to which the pulse width modulation control can be applied, according to the d-axis voltage command value. A rotor position estimation method for estimating a rotor position of an electric motor including a rotor having saliency,
Applying the d-axis voltage while the motor is stopped;
Detecting a q-axis current flowing in the motor when the d-axis voltage is applied;
Estimating the rotor position while the electric motor is stopped based on the q-axis current detected by the step of detecting the q-axis current,
The step of applying the d-axis voltage includes:
Setting a command value for the d-axis voltage to be applied to the motor;
Applying a pulse width modulation control to the inverter according to the d-axis voltage command value, thereby converting a DC voltage into an AC voltage and applying it to the motor;
Variably controlling the input voltage to the inverter according to the d-axis voltage command value,
The step of setting the command value of the d-axis voltage includes a maximum value within a current range in which a current flowing through the motor can maintain a salient pole ratio of the motor at an allowable value or more according to a magnetic saturation characteristic of the motor. The d-axis voltage command value is set so that
The step of variably controlling the input voltage to the inverter controls the input voltage to be a minimum value within the range of the input voltage to which the pulse width modulation control can be applied according to the d-axis voltage command value. The rotor position estimation method. The step of variably controlling the input voltage to the inverter includes:
Setting a command value of the input voltage;
Detecting the input voltage;
Feedback control of the input voltage so that the detected value of the input voltage matches the input voltage command value,
The rotor position estimating method according to claim 7, wherein the step of setting the command value of the input voltage sets the minimum value within the range of the input voltage to which the pulse width modulation control can be applied as the input voltage command value. .   The step of setting the command value of the input voltage is a modulation in which the modulation rate of the voltage conversion by the inverter calculated based on the d-axis voltage command value and the input voltage to the inverter can apply the pulse width modulation control. The rotor position estimation method according to claim 8, wherein the input voltage command value is set so as to be an upper limit value of the rate.   In the feedback control step, the detection value of the input voltage is added to a detection error in the step of detecting the input voltage with respect to the input voltage command value set by the step of setting the command value of the input voltage. The rotor position estimation method according to claim 8 or 9, wherein the input voltage is feedback-controlled so as to coincide with a value. The step of applying the d-axis voltage includes applying the d-axis voltage for each rotation angle while rotating the dq axes set as a rotation coordinate system of the electric motor by a predetermined angle,
The step of estimating the rotor position estimates the rotor position corresponding to the dq axis when the detected value of the q-axis current is closest to zero as the rotor position when the electric motor is stopped. The rotor position estimation method according to any one of 7 to 10.

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