JP5910296B2 - Induction machine control device - Google Patents

Description

  The present invention relates to a control device for an induction machine that operates AC voltage application means for applying an AC voltage to an induction machine in order to control a control amount of the induction machine based on an estimated secondary magnetic flux.

  For example, Patent Document 1 below proposes a method for calculating an offset amount that is an error of the secondary magnetic flux from the positive and negative peak values of the estimated value of the secondary magnetic flux and reducing the offset amount.

JP 2011-250630 A

  However, in order to detect the peak value, it is necessary to make the estimation period of the secondary magnetic flux extremely short, and the estimation period required for calculating the peak value with high accuracy particularly in the high rotation speed region of the induction machine. It is difficult to satisfy. In addition, when the torque or rotational speed of the induction machine varies, the estimated magnetic flux changes abruptly, and the accuracy of the offset amount calculated based on the peak value may be reduced.

  The present invention has been made in the process of solving the above-mentioned problems, and its object is to control the induction machine by operating an AC voltage application means for applying an AC voltage to the induction machine based on the estimated secondary magnetic flux. The object is to provide a new induction machine control device for controlling the quantity.

  Hereinafter, means for solving the above-described problems and the operation and effect thereof will be described.

According to a first aspect of the present invention, there is provided a secondary magnetic flux calculation means (28) for calculating a secondary magnetic flux component of the induction machine in a fixed coordinate system, and the induction machine based on the secondary magnetic flux calculated by the secondary magnetic flux calculation means. In order to control the control amount, the operating means (20) for operating the AC voltage applying means for applying an AC voltage to the induction machine, and the integration processing of the components of the fixed coordinate system calculated by the secondary magnetic flux calculating means And a correction means (S16) for correcting the calculation parameters used to generate an operation signal output from the operation means to the AC voltage application means based on an integration processing result by the integration processing means. S18, S32).

  The direct current component of the secondary magnetic flux in the fixed coordinate system is reflected in the integration processing result. For this reason, the DC component information of the secondary magnetic flux can be acquired from the integration processing result regardless of whether or not the peak value of the secondary magnetic flux can be sampled. In view of this point, the above invention includes an integration processing means.

  In addition, about the expansion of the concept regarding the following typical embodiment concerning this invention, it describes in the column of "other embodiment" after typical embodiment.

1 is a system configuration diagram according to a first embodiment. FIG. The flowchart which shows the procedure of the correction process of the secondary magnetic flux concerning the embodiment. The time chart which shows the effect of the embodiment. The time chart which shows the effect of the embodiment. The flowchart which shows the procedure of the correction | amendment process of the secondary magnetic flux concerning 2nd Embodiment. The flowchart which shows the procedure of the correction process of the secondary magnetic flux concerning 3rd Embodiment.

<First Embodiment>
Hereinafter, a first embodiment in which an induction machine control device according to the present invention is applied to an induction machine control device as an in-vehicle main machine will be described with reference to the drawings.

  The motor generator 10 shown in FIG. 1 is an induction machine. Specifically, it is a three-phase cage induction machine. The motor generator 10 is an in-vehicle main machine and is mechanically coupled to drive wheels (not shown).

  The motor generator 10 is connected to the high voltage battery 12 via the inverter INV. The inverter INV includes three sets of series connection bodies of switching elements S ¥ p, S ¥ n (¥ = u, v, w), and the connection points of these series connection bodies are U, V, Each is connected to the W phase. As these switching elements S ¥ # (¥ = u, v, w; # = p, n), an insulated gate bipolar transistor (IGBT) is used in the present embodiment. In addition, a diode D ¥ # is connected in antiparallel to each of these.

  In the present embodiment, the following is provided as detection means for detecting the state of the motor generator 10 and the inverter INV. First, a current sensor 14 for detecting currents iv and iw flowing through the V phase and the W phase of the motor generator 10 is provided. Moreover, the voltage sensor 16 which detects the input voltage (power supply voltage VDC) of the inverter INV is provided.

  The detection values of the various sensors are taken into the control device 20. The control device 20 generates and outputs an operation signal for operating the inverter INV based on the detection values of these various sensors. Here, the signal for operating the switching element S ¥ # of the inverter INV is the operation signal g ¥ #.

  The control device 20 operates the inverter INV so as to control the torque of the motor generator 10 to the torque command value Trq *. In the figure, processing relating to control to the torque command value Trq * is shown as a block diagram. This will be described below.

  The currents iv and iw detected by the current sensor 14 are converted into primary currents id1 and iq1, which are currents on the dq axis, in the dq converter 22. Here, the dq coordinate system is an orthogonal two-dimensional fixed coordinate system. Specifically, in the present embodiment, the d-axis is the positive direction of the U-phase of the stator of the motor generator 10, and the q-axis is the direction orthogonal to the direction.

  The primary currents id1 and iq1 are input to the γδ converter 24 and converted into primary currents iγ1 and iδ1 on the γδ axis. The γδ coordinate system is an orthogonal two-dimensional rotational coordinate system that rotates at a power supply angular frequency (hereinafter, primary frequency ω1) that is a rotational frequency of the output voltage vector of the inverter INV. Incidentally, the conversion process by the γδ conversion unit 24 is performed based on the power supply angle (hereinafter referred to as the primary phase θ1) which is the phase of the fundamental component of the output voltage vector of the inverter INV calculated by the process described later.

  In the present embodiment, the control system for controlling the torque command value Trq * is configured as a control system that performs feedback control of the primary currents iγ1 and iδ1 to the primary current command values iγ1 * and iδ1 *. At this time, the primary current command values iγ1 * and iδ1 * cannot be uniquely determined only by giving the torque command value Trq *. Therefore, in this embodiment, the secondary magnetic flux φδ2 of the δ axis is controlled to zero, and the primary current command value iγ1 * is based on the torque command value Trq * and the secondary magnetic flux command value φγ2 * of the γ-axis component. , Iδ1 * is generated.

  That is, the torque feedback control unit 26 calculates the primary current command value iδ1 * of the δ-axis component as an operation amount for feeding back the torque Trq estimated by the processing described later to the torque command value Trq *. More specifically, the primary current command value iδ1 * is calculated as the sum of the proportional element and the integral element of the difference between the torque command value Trq * and the torque Trq.

  On the other hand, the magnetic flux feedback control unit 28 uses the γ-axis component primary current command as an operation amount for feedback-controlling the secondary magnetic flux φγ2 of the γ-axis component estimated by the processing described later to the secondary magnetic flux command value φγ2 *. The value iγ1 * is calculated. Specifically, the primary current command value iγ1 * is calculated as the sum of the proportional elements of the difference between the secondary magnetic flux command value φγ2 * and the secondary magnetic flux φγ2 and the outputs of the integral elements.

  The δ-axis current feedback control unit 30 calculates a δ-axis primary voltage component as an operation amount for performing feedback control of the primary current iδ1 to the primary current command value iδ1 *. Specifically, the primary voltage component on the δ axis is calculated as the sum of the proportional elements and the integral elements of the difference between the primary current command value iδ1 * and the primary current iδ1.

  Further, the γ-axis current feedback control unit 32 calculates a γ-axis primary voltage component as an operation amount for performing feedback control of the primary current iγ1 to the primary current command value iγ1 *. Specifically, the primary voltage component of the γ-axis is calculated as the sum of the proportional element and the integral element of the difference between the primary current command value iγ1 * and the primary current iγ1.

  By adding the feedforward manipulated variable Δvγ1 output from the non-interference control unit 34 to the output value of the γ-axis current feedback control unit 32, the primary voltage command value vγ1 * of the γ-axis is calculated. On the other hand, the δ-axis primary voltage command value vδ1 * is calculated by adding the feedforward manipulated variable Δvδ1 output from the non-interference control unit 34 to the output value of the δ-axis current feedback control unit 30. Note that the feedforward manipulated variables Δvγ1 and Δvδ1 are well known and therefore will not be described here. The definitions of symbols used to express the feedforward manipulated variables Δvγ1 and Δvδ1 in the figure are as follows. It is written in the remarks column at the end of the book.

  The primary voltage command values vγ1 * and vδ1 * are converted into primary voltage command values vd * 1 and vq1 * on the dq axis by the dq converter 36. The primary voltage command values vd * 1 and vq1 * are converted by the uvw converter 38 into a U-phase voltage command value vu *, a V-phase voltage command value vv *, and a W-phase voltage command value vw *. Are input to the operation signal generator 40.

  The operation signal generation unit 40 performs a process of generating an operation signal g ¥ # for the inverter INV so that the output voltage of the inverter INV is a simulation of the voltage command values vu *, vv *, and vw *. In the present embodiment, in particular, based on a PWM signal generated based on a magnitude comparison between a signal in which the voltage command value v ¥ * (¥ = u, v, w) is normalized by the power supply voltage VDC and a triangular wave carrier, An operation signal g ¥ # is generated.

  Next, calculation (estimation) processing of parameters necessary for performing the above-described processing will be described.

  The primary currents iγ1 and iδ1 are input to the slip frequency estimation unit 42. The slip frequency estimating unit 42 receives the primary currents iγ1 and iδ1 and estimates the slip frequency ωs based on the following equation (c1).

ωs = iδ1 / (iγ1 · τ2) (c1)
The above equation (c1) is premised on the setting specific to the present embodiment. That is, since the secondary magnetic flux φδ2 of the δ-axis component is controlled to zero and the secondary magnetic flux command value φγ2 * of the γ-axis component is constant when the torque command value Trq * is constant, the torque command When the value Trq * is constant, the γ-axis primary current iγ1 is assumed to be constant and is calculated.

  On the other hand, the primary frequency calculation unit 44 calculates the primary frequency ω1 by differential calculation of the primary phase θ1 described later. The secondary frequency calculation unit 46 calculates the secondary frequency ω2 by subtracting the slip frequency ωs from the primary frequency ω1.

  In the secondary magnetic flux calculating means (magnetic flux observer 48), the primary currents id1, iq1, the primary voltage command values vd1 *, vq1 *, and the secondary frequency ω2 are input and the secondary magnetic fluxes φd2e, φq2e are estimated. . Here, the subscript “e” indicates an estimated value by the observer. The magnetic flux observer 48 is a minimum dimension observer. The derivation of the minimum dimension observer according to this embodiment is given in the remarks column at the end of this specification.

  On the other hand, the primary phase calculation unit 52 calculates the primary phase θ1 using the following equation (c2) based on the secondary magnetic fluxes φd2e and φq2e estimated by the magnetic flux observer 48.

θ1 = arctan (φq2e / φd2e) (c2)
The primary phase θ1 calculated in this way is used for coordinate conversion processing by the γδ conversion unit 24, the dq conversion unit 36, etc., and for calculation processing of the primary frequency ω1a by the primary frequency calculation unit 44.

  The torque estimation unit 56 receives the secondary magnetic fluxes φd2e and φq2e corrected by the magnetic flux correction unit 50 and the primary currents id1 and iq1, and estimates the torque Trq by the following equation (c3).

Trq = Pn · M · (iq1 · φd2e−id1 · φq2e) / L2 (c3)
In the above formula (c3), the number of pole pairs Pn, the secondary inductance L2, and the mutual inductance M are used. This torque Trq is a control amount for torque feedback control.

  The secondary magnetic fluxes φd2e and φq2e corrected by the magnetic flux correcting unit 50 are further converted into secondary magnetic fluxes φγ2 and φδ2 on the γδ axis by the γδ converter 54. Here, the secondary magnetic flux φγ2 of the γ-axis component serves as a control amount for magnetic flux feedback control and also serves as an input to the non-interference control unit 34.

  As can be seen from the above-described processing, in the present embodiment, sensorless control is performed that does not use a detection value by hardware means that detects rotation speed information of the motor generator 10. In this case, in particular, an offset error that causes the direct current components of the secondary magnetic fluxes φd2e and φq2e to deviate from zero is likely to occur. Therefore, in the present embodiment, the secondary magnetic fluxes φd2e and φq2e are corrected by the process shown in FIG.

  FIG. 2 shows a procedure of processing performed by the magnetic flux correction unit 50. This process is repeatedly executed, for example, at a predetermined cycle.

  In this series of processes, first, in step S10, the magnetic flux observer 48 estimates the secondary magnetic fluxes φd2e and φq2e. In the subsequent step S12, the primary phase calculation process by the primary phase calculation unit 52 described above is performed. In step S14, it is determined whether N cycles have elapsed based on the primary phase θ1 calculated in step S12. If a negative determination is made in step S14, each of the current secondary magnetic fluxes φd2e and φq2e is added to each of the magnetic flux integral values Inφd2 and Inφq2 in step S16. This processing constitutes an integration processing means in the present embodiment.

  On the other hand, when an affirmative determination is made in step S14, offset correction amounts Δφd and Δφq of the secondary magnetic fluxes φd2e and φq2e are calculated in step S18. The offset correction amounts Δφd and Δφq are operation amounts for feedback-controlling each of the magnetic flux integral values Inφd2 and Inφq2 to the target value of 0. In this embodiment, the offset correction amounts Δφd and Δφq are calculated as the sum of the outputs of the proportional element and the integral element for each of the values obtained by subtracting each of the magnetic flux integral values Inφd2 and Inφq2 from 0. After calculating the offset correction amounts Δφd and Δφq, the magnetic flux integral values Inφd2 and Inφq2 are initialized.

  When the processes of steps S16 and S18 are completed, in step S20, the current secondary magnetic fluxes φd2e and φq2e are corrected by adding the offset correction amounts Δφd and Δφq to the current secondary magnetic fluxes φd2e and φq2e. This process constitutes secondary magnetic flux correction means in the present embodiment.

  In addition, when the process of step S20 is completed, this series of processes is once complete | finished.

  FIG. 3 shows the effect of this embodiment.

  As shown in the figure, by starting the correction using the offset correction amounts Δφd and Δφq, the fluctuations in the estimated torque Trq and the primary frequency ω1 are reduced. Note that fluctuations in the torque Trq and the primary frequency ω1 due to the offset errors of the secondary magnetic fluxes φd2e and φq2e increase because the DC component in the fixed coordinate system is the AC component (pulsation component) in the rotating coordinate system. It is to become.

  FIG. 4 shows another effect of the present embodiment.

  As shown in the drawing, by starting the correction using the offset correction amounts Δφd and Δφq, the fluctuation amounts of the primary currents iγ1 and iδ1 are reduced. As the absolute values of the offset correction amounts Δφd and Δφq increase, the absolute values of the magnetic flux integral values Inφd2 and Inφq2 converge to zero.

  Hereinafter, some of the effects of this embodiment will be described.

  (1) The offset errors of the secondary magnetic fluxes φd2e and φq2e were calculated from the magnetic flux integral values Inφd2 and Inφq2. In this case, when sampling the secondary magnetic fluxes φd2e and φq2e of alternating current components that periodically change with the primary frequency ω1, the sampling phase hardly affects the calculation accuracy of the offset error. Therefore, the offset error can be calculated with high accuracy.

  (2) The offset error was calculated as the value of the integration process of the secondary magnetic fluxes φd2e and φq2e over a period N times the cycle of the primary frequency. Thereby, the calculation accuracy of the offset error can be further improved.

  (3) The offset correction amounts Δφd and Δφq are set as operation amounts for feedback control of the magnetic flux integral values Inφd2 and Inφq2 to zero. Thereby, the offset error can be reduced and controlled by the feedback controller.

  (4) The magnetic flux observer 48 is used as means for calculating the secondary magnetic fluxes φd2e and φq2e in the fixed coordinate system. Thereby, in the sensorless control system for estimating the secondary frequency ω2, the secondary magnetic fluxes φd2e and φq2e are used without using the primary phase θ1 estimated from the secondary frequency ω2 for estimating the secondary magnetic fluxes φd2e and φq2e. Can be estimated.

(5) A sensorless control system for estimating the secondary frequency ω2 is used. In this case, since an offset error is likely to occur in the secondary magnetic fluxes φd2e and φq2e in the fixed coordinate system, the utility value of the offset correction amounts Δφd and Δφq is particularly high.
<Second Embodiment>
Hereinafter, the second embodiment will be described with reference to the drawings with a focus on differences from the first embodiment.

  In this embodiment, the operation amount for reducing and controlling the offset error of the secondary magnetic fluxes φd2e and φq2e is changed to the estimated torque Trq.

  FIG. 5 shows a procedure of offset error reduction processing according to the present embodiment. This process is repeatedly executed by the control device 20, for example, at a predetermined control cycle. In the processing shown in FIG. 5, the same step numbers are assigned for convenience to those corresponding to the processing shown in FIG.

  In this series of processes, when an affirmative determination is made in step S14, the process proceeds to step S30. In step S30, the magnetic flux integral values Inφd2 and Inφq2 for N cycles are added to the magnetic flux integral values Inφd20 and Inφq20, and the magnetic flux integral values Inφd2 and Inφq2 are initialized. Here, the process of adding to the magnetic flux integral values Inφd20 and Inφq20 has a function of using the feedback controller as an integral controller in cooperation with the processes of steps S32 to S36 described later. The initial values of the magnetic flux integral values Inφd20 and Inφq20 are zero.

  On the other hand, when the processes in steps S16 and S30 are completed, in step S32, the magnetic flux integral values Inφd20 and Inφq20 determined in step S30 are converted into the magnetic flux integral values Inφγ2 and Inφδ2 in the rotating coordinate system. This process constitutes the rotating coordinate conversion means in the present embodiment.

  In the subsequent step S34, a torque correction amount ΔTrq that is an operation amount for performing feedback control of the magnetic flux integral value Inφγ2 on the γ-axis to zero is calculated. Here, the pulsation of the magnetic flux integral value Inφγ2 is an amount having a correlation with the offset error. This corresponds to the fact that the DC component in the fixed coordinate system becomes the AC component in the rotating coordinate system. Therefore, by performing feedback control of the magnetic flux integral value Inφγ2 to zero, it is possible to perform control for reducing the offset error of the secondary magnetic fluxes φd2e and φq2e. Specifically, in this embodiment, the torque correction amount ΔTrq is the sum of the proportional element and the output of the integral element with the magnetic flux integral value Inφγ2 as an input.

  In step S34, the torque Trq estimated by the torque estimation unit 56 is corrected by the torque correction amount ΔTrq. The corrected torque Trq becomes the feedback control amount in the torque feedback control unit 26. This process constitutes pulsation reducing means in the present embodiment.

In addition, when the process of step S32 is completed, this series of processes is once complete | finished.
<Third Embodiment>
Hereinafter, the third embodiment will be described with reference to the drawings with a focus on differences from the second embodiment.

  In this embodiment, the operation amount for reducing and controlling the offset error of the secondary magnetic fluxes φd2e and φq2e is changed to the estimated primary frequency ω1.

  FIG. 6 shows a procedure of offset error reduction processing according to this embodiment. This process is repeatedly executed by the control device 20, for example, at a predetermined control cycle. In the process shown in FIG. 6, the same step number is attached for convenience to the process corresponding to the process shown in FIG. 5.

In this series of processing, after step S32, in step S34a, a frequency correction amount Δω1 that is an operation amount for performing feedback control of the magnetic flux integral value Inφγ2 on the γ-axis to zero is calculated. Specifically, in the present embodiment, the frequency correction amount Δω1 is the sum of the proportional element and the output of the integral element having the magnetic flux integral value Inφγ2 as an input. In step S36a, the primary frequency ω1 calculated by the primary frequency calculation unit 44 is corrected by the frequency correction amount Δω1. A value obtained by subtracting the slip frequency ωs from the corrected primary frequency ω1 is input to the magnetic flux observer 48 as the secondary frequency ω2, and the corrected primary frequency ω1 is input to the non-interference control unit 34. It becomes.
<Other embodiments>
Each of the above embodiments may be modified as follows.

"Secondary magnetic flux calculation means"
The observer is not limited to the minimum-dimensional observer, but is exemplified in, for example, “Speed sensorless direct vector control of induction motor using adaptive secondary magnetic flux observer: Kubota et al., T. IEEE Japan Vol 111-D, No11, 91”. As shown, it may be the same dimensional observer.

  However, it is not limited to an observer.

“Secondary magnetic flux correction means (S20)”
The operation amount for performing feedback control of the magnetic flux integral values Inφd2 and Inφq2 to the target value (= 0) is not limited to being calculated as the sum of the outputs of the proportional element and the integral element. For example, it may be calculated as the sum of outputs of a proportional element, an integral element, and a derivative element.

  However, it is not limited to performing feedback control each time, and for example, the convergence value of the output of the integration element may be stored and used as a feedforward correction amount. Here, it is desirable to store the convergence value in a separate storage area for each area partitioned according to the primary current, the primary frequency ω1, and the like.

"Integration processing means (S16)"
It is not limited to calculating an integral value over a period of an integral multiple of the period of the primary phase θ1. For example, if the correction means calculates the average value by dividing the integral value by the integration period and uses this as the offset amount, it is necessary to lengthen the integration period to make it an integral multiple of the period of the primary phase θ1. In addition, it is possible to increase the calculation accuracy of the offset amount.

"About pulsation reduction means (S36, S36a)"
Instead of calculating the feedback operation amount in the process of step S34 of FIG. 5 and the process of step S34a of FIG. 6 as the sum of the outputs of the proportional element and the integral element, for example, the output value of the proportional element is used. It may be a thing. Further, the processing of steps S34 and S34a is not essential. Even in this case, the feedback controller is configured by the cooperation of the processes of steps S30, S32, and S36 (S36a).

  The feedback controller configured by the cooperation of the processes of steps S30, S32, and S36 (S36a) is not limited to the integral controller, and the sum of the outputs of the proportional controller and the integral controller is output as a feedback manipulated variable. You may do.

  In the second and third embodiments (FIGS. 5 and 6), each time affirmative determination is made in step S14, a process of adding the magnetic flux integral values Inφd2 and Inφq2 to the magnetic flux integral values Inφd20 and Inφq2, respectively, is performed. It does not have to be. In this case, the magnetic flux integral values Inφd2 and Inφq2 at the time when an affirmative determination is made in step S14 may be continuously used as the rotation coordinate conversion target in the subsequent processing of step S32. However, the present invention is not limited to this, and each time an affirmative determination is made in step S14, the feedback controller can be a proportional controller even if the latest magnetic flux integral values Inφd2 and Inφq2 are subject to rotational coordinate conversion.

  In the second embodiment (FIG. 5), the estimated torque Trq is set as an operation target (correction target) for feedback control of the pulsation component of the magnetic flux integral value Inφγ2 to zero, but is not limited thereto. For example, the torque command value Trq * may be the operation target. In short, the difference between the torque Trq serving as an input parameter of the torque feedback control unit 26 and the torque command value Trq * may be an operation target, and the difference itself may be corrected.

  The torque Trq and the primary frequency ω are not limited to be corrected by an operation amount for feedback-controlling the pulsation component of the magnetic flux integral value Inφγ2 of the γ-axis component to zero. The torque Trq and the primary frequency ω1 may be corrected by an operation amount for performing feedback control of the pulsation of the magnetic flux integral value Inφδ2 of the δ-axis component to zero.

  Further, the correction target (operation target) is not limited to the torque Trq or the primary frequency ω1. For example, the phase θ may be used.

"Rotating coordinate conversion means"
It is not limited to means for γδ conversion. For example, it may be one that rotates at a primary frequency and converts to a biaxial component that is in a primary independent relationship but not orthogonal.

"About operation means"
The control amount is not limited to torque, but may be rotation speed.

  The present invention is not limited to a sensorless control that includes secondary frequency estimation means for estimating the secondary frequency ω2 based on the secondary magnetic fluxes φd2e and φq2e. For example, even if a sensor for detecting the rotational speed of the rotor is provided and the detected value is used, the application of the present invention is effective in improving the estimation accuracy of the secondary magnetic fluxes φd2e and φq2e.

About AC voltage application means
The present invention is not limited to a DC / AC conversion circuit (inverter INV) including a switching element that connects a terminal of a rotating machine to each of a positive electrode and a negative electrode of a DC voltage source. For example, as described in Japanese Patent Application No. 2008-30825, a converter connected to each terminal of the rotating machine may be used.
<Remarks>
The state equation of the cage induction machine is expressed by the following equations (ca) and (cb).

ただし、１次インダクタンスＬ１，２次インダクタンスＬ２、１次抵抗Ｒ１，２次抵抗Ｒ２，相互インダクタンスＭを用いている。

However, primary inductance L1, secondary inductance L2, primary resistance R1, secondary resistance R2, and mutual inductance M are used.

  Here, the primary current vector I1 = (id1, iq1) is an amount that can be directly observed. If the estimated value of the secondary magnetic flux vector Φ2 is the secondary magnetic flux vector Φ2e, the following equation (cc) is established.

ただし、電流の微分演算を回避する上では中間変数ｘ「＝Φ２−ＧＩ１」を導入することが望ましい。

However, it is desirable to introduce an intermediate variable x “= Φ2−GI1” in order to avoid current differentiation.

  The observer gain G is determined by the following equations (cd) and (ce).

ただし、上記においてオブザーバの極の実部−αおよび虚部βを用いている。ここで、「α＞０」とすることで、２次磁束ベクトルΦ２ｅの各成分を真の値に収束させることができる。

However, in the above, the real part -α and the imaginary part β of the pole of the observer are used. Here, by setting “α> 0”, each component of the secondary magnetic flux vector Φ2e can be converged to a true value.

  DESCRIPTION OF SYMBOLS 10 ... Motor generator (one Embodiment of induction machine), 20 ... Control apparatus, 48 ... Magnetic flux observer, 50 ... Magnetic flux correction | amendment part.

Claims (5)

Secondary magnetic flux calculating means (48) for calculating the secondary magnetic flux component of the induction machine in the fixed coordinate system;
Operating means (20) for operating an AC voltage applying means for applying an AC voltage to the induction machine in order to control a control amount of the induction machine based on the secondary magnetic flux calculated by the secondary magnetic flux calculating means;
Integration processing means (S16) for integrating the components of the fixed coordinate system calculated by the secondary magnetic flux calculation means;
Correction means (S20, S36, S36a) for correcting calculation parameters used for generating an operation signal output from the operation means to the AC voltage application means based on an integration processing result by the integration processing means;
An induction machine control device comprising: The correction means includes secondary magnetic flux correction means (S20) for correcting the secondary magnetic flux calculated by the secondary magnetic flux calculation means based on the integration processing result,
2. The induction machine control device according to claim 1, wherein the operating means operates the AC voltage applying means based on the secondary magnetic flux corrected by the secondary magnetic flux correcting means. Phase calculating means (52) for calculating a primary phase which is a phase of a primary voltage based on the secondary magnetic flux corrected by the secondary magnetic flux correcting means;
The integration processing means, based on the primary phase calculated by the phase calculation means, sets the integration processing value in a period that is an integral multiple of the rotation period of the primary voltage as the integration processing result by the integration processing means. 3. The induction machine control device according to claim 2, wherein the control device inputs the correction means.   The secondary magnetic flux correction means corrects the secondary magnetic flux calculated by the secondary magnetic flux calculation means by an operation amount for feedback-controlling a value obtained by integrating the secondary magnetic flux to a target value. The induction machine control device according to claim 2 or 3. Primary voltage information estimating means for estimating at least one of a primary phase and a primary frequency that are phases of a primary voltage based on the secondary magnetic flux calculated by the secondary magnetic flux calculating means;
The correction means includes
Rotating coordinate conversion means for converting a value integrated by the integration processing means into a component of a rotating two-dimensional coordinate system rotating at a primary frequency;
Pulsation reducing means for correcting at least one of the components converted by the rotating coordinate conversion means to reduce and control pulsation components;
The induction machine control device according to claim 1, further comprising:

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