JP5656695B2 - Electric motor drive device and air conditioner - Google Patents

Description

  The present invention relates to an electric motor drive device and an air conditioner.

  Conventionally, for example, a converter circuit that boosts a DC voltage according to a DC voltage command value and outputs the boosted voltage to a smoothing circuit, and an inverter circuit that performs power conversion between DC power on the smoothing circuit and AC power that drives the motor, The torque command value of the motor is set according to the output request to the output, the minimum value of the voltage required for the torque output is calculated as the required minimum voltage, and multiple values within the range higher than the required minimum voltage and lower than the output upper limit voltage DC voltage command value candidates are estimated, the power loss in the converter circuit, inverter circuit, and motor is estimated for each DC voltage command value candidate, and the DC voltage command value candidate that minimizes the sum of these power losses is optimal A technique for setting a DC voltage command value as a correct voltage is disclosed (Patent Document 1).

  For example, when the torque of the motor is 0, the d-axis current corresponding to the motor current is composed of a fundamental wave component calculated from the motor equation and a ripple component calculated from the switching timing chart. Since the power component command value is related to the wave component and the inductance value and switching frequency of the motor are related to the ripple component, the copper loss of the motor is calculated from the formula based on these factors, and this copper loss is minimized. A technique for correcting an output voltage command value of an inverter circuit so as to be a value is disclosed (Patent Document 2).

JP 2007-325351 A JP 2007-14185 A

  However, in the technique described in Patent Document 1, the total power loss is minimized by setting the DC voltage output from the converter circuit. However, the loss of each converter circuit and inverter circuit is minimized by other factors. As a result, there is a problem that the total power loss cannot be minimized. In the technique described in Patent Document 2, the copper loss of the electric motor can be minimized by adjusting the switching frequency, but the iron loss, which is another element of the electric loss of the electric motor, cannot be minimized. There was a problem that.

  This invention is made | formed in view of the above, Comprising: It aims at providing the electric motor drive device which can make the total value of each electric power loss which generate | occur | produces in an inverter circuit and an electric motor smaller.

In order to solve the above-described problems and achieve the object, an electric motor driving device according to the present invention is supplied with a first AC voltage from an AC power source, and converts the first AC voltage into a second AC voltage. An electric motor driving device that outputs to an electric motor, wherein a converter circuit that converts the first AC voltage into a DC voltage, a smoothing circuit that smoothes the DC voltage, and an upper and lower circuit in which a semiconductor switch element and a diode are connected in antiparallel One or more pairs of arms are provided, and an inverter circuit that converts the DC voltage into the second AC voltage, a drive circuit that drives the semiconductor switch element, and a motor current that flows through the motor are detected. Based on the current detection circuit and the motor current, the switching generated by switching when the semiconductor switch element is turned on or turned off. Loss, conduction loss caused by conduction of the semiconductor switch element and the diode, iron loss occurring in the iron core of the motor, and copper loss occurring in the primary winding of the motor, calculating the switching loss, the conduction loss, A motor drive control unit that calculates a total loss of the iron loss and the copper loss and controls the total loss to be smaller, and the motor drive control unit includes a turn-on time of the semiconductor switch element and Switching speed control means for controlling at least one of the turn-off times, and the drive circuit restricts a gate current flowing through the semiconductor switch element at a rising edge of a drive pulse signal output from the electric motor drive control unit. 1 circuit element and the semiconductor switch element at the fall of the drive pulse signal A second circuit element for limiting the gate current generated, a turn-on time setting circuit for setting the turn-on time by controlling a first resistance element included in the first circuit element, and the second circuit element to control the second resistive elements contained, and the turn-off time setting circuit which sets the turn-off time, characterized by Rukoto equipped with.

  According to the present invention, there is an effect that the total value of each power loss generated in the inverter circuit and the electric motor can be further reduced.

FIG. 1 is a diagram of a configuration example of the electric motor drive device according to the first embodiment. FIG. 2 is a diagram showing an example of the relationship between the current and voltage flowing through the semiconductor switch element and the diode in a steady state. FIG. 3 is a timing chart showing the state of the semiconductor switch element and the period during which current flows through the semiconductor switch element and the diode. FIG. 4 is a schematic diagram of the switching pulse modulation factor, the motor current, and the collector-emitter saturation voltage. FIG. 5 is a diagram illustrating an example of the internal configuration of the drive circuit. FIG. 6 is a schematic diagram illustrating an example of fluctuations in the collector current and the collector-emitter voltage in a transient state when the semiconductor switch element is turned on and turned off. FIG. 7 is a diagram of a configuration example of the electric motor drive device according to the fourth embodiment. FIG. 8 is a schematic diagram illustrating an example of fluctuations in the collector current and the collector-emitter voltage in a transient state when the semiconductor switch element is turned on, and fluctuations in the current flowing through the diode and the voltage across the both ends. FIG. 9 is a diagram of a configuration example of the electric motor drive device according to the fifth embodiment. FIG. 10 is a diagram illustrating a configuration example of an air-conditioning apparatus according to the sixth embodiment.

  An electric motor drive device according to an embodiment of the present invention will be described below with reference to the accompanying drawings. In addition, this invention is not limited by embodiment shown below.

Embodiment 1 FIG.
FIG. 1 is a diagram of a configuration example of the electric motor drive device according to the first embodiment. As shown in FIG. 1, the electric motor drive device 1 according to the first exemplary embodiment is connected to an AC power source 2 and an electric motor 3, and is supplied with a first AC voltage from the AC power source 2 to drive the electric motor 3. The AC voltage is converted and output to the motor 3.

  The electric motor drive device 1 includes a converter circuit 4, a smoothing circuit 5, an inverter circuit 6, a current detection circuit 7, an electric motor drive control unit 10, and drive circuits 20a and 20b.

  The converter circuit 4 converts the first AC voltage supplied from the AC power supply 2 into a DC voltage based on the DC voltage command value from the electric motor drive control unit 10. The smoothing circuit 5 includes a smoothing capacitor 5 a that smoothes the output voltage of the converter circuit 4.

  The inverter circuit 6 includes semiconductor switch elements 6a and 6b and diodes 6c and 6d. The semiconductor switch elements 6a and 6b and the diodes 6c and 6d constitute a pair of upper and lower arms, and the smoothed DC power is converted into desired AC power. To be output to the electric motor 3. Note that the number of arms constituting the inverter circuit 6 varies depending on the configuration of the electric motor 3. For example, if the motor 3 is a three-phase AC motor, the inverter circuit 6 is a full-bridge circuit having an upper limit of three pairs of arms and a three-phase output. Further, the inverter circuit 6 may have any configuration as long as it drives the electric motor 3.

  The drive circuits 20a and 20b include a switching time control unit 15, and based on the drive pulse signal, the turn-on time command value, and the turn-off time command value from the electric motor drive control unit 10, the semiconductor switch elements 6a and 6b of the inverter circuit 6 are provided. Drive.

  The current detection circuit 7 detects a current flowing through the electric motor 3 and outputs it to the electric motor drive control unit 10. For example, when the inverter circuit 6 has a three-phase output, the current detection circuit 7 may detect all three-phase currents, or any two-phase current out of the three phases. May be detected to calculate another one-phase current. Alternatively, the current detection circuit 7 is provided between the smoothing circuit 5 and the inverter circuit 6, and the current flowing between the smoothing circuit 5 and the inverter circuit 6 is detected, and each current is detected according to the state of the semiconductor switch element of the inverter circuit 6. A phase current may be calculated. The current detection circuit 7 may have any configuration as long as it detects the current flowing through the motor 3. Hereinafter, the current flowing through the motor 3 is referred to as “motor current”.

  The electric motor drive control unit 10 includes a control element control unit 11, a DC voltage command generation unit 12, a switching frequency command generation unit 13, and a drive pulse signal generation unit 14.

  The control element control unit 11 outputs the output from the converter circuit 4 based on the operation command from the outside and the loss amounts of the inverter circuit 6 and the motor 3 calculated based on the motor current input from the current detection circuit 7. Voltage value, switching frequency of drive pulse signal for driving semiconductor switch elements 6a and 6b, switching pulse modulation factor indicating ratio of ON period of semiconductor switch elements 6a and 6b in one cycle of drive pulse signal, frequency of switching pulse modulation factor The control elements including the turn-on time and the turn-off time, which are times required for turning on and turning off the semiconductor switch elements 6a and 6b, are controlled, and the output voltage value information, the switching frequency information, the switching pulse modulation factor and the frequency are respectively controlled. Including switching pulse modulation degree information, tar On-time information, and outputs it as the turn-off time information. The control element control unit 11 functions as a converter circuit output voltage control unit when controlling the output voltage value, and functions as a switching frequency control unit when controlling the switching frequency. When controlling the frequency of the switching pulse modulation degree, it functions as switching pulse modulation degree control means, and when controlling the turn-on time and turn-off time, it functions as switching speed control means.

  The DC voltage command generation unit 12 generates a DC voltage command value based on the output voltage value information output from the control element control unit 11 and outputs it to the converter circuit 4.

  The switching frequency command generation unit 13 generates a switching frequency command value based on the switching frequency information output from the control element control unit 11 and outputs the switching frequency command value to the drive pulse signal generation unit 14.

  The drive pulse signal generation unit 14 generates a drive pulse command value based on the switching pulse modulation degree information output from the control element control unit 11 and the switching frequency command value output from the switching frequency command generation unit 13, and Based on the turn-on time information and the turn-off time information output from the control element control unit 11, a turn-on time command value and a turn-off time command value are generated, and the drive pulse command value and the turn-off time command value are sent to the drive circuits 20a and 20b. Output.

  Next, the operation of the electric motor drive device 1 according to the first embodiment will be described. As described above, the control element control unit 11 converts the converter circuit based on the operation command from the outside and the loss amounts of the inverter circuit 6 and the motor 3 calculated based on the motor current input from the current detection circuit 7. 4 outputs an output voltage value, a switching frequency of a driving pulse signal for driving the semiconductor switching elements 6a and 6b, a switching pulse modulation factor indicating a ratio of an ON period of the semiconductor switching elements 6a and 6b in one cycle of the driving pulse signal, a semiconductor The turn-on time and the turn-off time, which are times required for turning on and turning off the switch elements 6a and 6b, are controlled. In other words, in the present embodiment, the control element control unit 11 outputs the output voltage value, the switching frequency, the switching pulse modulation factor, the turn-on time so that the total loss obtained by adding up the loss amounts of the inverter circuit 6 and the electric motor 3 is reduced. And control each control element including turn-off time.

  Based on the motor current input from the current detection circuit 7, the control element control unit 11 conducts conduction loss, switching loss, and winding of the motor 3 generated in the semiconductor switch elements 6 a and 6 b and the diodes 6 c and 6 d of the inverter circuit 6. The copper loss occurring in the motor and the iron loss occurring in the motor 3 are calculated, and the total loss is obtained.

  First, the conduction loss calculation method will be described. In the present embodiment, the loss per second caused by the conduction of the semiconductor switch elements 6a and 6b and the diodes 6c and 6d is defined as the conduction loss. FIG. 2 is a diagram showing an example of the relationship between the current and voltage flowing through the semiconductor switch element and the diode in a steady state. FIG. 2A shows an example of Ic-Vce (sat) characteristics indicating the relationship between the collector current Ic of the semiconductor switch element and the collector-emitter saturation voltage Vce (sat), and FIG. 2 shows an example of an If-Vf characteristic indicating the relationship between the forward current If and the forward voltage Vf. When an on signal is applied to the semiconductor switch elements 6a and 6b and a collector current Ic flows, a collector-emitter saturation voltage Vce (sat) corresponding to the current value of Ic is generated. In the semiconductor switch elements 6a and 6b, a product of Ic and Vce (sat) is generated as a conduction loss. Similarly, when a forward current If flows through the diodes 6c and 6d, a forward voltage Vf corresponding to the current value of If is generated between the anode and the cathode, so that the product of If and Vf is present in the diodes 6c and 6d. Occurs as conduction loss.

  When current flows through the semiconductor switch elements 6a and 6b and the diodes 6c and 6d for the time T, the loss amount Ecs generated in the semiconductor switch elements 6a and 6b and the loss amount Ecd generated in the diodes 6c and 6d are respectively ( It can be expressed by the formulas 1) and (2).

  Ecs = Ic × Vce (sat) × T (1)

  Ecd = If × Vf × T (2)

  FIG. 3 is a timing chart showing the state of the semiconductor switch element and the period during which current flows through the semiconductor switch element and the diode. FIG. 3 (a) shows a timing chart when the motor current Ia is positive, and FIG. 3 (b) shows a timing chart when the motor current Ia is negative, each over one cycle of the drive pulse signal. Show. Here, the motor current Ia is positive (Ia> 0) when the direction of the current flowing through the motor 3 is the direction from the inverter circuit 6 to the motor 3, and the motor current Ia is negative (Ia <0). Suppose that the direction of the current flowing through the motor 3 is the direction from the motor 3 to the inverter circuit 6. Note that the reference numeral of the motor current Ia is the same in the following description. In the following description, a time corresponding to one cycle of the drive pulse signal is referred to as “switching cycle Tc”.

  Normally, in the control of the inverter circuit 6, a period during which the semiconductor switch element 6a is turned on and a period during which the semiconductor switch element 6b is turned on are provided during the switching cycle Tc. At this time, there is no period in which one of the semiconductor switch elements 6a and 6b is turned on and both are turned on simultaneously. Here, when the ratio of the ON period of the semiconductor switching element 6a in a certain switching period Tc, that is, in the example shown in FIG. 3A, the switching pulse modulation degree Da in a certain switching period Tc is “Duty”, the semiconductor switching element The ON period Tca of 6a and the ON period Tcb of the semiconductor switch element 6b can be expressed by the following equations (3) and (4), respectively.

  Tca = Tc × Duty (3)

  Tcb = Tc × (1-Duty) (4)

  When Ia is a positive value (Ia> 0) (see FIG. 3A), in the on period of the semiconductor switch element 6a, the semiconductor switch element 6a conducts and current flows, and in the on period of the semiconductor switch element 6b, The diode 6d conducts and current flows. Therefore, the loss amount Ecc (Ia> 0) generated in the inverter circuit 6 over the switching cycle Tc can be expressed by the following equation (5).

Ecc (Ia> 0) = Ic × Vce (sat) × Tca + If × Vf × Tcb
= Ic x Vce (sat) x Tc x Duty
+ If × Vf × Tc × (1-Duty) (5)

  When Ia is a negative value (Ia <0) (see FIG. 3B), the diode 6c conducts in the on period of the semiconductor switch element 6a and current flows, and in the on period of the semiconductor switch element 6b, the semiconductor switch Element 6b conducts and current flows. Therefore, the loss amount Ecc (Ia <0) generated in the inverter circuit 6 over the switching cycle Tc can be expressed by the following equation (6).

Ecc (Ia <0) = Ic * Vce (sat) * Tcb + If * Vf * Tca
= Ic * Vce (sat) * Tc * (1-Duty)
+ If × Vf × Tc × Duty (6)

  Here, Vce (sat) is determined by Ic based on the Ic-Vce (sat) characteristic shown in FIG. Vf is determined by If based on the If-Vf characteristic shown in FIG. Since Ic and If are both instantaneous values of the motor current Ia, the control element control unit 11 holds the Ic-Vce (sat) characteristic and If-Vf characteristic shown in FIG. Thus, Vce (sat) and Vf can be obtained from Ia. That is, the control element control unit 11 uses the switching pulse modulation degree Da controlled by itself, the motor current Ia input from the current detection circuit 7, and the Ic-Vce (sat) characteristic and If-Vf characteristic held in advance. The amount of loss for each switching period Tc can be obtained.

  Therefore, the loss amount for each switching period Tc is calculated one by one, added over an arbitrary period sufficiently long with respect to the switching period Tc, and then divided by that period, so that the semiconductor switch elements 6a and 6b and the diodes 6c and 6d are obtained. It is possible to obtain an average generated loss amount per second caused by the conduction, that is, a conduction loss generated in the inverter circuit 6.

  In the above calculation of the conduction loss, the loss amount for each switching period Tc is calculated one by one and the loss amount per second is obtained as the conduction loss. However, the conduction loss is estimated by estimating the loss amount over an arbitrary period. You may ask for it. In the following description, the amount of loss caused by conduction of the semiconductor switch elements 6a and 6b and the diodes 6c and 6d in one cycle Ta of the motor current Ia or the switching pulse modulation degree Da is obtained, and the frequency of the motor current Ia or the switching pulse modulation degree Da is obtained. A method of obtaining the conduction loss by multiplying fa will be described.

  FIG. 4 is a schematic diagram of the switching pulse modulation factor, the motor current, and the collector-emitter saturation voltage. In FIG. 4, the solid line indicates the motor current Ia, the broken line indicates the collector-emitter saturation voltage Vce (sat), and the alternate long and short dash line indicates the switching pulse modulation degree Da approximated by a sine wave. ing. The switching pulse modulation degree Da is equivalent to the duty in the equations (3) to (6) and represents the peak value or effective value in a simulated manner. In FIG. 4, the forward voltage Vf of the diodes 6c and 6d is not shown, but shows a tendency substantially equal to the collector-emitter saturation voltage Vce.

  If the instantaneous value of the motor current Ia can be expressed as a function of a sine wave, for example, the instantaneous value of the motor current Ia and the instantaneous collector-emitter saturation voltage Vce (sat) of the semiconductor switch elements 6a and 6b at that time The value and the instantaneous value of the forward voltage Vf of the diodes 6c and 6d can be expressed as a function or table of time t, respectively. Further, since the switching pulse modulation degree Da is an AC value having a certain phase difference from the motor current Ia due to the characteristics of the motor 3, it can be simplified by expressing it as a function by the phase difference of the instantaneous value of the motor current Ia.

  Therefore, IA (t) is the instantaneous value of the motor current Ia at time t, DA (t) is the instantaneous value of the switching pulse modulation degree Da at time t, and Vce (IA (t)) is the semiconductor at the instantaneous value IA (t). When the instantaneous value of the collector-emitter saturation voltage Vce (sat) of the switch elements 6a and 6b and Vf (IA (t)) are the instantaneous values of the forward voltages Vf of the diodes 6c and 6d at the instantaneous value IA (t), The amount of loss caused by the conduction of the semiconductor switch elements 6a and 6b and the diodes 6c and 6d in a half cycle in which Ia is positive with respect to one period of the motor current Ia can be expressed by the following equation (7): In the half cycle when is negative, the loss caused by conduction of the semiconductor switch elements 6a and 6b and the diodes 6c and 6d can be expressed by the following equation (8). Kill.

  Since the equations (7) and (8) are equations for calculating the amount of loss in each half cycle based on the motor current Ia, the motor current Ia is calculated by adding the equations (7) and (8). The amount of loss for one cycle can be obtained. Further, since the switching pulse modulation degree Da and the motor current Ia have a certain phase difference, the sum of the expressions (7) and (8) may be used as the loss amount for one cycle Ta of the switching pulse modulation degree Da. There is no. When the waveform in the half cycle in which the motor current Ia is positive and the waveform in the half cycle in which the motor current Ia is negative are symmetric, the calculation result of the equation (7) and the calculation result of the equation (8) Therefore, the loss amount corresponding to one cycle Ta of the motor current Ia or the switching pulse modulation degree Da can be obtained by doubling one of the calculation results. That is, the control element control unit 11 controls the switching pulse modulation degree Da and the period Ta of the switching pulse modulation degree Da that it controls, the motor current Ia input from the current detection circuit 7, and Ic−Vce (sat previously held. ) Characteristics and If-Vf characteristics, the loss amount for one period Ta of the switching pulse modulation degree Da can be obtained.

  Therefore, when the loss amount for one period Ta of the switching pulse modulation degree Da is obtained by the above-described equations (7) and (8), the loss amount is multiplied by the frequency fa of the switching pulse modulation degree Da, The amount of loss generated per second caused by the conduction of the semiconductor switch elements 6a and 6b and the diodes 6c and 6d, that is, the conduction loss generated in the inverter circuit 6 can be obtained. Further, since the phase of the motor current Ia is equal to the phase of the second AC voltage applied to the motor 3, the sum of the formulas (7) and (8) is 1 of the second AC voltage applied to the motor 3. Equal to the amount of loss over the period. Therefore, the average value of the loss amount over one cycle of the second AC voltage is obtained, and the obtained average value is used as the frequency of the second AC voltage, that is, switching pulse modulation having a constant phase difference from the second AC voltage. It is also possible to obtain the conduction loss generated in the inverter circuit 6 by multiplying the frequency fa of the degree Da.

  This conduction loss can be varied by controlling the switching pulse modulation degree Da. That is, the conduction loss can be further reduced if the switching pulse modulation degree Da is further reduced from the above-described equations (5), (6) or (7), (8).

  The conduction loss obtained by the above description is a conduction loss when the motor 3 is the single-phase AC motor shown in FIG. 1 and the inverter circuit 6 is composed of a pair of upper and lower arms. When the motor 3 is a three-phase AC motor, the inverter circuit 6 is a full-bridge type circuit having three pairs of arms and three-phase outputs. Therefore, the conduction loss calculated for each output phase is totaled. Calculating the conduction loss of the entire inverter circuit 6 or assuming that the conduction loss generated for each output phase is uniform in all arms and multiplying the conduction loss of the upper and lower pair of arms by the number of phases. What is necessary is just to calculate the conduction loss of the whole inverter circuit 6.

  Next, a switching loss calculation method will be described. In the present embodiment, the amount of loss per second caused by switching at the time of turn-on and turn-off of the semiconductor switch elements 6a and 6b is defined as a switching loss. FIG. 5 is a diagram illustrating an example of the internal configuration of the drive circuit. Since the internal configurations of the drive circuits 20a and 20b are the same, the example shown in FIG. 5 shows the internal configuration of the drive circuit 20a.

  As shown in FIG. 5, the drive circuit 20 a includes a switching time setting unit 15 including a turn-on time setting circuit 21 and a turn-off time setting circuit 22. When the semiconductor switch element 6a is, for example, an insulated gate bipolar transistor, the drive circuit 20a applies the gate-emitter voltage Vge to the gate terminal of the semiconductor switch element 6a and starts to flow the gate current Ig when switching at turn-on. Thus, the collector current Ic starts to flow, and in the switching at the turn-off, the collector current Ic is stopped by turning off the gate-emitter voltage Vge and finishing the flow of the gate current Ig. As shown in FIG. 5, since the semiconductor switch element 6a has a parasitic capacitance CG between the gate terminal and the collector terminal, if the gate current Ig starts to flow sharply or finishes flowing sharply, The high-frequency component of the current Ig passes through the parasitic inductance between the gate terminal and the collector terminal, the charge accumulated in the parasitic capacitance CG is drawn into the gate current Ig, the voltage across the parasitic capacitance CG vibrates, and the collector-emitter There is a risk of generating an excessive surge voltage or vibration voltage.

  Generally, in order to limit such a steep fluctuation of the gate current Ig, a current limiting circuit element is provided in series with the gate terminal. In the present embodiment, as shown in FIG. 5, a circuit element 23a for limiting the amount of fluctuation of the gate current Ig at turn-on and a circuit element 23b for limiting the amount of fluctuation of the gate current Ig at turn-off are connected in parallel. Configured. The circuit elements 23a and 23b are each configured to include a resistance element, and each of the circuit elements 23a and 23b includes a rectifying element such as a diode in addition to the resistance element because the start and end of the flow of the gate current are distinguished and limited. .

  FIG. 6 is a schematic diagram illustrating an example of fluctuations in the collector current and the collector-emitter voltage in a transient state when the semiconductor switch element is turned on and turned off. FIG. 6A shows the fluctuation of the collector current Ic and the collector-emitter voltage VCE at the turn-on time, and FIG. 6B shows the fluctuation of the collector current Ic and the collector-emitter voltage VCE at the turn-off time. Yes.

  As shown in FIG. 6, by limiting the amount of fluctuation of the gate current Ig, the time required to change from 0 to a constant current value at the beginning of the collector current Ic, that is, the semiconductor switch elements 6a and 6b A turn-on time Ton, which is a time required for turn-on, and a time required to change from a constant current value to 0 at the end of the flow of Ic, that is, a turn-off time Toff, which is a time required for turning off the semiconductor switch elements 6a and 6b. Each is provided. Further, the collector-emitter voltage VCE when the semiconductor switch element 6a is in the OFF state is the same value as the output voltage value VDC output from the converter circuit 4.

  The turn-on time setting circuit 21 adjusts the rise time of the drive pulse signal by changing the resistance element included in the circuit element 23 a based on the turn-on time command value output from the drive pulse signal generation unit 14. Time Ton is set. The turn-off time setting circuit 22 adjusts the fall time of the drive pulse signal by changing the resistance element included in the circuit element 23b based on the turn-off time command value output from the drive pulse signal generation unit 14. The turn-off time Toff is set.

  As shown in FIG. 6, the collector current Ic and the collector-emitter voltage VCE during the turn-on time Ton (hereinafter referred to as “turn-on period”) and the turn-off time Toff (hereinafter referred to as “turn-off period”). The fluctuation can be approximated approximately linearly. In the transient state of these turn-on periods and turn-off periods, the collector current Ic and the collector-emitter voltage VCE are not zero, so that the semiconductor switch elements 6a and 6b are affected by the collector current Ic and the collector-emitter voltage VCE. Power loss occurs. Here, assuming that the loss amount in the turn-on period is Eon and the loss amount in the turn-off period is Eoff, the loss amount Eon generated by one turn-on operation can be expressed by the following equation (9). The loss amount Eoff generated by the operation can be expressed by the following equation (10).

  Here, the collector current Ic is an instantaneous value of the motor current Ia, and the output voltage value VDC is an output voltage value output from the converter circuit 4 based on a DC voltage command value output from the DC voltage command generator 12. is there. That is, the control element control unit 11 obtains the loss amounts Eon and Eoff from the output voltage value VDC controlled by itself, the turn-on time Ton and the turn-off time Toff, and the motor current Ia input from the current detection circuit 7. it can.

  As shown in FIG. 3, when the motor current Ia is a positive value (Ia> 0) in the period of the switching period Tc, the semiconductor switch element 6a is turned on and off once, and the motor current Ia Is a negative value (Ia <0), turn-on and turn-off occur once each in the semiconductor switch element 6b. Therefore, in 1 second, the number of times of the switching period Tc, that is, Eon and Eoff of the switching frequency fc are generated. Generally, in order to control the electric motor 3 stably, the switching frequency fc needs to be a frequency sufficiently higher than the frequency fa of the switching pulse modulation degree Da. As described above, when the switching frequency fc is sufficiently higher than the frequency fa of the switching pulse modulation degree, the generated loss per second caused by switching when the semiconductor switch elements 6a and 6b are turned on and turned off. The quantity is approximately proportional to the switching frequency fc.

  Therefore, the loss amount for each switching period Tc obtained by the above equations (9) and (10) is calculated one by one, added over an arbitrary period sufficiently long with respect to the switching period Tc, and divided by that period. Thus, it is possible to obtain an average generated loss amount per second generated by switching at the turn-on time and turn-off time of the semiconductor switch elements 6a and 6b, that is, a switching loss generated in the inverter circuit 6.

  Alternatively, since the switching loss is proportional to the switching frequency fc, the switching loss can be obtained by multiplying the amount of loss for one cycle of the switching cycle Tc by the switching frequency fc.

  This switching loss can be varied by controlling the output voltage value VDC, the switching frequency fc, the turn-on time Ton, and the turn-off time Toff. That is, if the output voltage value VDC is lower and the turn-on time Ton and the turn-off time Toff are further shortened from the equations (9) and (10), switching loss can be further reduced. Alternatively, since the switching loss is proportional to the switching frequency fc, the switching loss can be further reduced even if the switching frequency fc is made smaller.

  The switching loss obtained by the above description is a switching loss when the motor 3 is the single-phase AC motor shown in FIG. 1 and the inverter circuit 6 is composed of a pair of upper and lower arms. When the motor 3 is a three-phase AC motor, the inverter circuit 6 is a full-bridge circuit having three pairs of arms and three-phase outputs, so the switching loss calculated for each output phase is totaled. The switching loss of the entire inverter circuit 6 is calculated, or the switching loss generated for each output phase is assumed to be uniform in all arms, and the switching loss of the upper and lower pair of arms is multiplied by the number of phases. What is necessary is just to calculate the switching loss of the whole inverter circuit 6.

  Next, a method for calculating the copper loss of the electric motor 3 will be described. The copper loss Wa generated in the primary winding of the electric motor 3 can be expressed by the following equation (11) from the effective value Iae of the electric motor current Ia and the resistance value Ra of the primary winding.

Wa = Iae ² × Ra (11)

  The effective value Iae of the motor current Ia can be obtained from the motor current Ia input from the current detection circuit 7. The resistance value Ra of the primary winding of the electric motor 3 is a constant unique to the electric motor 3 and is previously held by the control element control unit 11. That is, the control element control unit 11 calculates the copper loss Wa generated in the primary winding of the motor 3 from the motor current Ia input from the current detection circuit 7 and the resistance value Ra of the primary winding of the motor 3 held in advance. Can be sought.

  Next, a method for calculating the iron loss of the electric motor 3 will be described. The iron loss that occurs in the iron core of the motor 3 is a loss that is generated as energy in the iron core and lost when it is magnetized by flowing an alternating current through the coil wire surrounding the iron core. There is.

  Hysteresis loss is a magnetization curve drawn when the magnetic field and magnetic flux fluctuate due to the alternating magnetic field generated due to the alternating current of the coil conductor, and the area surrounded by the hysteresis curve, that is, the magnetic field and The product of the magnetic flux density is a loss that is generated as energy in the iron core and lost as heat. In addition to the fundamental wave component due to the motor current Ia, the alternating magnetic field also acts as a switching frequency component of a ripple current generated by the switching operation of the semiconductor switch elements 6a and 6b of the inverter circuit 6.

  This hysteresis loss is the cumulative total per second of the area surrounded by the hysteresis curve. Since the number of times per second for drawing the hysteresis curve is equal to the frequency of the current, the hysteresis loss due to the fundamental wave component of the motor current Ia is proportional to the frequency fa of the switching pulse modulation degree Da, and the hysteresis due to the switching frequency component of the ripple current The loss is proportional to the switching frequency fc. However, for the switching frequency component of the ripple current, the peak value of the ripple current is inversely proportional to the switching frequency, and the amount of change in the magnetic field and magnetic flux is both proportional to the peak value of the ripple current. The area is inversely proportional to the square of the switching frequency fc.

  Therefore, the hysteresis loss Wh is represented by the following (12) by the frequency fa of the switching pulse modulation degree Da, the switching frequency fc, the maximum magnetic flux density Bm that is a constant specific to the electric motor 3, and the coefficients kh and khc determined by the characteristics of the iron core material. It can be expressed by a formula.

Wh = (kh × fa + khc × fc / fc ² ) × Bm ²
= (Kh × fa + khc / fc) × Bm ² (12)

  Eddy current loss refers to the Joule heat loss that occurs in the iron core due to the eddy current, and the current that flows through the iron core to surround the magnetic flux when the magnetic flux passes through the iron core and changes due to alternating current. Say. Since the number of eddy currents flowing in the iron core per second is equal to the frequency fa of the switching pulse modulation degree Da, the cumulative number of eddy currents per second is proportional to the frequency fa of the switching pulse modulation degree Da. The loss is proportional to the square of the frequency fa of the switching pulse modulation degree Da.

  Therefore, the eddy current loss We can be expressed by the following equation (13) by the frequency fa of the switching pulse modulation degree Da, the maximum magnetic flux density Bm that is a constant specific to the electric motor 3, and the coefficient ke determined by the thickness of the iron core. it can.

We = ke × fa ² × Bm ² (13)

  The frequency fa of the switching pulse modulation degree Da is a frequency of the switching pulse modulation degree controlled by the control element control unit 11 itself, and the switching frequency fc is one of the parameters controlled by the control element control unit 11 itself. The coefficients kh, khc, ke and the maximum magnetic flux density Bm are constants specific to the electric motor 3, and are previously held by the control element control unit 11. That is, the control element control unit 11 uses the frequency fa and the switching frequency fc of the switching pulse modulation degree Da controlled by itself and the coefficients kh, khc, ke, and the maximum magnetic flux density Bm stored in advance in the iron core of the motor 3. The generated hysteresis loss Wh and eddy current loss We can be obtained.

  This iron loss can be changed by controlling the switching frequency fc and the frequency fa of the switching pulse modulation degree Da. In other words, the hysteresis loss Wh included in the iron loss can be further reduced by increasing the switching frequency fc from the above equation (12), and the switching pulse modulation from the above equations (12) and (13). If the frequency fa of the degree Da is made smaller, both the hysteresis loss Wh and the eddy current loss We included in the iron loss can be further reduced.

  As described above, if the switching pulse modulation degree Da is made smaller, the conduction loss can be further reduced, the output voltage value VDC is made lower, the switching frequency fc is made smaller, and the turn-on time Ton and the turn-off time Toff are made shorter. Thus, the switching loss can be further reduced. If the switching frequency fc is further increased, the hysteresis loss included in the iron loss can be further reduced, and if the frequency fa of the switching pulse modulation degree Da is further decreased. Both the hysteresis loss Wh and the eddy current loss We included in the iron loss can be further reduced. However, when each of these control elements is controlled individually, the motor current Ia increases and the individual loss including the copper loss increases. The amount or the sum of these losses may increase. Therefore, the control element control unit 11 is a total loss of the conduction loss, the switching loss, the copper loss, and the iron loss calculated by the above-described calculation formulas within a range in which the operation of the electric motor 3 in accordance with the operation command from the outside is possible. Each control element is controlled so that becomes smaller. When switching loss is controlled, it is not necessary to control all of output voltage value VDC, turn-on time Ton, turn-off time Toff, and switching frequency fc, and one or more control elements are selected from these. You may control.

  The control element control unit 11 needs to control each control element within a range in which the electric motor 3 can be operated in accordance with an operation command from the outside. Therefore, there is a lower limit value in the control range of each control element. To do. For example, the lower limit value of the output voltage value VDC is determined in consideration of the control element of the switching pulse modulation degree Da, and the turn-on time Ton and the turn-off time Toff are the characteristics of the semiconductor switch elements 6a and 6b and the control element of the switching frequency fc. The lower limit value is determined based on the balance between the two, and the lower limit value of the switching frequency fc is determined by the control characteristics of the electric motor 3. Therefore, it is necessary to control each control element in consideration of the lower limit value of the control range of each control element.

  As described above, according to the motor drive device of the first embodiment, the conduction loss caused by the conduction of the semiconductor switch element and the diode of the inverter circuit, the switching caused by the switching at the turn-on and turn-off of the semiconductor switch element of the inverter circuit Calculate the loss, the copper loss that occurs in the primary winding of the motor, and the iron loss that occurs in the iron core of the motor, and the output voltage value of the converter circuit and the switching pulse modulation degree of the inverter circuit so that the total loss becomes smaller Since the frequency of the switching pulse modulation, the switching frequency, the turn-on time, and the turn-off time are controlled, the total value of each power loss generated in the inverter circuit and the electric motor can be further reduced.

  In Embodiment 1 described above, the turn-on time setting circuit and the turn-off time setting circuit included in the switching time setting unit are configured as part of the drive circuit, but are configured as part of the drive pulse signal generation unit. Is also possible. In this case, the drive pulse signal generation unit generates the drive pulse command value based on the switching pulse modulation degree information output from the control element control unit and the switching frequency command value output from the switching frequency command generation unit. The turn-on time command value and the turn-off time command value are generated based on the turn-on time information and the turn-off time information output from the control element control unit, and the rise time and fall time of the drive pulse signal are set in the switching time setting unit. It may be adjusted and output to the drive circuit.

Embodiment 2. FIG.
Since the configuration of the electric motor drive device according to the second embodiment is the same as the configuration shown in FIG. 1 described in the first embodiment, detailed description of each component will be omitted.

  In the first embodiment, conduction loss caused by conduction of semiconductor switching elements 6a and 6b and diodes 6c and 6d of inverter circuit 6, switching loss caused by switching of semiconductor switching elements 6a and 6b of inverter circuit 6 at turn-on and turn-off. The copper loss generated in the primary winding of the motor 3 and the iron loss generated in the iron core of the motor 3 are calculated, and the output voltage value VDC of the converter circuit 4 and the inverter circuit 6 The control elements of the switching pulse modulation degree Da, the frequency fa of the switching pulse modulation degree Da, the switching frequency fc, the turn-on time Ton, and the turn-off time Toff are controlled. For example, hysteresis included in the iron loss of the motor 3 Switching to reduce losses Increasing the wave number fc, switching loss of the inverter circuit 6 increases, resulting temperature of the inverter circuit 6 rises as becomes overheated.

  Further, as described in the first embodiment, since each control element has a lower limit value, a lower limit value also exists in the switching loss of the inverter circuit 6. Even if all these control elements are set to the lower limit values, depending on the control characteristics of the motor 3, the motor current Ia may increase, and the sum of the conduction loss and the switching loss may increase.

  Therefore, in the second embodiment, in addition to the control of the first embodiment, when the upper limit temperature of the inverter circuit 6 is set in advance and the temperature of the inverter circuit 6 exceeds the upper limit temperature, the conduction loss and the switching loss are Each control element is controlled so that the sum of the values becomes smaller.

  As described above, according to the motor drive device of the second embodiment, in addition to the control of the first embodiment, when the inverter circuit temperature exceeds the preset upper limit temperature, the conduction loss and the switching loss are reduced. Since the output voltage value of the converter circuit, the switching pulse modulation degree of the inverter circuit, the switching frequency, the turn-on time, and the turn-off time are controlled so that the total becomes smaller, the temperature rise of the inverter circuit is suppressed, and the inverter It is possible to prevent the circuit from being overheated.

Embodiment 3 FIG.
Since the configuration of the electric motor drive device according to the third embodiment is the same as the configuration shown in FIG. 1 described in the first embodiment, detailed description of each component will be omitted.

  In the first embodiment, conduction loss caused by conduction of semiconductor switch elements 6a and 6b and diodes 6c and 6d of the inverter circuit, switching loss caused by switching at turn-on and turn-off of semiconductor switch elements 6a and 6b of the inverter circuit, electric motor 3 is calculated, and the output voltage value VDC of the converter circuit 4 and the switching pulse of the inverter circuit 6 are calculated so that the total loss becomes smaller. The control element of the modulation factor Da, the frequency fa of the switching pulse modulation factor Da, the switching frequency fc, the turn-on time Ton, and the turn-off time Toff is controlled. In order to control a plurality of control elements, The number of combinations increases significantly, reducing the total loss Ri calculation amount increases required in order to reduce, even a long time to complete the control of the control element. As a result, the total amount of loss that occurs until the control of the control element is completed increases.

  Therefore, in the third embodiment, the conduction loss, the switching loss, the copper loss, and the iron loss are calculated by the calculation methods described in the first embodiment, and the one with the larger loss amount is extracted, Each control element is controlled so as to reduce the loss amount.

  For example, when the ratio of the conduction loss to the total loss is high, the switching pulse modulation degree Da is controlled. For example, when the ratio of the switching loss to the total loss is high, the output voltage value VDC, the switching frequency fc, the turn-on time Ton, and the turn-off time Toff are controlled. For example, when the ratio of the iron loss to the total loss is high, the frequency fa and the switching frequency fc of the switching pulse modulation degree Da are controlled. Further, since the copper loss is almost uniquely determined by the motor current Ia, when the ratio of the copper loss to the total loss is high, the control element that controls the loss having the next highest ratio to the total loss is controlled. .

  A loss with a large loss amount has a large loss reduction effect obtained by controlling a control element that fluctuates the loss. Therefore, the effect of reducing the total loss can be sufficiently obtained.

  As described above, according to the motor drive device of the third embodiment, the conduction loss, the switching loss, the copper loss, and the iron loss are calculated by each calculation method described in the first embodiment, and the loss is further increased. Since a large amount is extracted and the control element that fluctuates the loss is controlled, the time until the control of the control element is completed can be shortened. In addition, since a loss that is a high proportion of the total loss has a large loss reduction effect obtained by controlling the control element that fluctuates the loss, the reduction effect of the total loss can be sufficiently obtained. It is possible to reduce the total amount of loss that occurs until the process is completed.

  In the third embodiment described above, a loss having a large loss amount other than the copper loss is extracted, and the control element that fluctuates the loss is controlled. However, the loss extracted at this time is the largest other than the copper loss. Only the loss with a large loss amount may be extracted, or the top two losses may be extracted. For example, when the top two losses are conduction loss and iron loss, the control elements to be controlled are limited to the switching pulse modulation degree Da, the frequency fa of the switching pulse modulation degree Da, and the switching frequency fc. Compared with the case of controlling the elements, it is possible to expect a shortening time until the control of the control element is completed and a reduction effect of the total loss generated until the control of the control element is completed.

Embodiment 4 FIG.
FIG. 7 is a diagram of a configuration example of the electric motor drive device according to the fourth embodiment. In addition, the same code | symbol is attached | subjected to the component which is the same as that of Embodiment 1 thru | or 3, or equivalent, and the detailed description is abbreviate | omitted.

  As shown in FIG. 7, in the electric motor drive device 1 according to the fourth embodiment, the diode 6c of the inverter circuit 6 of the electric motor drive device 1 according to the first embodiment shown in FIG. 6d is replaced with a diode made of a wide band gap semiconductor, for example, Schottky barrier diodes 6cw and 6dw made of silicon carbide (SiC).

  For example, in the electric motor drive device 1 according to the first to third embodiments shown in FIG. 1, when the semiconductor switch element 6b is suddenly turned on when a current flows through the diode 6c of the inverter circuit 6, or the diode 6d If the semiconductor switch element 6a is suddenly turned on while a current is flowing, a reverse recovery loss amount Err is generated when the current flowing through the diodes 6c and 6d is commutated to the semiconductor switch elements 6a and 6b.

  In the first embodiment, when the semiconductor switch elements 6a and 6b are turned on, in order to further reduce the turn-on loss Eon generated in the inverter circuit 6, from the equation (9) described in the first embodiment, Although it is desirable to shorten the turn-on time Ton output from the drive pulse signal generation unit 14, generally, when the time change rate dI / dt of the current flowing through the diode increases, the peak value of the reverse recovery current of the diode increases. The reverse recovery loss amount Err also increases. That is, when the turn-on time Ton is shortened, the reverse recovery loss amount Err is generated and added to the switching loss, so that the loss reduction effect cannot be sufficiently obtained. For this reason, the control range of the turn-on time Ton is limited to the reverse recovery time of the diode. Further, if the turn-on time Ton is too short, the peak value of the reverse recovery current becomes excessive, which is not preferable.

  In the fourth embodiment, in order to further reduce the reverse recovery loss amount Err, the diodes 6cw and 6dw constituting the inverter circuit 61 are configured by wide band gap (WBG) type semiconductors (for example, SiC) having a short reverse recovery time. The arranged Schottky barrier diode is arranged. By reducing the reverse recovery time, the turn-on time Ton can be further shortened, and the switching loss can be further reduced. Further, the reverse recovery loss amount Err becomes relatively small, and the influence on the switching loss can be further reduced.

  WBG semiconductor is a general term for semiconductors with a large energy bandwidth compared to silicon (Si) semiconductors that are widely used as semiconductors for inverter circuits and converter circuits in electrical equipment for industrial and home appliances. There are gallium nitride (GaN) -based materials, silicon carbide (SiC), diamond, and the like as power semiconductors. A WBG semiconductor has a characteristic that the accumulation of reverse recovery charge is extremely small compared to a Si-based semiconductor. Therefore, when a diode composed of a WBG semiconductor is used, the reverse of the diode when the semiconductor switch element is turned on. The occurrence of reverse recovery current and vibration current due to the recovery operation is extremely small, and the reverse recovery loss amount Err associated therewith is also extremely small.

  FIG. 8 is a schematic diagram illustrating an example of fluctuations in the collector current and the collector-emitter voltage in a transient state when the semiconductor switch element is turned on, and fluctuations in the current flowing through the diode and the voltage across the both ends. In the example shown in FIG. 8, the reverse recovery operation of the diode is schematically taken into consideration. When the turn-on time Ton is changed by the circuit element 23a being changed by the turn-on time setting circuit 21 shown in FIG. 5 described in the first embodiment, the time change rate of the collector current Ic and the current If flowing through the diode, that is, dI / dt Also changes.

  In general, when the turn-on time Ton of the semiconductor switch element is shortened and the time change rate dI / dt of the collector current Ic and the current If flowing through the diode is increased, the peak value of the reverse recovery current flowing through the diode is also increased. ing. As a result, not only the reverse recovery loss amount Err of the diode increases, but further shortening the turn-on time Ton is not preferable because the peak value of the reverse recovery current becomes excessive and a large amount of energy is given to the diode 6d.

  In the present embodiment, the diodes 6cw and 6dw of the inverter circuit 61 are made of WBG semiconductor, so that the reverse recovery loss amount Err itself generated in the diodes 6cw and 6dw is reduced, and the reverse of the diodes 6cw and 6dw. Since the recovery charge accumulation is extremely small, even if the turn-on time Ton is shortened and the time change rate dI / dt of the current If flowing through the diode is increased, the reverse recovery loss Err does not increase remarkably. The peak value of the recovery current does not become excessive. For this reason, the control range of the turn-on time Ton can be expanded in the direction in which the turn-on time Ton becomes smaller, the loss amount Eon at the turn-on time of the semiconductor switch elements 6a and 6b can be made smaller, and reverse recovery is possible. Since the loss amount Err also does not increase significantly, the effect of reducing the switching loss including the reverse recovery loss amount Err is great.

  The diodes 6cw and 6dw constituting the inverter circuit 61 may be fast recovery type diodes having a short reverse recovery time, but are more preferably constituted by the WBG semiconductor described in the present embodiment.

  As described above, according to the motor drive device of the fourth embodiment, since the diode of the inverter circuit is configured by the wide band gap semiconductor, in addition to the amount of reverse recovery loss generated by the diode being reduced, The control range of the turn-on time of the semiconductor switch element can be expanded. As a result, the turn-on time can be shortened to reduce the loss amount at turn-on, and further, the reverse recovery loss amount does not increase, so that the effect of reducing the switching loss generated in the inverter circuit can be further increased. it can.

Embodiment 5 FIG.
FIG. 9 is a diagram of a configuration example of the electric motor drive device according to the fifth embodiment. In addition, the same code | symbol is attached | subjected to the component which is the same as that of Embodiment 1 thru | or 4, or equivalent, and the detailed description is abbreviate | omitted.

  As shown in FIG. 9, in the electric motor drive device 1 according to the fifth embodiment, the semiconductor switch element constituting the inverter circuit 62 is a semiconductor of the inverter circuit 6 of the electric motor drive device 1 according to the first embodiment shown in FIG. 1. The switch elements 6a and 6b are replaced with semiconductor switch elements (for example, metal oxide semiconductor field effect transistors configured with silicon carbide (SiC)) 6aw and 6bw configured with the WBG semiconductor described in the fourth embodiment.

  In the fourth embodiment, in addition to reducing the reverse recovery loss amount Err generated in the diodes 6cw and 6dw of the inverter circuit 61, the control range of the turn-on time Ton is expanded in a direction in which the turn-on time Ton becomes smaller. Accordingly, the loss amount Eon at turn-on is also reduced, but in the fifth embodiment, in order to further reduce the loss amount generated in the inverter circuit 62, the semiconductor switch element that constitutes the inverter circuit 62 is configured by a WBG semiconductor. Semiconductor switch elements 6aw and 6bw are arranged.

  Since the semiconductor switch elements 6aw and 6bw made of the WBG semiconductor have much less charge accumulation than the semiconductor switch elements made of a narrow band gap semiconductor such as a Si-based semiconductor, the turn-on time Ton and the turn-off time Toff The control range can be expanded in the direction of shortening the turn-on time Ton and the turn-off time Toff. Therefore, from the equations (9) and (10) described in the first embodiment, it is possible to reduce both the loss amount Eon at the turn-on time and the loss amount Eoff at the turn-off time of the semiconductor switch elements 6aw and 6bw.

  Further, by reducing the turn-on time Ton and the turn-off time Toff to reduce the turn-on loss Eon and the turn-off loss Eoff, even if the switching frequency fc of the semiconductor switch elements 6aw and 6bw is increased, switching is performed. The switching loss proportional to the frequency fc does not increase significantly. On the other hand, by increasing the switching frequency fc, the hysteresis loss included in the iron loss of the electric motor 3 can be further reduced from the equation (12) described in the first embodiment.

  Furthermore, as described with reference to FIG. 3 in the first embodiment, normally, in the control of the inverter circuit 6, the turn-on of the semiconductor switch element 6a of the upper arm and the turn-off of the semiconductor switch element 6b of the lower arm are simultaneously performed. The semiconductor switch element 6a is turned off and the semiconductor switch element 6b is turned on at the same time. When the turn-on time Ton and the turn-off time Toff are provided, the turn-on of one semiconductor switch element and the turn-off of the other semiconductor switch element are performed. If both are simultaneously performed, a period in which both semiconductor switch elements are simultaneously turned on occurs, the voltage across the smoothing circuit 5 is short-circuited, and an excessive current flows through both semiconductor switch elements. In order to prevent an excessive current accompanying the turn-on and turn-off of the semiconductor switch, a dead time Td is usually provided between the turn-off period and the turn-on period so that a period in which both semiconductor switch elements are simultaneously turned on does not occur. To do.

  By providing this dead time Td, the substantial ratio of the ON period of the semiconductor switch element in the switching cycle Tc is the dead time Td corresponding to the direction of the motor current Ia (Ia> 0 or Ia <0). Due to the fluctuation, an error occurs in the voltage output from the inverter circuit to the electric motor. In addition, since the ripple current generated by the switching operation is shifted by Td, the hysteresis loss included in the iron loss of the motor is also increased. Various methods have been disclosed for correcting an error corresponding to the dead time Td of the output voltage of the inverter circuit by providing the dead time Td. However, for correcting a deviation of the ripple current due to the dead time Td, An effective method is not disclosed.

  In the present embodiment, the inverter circuit 62 is composed of semiconductor switch elements 6aw and 6bw made of WBG semiconductor, and the turn-on time Ton and the turn-off time Toff are shortened, thereby reducing the dead time Td of the semiconductor switch elements 6aw and 6bw. It can be made smaller. As a result, the deviation of the dead time Td generated in the ripple current is reduced, and an increase in hysteresis loss due to the dead time Td can be suppressed.

  As described above, according to the motor drive device of the fifth embodiment, the semiconductor switch element of the inverter circuit is configured with a wide band gap semiconductor, so that the control range of the turn-on time and the turn-off time of the semiconductor switch element is The turn-on time and the turn-off time can be expanded in a shorter direction. As a result, it is possible to reduce both the amount of loss when the semiconductor switch element is turned on and when it is turned off, and the effect of reducing the switching loss generated in the inverter circuit can be further increased.

  Further, since the switching frequency can be increased, it contributes to the reduction of hysteresis loss included in the iron loss of the electric motor.

  Furthermore, since the turn-on time and the turn-off time can be further shortened, the dead time provided between the turn-off period and the turn-on period of the semiconductor element can be reduced, and an increase in hysteresis loss due to the dead time is suppressed. be able to.

Embodiment 6 FIG.
In the sixth embodiment, an example in which the electric motor drive device of the first to fifth embodiments is applied to an air conditioner will be described. FIG. 10 is a diagram illustrating a configuration example of an air-conditioning apparatus according to the sixth embodiment. Note that the internal configuration of the electric motor drive device 1 is the same as the configuration described in the first to fifth embodiments, and thus the description thereof is omitted here.

  As shown in FIG. 10, the air conditioner 30 according to the sixth embodiment includes an outdoor unit 31 and an indoor unit 32, and performs heat exchange between the outdoor unit 31 and the indoor unit 32. The heat exchange medium pipes 33 and 34 for circulating the heat exchange medium are connected. The outdoor unit 31 includes an electric motor driving device 1 and an electric motor 3 that is a compressor, and AC power is supplied from an external AC power source 2.

  In general, in an air conditioner, the frequency and mechanical output of the electric motor are not constant, the air temperature around the outdoor unit, the air temperature around the indoor unit, the setting of the operation mode of the air conditioner, or the air temperature that the indoor unit should harmonize with Varies depending on the set value.

  That is, by controlling the command value of the frequency or output torque of the motor based on the operation command from the outside, the change of the air temperature of the outdoor unit and the indoor unit, or the change of the operation condition such as the temperature and pressure of the heat exchange medium The motor current fluctuates, and the conduction loss and switching loss generated in the semiconductor switch elements and diodes of the inverter circuit, the copper loss generated in the primary winding of the motor, and the iron loss generated in the iron core of the motor also fluctuated.

  Therefore, in the air conditioner 30 according to the sixth embodiment, by applying the electric motor drive device 1 according to the first to fifth embodiments, even when the operation condition of the air conditioner 30 is changed, the power loss is appropriately reduced. To be.

  For example, when the motor drive device 1 according to the first embodiment is applied to the air conditioner 30, the control element control unit 11 controls the command value of the frequency of the motor 3 or the output torque based on the operation command from the outside. Based on the motor current Ia detected by the current detection circuit 7 when the motor current Ia fluctuates due to a change in the air temperature of the outdoor unit 31 and the indoor unit 32 or a change in the temperature or pressure of the heat exchange medium. According to the calculation methods described in the first embodiment, the conduction loss caused by the conduction of the semiconductor switch elements 6a and 6b and the diodes 6c and 6d of the inverter circuit 6, the turn-on time and the turn-off time of the semiconductor switch elements 6a and 6b of the inverter circuit 6 Switching loss caused by switching, copper loss generated in the primary winding of the motor 3, and motor 3 The iron loss generated in the iron core is calculated, and the output voltage value VDC of the converter circuit 6, the switching pulse modulation degree Da of the inverter circuit 6, the frequency fa of the switching pulse modulation degree Da, the switching frequency so that the total loss becomes smaller. By controlling fc, turn-on time Ton, and turn-off time Toff, the total value of each power loss generated in the inverter circuit and the electric motor is further reduced.

  For example, when the electric motor drive device 1 according to the second embodiment is applied to the air conditioner 30, the control element control unit 11 sets the temperature of the inverter circuit 6 in advance in addition to the control of the first embodiment. When the set upper limit temperature is exceeded, the output voltage value VDC of the converter circuit 6, the switching pulse modulation degree Da, the switching frequency fc, the turn-on time Ton of the inverter circuit 6 so that the sum of the conduction loss and the switching loss becomes smaller. By controlling the turn-off time Toff, the temperature rise of the inverter circuit 6 is suppressed and the inverter circuit 6 is prevented from being overheated.

  For example, when the motor drive device 1 according to the third embodiment is applied to the air conditioner 30, the control element control unit 11 performs conduction loss and switching loss by the calculation methods described in the first embodiment. By calculating the copper loss and iron loss, extracting the loss with a large amount of loss and controlling the control element that fluctuates the loss, the time to complete control of the control element is shortened and the control element The total amount of loss that occurs until the control is completed is reduced.

  For example, when the motor drive device 1 according to the fourth embodiment is applied to the air conditioner 30, the diodes 6cw and 6dw of the inverter circuit 61 are formed of WBG semiconductors, so that the diodes 6cw and 6dw are generated. The reverse recovery loss amount Err is reduced and the turn-on time Ton of the semiconductor switch elements 6a and 6b is shortened to reduce the loss amount at the turn-on time, thereby further increasing the effect of reducing the switching loss generated in the inverter circuit 61.

  For example, when the motor drive device 1 according to the fifth embodiment is applied to the air conditioner 30, the semiconductor switch elements 6 aw and 6 bw of the inverter circuit 62 are configured by WBG semiconductors, thereby providing the semiconductor switch element 6 aw. , 6 bw turn-on time and turn-off time are shortened to reduce both the loss amount at the turn-on time and the turn-off time, and the effect of reducing the switching loss generated in the inverter circuit 62 is further increased. In addition, the dead time Td provided between the turn-off period and the turn-on period of the semiconductor elements 6aw and 6bw is reduced to suppress an increase in hysteresis loss due to the dead time.

  As described above, according to the air conditioner of the sixth embodiment, by applying the motor driving device of the first to fifth embodiments, the motor frequency or the output torque command based on the operation command from the outside. Even if the motor current fluctuates due to control of values, changes in the air temperature around the outdoor unit and the indoor unit, or changes in operating conditions such as changes in the temperature and pressure of the heat exchange medium, the above-described first to first embodiments 5 can be obtained, and the power loss can be appropriately reduced.

  It should be noted that the effect obtained by using the diode or the semiconductor switch element formed of the WBG semiconductor described in the fourth embodiment or the fifth embodiment is not limited to the effect described in each embodiment.

  For example, a diode or a semiconductor switch element made of a WBG semiconductor has a high voltage resistance and a high allowable current density, so that the diode and the semiconductor switch element can be miniaturized. By using elements, it is possible to reduce the size of an inverter circuit incorporating these elements.

  In addition, since the diode and the semiconductor switching element made of the WBG semiconductor have high heat resistance, the upper limit temperature of the inverter circuit described in Embodiment 2 can be set to a higher temperature.

  Note that the configuration shown in the above embodiment is an example of the configuration of the present invention, and can be combined with another known technique, and a part thereof is omitted without departing from the gist of the present invention. Needless to say, it is possible to change the configuration.

DESCRIPTION OF SYMBOLS 1 Motor drive device 2 AC power supply 3 Electric motor 4 Converter circuit 5 Smoothing circuit 5a Smoothing capacitor 6, 61, 62 Inverter circuit 6a, 6b Semiconductor switch element 6aw, 6bw Semiconductor switch element 6c, 6d diode comprised by wide band gap semiconductor 6 cw, 6 dw Diodes configured with wide band gap semiconductors 7 Current detection circuit 10 Motor drive control unit 11 Control element control unit 12 DC voltage command generation unit 13 Switching frequency command generation unit 14 Drive pulse signal generation unit 15 Switching time setting unit 20a , 20b Drive circuit 21 Turn-on time setting circuit 22 Turn-off time setting circuit 23a, 23b Circuit element 30 Air conditioner 31 Outdoor unit 32 Indoor unit 33, 34 Heat exchange medium piping

Claims (11)

A first AC voltage is supplied from an AC power source, the first AC voltage is converted into a second AC voltage, and the motor drive device outputs the electric voltage to the motor;
A converter circuit for converting the first AC voltage into a DC voltage;
A smoothing circuit for smoothing the DC voltage;
An inverter circuit configured to include one or a plurality of upper and lower arms to which a semiconductor switch element and a diode are connected in anti-parallel, and that converts the DC voltage into the second AC voltage;
A drive circuit for driving the semiconductor switch element;
A current detection circuit for detecting a motor current flowing in the motor;
Based on the motor current, switching loss caused by switching of the semiconductor switch element at turn-on or turn-off, conduction loss caused by conduction of the semiconductor switch element and the diode, iron loss generated in the iron core of the motor, and A motor drive that calculates a copper loss occurring in the primary winding of the motor, calculates a total loss of the switching loss, the conduction loss, the iron loss, and the copper loss, and controls the total loss to be smaller A control unit;
With
The electric motor drive control unit includes switching speed control means for controlling at least one of a turn-on time and a turn-off time of the semiconductor switch element ,
The drive circuit is
A first circuit element that limits a gate current that flows to the semiconductor switch element at the time of a rise of a drive pulse signal output from the electric motor drive control unit;
A second circuit element for limiting a gate current flowing in the semiconductor switch element at the fall of the drive pulse signal;
A turn-on time setting circuit for setting the turn-on time by controlling a first resistance element included in the first circuit element;
A turn-off time setting circuit for setting the turn-off time by controlling a second resistance element included in the second circuit element;
Electric motor drive device according to claim Rukoto equipped with.   The first circuit element is:
  A first rectifying element connected in a forward direction with respect to a gate current flowing through the semiconductor switch element when the drive pulse signal is turned on in series with the first resistance element;
  The second circuit element is:
  A second rectifying element connected in a forward direction with respect to a gate current flowing through the semiconductor switch element when the drive pulse signal is turned off in series with the second resistance element;
  The turn-on time setting circuit includes:
  Based on the turn-on time command value output from the electric motor drive control unit, by changing the first resistance element, the rise time of the drive pulse signal is adjusted to set the turn-on time,
  The turn-off time setting circuit includes:
  Based on the turn-off time command value output from the motor drive control unit, the turn-off time is set by adjusting the fall time of the drive pulse signal by changing the second resistance element.
  The electric motor drive device according to claim 1. Said motor drive control unit, the motor driving apparatus according to claim 1 or 2, further comprising a switching frequency control means for controlling the switching frequency of the semiconductor switching element. Said motor drive control unit, the electric motor drive device according to any one of claims 1 to 3, further comprising a converter circuit output voltage control means for controlling an output voltage value of said converter circuit. It said motor drive control unit, the electric motor drive device according to claim 1, further comprising a switching pulse modulation degree control means for controlling the switching pulse modulation to any one of 4 of the semiconductor switching element. Said motor drive control unit, the calculated average value over one period of the second AC voltage, motor drive according to any one of claims 1 to 5, characterized in that for calculating the conduction loss apparatus. When the temperature of the inverter circuit exceeds a preset upper limit temperature of the inverter circuit, the motor drive control unit calculates a total loss of the conduction loss and the switching loss, and the total loss is smaller. It controls so that it may become. The electric motor drive device as described in any one of Claim 1 to 6 characterized by the above-mentioned. The diode, motor driving device according to any one of claims 1 to 7, characterized in that it is formed by a wide band gap semiconductor. It said semiconductor switching element, motor driving device according to any one of claims 1 8, characterized in that it is formed by a wide band gap semiconductor.   10. The electric motor driving device according to claim 8, wherein the wide band gap semiconductor is a gallium nitride-based material, silicon carbide, or diamond. An air conditioner comprising the electric motor drive device according to any one of claims 1 to 10.

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