JP5460548B2 - DC-DC converter and control method thereof - Google Patents

Description

  The present invention relates to a DC-DC converter and a control method thereof, and more particularly to a DC-DC converter and a control method thereof capable of appropriately executing a low-loss one-side drive type V2 control.

In recent years, HEVs (Hybrid Electric Vehicles) having an electrically driven motor as one of the power sources are becoming popular.
Such a HEV is equipped with a PDU (Power Drive Unit) system in order to supply electric power to the motor for driving.

In many cases, a PDU system is provided with a DC (Direct Current) -DC converter (see Patent Documents 1 and 2).
In the DC-DC converter, a battery is connected to the input side (primary side), and an inverter for driving the motor is connected to the output side (secondary side) as a load.
The DC-DC converter can execute an operation (hereinafter, referred to as “power running operation”) that causes the load to power the power by boosting the secondary side voltage and discharging the battery. Conversely, the DC-DC converter can perform an operation of charging the battery (hereinafter referred to as “regenerative operation”) by stepping down the secondary side voltage and regenerating power from the load.
Note that a DC-DC converter capable of supplying power in both directions on the primary side and the secondary side, that is, a power running operation and a regenerative operation in this way is hereinafter referred to as a “bidirectional DC-DC converter”.

Such control for increasing or decreasing the secondary voltage V2 of the bidirectional DC-DC converter is referred to as “V2 control”.
V2 control is realized by controlling each drive of two switching elements connected in series, and there are two types of methods in which the control of each drive of these two switching elements is different from each other, that is, one-side drive method and Broadly divided into complementary drive systems.
In the one-side drive method, generally, only one of the two switching elements is driven by a power running operation (the other prohibits driving), and only the other is driven by a regenerative operation (one prohibits driving). Refers to the method.
The complementary driving method is a method of driving two switching elements in both the power running operation and the regenerative operation. The complementary driving method is widely used because V2 control is easier than the one-side driving method.

JP 2009-33878 A JP 2008-83648 A

However, in the V2 control, when the power decreases due to a light load or the like, the circuit current of the bidirectional DC-DC converter may reach 0 [A]. The V2 control in such a case is hereinafter referred to as “V2 control in a low power range”.
In the V2 control in the low power range by the complementary driving method, the circuit current continues to increase or decrease as it increases or decreases and reaches 0 [A]. For this reason, the circuit current flows backward, but such a backward flowing circuit current is an excess current for the V2 control, and the loss increases accordingly.

On the other hand, in the V2 control in the low power range by the conventional one-side drive method, when the circuit current increases or decreases and reaches 0 [A], 0 [A] is maintained as it is. Accordingly, in the one-side driving method, the circuit current does not flow backward unlike the complementary driving method, and thus the loss is reduced accordingly.
For this reason, it is preferable to adopt a low-loss one-side drive method as the V2 control of the bidirectional DC-DC converter. However, if the general handling similar to the complementary drive method is performed, appropriate V2 control cannot be performed. A technique capable of realizing V2 control is required.

  The present invention has been made in view of such a situation, and provides a bidirectional DC-DC converter and a control method thereof capable of appropriately executing low-loss single-side drive V2 control. Objective.

The DC-DC converter of the present invention (for example, the bidirectional DC-DC converter 12 in the embodiment)
A DC power source (for example, the battery 11 in the embodiment) is connected to the primary side, and a load (for example, the load 13 including the motor 22 in the embodiment) is connected to the secondary side. The first switching element (for example, the switching element SL in the embodiment) ) And a second switching element (for example, the switching element SH in the embodiment) having a series connection (for example, the power supply circuit section 51 in the embodiment),
A power running operation for powering the load from the DC power source to the load by controlling driving of the first switching element and the second switching element and maintaining a secondary voltage at an arbitrary value, or the load A control unit (for example, the control unit 52 in the embodiment) for realizing a regenerative operation for regenerating power from the DC power source to
A DC-DC converter comprising:
As drive modes of the first switching element and the second switching element,
A first drive mode that maintains the secondary voltage at an arbitrary value by driving the first switching element with a pulse of the first DUTY and prohibiting the driving of the second switching element (for example, SL driving in the embodiment) Mode) and
A second drive mode for maintaining the secondary side voltage at an arbitrary value by prohibiting the drive of the first switching element and driving the second switching element with a pulse of the second DUTY (for example, SH drive in the embodiment) Mode) and
Have
The controller is
Generating the first DUTY command (for example, BOOSTDUTY ^{* in the} embodiment) and the second DUTY command (for example, BUCKDUTY ^{* in the} embodiment);
In the power running operation, the first drive mode is executed when the first DUTY command is 0 or more, and the second drive mode is executed when the first DUTY command is less than 0,
In the regenerative operation, when the second DUTY command is 0 or more, the second drive mode is executed, and when the second DUTY command is less than 0, the first drive mode is executed.
It is characterized by that.

According to the present invention, in the power running operation, when the first DUTY command is lowered due to load fluctuation or the like, the first DUTY command is executed in the first drive mode when the first DUTY command is 0 or more, and the first DUTY command reaches 0. However, if the operation of the first DUTY becomes less than 0 without stopping the operation, the operation is executed in the second drive mode.
Similarly, in the regenerative operation, when the second DUTY command is lowered due to load fluctuation or the like, it is executed in the second drive mode when the second DUTY command is 0 or more, and the operation is performed even if the second DUTY command reaches 0. If the second DUTY command is less than 0 without stopping, the first drive mode is executed.
In this way, it is possible to appropriately execute the V2 control of the one-side drive method.
Specifically, as will be described later with reference to FIG. 9, since the V2 control of the one-side drive method can be appropriately executed, an unnecessary ripple voltage is suppressed and a stable secondary voltage V can be obtained. become.

in this case,
The controller is
Further, it is determined that the case where the load current is 0 or more is the case of the power running operation, and the case where the load current is less than 0 is determined as the case of the regenerative operation,
When the load current is 0 or more,
When the first DUTY command is 0 or more, as the first drive mode, the first switching element is driven using the first DUTY command, and the driving of the second switching element is prohibited,
When the first DUTY command is less than 0, as the second drive mode, the driving of the first switching element is prohibited and the sign inversion value of the first DUTUY command is used as the second DUTY command. 2 Drive the switching element,
When the load current is less than 0,
When the second DUTY command is equal to or greater than 0, as the second drive mode, the driving of the first switching element is prohibited, and the second switching element is driven using the second DUTY command,
When the second DUTY command is less than 0, as the first drive mode, the first switching element is driven using the sign inversion value of the second DUTUY command as the first DUTY command, and the second switching Prohibit driving of the element,
You may do it.

  The DC-DC converter control method of the present invention is a control method for the above-described DC-DC converter of the present invention. Therefore, it is possible to achieve the same effect as the above-described DC-DC converter of the semiconductor element of the present invention.

  According to the present invention, it is possible to appropriately execute V2 control of a low-loss one-side drive method.

It is a figure which shows schematic structure of one Embodiment of the vehicle-mounted PDU system containing the bidirectional | two-way DC-DC converter which concerns on this invention. It is a figure which shows schematic structure of the common vehicle-mounted PDU system containing a general bidirectional DC-DC converter. It is a figure which shows the comparison of the complementary drive system in a bidirectional | two-way DC-DC converter, and a single side drive system. It is a figure which shows an example of the current waveform of a reactor in case V2 control is performed by each of a complementary drive system and the conventional one side drive system. It is a figure explaining the circuit operation | movement of a bidirectional | two-way DC-DC converter in case V2 control in a low electric power area is performed by each of a complementary drive system and the conventional one side drive system. It is a figure explaining DUTY applied as a feedforward term by V2 control of a continuous region. It is a figure explaining DUTY applied as a feedforward term by V2 control of the low electric power area in power running operation. It is a figure explaining DUTY applied as a feedforward term by V2 control of the low electric power area in regeneration operation. It is a figure which shows the comparison with the conventional method in which the tracking delay subject generate | occur | produces as a method of V2 control in the low electric power area by the one side drive system, and the switching element drive mode conversion method of this embodiment. It is a block diagram which shows the detailed structure of the bidirectional | two-way DC-DC converter of FIG. It is a flowchart explaining the flow of the V2 control process which the control part of the bidirectional | two-way DC-DC converter of FIG. 10 performs. 1 is a diagram showing a schematic configuration of an embodiment of an in-vehicle PDU system including a DC-DC converter according to the present invention, which is different from FIG.

  Hereinafter, embodiments of the present invention will be described with reference to the drawings.

  FIG. 1 is a diagram showing a schematic configuration of an embodiment of an in-vehicle PDU system including a DC-DC converter according to the present invention.

  The in-vehicle PDU system includes a battery 11, a bidirectional DC-DC converter 12, and a load 13. The in-vehicle PDU system is mounted on, for example, an HEV.

  The battery 11 is a DC power source that can be charged and discharged.

In the bidirectional DC-DC converter 12, the primary side (input side) is connected to the battery 11, and the secondary side (output side) is connected to the load 13.
The bidirectional DC-DC converter 12 can perform an operation of causing the load 13 to power, that is, a power running operation, by boosting the secondary side voltage V2 and discharging the battery 11. Conversely, the bidirectional DC-DC converter 12 can perform an operation of charging the battery 11, that is, a regenerative operation, by reducing the secondary voltage V <b> 2 and regenerating power from the load 13.

The load 13 includes an inverter 21, a motor 22, and a tire 23.
The inverter 21 drives the motor 22 by converting the direct current output (secondary voltage V <b> 2) of the bidirectional DC-DC converter 12 into alternating current and supplying the alternating current to the motor 22.
That is, the motor 22 is a synchronous motor driven by, for example, alternating current, and is driven by the alternating current output of the inverter 21.
The tire 23 rotates with the rotation of the shaft of the motor 22 and advances HEV (not shown).

  Here, in order to facilitate understanding of the present invention, a circuit configuration of a general bidirectional DC-DC converter and an outline of a V2 control method will be described with reference to FIGS. .

  FIG. 2 is a diagram showing a schematic configuration of a general in-vehicle PDU system including a general bidirectional DC-DC converter.

  A general in-vehicle PDU system includes a battery 31, a general bidirectional DC-DC converter 32, and a load 33.

  Since each of the battery 31 and the load 33 has basically the same configuration and function as each of the battery 11 and the load 13 in FIG. 1, description thereof is omitted here.

  The general bidirectional DC-DC converter 32 includes a power supply circuit unit 41 and a CPU (Central Processing Unit) 42.

The power supply circuit unit 41 includes switching elements SH and SL, capacitors C1 and C2, a reactor L, and resistors R1 to R4.
The switching elements SH and SL are connected in series, and a load 33 is connected to both ends of the series connection. A capacitor C2 and a series connection of resistors R3 and R4 are also connected to both ends of the series connection.
The positive end of the battery 31 is connected to the connection point of the switching elements SH and SL via the reactor L. The negative end of the battery 31 is connected to the end on the switching element SL side of both ends of the series connection of the switching elements SH and SL. A capacitor C1 and a series connection of resistors R1 and R2 are connected between the positive electrode end and the negative electrode end of the battery 31, respectively.

The CPU 42 performs V2 control, that is, variable control of the secondary side voltage V2, which is the output voltage of the power supply circuit unit 41.
There are roughly two types of V2 control methods, a complementary drive method and a one-side drive method adopted in the present invention.
Hereinafter, the complementary driving method and the one-side driving method will be described while clarifying the differences.

  FIG. 3 is a diagram showing a comparison between a complementary driving method and a one-side driving method in a bidirectional DC-DC converter.

As shown in FIG. 3, a PWM (Pulse Width Modulation) driving method is employed as a driving method for the switching elements SH and SL of the power supply circuit unit 41.
That is, each of the switching elements SH and SL is configured by a pair of a semiconductor element such as an IGBT (Insulated Gate Bipolar Transistor) and a FWD (Free Wheeling Diode).
The switching elements SH and SL having such a configuration can be turned off and on by a supplied drive signal. Here, the off-switching means that the switching elements SH and SL are switched from a conduction state (also referred to as “on state”) to a cutoff state (also referred to as “off state”). On-switching means that the switching elements SH and SL are switched from a cut-off state (off state) to a conduction state (on state).
In this case, each drive signal of the switching elements SH and SL is given as a pulse signal, and on switching is performed when the pulse signal is switched to a high level, and off switching is performed when the pulse signal is switched to a low level. .
A method for driving the switching elements SH and SL by supplying such a pulse signal to the switching elements SH and SL while changing the DUTY of the pulse width is a PWM driving method.
Hereinafter, in conjunction with the description of FIG. 3, the drive signal supplied to the switching element SH by the PWM drive method is referred to as “H pulse”, and the drive signal supplied to the switching element SL is referred to as “L pulse”.

As shown in FIG. 3, one of the differences between the complementary drive method and the one-side drive method is that the PWM drive method is different.
That is, in the complementary drive method, both the H pulse and the L pulse are output in the inverted state regardless of whether the operation is a power running operation or a regenerative operation. In other words, in the complementary drive system, each of the switching elements SH and SL keeps the relationship that when one of the switching elements SH and SL is on, the other is off. Both are driven.
On the other hand, in the conventional one-side drive method, only the L pulse is output in the power running operation (output of the H pulse is prohibited), while only the H pulse is output in the regenerative operation. Output (L pulse output is prohibited). That is, in the conventional one-side drive system, in the case of a power running operation, only the switching element SL is driven, and the switching element SH is prohibited from being driven. On the other hand, in the regenerative operation, only the switching element SH is driven, and the driving of the switching element SL is prohibited.
Note that the reason for specifying “conventional one-side drive method” is that, in the one-side drive method (switching element drive mode conversion method described later) of the present embodiment, even when a predetermined condition is satisfied even during powering operation, This is because the switching element SH is driven. Similarly, in the one-side drive method of the present embodiment, the switching element SL is driven when a predetermined condition is satisfied even during the regenerative operation.

  As shown in FIG. 3, the complementary driving method does not require an explicit instruction to switch between the power running operation and the regenerative operation. On the other hand, in the one-side drive method, such an explicit instruction is necessary. This is also one of the differences between the complementary drive method and the one-side drive method.

FIG. 4 shows an example of the waveform of the reactor L current (hereinafter referred to as “reactor current”) when V2 control is executed by each of the complementary driving method and the (conventional) one-side driving method. Show.
In FIG. 4, the vertical axis represents the reactor current [A] when the V2 control by the power running operation is being executed. The horizontal axis indicates time [s].
In FIG. 4, a waveform WC shows the time transition of the reactor current when the power is large. That is, when the power is large, the general bidirectional DC-DC converter 32 operates in a so-called continuous mode, so the time transition of the reactor current is almost the same in both the complementary driving method and the (conventional) one-side driving method. Only one waveform WC is shown in FIG.
However, when the power decreases, the minimum value of the reactor current reaches 0 [A]. Then, when the reactor current further decreases, thereafter, different time transitions are shown in the complementary driving method and the (conventional) one-side driving method. Note that the V2 control in such a case is referred to as “V2 control in a low power range” as described above.
That is, in the case of V2 control in the low power region, the time transition of the reactor current in the complementary drive system is as shown by the waveform WL2, and there is a section where the current flows backward. On the other hand, the time transition of the reactor current in the (conventional) one-side drive method is maintained at 0 [A] without dropping below the current when the current reaches 0 [A] as in the waveform WL1.

FIG. 5 is a diagram for explaining the circuit operation of the bidirectional DC-DC converter when V2 control (powering operation) is performed in the low power range by each of the complementary drive method and the (conventional) one-side drive method. It is.
In the table of FIG. 5, the first column in the vertical direction indicates the item name, and the second column in the vertical direction indicates the current waveform and circuit operation in the case of the complementary drive method. The three columns show the current waveform and circuit operation in the case of the (conventional) one-side drive method.

  First, the circuit operation of the bidirectional DC-DC converter when the V2 control in the low power region is executed by the complementary drive method will be described using the second column in the table of FIG.

  In the first row and the second column in the table of FIG. 5, a waveform WL2 showing the time transition of the reactor current in the case of the complementary drive method is shown, and the waveform for one cycle is divided into periods T1 to T4. ing.

  In the second column of each of the second to fifth rows in the table of FIG. 5, an operation example of the power supply circuit unit 41 (FIG. 2) in each of the periods T1 to T4 in the case of the complementary driving method is shown. Has been.

In the second column of each of the second to fifth rows, it is assumed that the following preconditions are satisfied.
For the convenience of the size of the drawing, the reference numerals of the elements of the power supply circuit unit 41 are omitted as appropriate.
Each of “ON” or “OFF” shown on the right side of the switching element SH or SL indicates “on state” or “off state”.
A round arrow indicates the current flowing in that direction.
Each of switching elements SL and SH is assumed to be composed of a pair of IGBT and FWD.
These preconditions also apply to the third column of each of the second to fifth rows for the one-side drive method described later.

  In the periods T1 and T2 in the case of the complementary driving method, the switching element SH is in an off state and the switching element SL is in an on state. For this reason, the reactor current increases during the periods T1 and T2.

Specifically, immediately before the start of the period T1, with the reactor current being less than 0 [A], the switching element SH is switched from the on state to the off state, and the switching element SL is switched from the off state to the on state. Therefore, in the period T1, focusing on the switching element SL, the L pulse is supplied, but the current does not yet flow on the IGBT side, but the commutation current flows on the FWD side.
For this reason, as indicated by an arrow, the current of the power supply circuit unit 41 flows in the order of the negative electrode end of the battery 31 → the FWD side of the switching element SL → the reactor L → the positive electrode end of the battery 31, and as a result, the reactor current increases. However, it is less than 0 [A].

Thereafter, when the reactor current rises to 0 [A], the period T2 starts. In the period T2, a current flows to the IGBT side of the switching element SL by the L pulse.
For this reason, the current of the power supply circuit section 41 flows in the order of the positive electrode end of the battery 31 → the reactor L → the IGBT side of the switching element SL → the negative electrode end of the battery 31 as shown by the arrow. It continues to rise after reaching A].

After that, at the end of the period T2, the switching element SL is switched from the on state to the off state, and the switching element SH is switched from the off state to the on state.
Therefore, in the subsequent periods T3 and T4 in the case of the complementary driving method, the switching element SH is in the on state and the switching element SL is in the off state. For this reason, in the periods T3 and T4, the reactor current decreases.

Specifically, in the period T3, when attention is paid to the switching element SH, since it is immediately after switching, the H pulse is supplied, but the current does not yet flow to the IGBT side, but the commutation current is to the FWD side. It is in a flowing state.
For this reason, as indicated by the arrow, the current of the power supply circuit unit 41 flows in the order of the positive electrode end of the battery 31 → the reactor L → the FWD side of the switching element SH → the load 33 → the negative electrode end of the battery 31. Current continues to fall.

Thereafter, when the reactor current decreases to 0 [A], the period T4 starts. In the period T4, a current flows to the IGBT side of the switching element SH by the H pulse.
For this reason, the current of the power supply circuit section 41 flows in the order of the load 33 → the IGBT side of the switching element SH → the reactor L → the positive terminal of the battery 31 → the negative terminal of the battery 31 → the load 33, as shown by the arrows. The reactor current continues to decrease after reaching 0 [A].

As described above, the circuit operation of the bidirectional DC-DC converter in the case where the V2 control (power running operation) in the low power region is executed by the complementary drive method using the second column in the table of FIG. 5 has been described.
Next, using the third column in the table of FIG. 5, the circuit of the bidirectional DC-DC converter when the V2 control (powering operation) in the low power region is executed by the (conventional) one-side drive method The operation will be described.

In the first row and the third column in the table of FIG. 5, a waveform WL2 indicating the time transition of the reactor current in the case of the (conventional) one-side drive method is shown, and the waveform for one period is represented by the periods T1 to T1. It is divided into T3.
It should be noted that the periods T1 to T3 in the (conventional) one-side drive method are mutually independent periods from the periods T1 to T4 in the complementary drive method and do not coincide with each other as an absolute time.

In the third column of each of the second to fifth rows in the table of FIG. 5, the power supply circuit section 41 (FIG. 2) in each of the periods T1 to T3 in the case of the (conventional) one-side drive method. An example of operation is shown.
FIG. 5 shows an example of the power running operation. In the (conventional) one-side drive method, since the output of the H pulse is prohibited, the switching element SH is always in the off state, and only the switching element SL is caused by the L pulse. Repeats switching between the on state and the off state.

The period T1 in the case of the (conventional) one-side drive method is a period in which the switching element SL is in the ON state.
For this reason, the current of the power supply circuit section 41 flows in the order of the positive electrode end of the battery 31 → the reactor L → the IGBT side of the switching element SL → the negative electrode end of the battery 31 as shown by the arrow. A] continues to rise.

After that, at the end of the period T1, the switching element SL is switched from the on state to the off state, and the period T2 starts.
Here, when attention is paid to the switching element SH, as described above, the commutation current flows on the FWD side during the period T2 because the switching element SH is always in the OFF state.
For this reason, as indicated by the arrow, the current of the power supply circuit unit 41 flows in the order of the positive electrode end of the battery 31 → the reactor L → the FWD side of the switching element SH → the load 33 → the negative electrode end of the battery 31. Current continues to fall.

  Thereafter, when the reactor current decreases to 0 [A], the period T3 starts. In the period T3, since the switching elements SH and SL are both in the off state and the commutation current flows, the current does not flow through the power supply circuit unit 41. As a result, the reactor current maintains 0 [A].

As described above, in the V2 control in the low power range, the complementary driving method causes a larger loss because extra current flows in the period T1, the period T4, and the like. In other words, in the V2 control in the low power range, the (conventional) one-side drive method does not cause such extra current to flow, so the loss is reduced and higher efficiency can be realized. become.
Therefore, the present inventors decided to adopt the one-side drive method as the V2 control method of the bidirectional DC-DC converter 12 of FIG.

However, as can be seen from the fact that the conventional single-side drive method is applied to the bidirectional DC-DC converter 12 as it is, it can be understood from the fact that the conventional single-side drive method is referred to in the above-described description, the following first one is obtained. A problem and a second problem occur.
The first problem is that when a general arithmetic expression used in a complementary drive system as a DUTY arithmetic expression is applied to a conventional one-side drive system as it is, in V2 control in a low power region such as at light load, The problem is that the secondary voltage V2 rises or falls unnecessarily. Hereinafter, such a first problem is referred to as a “DUTY problem”.
The second problem is that in the V2 control using the conventional one-side drive method, when the load fluctuates, a follow-up delay of the feedback control employed in the V2 control occurs. Such a second problem is hereinafter referred to as a “follow-up delay problem”.

Therefore, the present inventors have invented the following method in order to solve the DUTY problem.
That is, in V2 control when the reactor current continues without reaching 0 [A] (hereinafter referred to as “continuous V2 control”), DUTY is calculated based on the boost ratio. On the other hand, in the V2 control in the low power region, the maximum value (absolute value) of the inductor current of the circuit is predicted, and DUTY is calculated based on the prediction result.
Such a method is hereinafter referred to as a “continuous region / low power region DUTY switching method”.

Furthermore, the present inventors have invented the following technique in order to solve the follow-up delay problem.
That is, as a driving mode of the switching elements SH and SL, a mode in which the switching element SH is driven (H pulse is output) and the driving of the switching element SL is prohibited (L pulse output is prohibited) is as follows. This is called “SH drive mode”. On the other hand, the mode in which the switching element SL is driven (L pulse is output) and the switching element SH is prohibited from being driven (H pulse output is prohibited) is hereinafter referred to as “SL drive mode”. .
In addition, as the command for the SH drive mode or the SL drive mode, DUTY is not used as it is, and a correction value after PID compensation is performed using the DUTY as a feedforward term is used. Therefore, hereinafter, DUTY used as a command after PID compensation is performed will be referred to as “DUTY ^* ” and specifically referred to as “command DUTY ^* ” in order to clearly distinguish it from DUTY used as a feedforward term.
In this case, in the conventional one-side drive type V2 control, in the power running operation, the SL drive mode is continued while the command DUTY ^* is positive (+), and when the command DUTY ^* becomes 0, the power running operation itself stops. It was. In the regenerative operation, the SH drive mode is continued while the command DUTY ^* is positive (+). When the command DUTY ^* becomes 0, the regenerative operation is stopped.
For this reason, in the conventional one-side drive method, a follow-up delay problem has occurred (detailed causes will be described later).
On the other hand, in the V2 control of the one-side drive method to which the technique invented by the present inventors is applied, when the command DUTY ^* is positive (+) in the power running operation, the SL drive mode (conventional one-side drive) When the command DUTY ^* is negative (-), the SH drive mode is applied. That is, when the command DUTY ^* is switched from positive (+) to negative (-), the SL drive mode is switched to the SH drive mode, and when the command DUTY ^* is switched again from negative (-) to positive (+), the SH drive mode is switched. To SL drive mode.
Further, in the V2 control of the one-side drive method to which the technique invented by the present inventors is applied, in the regenerative operation, when the command DUTY ^* is positive (+), the SH drive mode (the same as the conventional one-side drive method) Mode) is applied, and when the command DUTY ^* is negative (-), the SL drive mode is applied. That is, when the command DUTY ^* is switched from positive (+) to negative (−), the SH drive mode is switched to the SL drive mode, and when the command DUTY ^* is switched again from negative (−) to the positive (+), the SL drive mode is switched. To the SH drive mode.
Such a method invented by the present inventors is hereinafter referred to as a “switching element drive mode conversion method”.

Further, details of each of the continuous area / low power area DUTY switching method and the switching element drive mode conversion method will be described individually in that order.
First, the details of the continuous area / low power area DUTY switching method will be described with reference to FIGS.

  FIG. 6 is a diagram for explaining DUTY applied as a feedforward term in the continuous region V2 control during the power running operation.

As preconditions of FIG. 6, IL indicates a reactor current, and VL indicates a voltage of the reactor L.
V1 indicates a primary side voltage of the bidirectional DC-DC converter 12, and V2 indicates a secondary side voltage of the bidirectional DC-DC converter 12.
T indicates a switching cycle (switching cycle between an on state and an off state) for the switching element SH or SL, that is, a cycle of an H pulse or an L pulse. Ton indicates an on-state period (hereinafter referred to as “on period”), and Toff indicates an off-state period (hereinafter referred to as “off period”).
Note that the assumptions in this paragraph apply to FIGS. 7 and 8 as well.

ここで、フィードフォワード項として用いるＤＵＴＹは、１周期Ｔに対する、オン期間Ｔｏｎの割合で示される。従って、当該ＤＵＴＹは、次の式（２）で表わされる。

ここで、（Ｖ１／Ｖ２）は昇圧比を示している。従って、式（２）は、昇圧比に基づいてフィードフォワード項たるＤＵＴＹを演算する式であるといえる。そこで、以下、式（２）を、「昇圧比の式（２）」と呼ぶ。 Here, since the average voltage VL of reactor L is 0 when considered in one cycle T, the following equation (1) is established from the relationship shown in FIG.

Here, DUTY used as a feedforward term is represented by the ratio of the on period Ton to one period T. Therefore, the DUTY is expressed by the following equation (2).

Here, (V1 / V2) indicates the step-up ratio. Therefore, it can be said that the expression (2) is an expression for calculating DUTY which is a feedforward term based on the step-up ratio. Therefore, hereinafter, the equation (2) is referred to as “step-up ratio equation (2)”.

The step-up ratio equation (2) itself has been adopted as an equation for calculating DUTY as a feed-forward term not only in the one-side drive method of the present embodiment but also in the general complementary drive method V2 control.
Here, assuming that the voltage of the battery 11 is constant, the measured primary voltage is substituted for V1, and the command value of the secondary voltage (hereinafter referred to as “secondary voltage command value”) is used. By using the expression (2) of the step-up ratio substituted for V2, DUTY that is a feedforward term can be easily calculated.
As described above with reference to FIG. 4, in the continuous region V2 control, the waveform of the reactor current IL does not change in either the complementary driving method or the one-side driving method (see the waveform WC in FIG. 4). Therefore, even in the continuous area / low power area DUTY switching method of the present embodiment, in the continuous area V2 control, DUTY which is a feedforward term is calculated by the expression (2) of the step-up ratio.

However, as described above with reference to FIG. 4, in the V2 control in the low power range, the waveform of the reactor current IL in the one-side drive method (see the waveform WL1 in FIG. 4) is that in the complementary drive method (the waveform in FIG. 4). Unlike WL2, the waveform is intermittent with a period of 0 [A].
For this reason, in the one-side drive method, when the boost ratio formula (2) is used as it is in the V2 control in the low power region, the secondary side voltage V2 rises or falls unnecessarily with respect to the command value. This is the cause of the above-described DUTY problem.
Therefore, in the continuous region / low power region DUTY switching method, the inductor current (reactor current) of the circuit is not used in the V2 control in the low power region without using the step-up ratio equation (2) so that the DUTY problem does not occur. A formula based on the prediction result of the maximum value (absolute value) of IL) is used to calculate DUTY as a feedforward term.
Hereinafter, with reference to FIG. 7 and FIG. 8, a calculation formula of DUTY applied as a feedforward term in the V2 control in the low power region in the continuous region / low power region DUTY switching method will be described.

また、式（１）より、オフ期間Ｔｏｆｆは、次の式（４）で表わされる。

ここで、図７のリアクトル電流ＩＬの波形のうち斜線部分が、双方向ＤＣ−ＤＣコンバータ１２の出力の両端に接続されるコンデンサＣ２（図１０参照）に対して、充電される電流を示している。このようなコンデンサＣ２の充電電流（斜線部分）の平均と、負荷１３（図１や図１０参照）の電流とが同一値になると、２次側電圧Ｖ２が安定になる。即ち、２次側電圧Ｖ２が安定になるためには、次の式（５）が成立する必要がある。

式（５）において、Ｉｏｕｔが、負荷１３の電流を示している。
よって、式（５）に対して、式（３）及び式（４）を代入して、２次側電圧Ｖ２について解くと、次の式（６）が得られる。

FIG. 7 is a diagram for explaining DUTY applied as a feedforward term in the V2 control in the low power range in the power running operation.
The maximum value Imax of the reactor current IL shown in FIG. 7 is expressed by the following equation (3).

Further, from the equation (1), the off period Toff is expressed by the following equation (4).

Here, the hatched portion in the waveform of the reactor current IL in FIG. 7 indicates the current charged to the capacitor C2 (see FIG. 10) connected to both ends of the output of the bidirectional DC-DC converter 12. Yes. When the average of the charging current (shaded portion) of the capacitor C2 and the current of the load 13 (see FIGS. 1 and 10) have the same value, the secondary side voltage V2 becomes stable. That is, in order for the secondary side voltage V2 to become stable, the following equation (5) needs to be satisfied.

In Expression (5), Iout represents the current of the load 13.
Therefore, when Expression (3) and Expression (4) are substituted into Expression (5) and solved for the secondary side voltage V2, the following Expression (6) is obtained.

  Here, it is assumed that the step-up ratio equation (2) is used as it is in the low power V2 control in the one-side drive method. Under this assumption, the DUTY calculated as the feedforward term by the boost ratio equation (2) is increased, and as a result, the ON period Ton is also increased. In Equation (6), assuming that the secondary side voltage V2 is a variable and all other values are constants, the secondary side voltage V2 increases as the square of the on-period Ton increases. Go. This is the cause of the above-described DUTY problem.

式（７）は、式（６）におけるＶ２を、２次側電圧指令値Ｖ２^＊に置き換えた式から得られたものであり、式（６）は、上述の如くリアクトル電流ＩＬの最大値Ｉｍａｘの予測式から得られたものである。よって、式（７）は、リアクトル電流ＩＬの最大値Ｉｍａｘ、即ち回路のインダクタ電流の最大値（絶対値）の予測結果に基づいて、ＤＵＴＹが算出される式となっている。 Therefore, in the continuous area / low power area DUTY switching method, in order to prevent the DUTY problem from occurring, in the V2 control in the low power area during the power running operation, the boost ratio formula (2) is not applied, and Equation (7) is applied to calculate DUTY as a feedforward term.

Expression (7) is obtained from an expression in which V2 in Expression (6) is replaced with the secondary side voltage command value V2 ^* . As described above, Expression (6) represents the maximum value Imax of the reactor current IL. It was obtained from the prediction formula. Therefore, Expression (7) is an expression for calculating DUTY based on the prediction result of the maximum value Imax of the reactor current IL, that is, the maximum value (absolute value) of the inductor current of the circuit.

In the above, with reference to FIG. 7, the DUTY applied as a feedforward term in the V2 control in the low power range in the power running operation has been described.
Next, with reference to FIG. 8, DUTY applied as a feedforward term in the low power V2 control in the regenerative operation will be described.

  FIG. 8 is a diagram for explaining DUTY applied as a feedforward term in the V2 control in the low power range in the regenerative operation.

ここで、（Ｖ１／Ｖ２）は昇圧比を示している。従って、式（８）は、昇圧比に基づいてフィードフォワード項たるＤＵＴＹを演算する式であるといえる。そこで、以下、式（８）を、「昇圧比の式（８）」と呼ぶ。 Here, although not illustrated, DUTY applied as a feedforward term in the continuous region V2 control in the regenerative operation is expressed by the following equation (8).

Here, (V1 / V2) indicates the step-up ratio. Therefore, it can be said that Expression (8) is an expression for calculating DUTY as a feedforward term based on the step-up ratio. Therefore, hereinafter, the equation (8) is referred to as “step-up ratio equation (8)”.

ここで、図８のリアクトル電流ＩＬの波形のうち斜線部分が、双方向ＤＣ−ＤＣコンバータ１２の出力の両端に接続されるコンデンサＣ２（図１０参照）に対して、充電される電流を示している。このようなコンデンサＣ２の充電電流（斜線部分）の平均と、負荷１３（図１や図１０参照）の電流とが同一値になると、２次側電圧Ｖ２が安定になる。即ち、２次側電圧Ｖ２が安定になるためには、次の式（１０）が成立する必要がある。

よって、式（１０）に対して、式（９）を代入して、２次側電圧Ｖ２について解くと、次の式（１１）が得られる。

The minimum value Imin (maximum absolute value) of the reactor current IL shown in FIG. 8 is expressed by the following equation (9).

Here, the hatched portion of the waveform of the reactor current IL in FIG. 8 indicates the current charged to the capacitor C2 (see FIG. 10) connected to both ends of the output of the bidirectional DC-DC converter 12. Yes. When the average of the charging current (shaded portion) of the capacitor C2 and the current of the load 13 (see FIGS. 1 and 10) have the same value, the secondary side voltage V2 becomes stable. That is, in order for the secondary side voltage V2 to be stable, the following equation (10) needs to be established.

Therefore, the following equation (11) is obtained by substituting equation (9) for equation (10) and solving for the secondary side voltage V2.

  Here, it is assumed that the step-up ratio equation (2) is used as it is in the low power V2 control in the one-side drive method. In this case, DUTY calculated as a feed-forward term by the boost ratio equation (2) is increased, and as a result, the ON period Ton is also increased. In equation (11), assuming that the secondary side voltage V2 is a variable and all other values are constants, the secondary side voltage V2 becomes smaller by the square of the on-period Ton as the constant becomes larger. End up. This is the cause of the above-described DUTY problem.

式（１２）は、式（１１）におけるＶ２を、２次側電圧指令値Ｖ２^＊に置き換えた式から得られたものであり、式（１１）は、上述の如くリアクトル電流ＩＬの最小値Ｉｍｉｎ（絶対値の最大値）の予測式から得られたものである。よって、式（１２）は、リアクトル電流ＩＬの最小値Ｉｍｉｎ（絶対値の最大値）、即ち回路のインダクタ電流の最大値（絶対値）の予測結果に基づいて、フィードフォワード項たるＤＵＴＹが算出される式となっている。 Therefore, in the continuous area / low power area DUTY switching method, the V2 control in the low power area during the regenerative operation does not apply the step-up ratio equation (2) so that the DUTY problem does not occur. Equation (12) is applied to calculate DUTY as a feedforward term.

The expression (12) is obtained from an expression in which V2 in the expression (11) is replaced with the secondary side voltage command value V2 ^* , and the expression (11) is the minimum value Imin of the reactor current IL as described above. It is obtained from the prediction formula (maximum absolute value). Therefore, the equation (12) is calculated based on the prediction result of the minimum value Imin (maximum value of the absolute value) of the reactor current IL, that is, the maximum value (absolute value) of the inductor current of the circuit. It is a formula.

As described above, in the one-side drive method to which the continuous region / low power region DUTY switching method is applied, the V2 control in the continuous region is generally used in the complementary drive method as a calculation formula for the DUTY as a feedforward term. The boost ratio formula is applied. Specifically, the formula (2) of the boost ratio is applied in the V2 control in the low power region during the power running operation, and the formula (9) of the boost ratio is applied in the V2 control in the low power region during the regenerative operation. Applied.
On the other hand, in the V2 control in the low power range, the formula of the boost ratio is not applied as the calculation formula of DUTY which is a feedforward term, and based on the prediction result of the maximum value (absolute value) of the circuit inductor current. The calculation formula is applied. Specifically, Equation (7) is applied in the V2 control in the low power region during the power running operation, and Equation (12) is applied in the V2 control in the low power region during the regeneration operation.

Thus, the V2 control of the one side drive system can be appropriately executed by applying the continuous area / low power area DUTY switching method.
Specifically, it is possible to solve the DUTY problem, that is, the problem that the secondary voltage V2 is unnecessarily increased or decreased in the V2 control in a low power range such as a light load. That is, unnecessary voltage fluctuations at light loads can be significantly suppressed. As a result, it is possible to reduce the loss in the V2 control in the low power range and to suppress the heat generation of the circuit, so that the cooling device (not shown) for the bidirectional DC-DC converter 12 can be downsized, and consequently This makes it possible to reduce the volume and cost of the entire in-vehicle PDU system.

The details of the continuous area / low power area DUTY switching method have been described above with reference to FIGS. 6 to 8.
Next, a switching element drive mode conversion method will be described with reference to FIG.
FIG. 9 is a diagram showing a comparison between a conventional method in which a tracking delay problem occurs as a method of V2 control in a low power region in the one-side drive method and the switching element drive mode conversion method of the present embodiment. .

FIG. 9A shows a timing chart of the command DUTY ^* in the V2 control during the power running operation in the one-side drive method and when the secondary side voltage V2 is larger than the secondary side voltage command value V2 ^*. ing. A solid curve Dn indicates a time transition of the command DUTY ^* by the switching element drive mode conversion method of the present embodiment, and a dotted curve Dp indicates a time transition of the command DUTY ^* by the conventional method.
FIG. 9B shows a timing chart of the secondary side voltage V2 (actually measured value) when feedback control is performed according to the command DUTY ^* of FIG. A solid curve N2 indicates a time transition of the secondary side voltage V2 (actually measured value) by the switching element drive mode conversion method of the present embodiment, and a dotted line curve P2 indicates the secondary side voltage V2 (by the conventional method). It shows the time transition of the measured value.

In the conventional method, the command DUTY ^* indicated by the curve Dp decreases during the period Ta in FIG.
In the period Tb, when the L pulse (step-up PWM triangular wave) becomes 0 and the command DUTY ^* indicated by the curve Dp reaches 0, it remains 0 as it is. However, since the discharge of the capacitor C2 connected to both ends of the load (load 33 in the example of FIG. 2) does not proceed, as shown in FIG. 9B, the secondary voltage V2 (measured value) indicated by the curve P2 is shown. Will not go down.
Thereafter, the discharge of the capacitor C2 gradually proceeds, and accordingly, the secondary side voltage V2 (measured value) indicated by the curve P2 gradually decreases and approaches the secondary side voltage command value V2 ^* as shown in FIG. 9A. In the period Tc, the command DUTY ^* indicated by the curve Dp increases.
In the period Td in FIG. 9A, when the secondary side voltage V2 (actually measured value) indicated by the curve P2 substantially coincides with the secondary side voltage command value V2 ^* , the command DUTY ^* indicated by the curve Dp is also stabilized.
Thus, in the V2 control in the low power range, the main cause of the feedback control follow-up delay, that is, the cause of the conventional follow-up delay issue is the both ends of the load (load 33 in the example of FIG. 2). The discharge of the capacitor C2 connected to the terminal does not proceed.

  Therefore, the inventors have invented a switching element drive mode conversion method capable of solving the conventional tracking delay problem as shown in FIG. 9B.

That is, in the switching element drive mode conversion method of this embodiment, the command DUTY ^* indicated by the curve Dn decreases during the period TA in FIG. At this stage, an L pulse (step-up PWM triangular wave) is output and the SL drive mode is set. Up to this point, the operation is similar to the period Ta in the conventional method.
However, in the period Tb, when the L pulse (step-up PWM triangular wave) becomes 0 and the command DUTY ^* indicated by the curve Dn becomes negative (−), the H pulse (step-down PWM triangular wave) is output. Switch to drive mode. In this case, since the current of the power supply circuit unit 51 (see FIG. 10) flows in the direction of regeneration toward the battery 31, the discharge of the capacitor C2 connected to both ends of the load (load 13 in the example of FIG. 1) suddenly Proceed to Along with this, the secondary side voltage V2 (actually measured value) indicated by the curve N2 also decreases significantly.
As a result, when the secondary side voltage V2 (actually measured value) approaches the secondary side voltage command value V2 ^* , the command DUTY ^* indicated by the curve Dn increases during the period TC in FIG. 9A.
In the period TD in FIG. 9A, when the secondary side voltage V2 (actually measured value) indicated by the curve N2 substantially coincides with the secondary side voltage command value V2 ^* , the command DUTY ^* indicated by the curve Dn is also stabilized.
Thus, the negative from the command DUTY ^* is positive (+) (-) transitions to switched from SL drive mode to the SH driving mode. As a result, since the discharge of the capacitor C2 proceeds rapidly, the secondary side voltage V2 (actually measured value) indicated by the curve N2 can be quickly lowered to approach the secondary side voltage command value V2 ^* .

Although illustration and detailed description are omitted, even in the V2 control during the regenerative operation in the one-side drive method, by applying the switching element drive mode conversion method of the present embodiment, the power drive operation and the SL drive mode can be performed. The same operation is realized except that the switching between the SH drive mode and the SH drive mode is reversed. As a result, since charging of the capacitor C2 proceeds rapidly, the secondary side voltage V2 (actually measured value) can be quickly lowered to approach the secondary side voltage command value V2 ^* .

In other words, the switching element drive mode conversion method is as follows.
That is, when the current Iout of the load (load 13 in the example of FIG. 1) is 0 or more (when Iout ≧ 0), it is determined that the powering operation is being performed, and when the load current Iout is less than 0 (when Iout <0) ), It can be determined that the regenerative operation.
Further, the command DUTY ^* is roughly divided into a SL drive command DUTY ^* (hereinafter referred to as “BOOSTDUTY ^* ”) and a SH drive command DUTY ^* (hereinafter referred to as “BUCKDUTY ^* ”). it can.
In the case of Iout ≧ 0 (in the case of powering operation), when BOOSTDUTY ^* ≧ 0, the SL drive mode is set, and BOOSTDUTY ^* is output. As a result, the switching element SL is driven. At this time, since BUCKDUTY ^* = 0, the driving of the switching element SH is prohibited.
On the other hand, even for Iout ≧ 0 (the case of power running operation), when the BOOSTDUTY ^* <0, becomes SH driving mode, -BOOSTDUTY ^* value is outputted is substituted as BUCKDUTY ^*, as a result, the switching The element SH is driven. At this time, since BOOSTDUTY ^* = 0, the driving of the switching element SL is prohibited.
On the other hand, when Iout <0 (in the case of regenerative operation), when BUCKDUTY ^* ≧ 0, the SH drive mode is set, and BUCKDUTY ^* is output, and as a result, the switching element SH is driven. At this time, since BOOSTDUTY ^* = 0, the driving of the switching element SL is prohibited.
On the other hand, even for Iout <0 (the case of regenerative operation), when the BUCKDUTY ^* <0, becomes SL drive mode, -BUCKDUTY ^* value is outputted is substituted as BOOSTDUTY ^*, as a result, the switching The element SL is driven. At this time, since BUCKDUTY ^* = 0, the driving of the switching element SH is prohibited.

Thus, by applying the switching element drive mode conversion method, the V2 control of the one-side drive method can be appropriately executed.
Specifically, it is possible to solve the follow-up delay problem, that is, the problem that the follow-up delay of the feedback control and the ripple voltage occur when the load and voltage fluctuate. As a result, the V2 control of the one-side drive method can be appropriately executed, so that an unnecessary ripple voltage is suppressed and a stable secondary voltage V2 can be obtained.

  The details of the continuous region / low power region DUTY switching method and the switching element drive mode conversion method of the present embodiment have been described above.

  FIG. 10 is a configuration diagram showing a detailed configuration of the bidirectional DC-DC converter 12 of FIG. 1 to which the continuous / low power range DUTY switching method and the switching element drive mode conversion method are applied.

  The bidirectional DC-DC converter 12 includes a power supply circuit unit 51 and a control unit 52.

In FIG. 10, among the constituent elements (elements) of the power circuit section 51, the same constituent elements (elements) as the general power circuit section 41 in FIG. Will be omitted as appropriate.
The power supply circuit unit 51 is different from the general power supply circuit unit 41 in that a capacitor C3 is further connected to both ends of the load 13 and a current detection unit 61 is provided. The point of is the same.

  The control unit 52 includes an ECU (Electronic Control Unit) 71, a DUTY calculation unit 72, A / D (Analog to Digital) conversion units 73 to 75, a deviation calculation unit 76, a PID compensation unit 77, and a DUTY command calculation. Part 78, PWM output part 79, and gate driver 80.

The ECU 71 controls processing of the entire control unit 52. For example, the ECU 71 generates data of the secondary side voltage command value V2 ^* and supplies the data to the DUTY calculation unit 72 and the deviation calculation unit 76.

In the DUTY calculation unit 72, the data of the current Iout of the load 13 from the A / D conversion unit 73, the data of the primary side voltage V1 from the A / D conversion unit 74, and the secondary side voltage command value from the ECU 71 are displayed. V2 ^* data is supplied respectively.
The DUTY computing unit 72 computes DUTY as a feedforward term based on these data, and supplies the data to the DUTY computing unit 72.

The A / D converter 73 converts the current Iout detected as an analog signal by the current detector 61 into a digital signal (data) and supplies the digital signal (data) to the DUTY calculator 72.
The A / D conversion unit 74 converts an analog signal indicating a divided voltage of the resistor R1 and the resistor R2 connected in series to both ends (primary side) of the battery 11 into a digital signal (data), and converts it to 1 The data is supplied to the DUTY calculation unit 72 as data of the secondary voltage V1.
The A / D conversion unit 75 converts an analog signal indicating a divided voltage of the resistor R3 and the resistor R4 connected in series to both ends (secondary side) of the load 13 into a digital signal (data), and converts it to 2 The data is supplied to the deviation calculator 76 as data of the secondary voltage V2.

The deviation calculation unit 76 calculates a difference between the secondary side voltage command value V2 ^* supplied as data from the ECU 71 and the secondary side voltage V2 supplied as data from the A / D conversion unit 75, thereby providing feedback. The control deviation is obtained and the data is supplied to the PID compensation unit 77.

  The PID compensation unit 77 calculates a so-called feedback term based on the deviation supplied as data from the deviation calculation unit 76 in order to perform PID compensation for the feedforward term calculated by the DUTY calculation unit 72, and the data is calculated as DUTY. This is supplied to the command calculation unit 78.

The DUTY command calculation unit 78 obtains the command DUTY ^* by adding the feedforward term (DUTY) supplied as data from the DUTY calculation unit 72 and the feedback term supplied as data from the PID compensation unit 77, The data is supplied to the PWM output unit 79.

The PWM output unit 79 outputs an L pulse (step-up PWM triangular wave) or an H pulse (step-down PWM triangular wave) to the gate driver 80 based on the command DUTY ^* supplied from the DUTY command calculation unit 78.
Specifically, the L pulse (step-up PWM triangular wave) is output to the gate driver 80 with the pulse width varied according to the value of BOOSTDUTY ^* . On the other hand, the pulse width of the H pulse (step-down PWM triangular wave) is varied according to the value of BUCKDUTY ^* and output to the gate driver 80.

When an L pulse (step-up PWM triangular wave) is output from the PWM output unit 79, the gate driver 80 generates a gate signal Vge_Lo based on the L pulse and supplies it to the switching element SL (the gate of the IGBT). The switching element SL is driven based on the gate signal Vge_Lo and appropriately switches between an on state and an off state.
Further, when an H pulse (step-down PWM triangular wave) is output from the PWM output unit 79, the gate driver 80 generates a gate signal Vge_Hi based on the H pulse, and supplies the gate signal Vge_Hi to the switching element SH (IGBT gate). The switching element SH is driven based on the gate signal Vge_Hi and switches between an on state and an off state as appropriate.

A process in which the control unit 52 having the above configuration performs V2 control on the power supply circuit unit 51 is hereinafter referred to as “V2 control process”.
Next, the flow of such a V2 control process will be described with reference to FIG.

FIG. 11 is a flowchart for explaining the flow of the V2 control process.
The V2 control process is repeatedly executed at predetermined time intervals, for example, every time the ECU 71 generates data of the secondary side voltage command value V2 ^* while the V2 control is executed.

  In step S <b> 1, the control unit 52 acquires data of each actual measurement value of the power supply circuit unit 51, that is, each data of the primary side voltage V <b> 1, the secondary side voltage V <b> 2, and the load 13 current Iout.

In step S2, the DUTY calculation unit 72 of the control unit 52 calculates DUTY as a feedforward term.
Here, the continuous area / low power area DUTY switching method is applied to the V2 control processing.
Therefore, the DUTY calculation unit 72 recognizes whether the V2 control is currently executed in the continuous region or the V2 control is executed in the low power region. The recognition method is not particularly limited, and for example, a recognition method of recognizing whether it is a continuous region or a low power region based on the value of the current Iout of the load 13 can be employed.
When recognized as V2 control in the continuous region, DUTY as a feedforward term for BOOSTDUTY ^* is calculated according to the boost ratio equation (2), while DUTY as a feedforward term for BUCKDUTY ^* is , And is calculated according to the boost ratio equation (8).
On the other hand, when it is recognized that it is a low power range, DUTY as a feedforward term for BOOSTDUTY ^* is calculated according to Equation (7), while DUTY as a feedforward term for BUCKDUTY ^* is It calculates according to Formula (12).

  In step S3, the PID compensation unit 77 of the control unit 52 calculates a feedback term.

In step S4, the DUTY command calculation unit 78 of the control unit 52 adds the feedforward term (DUTY) calculated in the process of step S2 and the feedback term calculated in the process of step S3, thereby adding a command DUTY. ^* Is calculated.
As described above, as the command DUTY ^*, and BOOSTDUTY ^* for SL drive, BUCKDUTY ^* and is calculated for SH driving.

  In step S5, the DUTY command calculation unit 78 determines whether or not the current Iout of the load 13 is 0 or more (Iout ≧ 0?).

  When the current Iout of the load 13 is 0 or more, that is, during the power running operation, it is determined as YES in Step S5, and the process proceeds to Step S6.

In step S6, the DUTY command calculation unit 78 determines whether or not BOOSTDUTY ^* is 0 or more (BOOSTDUTY ^* ≧ 0?).

When BOOSTDUTY ^* is 0 or more, it is determined as YES in step S6, and the process proceeds to step S7.
In step S7, the DUTY command calculation unit 78 outputs BOOSTDUTY ^* and sets BUCKDUTY ^* = 0 (inhibits output) by setting the SL drive mode.
In step S14, the PWM output unit 79 of the control unit 52 performs PWM output. That is, in this case, an L pulse (a step-up PWM triangular wave) whose pulse width is varied according to the value of BOOSTDUTY ^* is output, and as a result, the switching element SL is driven. On the other hand, since the output of the H pulse (step-down PWM triangular wave) is prohibited, the driving of the switching element SH is also prohibited.
Thereby, the V2 control process ends.

On the other hand, if BOOSTDUTY ^* is less than 0, it is determined as NO in step S6, and the process proceeds to step S8.
In step S8, the DUTY command calculation unit 78 enters the SH drive mode.
In step S9, the DUTY command calculation unit 78 recalculates the command DUTY ^* .
That is, the DUTY command calculation unit 78 re-substitutes -BOOSTDUTY ^* as BUCKDUTY ^* and outputs it, and sets BOOSTDUTY ^* = 0 (prohibits output).
In step S14, the PWM output unit 79 performs PWM output. That is, in this case, an H pulse (step-down PWM triangular wave) whose pulse width is varied according to the value of BUCKDUTY ^* is output, and as a result, the switching element SH is driven. On the other hand, since the output of the L pulse (step-up PWM triangular wave) is prohibited, the driving of the switching element SL is also prohibited.
Thereby, the V2 control process ends.

The series of processes when the current Iout of the load 13 is 0 or more, that is, during the power running operation has been described above.
Next, a series of processes when the current Iout of the load 13 is less than 0, that is, during the regenerative operation will be described.
In this case, it is determined as NO in Step S5, and the process proceeds to Step S10.

In step S10, the DUTY command calculation unit 78 determines whether or not BUCKDUTY ^* is 0 or more (BUCKDUTY ^* ≧ 0?).

When BUCKDUTY ^* is 0 or more, it is determined as YES in Step S10, and the process proceeds to Step S13.
In step S13, the DUTY command calculation unit 78 outputs BUCKDUTY ^* and sets BOOSTDUTY ^* = 0 (inhibits output) by setting the SH drive mode.
In step S14, the PWM output unit 79 of the control unit 52 performs PWM output. That is, in this case, an H pulse (step-down PWM triangular wave) whose pulse width is varied according to the value of BUCKDUTY ^* is output, and as a result, the switching element SH is driven. On the other hand, since the output of the L pulse (step-up PWM triangular wave) is prohibited, the driving of the switching element SL is also prohibited.
Thereby, the V2 control process ends.

On the other hand, if BUCKDUTY ^* is less than 0, it is determined as NO in step S10, and the process proceeds to step S11.
In step S11, the DUTY command calculation unit 78 is set to the SL drive mode.
In step S12, the DUTY command calculation unit 78 recalculates the command DUTY ^* .
That is, the DUTY command calculation unit 78 re-substitutes -BUCKDUTY ^* as BOOSTDUTY ^* and outputs it, and sets BUCKDUTY ^* = 0 (prohibits output).
In step S14, the PWM output unit 79 performs PWM output. That is, in this case, an L pulse (a step-up PWM triangular wave) whose pulse width is varied according to the value of BUCKDUTY ^* is output, and as a result, the switching element SL is driven. On the other hand, since the output of the H pulse (step-down PWM triangular wave) is prohibited, the driving of the switching element SH is also prohibited.
Thereby, the V2 control process ends.

  According to this embodiment described above, the following effects (1) and (2) can be obtained.

(1) The bidirectional DC-DC converter 12 of the present embodiment can appropriately execute the V2 control of the one-side drive method by applying the continuous region / low power region DUTY switching method.
Specifically, it is possible to solve the DUTY problem, that is, the problem that the secondary voltage V2 is unnecessarily increased or decreased in the V2 control in a low power range such as a light load. That is, unnecessary voltage fluctuations at light loads can be significantly suppressed. As a result, it is possible to reduce the loss in the V2 control in the low power range and to suppress the heat generation of the circuit, so that the cooling device (not shown) for the bidirectional DC-DC converter 12 can be downsized, and consequently This makes it possible to reduce the volume and cost of the entire in-vehicle PDU system.

(2) The bidirectional DC-DC converter 12 of the present embodiment can appropriately execute the V2 control of the one-side drive method by applying the switching element drive mode conversion method.
Specifically, it is possible to solve the follow-up delay problem, that is, the problem that the follow-up delay of feedback control occurs when the load fluctuates. As a result, since the V2 control of the one-side drive method can be appropriately executed, unnecessary ripple voltage is suppressed, and a stable secondary voltage V can be obtained.

Note that each of the continuous area / low power area DUTY switching method and the switching element drive mode conversion method is an independent method and can be applied alone. In that case, by applying the continuous region / low power region DUTY switching method alone, the above-described effect (1) can be obtained, and by applying the switching element drive mode conversion method alone, the above-described effect can be obtained. The effect of (2) can be achieved.
However, by combining the effects (1) and (2) described above, the V2 control of the one-side drive method can be more appropriately executed. Therefore, it is preferable to apply the continuous area / low power area DUTY switching method and the switching element drive mode conversion method in combination as in the present embodiment.

It should be noted that the present invention is not limited to the above-described embodiment, and modifications, improvements, and the like within the scope that can achieve the object of the present invention are included in the present invention.
For example, the continuous region / low power region DUTY switching method and the switching element drive mode conversion method include a DC-DC converter 101 that becomes a step-down converter as shown in FIG. The present invention can also be applied to a DC-DC converter 102 that is a buck-boost converter as shown in FIG.

  By the way, the above-described series of processes can be executed by hardware or can be executed by software.

When a series of processing is executed by software, a program constituting the software is installed on a computer or the like from a network or a recording medium.
The computer may be a computer incorporated in dedicated hardware. The computer may be a computer capable of executing various functions by installing various programs, for example, a general-purpose personal computer.
Although not shown, the recording medium including such a program is not only constituted by a removable medium distributed separately from the apparatus main body in order to provide a program to the user, but also in a state of being incorporated in the apparatus main body in advance. It consists of a recording medium provided to the user.
The removable medium is composed of, for example, a magnetic disk (including a floppy disk), an optical disk, a magneto-optical disk, or the like. The optical disk is composed of, for example, a CD-ROM (Compact Disk-Read Only Memory), a DVD (Digital Versatile Disk), or the like. The magneto-optical disk is configured by an MD (Mini-Disk) or the like. Moreover, the recording medium provided to the user in a state of being pre-installed in the apparatus main body is configured by, for example, a memory or a hard disk in which a program is recorded.

  In the present specification, the step of describing the program recorded on the recording medium is not limited to the processing performed in time series along the order, but is not necessarily performed in time series, either in parallel or individually. The process to be executed is also included.

DESCRIPTION OF SYMBOLS 11 Battery 12 Bidirectional DC-DC converter 13 Load 51 Power supply circuit part 52 Control part 71 ECU
72 DUTY Calculation Unit 73 A / D Conversion Unit 74 A / D Conversion Unit 75 A / D Conversion Unit 76 Deviation Calculation Unit 77 PID Compensation Unit 78 DUTY Command Calculation Unit 79 PWM Output Unit 80 Gate Driver

Claims (3)

A DC power source is connected to the primary side, a load is connected to the secondary side,
A circuit unit having a series connection of a first switching element and a second switching element;
A power running operation for powering the load from the DC power source to the load by controlling driving of the first switching element and the second switching element and maintaining a secondary voltage at an arbitrary value, or the load A control unit for realizing a regenerative operation for regenerating power from the DC power source to
A DC-DC converter comprising:
As drive modes of the first switching element and the second switching element,
A first drive mode for maintaining the secondary side voltage at an arbitrary value by driving the first switching element by a pulse of the first DUTY and prohibiting the driving of the second switching element;
A second driving mode for maintaining the secondary voltage at an arbitrary value by prohibiting driving of the first switching element and driving the second switching element with a pulse of a second DUTY;
Have
The controller is
Generating the first DUTY command and the second DUTY command;
In the power running operation, the first drive mode is executed when the first DUTY command is 0 or more, and the second drive mode is executed when the first DUTY command is less than 0,
In the regenerative operation, when the second DUTY command is 0 or more, the second drive mode is executed, and when the second DUTY command is less than 0, the first drive mode is executed.
DC-DC converter. The controller is
Further, it is determined that the case where the load current is 0 or more is the case of the power running operation, and the case where the load current is less than 0 is determined as the case of the regenerative operation,
When the load current is 0 or more,
When the first DUTY command is 0 or more, as the first drive mode, the first switching element is driven using the first DUTY command, and the driving of the second switching element is prohibited,
When the first DUTY command is less than 0, as the second drive mode, the driving of the first switching element is prohibited and the sign inversion value of the first DUTUY command is used as the second DUTY command. 2 Drive the switching element,
When the load current is less than 0,
When the second DUTY command is equal to or greater than 0, as the second drive mode, the driving of the first switching element is prohibited, and the second switching element is driven using the second DUTY command,
When the second DUTY command is less than 0, as the first drive mode, the first switching element is driven using the sign inversion value of the second DUTUY command as the first DUTY command, and the second switching Prohibit driving of the element,
The DC-DC converter according to claim 1. A DC power source is connected to the primary side, a load is connected to the secondary side,
A circuit unit having a series connection of a first switching element and a second switching element;
A power running operation for powering the load from the DC power source to the load by controlling driving of the first switching element and the second switching element and maintaining a secondary voltage at an arbitrary value, or the load A control unit for realizing a regenerative operation for regenerating power from the DC power source to
A method for controlling a DC-DC converter comprising:
As drive modes of the first switching element and the second switching element,
A first drive mode for maintaining the secondary side voltage at an arbitrary value by driving the first switching element by a pulse of the first DUTY and prohibiting the driving of the second switching element;
A second driving mode for maintaining the secondary voltage at an arbitrary value by prohibiting driving of the first switching element and driving the second switching element with a pulse of a second DUTY;
Have
The control unit is
Generating the first DUTY command and the second DUTY command;
In the power running operation, the first drive mode is executed when the first DUTY command is 0 or more, and the second drive mode is executed when the first DUTY command is less than 0,
In the regenerative operation, when the second DUTY command is 0 or more, the second drive mode is executed, and when the second DUTY command is less than 0, the first drive mode is executed.
A method for controlling a DC-DC converter.

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