JP5078562B2 - Signal processing apparatus, signal processing method, signal processing program, and recording medium therefor - Google Patents

Description

  The present invention relates to a signal processing device, a signal processing method, a signal processing program, and a recording medium on which the signal processing program is recorded.

  2. Description of the Related Art Conventionally, FM receivers that receive and process FM (Frequency Modulation) broadcast waves, which are composite signals including a predetermined signal such as a pilot signal that serves as a reference for demodulation processing, have been widely used. In such an apparatus, in order to reproduce the predetermined signal or generate a signal having a frequency that is an integral multiple of the frequency of the predetermined signal, a phase locked loop (PLL) technique is generally employed ( Patent Document 1; hereinafter referred to as “Conventional Example 1”).

  Apart from the technique of Conventional Example 1, a technique for reproducing the predetermined signal or generating a signal having a frequency that is an integral multiple of the frequency of the predetermined signal without adopting the PLL technique has been proposed. (See Patent Document 2; hereinafter referred to as “Conventional Example 2”). In the conventional technique, a phase difference between a base signal generated internally and the predetermined signal is obtained. Then, the predetermined signal is reproduced by adding the obtained phase difference to the base signal.

JP 2000-13339 A JP 2006-528451 Gazette

  Although the technique of the conventional example 1 employs the PLL technique, the PLL is generally difficult to design. Further, when the PLL method is used, it is difficult to synchronize with the predetermined signal with good follow-up in a situation where the reception environment changes as in the case of mobile reception of FM broadcast waves, for example. .

  Further, in the technique of Conventional Example 2 described above, a reference signal for matching the phase of the predetermined signal is necessary. However, when the frequency and phase of the predetermined signal fluctuate due to the occurrence of multipath, it is not always easy to generate a reference signal for synchronizing with the predetermined signal with high accuracy.

  For this reason, there is a need for a new technique that can easily and quickly generate a signal synchronized with a predetermined signal. In such a new technique, it is desirable that the technique can reduce the calculation amount as much as possible. Meeting this requirement is one of the problems to be solved by the present invention.

  The present invention has been made in view of the above, and an object of the present invention is to provide a signal processing apparatus capable of quickly generating a signal that is accurately synchronized with an input signal while reducing the amount of calculation.

The invention according to claim 1 is a signal processing device that performs processing related to an input signal that is sampled at a first sample rate and includes a predetermined signal in a predetermined frequency band, and reflects the phase of the predetermined signal and is orthogonal to each other. And orthogonal signal generation means for generating a first signal and a second signal sampled at a second sample rate lower than the first sample rate; the amplitude of the first signal at the same time point and the second signal After calculating the ratio with the amplitude of the two signals, the phase of the signal reflecting the predetermined signal is changed based on the time change of the calculated ratio without feeding back the calculated ratio to the orthogonal signal generating means. a phase calculation means for calculating; based on the calculation result of the phase calculation means, and when it is sampled at a higher third sample rate higher than the second sample rate Up-sampling means and for calculating a phase and the like; while reflecting the calculation result of the phase calculation means, a reference signal generating means for generating a reference signal at said predetermined signal and a predetermined relationship; wherein the reference signal generating means Is a signal processing device that generates the reference signal based on a calculation result by the up-sampling means .

The invention according to claim 12 is a signal processing method for performing processing related to an input signal sampled at a first sample rate and including a predetermined signal in a predetermined frequency band, which reflects the phase of the predetermined signal and is orthogonal to each other. A quadrature signal generating step of generating a first signal and a second signal sampled at a second sample rate lower than the first sample rate; and an amplitude of the first signal and the first signal at the same time point The ratio of the amplitude of the two signals is calculated, and the phase of the signal reflecting the predetermined signal is calculated based on the time change of the calculated ratio without using the calculated ratio in the orthogonal signal generation step. a phase calculation step of calculating; based on the calculation result in the phase calculating step, the the case where it is sampled at a higher third sample rate higher than the second sample rate Upsampling step of calculating the phase of; reflecting the calculation result of the phase calculating step, the predetermined signal and the reference signal generating step of generating a reference signal in a predetermined relationship; equipped with, in the reference signal generating step, The signal processing method is characterized in that the reference signal is generated based on a calculation result in the upsampling step .

According to a thirteenth aspect of the present invention, there is provided a signal processing program for causing a calculation means to execute the signal processing method according to the twelfth aspect.

A fourteenth aspect of the present invention is a recording medium in which the signal processing program according to the thirteenth aspect is recorded so as to be readable by an arithmetic means.

  Hereinafter, embodiments of the present invention will be described with reference to the accompanying drawings. In the following description, the same or equivalent elements are denoted by the same reference numerals, and redundant description is omitted.

[First Embodiment]
First, a first embodiment of the present invention will be described with reference to FIGS.

<Configuration>
FIG. 1 is a block diagram illustrating a schematic configuration of a signal processing device 100A according to the first embodiment. The signal processing device 100A is a device that receives and processes the signal SIA output from the signal source 910A via the input terminal 191 and outputs the processed signal as a processing result signal MSA from the output terminal 192.

In the first embodiment, the signal SIA, as shown in FIG. 2, the pilot signal PS of the angular frequency _{ω c (αsin (ω c t} + φ 0)), the frequency band of the angular frequency ω _c ~3ω _c It is assumed that the signal component SG2 is included. Here, the signal component SG2 is by the band of the signal of the angular frequency 0~ω _c, 2 times the angular frequency of the signal _{(αsin [2 (ω c t} + φ 0)]) of the pilot signal PS in the amplitude modulated signal It shall be. In the first embodiment, the signal processing device 100A generates a reference signal BSA (∝sin [2 (ω _c t + φ ₀ )]) having an angular frequency of 2ω _c based on the pilot signal PS, and the reference signal BSA is generated. By utilizing this, the amplitude modulation signal SG2 is demodulated.

In the first embodiment, as shown in FIG. 2, the signal SIA includes the signal component SG1 in the frequency band of the angular frequency 0 to ω _c , but the object to be processed by the signal processing apparatus 100A Is not a signal component. Therefore, in FIG. 2, the signal component SG1 is expressed by a broken line. In the first embodiment, the signal SIA is a digital signal sampled and digitized at a predetermined sample rate f _SM1 (> 6ω _c / (2π)).

In the first embodiment, in order to simplify the description, it is assumed that the pilot signal PS included in the signal SIA is expressed by the following equation (1).
PS (t) ∝sin [θ _C (t)] = sin (ω _C t + φ ₀ ) (1)
Here, the value φ ₀ is the phase of pilot signal PS when time t = 0.

Note that when the signal processing device 100A is mounted on a moving body such as a vehicle, the angular frequency ω _C and the initial phase value φ ₀ change slightly.

  Examples of the signal represented by the equation (1) include a pilot signal in a composite signal that is a signal obtained by FM detection of an FM broadcast wave.

  Returning to FIG. 1, the signal processing device 100A includes an orthogonal signal generation unit 110A as an orthogonal signal generation unit and a phase calculation unit 120A as a phase calculation unit. The signal processing device 100A includes an upsampling unit 125 as an upsampling unit, a reference signal generation unit 130A as a reference signal generation unit, and a signal processing unit 140 as a first type signal processing unit.

The orthogonal signal generation unit 110A is a signal in a band including the angular frequency ω _C from the signal SIA received via the input terminal 191, and the components of the angular frequency ω _C are orthogonal to each other, and the sample rate f _SM2 Two signals PWA ₁ and PWA ₂ are generated. As shown in FIG. 3, the orthogonal signal generation unit 110A having such a function includes an orthogonalization unit 112A as an orthogonalization unit, filters (FIL) 113A ₁ and 113A ₂ as a filter unit, and a downsampling unit. A downsampling unit 115A.

Based on the component of the angular frequency ω _C included in the signal SIA (that is, the pilot signal PS), the orthogonalizing unit 112A is orthogonal to each other and two signals OSA reflecting the phase θ _C (t) of the pilot signal PS ₁ and OSA ₂ are generated. Orthogonalization section 112A having such a function, as shown in FIG. 4, and a 90 ° phase shifter 119 for the signal of the angular frequency omega _C.

In the orthogonalizing unit 112A configured as described above, a signal in phase with the received signal SIA is output to the FIL 113A ₁ as the signal OSA ₁ of the sample rate f _SM1 . On the other hand, in the orthogonalizing unit 112A, the phase of the received signal SIA is shifted by 90 ° with respect to the component of the angular frequency ω _C , and is output toward the FIL 113A ₂ as the signal OSA ₂ of the sample rate f _SM1 .

Returning to FIG. 3, the FIL 113A ₁ is configured as a digital filter such as an infinite impulse response (IIR) filter. The FIL 113A ₁ receives the signal OSA ₁ from the orthogonalizing unit 112A. The FIL 113A ₁ selectively passes the component of the angular frequency ω _C in the signal OSA ₁ , that is, the signal component corresponding to the pilot signal PS, toward the downsampling unit 115A as the signal PSA ₁ of the sample rate f _SM1. Output.

Note that a fixed phase shift Δθ associated with the filtering operation may occur through the FIL 113A ₁ . In the first embodiment, it is assumed that such a phase shift Δθ occurs. This phase shift Δθ is determined by the configuration of the FIL 113A ₁ and is determined at the design stage of the FIL 113A ₁ .

In the first embodiment, the signal PSA ₁ output from the FIL 113A ₁ is expressed by the following equation (2).
PSA ₁ (t) = A (t) · sin [θ _S (t)]
= A (t) · sin [θ _C (t) −Δθ]
= A (t) · sin [(ω _C t + φ ₀ ) −Δθ] (2)
Here, A (t) represents the amplitude value of the pilot signal PS.

The FIL 113A ₂ is configured in the same manner as the FIL 113A ₁ . The FIL 113A ₂ receives the signal OSA ₂ from the orthogonalizing unit 112A. Then, FIL 113A ₂ selectively passes the component of angular frequency ω _C in signal OSA ₂ , that is, the signal component corresponding to pilot signal PS, and directs signal PSA ₂ of sample rate f _SM1 to downsampling unit 115A. Output.

The signal PSA ₂ output from the FIL 113A ₂ is expressed by the following equation (3).
PSA ₂ (t) = A (t) · cos [θ _S (t)]
= A (t) · cos [θ _C (t) −Δθ]
= A (t) · cos [(ω _C t + φ ₀ ) −Δθ] (3)

Note that the FILs 113A ₁ and 113A ₂ also function as a decimation filter for reducing the sample rate in a downsampling unit 115A described later.

The downsampling unit 115A generates the signals PWA ₁ and PWA ₂ sampled at the sample rate f _SM2 by reducing the sample rates f _SM1 of the signals PSA ₁ and PSA ₂ . In the first embodiment, the angular frequency of the pilot signal PS whose phase is calculated by the phase calculation unit 120A is ω _C , and the sample rate is (2ω _C ) / (2π) for reproduction of the signal of the angular frequency ω _C. The downsampling unit 115A reduces the sample rate from the sample rate f _SM1 to the sample rate f _SM2 (= f _SM1 / 3> (2ω _C ) / (2π)). ing.

More specifically, the downsampling unit 115A, the data sequence of signals _{PSA 1 (..., PSA 1 (} t 0), PSA 1 (t 1), ..., PSA 1 (t p), ...) from 3 Tsugoto , PSA ₁ (t ₀ ), PSA ₁ (t ₃ ),..., PSA ₁ (t _3p ,...) _Are generated to generate the signal PWA ₁ . Further, down-sampling unit 115A, the data sequence of the signal _{PSA 2 (..., PSA 2 (} t 0), PSA 2 (t 1), ..., PSA 2 (t p), ...) the data selection every third from conducted, data sequence _{(..., PSA 2 (t 0} ), PSA 2 (t 3), ..., PSA 2 (t 3p), ...) by generating, for generating a signal PWA _2. The signals PWA ₁ and PWA ₂ thus generated are output toward the phase calculation unit 120A.

Returning to FIG. 1, the phase calculation unit 120A calculates the phase θ _C (t) of the pilot signal PS based on the signal PWA ₁ and the signal PWA ₂ from the orthogonal signal generation unit 110A. In this calculation, the phase calculation unit 120A calculates, for example, the phase θ _C by correcting the amount of the phase shift Δθ described above after performing an arithmetic operation such as arctan on the signal PSA ₁ and the signal PSA ₂ .

The phase θ _C calculated in this way is output from the phase calculation unit 120A to the upsampling unit 125 as the signal PHW of the sample rate f _SM2 .

The upsampling unit 125 increases the sample rate f _SM2 of the signal PHW from the phase calculation unit 120A, and generates the signal PHA sampled at the sample rate f _SM1 . That is, the upsampling unit 125 returns the sample rate reduced by the downsampling unit 115 to a value before reduction.

More specifically, the upsampling unit 125 performs data PHW (t _3q ) in the data string (..., PHW (t ₀ ), PHW (t ₃ ),..., PHW (t _3p ),. By inserting the data PHW (t _3q + Δt) and the data PHW (t _3q + 2Δt) between the data PHW (t _{3 (q + 1)} ), the signal PHA is generated.

Here, the time Δt is expressed by the following equation (4).
Δt = 2π / 3ω _SM2 (4)

  The signal PHA generated in this way is output from the upsampling unit 125 toward the reference signal generation unit 130A.

Based on the signal PHA of the sample rate f _SM1 from the upsampling unit 125, the reference signal generation unit 130A uses the reference signal BSA (∝sin [2 (ω _c t + φ ₀ )]) of the sample rate f _SM1 of the angular frequency 2ω _C. Is generated. The reference signal generation unit 130A having such a function includes a phase processing unit 131A and a signal generation unit 132, as shown in FIG.

The phase processing unit 131A receives the signal PHA from the upsampling unit 125 and processes the phase θ _C (t) indicated by the signal PHA. In the first embodiment, the phase processing unit 131A calculates the phase θ _M (t) by the following equation (5).
θ _M (t) = 2θ _C (t) = 2 (ω _C t + φ ₀ ) (5)

That is, in the first embodiment, the phase processing unit 131A calculates the phase θ _M (t) whose angular frequency is twice in synchronization with the pilot signal PS. The phase θ _M (t) calculated in this way is output to the signal generator 132 as the phase processing signal MPA.

The signal generator 132 generates the reference signal BSA based on the phase processing signal MPA from the phase processing unit 131A. In the first embodiment, the signal generation unit 132 includes a sine value table in which amplitude values corresponding to phase values are registered, and a sine wave signal having a phase θ _M (t) indicated by the phase processing signal MPA. Is generated as a reference signal BSA.

This reference signal BSA is expressed by the following equation (6).
BSA (t) = C ₀ · sin [θ _M (t)]
= C ₀ · sin [2θ _C (t)]
= C ₀ · sin [2 (ω _C t + φ ₀ )] (6)
Here, C ₀ is a constant.

  The reference signal BSA generated in this way is output from the reference signal generation unit 130A to the signal processing unit 140.

Returning to FIG. 1, the signal processing unit 140 receives the signal SIA of the sample rate f _SM1 via the input terminal 191 and the reference signal BSA of the sample rate f _SM1 from the reference signal generation unit 130A. Then, the signal processing unit 140, a signal component SG2 is an amplitude modulated signal in the signal SIA, and processed using a reference signal BSA, demodulates the band of the signal of the angular frequency 0~ω _c. As shown in FIG. 6, the signal processing unit 140 having such a function includes a multiplication unit 141 and a low-pass filter (LPF) 142.

Multiplier 141 receives signal SIA at the A terminal and reference signal BSA at the B terminal. Then, the multiplication result of the reception result at the A terminal and the reception result at the B terminal is output from the C terminal to the LPF 142 as the mixed signal MXA. Here, the signal component SG2 in the signal SIA has an angular frequency component of angular frequencies ω _{C to} 3ω _C , and the reference signal BSA has only a component of the angular frequency 2ω _C. Therefore, the signal component corresponding to the signal component SG2 in the mixing signal MXA is a component of the angular frequency 0～Omega _C, it is the angular frequency converted into a component corresponding to the angular frequency 3ω _C ~5ω _C.

The LPF 142 is configured as a low-pass filter such as a finite impulse response (FIR) filter. The LPF 142 selectively passes the signal component of the angular frequency 0 to ω _C in the mixed signal MXA from the multiplication unit 141 and outputs the processed signal MSA of the sample rate f _SM1 to the outside through the output terminal 192. To do.

<Operation>
Next, a signal processing operation in the signal processing device 100A configured as described above will be described.

When the signal SIA from the signal source 910A is received by the signal processing device 100A via the input terminal 191, the signal processing device 100A supplies the signal SIA to the orthogonal signal generation unit 110A and the signal processing unit 140 ( (See FIG. 1). In the orthogonal signal generation unit 110A that receives the supply of the signal SIA, first, the orthogonalization unit 112A generates _two signals OSA ₁ and OSA _{2 in} which the components of the angular frequency ω _C included in the signal SIA are orthogonal to each other, and the FIL 113A ₁ , sent to 113A ₂ (see FIG. 4).

Subsequently, FIL113A ₁ which has received the signal OSA ₁ are components of the angular frequency omega _C in the signal OSA _1, i.e., a signal component corresponding to the pilot signal PS selectively passes, directed as a signal PSA ₁ to down-sampling section 115A Output. Further, FIL113A ₂ which has received the signal OSA ₂ is a component of the angular frequency omega _C in the signal OSA _2, i.e., selectively pass a signal component corresponding to the pilot signal PS, directed as a signal PSA ₂ to down-sampling unit 115A (See FIG. 3).

Here, the signal PSA ₁ has a waveform represented by the above-described equation (2). On the other hand, the signal PSA ₂ has a waveform represented by the above-described equation (3).

Downsampling unit 115A which receives the signals PSA _1, PSA ₂ is as described above, by reducing the respective sample rate f _SM1 signals PSA _1, PSA _2, sample rate f _SM2 signal PWA ₁ sampled at , PWA ₂ is generated. The signals PWA ₁ and PWA ₂ thus generated are sent to the phase calculation unit 120A (see FIG. 3).

Phase calculation section 120A having received the signal PWA _1, PWA _2, as described above, the signal PWA ₁ and the signal PWA _2, and based on the phase shift [Delta] [theta], and calculates the phase theta _C of the pilot signal PS. The phase θ _C calculated in this way is sent to the upsampling unit 125 as a signal PHW (see FIG. 1).

Upon receiving the signal PHW, the upsampling unit 125 increases the sample rate f _SM2 of the signal PHW as described above, and generates the signal PHA sampled at the sample rate f _SM1 . The signal PHA generated in this way is sent to the reference signal generation unit 130A (see FIG. 1).

Reference signal generator 130A which receives the signal PHA sample rate f _SM1, based on the signals PHA, to generate a reference signal BSA sample rate f _SM1. Here, the reference signal BSA is a signal having an angular frequency of 2ω _{C in} synchronization with the pilot signal PS. In generating the reference signal BSA, in the reference signal generation unit 130A, first, the phase processing unit 131A calculates the phase θ _M (t) having the angular frequency 2ω _C by the above-described equation (5), and the signal generation unit 132 (see FIG. 5). Signal generating unit 132 which received the phase theta _M a (t) refers to the internal sine table based on the phase theta _M (t), it is a sine wave signal represented by the above-described (6) reference A signal BSA is generated. The reference signal BSA generated in this way is sent to the signal processing unit 140 (see FIG. 5).

Upon receiving the reference signal BSA of the sample rate f _SM1 from the reference signal generation unit 130A and the signal SIA of the sample rate f _SM1 from the signal source 910A, the signal processing unit 140 uses the signal component SG2 that is an amplitude modulation signal as a reference. Processing is performed using the signal BSA, and the signal is demodulated into a signal in the band of the angular frequency 0 to ω _c . In the processing, in the signal processing unit 140, first, the multiplication unit 141 multiplies the signal SIA and the reference signal BSA to generate a mixed signal MXA. Here, as described above, the signal component SG2 in the signal SIA has the angular frequency component of the angular frequency ω _{C to} 3ω _C , and the reference signal BSA has only the component of the angular frequency 2ω _C , so that signal component corresponding to the signal component SG2 in the signal MXA is a component of the angular frequency 0～Omega _C, is the angular frequency converted into a component corresponding to the angular frequency 3ω _C ~5ω _C. The mixed signal MXA having the sample rate f _SM1 generated in this way is sent to the LPF 142 (see FIG. 6).

Receiving the mixed signal MXA, the LPF 142 selectively passes the signal component of the angular frequency 0 to ω _C. As a result, as shown in FIG. 7 corresponding to the signal component SG2, a processing result signal MSA in the frequency band of the angular frequency 0 to ω _C of the sample rate f _SM1 is generated, and is output to the external via the output terminal 192. Is output.

As described above, in the first embodiment, the pilot signal PS is used from the signal SIA including the pilot signal PS having the angular frequency ω _C without using a feedback loop in the PLL system or a base signal that is a reference for phase measurement. The phase θ _{C of} is derived. Therefore, it is possible to generate the reference signal BSA that is easily and quickly synchronized with the pilot signal PS.

In the first embodiment, the sample rate f _SM2 of the signals PWA ₁ and PWA ₂ used for calculating the phase of the pilot signal PS is reduced from the sample rate f _{SM1 of} the signal SIA received at the input terminal 191. Therefore, the number of calculations with a large calculation amount such as arctan calculation in the phase calculation unit 120B can be reduced. Therefore, the total calculation amount can be reduced.

In the first embodiment, the phase signal PHW obtained at the reduced sample rate f _SM2 is increased to the sample rate f _SM1 . Here, since the target for increasing the sample rate f _SM2 is the phase, it is possible to easily obtain the phase signal PHA increased to the sample rate f _SM1 without degrading the accuracy. Therefore, it is possible to obtain a reference signal BSA sample rate f _SM1 suitable for processing of the signal SIA sample rate f _SM1.

  In the first embodiment, since the signal component SG2 in the signal SIA is processed using the reference signal BSA, desired processing can be performed on the signal component SG2 quickly and accurately.

Further, in the first embodiment, when there is a phase shift Δθ in FIL113A _1, 113A ₂ is calculated in consideration of this, since the calculation of the phase theta _C of the pilot signal PS, precisely phase theta _C can do. Therefore, it is possible to generate the reference signal BSA that is accurately synchronized with the pilot signal PS.

In the first embodiment, filters that do not cause a phase shift can be employed as the FILs 113A ₁ and 113A ₂ . In this case, it is not necessary to calculate the phase compensation corresponding to the FILs 113A ₁ and 113A ₂ in the phase calculation unit 120A.

[Second Embodiment]
Next, a second embodiment of the present invention will be described with reference mainly to FIGS.

FIG. 8 is a block diagram illustrating a schematic configuration of a signal processing device 100B according to the second embodiment. Note that the signal processing device 100B receives and processes the signal SIA of the sample rate f _SM1 output from the signal source 910A via the input terminal 191 and outputs the same as the signal processing device 100A in the first embodiment. This is a device that outputs from a terminal 192 as a processing result signal MSA at a sample rate f _SM1 .

  As shown in FIG. 8, the signal processing device 100B includes an orthogonal signal generation unit 110B instead of the orthogonal signal generation unit 110A and a phase calculation instead of the phase calculation unit 120A, as compared with the signal processing device 100A described above. Only the point provided with the part 120B is different. Hereinafter, description will be made mainly focusing on these differences.

The quadrature signal generation unit 110B generates two signals PWB ₁ and PWB ₂ that are orthogonal to each other and sampled at the sample rate f _SM2 from the signal SIA received via the input terminal 191. As shown in FIG. 9, the orthogonal signal generating unit 110B having such a function includes a band limiting filter 111B as a band limiting unit, an orthogonalizing unit 112B as an orthogonalizing unit, and a filter (FIL) 113B as a filtering unit. ₁ and 113B ₂ and a downsampling unit 115A as downsampling means.

The band limiting filter 111B is configured as a digital filter such as an infinite impulse response (IIR) filter. The band limiting filter 111B selectively passes a signal component in a predetermined frequency band including the angular frequency ω _C in the signal SIA, is limited in frequency band, and is orthogonalized as a band limited signal LSB sampled at the sample rate f _SM1 . Output to the unit 112B.

In the second embodiment, the band limiting filter 111B has a component folded at an angular frequency of 0 as a result of multiplication of a sine wave (∝sin (ω _SH · t)) performed at the time of angular frequency conversion in the orthogonalizing unit 112B. The frequency band of the signal to be passed so that it does not overlap (or is sufficiently suppressed) with the result of the angular frequency shift of the pilot signal PS (in this second embodiment, the angular frequency (ω _SH −ω _C )). Limit. In addition, the band limiting filter 111B is configured so that the component folded at the angular frequency corresponding to ½ of the sampling rate does not overlap (or is sufficiently suppressed) with the result of the angular frequency shift of the pilot signal PS. Limit the frequency band of the signal to pass through. The band limiting filter 111B having such a function can be realized as an IIR band pass filter, for example.

Note that a phase shift may occur due to a filter delay through the band limiting filter 111B. In the second embodiment, the following description will be given on the assumption that the phase shift Δθ ₁ due to the filter delay occurs in the band limiting filter 111B.

In the second embodiment, the signal component PS ′ corresponding to the pilot signal PS in the band limited signal LSB output from the band limiting filter 111B configured as described above is expressed by the following equation (7). It has become.
PS ′ (t) ∝sin [θ _C (t) −Δθ ₁ ]
= Sin [(ω _C t + φ ₀ ) −Δθ ₁ ] (7)

Returning to FIG. 9, the orthogonalizing unit 112B is based on the band limited signal LSB, and is orthogonal to each other and two signals OSB ₁ and OSB ₂ having a sample rate f _SM1 reflecting the phase θ _C (t) of the pilot signal PS. Is generated. As shown in FIG. 10, the orthogonalizing unit 112B having such a function includes an oscillating unit 210 and multiplying units 220 ₁ and 220 ₂ .

The oscillation unit 210 generates a signal OTS ₁ to be supplied to the multiplication unit 220 ₁ and a signal OTS ₂ to be supplied to the multiplication unit 220 ₂ . In the second embodiment, the signal OTS ₁ and the signal OTS ₂ are represented by the following equations (8) and (9).
OTS ₁ (t) = B ₀ · cos (ω _SH · t) (8)
OTS ₂ (t) = B ₀ · sin (ω _SH · t) (9)
Here, B ₀ is a constant.

In the present embodiment, the angular frequency ω _SH is set to a predetermined value larger than 3ω _C. The value of the angular frequency ω _SH is determined from a comprehensive point of view combined with the band limit specification by the band limit filter 111B described above, from the viewpoint of preventing noise components from being mixed into the angular frequency shift result of the pilot signal PS. .

Multiplier 220 ₁ receives band limited signal LSB at the A terminal and receives signal OTS ₁ from oscillating section 210 at the B terminal. Then, the multiplication result of the reception result at the A terminal and the reception result at the B terminal is output from the C terminal to the FIL 113B ₁ as the signal OSB ₁ of the sample rate f _SM1 . Here, pilot signal PS in signal SIA has angular frequency ω _C , and signal OTS ₁ has only a component of angular frequency ω _SH . For this reason, the signal component corresponding to the pilot signal PS in the signal OSB ₁ is subjected to angular frequency conversion into two components of the angular frequency (ω _SH −ω _C ) and the angular frequency (ω _SH + ω _C ).

Multiplier 220 ₂ receives band-limited signal LSB at the A terminal and receives signal OTS ₂ from oscillating unit 210 at the B terminal. Then, the multiplication result of the reception result at the A terminal and the reception result at the B terminal is output from the C terminal to the FIL 113B ₂ as the signal OSB ₂ of the sample rate f _SM1 . Here, pilot signal PS in signal SIA has an angular frequency ω _C , and signal OTS ₂ has only a component of angular frequency ω _SH . For this reason, the signal component corresponding to the pilot signal PS in the signal OSB _{2 has two} angular components (angular frequency (ω _SH −ω _C ) and angular frequency (ω _SH + ω _C )) as in the case of the signal OSB _1. Frequency conversion is performed.

Returning to FIG. 9, the FIL 113B ₁ is configured as a digital filter such as a finite impulse response (FIR) filter. The FIL 113B ₁ receives the signal OSB ₁ from the orthogonalizing unit 112B. Then, FIL 113B ₁ selectively passes _one of the components of angular frequency (ω _SH −ω _C ) in signal OSB ₁ , that is, the signal component corresponding to pilot signal PS, as signal PSB ₁ of sample rate f _SM1. Output to the downsampling unit 115A.

In the second embodiment, a FIR low-pass filter that does not cause a phase shift is employed as the FIL 113B ₁ .

The signal PSB ₁ output from the FIL 113B ₁ is expressed by the following equation (10).
PSB ₁ (t) ∝A (t) · sin [θ _S (t)]
= A (t) · sin [ω _SH t−θ _C (t) + Δθ ₁ ]
= A (t) · sin [ω _SH t− (ω _C t + φ ₀ ) + Δθ ₁ ] (10)
Here, A (t) represents the amplitude value of the pilot signal PS.

The FIL 113B ₂ is configured in the same manner as the FIL 113B ₁ . The FIL 113B ₂ receives the signal OSB ₂ from the orthogonalizing unit 112B. Then, the FIL 113B ₂ selectively passes one of the components of the angular frequency (ω _SH −ω _C ) in the signal OSB ₂ , that is, the signal component corresponding to the pilot signal PS, as the signal PSB ₂ of the sample rate f _SM1. Output to the downsampling unit 115A.

The signal PSB ₂ output from the FIL 113B ₂ is expressed by the following equation (11).
PSB ₂ (t) ∝A (t) · cos [θ _S (t)]
= A (t) · cos [ω _SH t−θ _C (t) + Δθ ₁ ]
= A (t) · cos [ω _SH t− (ω _C t + φ ₀ ) + Δθ ₁ ] (11)

Note that the FILs 113B ₁ and 113B ₂ also function as a decimation filter for reducing the sample rate in the downsampling unit 115A described later, similarly to the FILs 113A ₁ and 113A ₂ of the first embodiment described above.

As in the case of the signals PSA ₁ and PSA ₂ in the first embodiment described above, the downsampling unit 115A reduces the respective sample rates f _SM1 of the signals PSB ₁ and PSB ₂ and samples them at the sample rate f _SM2. The signals PWB ₁ and PWB ₂ are generated. More specifically, the downsampling unit 115A, the data sequence of the signal _{PSB 1 (..., PSB 1 (} t 0), PSB 1 (t 1), ..., PSB 1 (t p), ...) from 3 Tsugoto , PSB ₁ (t ₀ ), PSB ₁ (t ₃ ),..., PSB ₁ (t _3p ,...) _Are generated, thereby generating the signal PWB ₁ . Further, down-sampling unit 115A, the data sequence of the signal _{PSB 2 (..., PSB 2 (} t 0), PSB 2 (t 1), ..., PSB 2 (t p), ...) the data selection every third from conducted, data sequence _{(..., PSB 2 (t 0} ), PSB 2 (t 3), ..., PSB 2 (t 3p), ...) by generating, for generating a signal PWB _2. The signals PWB ₁ and PWB ₂ generated in this way are output to the phase calculation unit 120B.

Returning to FIG. 8, the phase calculation unit 120B calculates the phase θ _C (t) of the pilot signal PS based on the signal PWB ₁ and the signal PWB ₂ from the quadrature signal generation unit 110B. For this calculation, the phase calculation unit 120B, for example, performs an arctan calculation on the signal PSB ₁ and the signal PSB ₂ and corrects the amount of time change due to the phase shift Δθ ₁ and the angular frequency ω _SH described above. The phase θ _C is calculated.

The calculated θ _C (t) is output from the phase calculation unit 120B to the upsampling unit 125 as the signal PHW.

<Operation>
Next, a signal processing operation in the signal processing device 100B configured as described above will be described.

When the signal SIA from the signal source 910A is received by the signal processing device 100B via the input terminal 191, the signal SIA is supplied to the orthogonal signal generation unit 110B and the signal processing unit 140 in the signal processing device 100B ( (See FIG. 8). In the orthogonal signal generation unit 110B that receives the supply of the signal SIA, first, the band limiting filter 111B selectively passes signal components in a predetermined frequency band including the angular frequency ω _C in the signal SIA, and the frequency band is limited. The band-limited signal LSB is sent to the orthogonalization unit 112B (see FIG. 9). As a result, even when the signal SIA has a wide signal component in a high angular frequency region as shown by a two-dot chain line in FIG. 11, for example, as shown in FIG. Is limited.

Upon receiving the band limited signal LSB, the orthogonalizing unit 112B generates _two signals OSB ₁ and OSB ₂ that are orthogonal to each other based on the band limited signal LSB, and sends them to the FILs 113B ₁ and 113B ₂ (see FIG. 9). Here, as shown in FIG. 13, each of the signals OSB _j (j = 1, 2) has a signal component PSM _j having an angular frequency (ω _SH −ω _C ) as a signal component corresponding to the pilot signal PS and The signal component PSP _j of the angular frequency (ω _SH + ω _C ) is included.

In the second embodiment, the orthogonalizing unit 112B performs the angular frequency conversion of the band limited signal LSB. Therefore, as shown by the two-dot chain line in FIG. 11 described above, the signal component PSM _j and the signal component PSP _j that can be generated when the angular frequency conversion of the signal SIA having a wide signal component in the high angular frequency region is performed. (See the alternate long and short dash line in FIG. 13).

Subsequently, down FIL113B ₁ which has received the signal OSB ₁ is the component of the angular frequency of the signal _{OSB 1 (ω SH -ω C)} , i.e., selectively pass a signal component corresponding to the pilot signal PS, as the signal PSB ₁ Output toward the sampling unit 115A. Also, down FIL113B ₂ which has received the signal OSB ₂ is the component of the angular frequency (ω _SH -ω _C) in the signal OSB _2, i.e., selectively pass a signal component corresponding to the pilot signal PS, as the signal PSB ₂ Output toward the sampling unit 115A (see FIG. 9).

Here, the signal PSB ₁ has a waveform represented by the above-described equation (10). On the other hand, the signal PSB ₂ has a waveform represented by the above-described equation (11).

Downsampling unit 115A which receives the signal PSB _1, PSB ₂ is as described above, by reducing the respective sample rate f _SM1 signals PSB _1, PSB _2, sample rate f _SM2 signal PWB ₁ sampled at , PWB ₂ is generated. The signals PWB ₁ and PWB ₂ generated in this way are output to the phase calculation unit 120B.

The phase calculation unit 120B that has received the signals PWB ₁ and PWB ₂ calculates the phase θ _C of the pilot signal PS based on the signal PWB ₁ and the signal PWB ₂ , the phase shift Δθ ₁ and the angular frequency ω _SH as described above. calculate. The phase θ _C calculated in this way is sent to the upsampling unit 125 as a signal PHW (see FIG. 8).

Thereafter, as in the case of the signal processing device 100A of the first embodiment, the upsampling unit 125 that has received the signal PHW increases the sample rate f _SM2 of the signal PHW, and the signal PHA sampled at the sample rate f _SM1 Is sent to the reference signal generator 130A (see FIG. 8). Subsequently, the reference signal generation unit 130A that has received the signal PHA generates a reference signal BSA having a double angular frequency in synchronization with the pilot signal PS based on the signal PHA, and sends the reference signal BSA to the signal processing unit 140 (FIG. 8). reference). Then, the signal processing unit 140 that receives the reference signal BSA from the reference signal generation unit 130A and the signal SIA from the signal source 910A processes the signal component SG2, which is an amplitude modulation signal, using the reference signal BSA. Then, the signal is demodulated into a signal in a band of the angular frequency 0 to ω _c to generate a processing result signal MSA. Then, the generated machining result signal MSA is output to the outside via the output terminal 192.

As described above, in the second embodiment, as in the case of the first embodiment, the signal SIA including the pilot signal PS of the angular frequency ω _{C is used} as a reference for the feedback loop and phase measurement in the PLL system. The phase θ _C of the pilot signal PS is derived without using the base signal. Therefore, it is possible to generate the reference signal BSA that is easily and quickly synchronized with the pilot signal PS.

In the second embodiment, as in the first embodiment, the signal SIA received at the input terminal 191 receives the sample rate f _SM2 of the signals PWB ₁ and PWB ₂ used for calculating the phase of the pilot signal PS. Since the sample rate is reduced from the sample rate f _SM1, the number of calculations with a large calculation amount such as arctan calculation in the phase calculation unit 120B can be reduced. Therefore, the total calculation amount can be reduced.

In the second embodiment, the phase signal PHW obtained at the reduced sample rate f _SM2 is increased to the sample rate f _SM1 as in the case of the first embodiment. Here, since the target for increasing the sample rate f _SM2 is the phase, it is possible to easily obtain the phase signal PHA increased to the sample rate f _SM1 without degrading the accuracy. Therefore, it is possible to obtain a reference signal BSA sample rate f _SM1 suitable for processing of the signal SIA sample rate f _SM1.

  In the second embodiment, as in the case of the first embodiment, since the signal component SG2 in the signal SIA is processed using the reference signal BSA, the signal component SG2 can be quickly and accurately processed. Desired processing can be performed.

Further, in the second embodiment, if there is a phase shift [Delta] [theta] ₁ in the band limiting filter 111B, taking into account this calculation is performed to calculate the phase theta _C of the pilot signal PS, precisely phase theta _C can do. Therefore, it is possible to generate the reference signal BSA that is accurately synchronized with the pilot signal PS.

  Further, in the second embodiment, since the orthogonalization unit 112B performs the orthogonalization while performing the frequency conversion, the amount of calculation can be reduced as compared with the case where the orthogonalization is performed using the Hilbert transform or the like.

  In the second embodiment, since the band limited signal LSB obtained by band limiting the signal SIA by the band limiting filter 111B is orthogonalized while performing angular frequency conversion, noise is mixed into the angular frequency converted result of the pilot signal PS. Can be reduced.

In the second embodiment, it is assumed that no phase shift occurs through the FIL 113B ₁ and the FIL 113B ₂ . On the other hand, for example, when the phase shift Δθ ₂ is generated through the FIL 113B ₁ and the FIL 113B ₂ , the phase compensation for the phase shift Δθ ₂ is performed as in the first embodiment. The phase calculation unit 120B may perform this.

  In the second embodiment, a filter that does not cause a phase shift may be employed as the band limiting filter 111B. In this case, it is not necessary to calculate the phase compensation corresponding to the band limiting filter 111B in the phase calculation unit 120B.

[Third Embodiment]
Next, a third embodiment of the present invention will be described with reference mainly to FIGS.

  FIG. 14 is a block diagram illustrating a schematic configuration of a signal processing device 100C according to the third embodiment. Note that the signal processing device 100C receives and processes the signal SIA output from the signal source 910A via the input terminal 191 and processes the result from the output terminal 192, similarly to the signal processing device 100B in the second embodiment. It is a device that outputs as signal MSA.

As shown in FIG. 14, the signal processing device 100C is different from the signal processing device 100B only in that an orthogonal signal generation unit 110C is provided instead of the orthogonal signal generation unit 110B. As shown in FIG. 15, in the orthogonal signal generation unit 110C, the downsampling unit 115A is disposed between the orthogonalization unit 112B and the FIL 113B ₁ and FIL 113B _{2 as} compared to the orthogonal signal generation unit 110B. Only the point is different. Hereinafter, the description will be given mainly focusing on such differences.

The down-sampling unit 115A in the third embodiment generates the signals OWB ₁ and OWB ₂ sampled at the sample rate f _SM2 by reducing the sample rate f _SM1 of the orthogonalized signals OSB ₁ and OSB ₂ . More specifically, the present down-sampling section 115A in the third embodiment, the signal OSB ₁ data string _{(..., OSB 1 (t 0} ), OSB 1 (t 1), ..., OSB 1 (t p), ..) Is selected every three, and a data string (..., OSB ₁ (t ₀ ), OSB ₁ (t ₃ ),..., OSB ₁ (t _3p ),. Generate ₁ Further, down-sampling unit 115A, the data string signal _{OSB 2 (..., OSB 2 (} t 0), OSB 2 (t 1), ..., OSB 2 (t p), ...) the data selection every third from Then, a signal OWB ₂ is generated by generating a data string (..., OSB ₂ (t ₀ ), OSB ₂ (t ₃ ),..., OSB ₂ (t _3p ),. The signals OWB ₁ and OWB ₂ thus generated are output toward the FILs 113B ₁ and 113B ₂ .

As a result, the signals PWB ₁ and PWB ₂ are generated by passing the signals OWB ₁ and OWB ₂ through the FIL 113 B ₁ and FIL 113 B ₂ . That is, the orthogonal signal generation unit 110C generates the signals PWB ₁ and PWB ₂ based on the signal SIA from the signal source 910A as in the case of the orthogonal signal generation unit 110B.

  In the third embodiment, the band limiting filter 111B also functions as a decimation filter for reducing the sample rate in the downsampling unit 115A.

<Operation>
Next, a signal processing operation in the signal processing device 100C configured as described above will be described.

When the signal SIA from the signal source 910A is received by the signal processing device 100C via the input terminal 191, the signal SIA is supplied to the orthogonal signal generation unit 110C and the signal processing unit 140 in the signal processing device 100C ( (See FIG. 14). In the orthogonal signal generation unit 110C that receives the supply of the signal SIA, as described above, the band limiting filter 111B, the orthogonalization unit 112B, the downsampling unit 115A, and the FILs 113B ₁ and 113B ₂ perform processing in sequence, and the signal PWB ₁ , PWB ₂ is generated.

  Thereafter, similarly to the case of the second embodiment, the processing by the phase calculation unit 120B, the upsampling unit 125, the reference signal generation unit 130A, and the signal processing unit 140 is sequentially performed, and the processing result signal MSA is generated. The machining result signal MSA generated in this way is output to the outside via the output terminal 192.

  As described above, according to the third embodiment, the same effects as those of the second embodiment described above can be obtained.

[Fourth Embodiment]
Next, a fourth embodiment of the present invention will be described with reference mainly to FIGS.

  FIG. 16 is a block diagram illustrating a schematic configuration of a signal processing device 100D according to the fourth embodiment. Note that the signal processing device 100D receives and processes the signal SIA output from the signal source 910A via the input terminal 191 and processes the result from the output terminal 192, similarly to the signal processing device 100B in the second embodiment. It is a device that outputs as signal MSA.

  As shown in FIG. 16, the signal processing device 100D is different from the signal processing device 100B only in that it includes an orthogonal signal generation unit 110D instead of the orthogonal signal generation unit 110B. As shown in FIG. 17, the orthogonal signal generation unit 110D is arranged between the band limiting filter 111B and the orthogonalization unit 112B, instead of the downsampling unit 115A, as compared to the orthogonal signal generation unit 110B. The only difference is that the downsampling unit 115D is provided. Hereinafter, the description will be given mainly focusing on such differences.

The downsampling unit 115D in the fourth embodiment reduces the sample rate f _SM1 of the signal LSB and generates the signal LSW sampled at the sample rate f _SM2 . More specifically, the down-sampling unit 115D according to the fourth embodiment is configured so that the signal LSB data string (..., LSB (t ₀ ), LSB (t ₁ ),..., LSB (t _p ),. The data LSW is generated by selecting each data and generating a data string (..., LSB (t ₀ ), LSB (t ₃ ),..., LSB (t _3p ),. The signal LSW of the sample rate f _SM2 generated in this way is output toward the orthogonalizing unit 112B.

As a result, the orthogonalization unit 112B that has received the signal LSW generates orthogonalized signals OWB ₁ and OWB ₂ having the sample rate f _SM2 . By orthogonalizing signal OWB ₁ is via the FIL113B _1, together with the signal PWB ₁ is generated, orthogonalized signal OWB ₂ is by through FIL113B _2, signal PWB ₂ is generated.

That is, the orthogonal signal generation unit 110D generates the signals PWB ₁ and PWB ₂ based on the signal SIA from the signal source 910A as in the case of the orthogonal signal generation unit 110B.

  In the fourth embodiment, the band limiting filter 111B also functions as a decimation filter for reducing the sample rate in the downsampling unit 115D.

<Operation>
Next, a signal processing operation in the signal processing device 100D configured as described above will be described.

When the signal SIA from the signal source 910A is received by the signal processing device 100D via the input terminal 191, the signal SIA is supplied to the orthogonal signal generation unit 110D and the signal processing unit 140 in the signal processing device 100D ( FIG. 16). In the orthogonal signal generation unit 110D that receives the supply of the signal SIA, the processing is sequentially performed by the band limiting filter 111B, the downsampling unit 115D, the orthogonalization unit 112B, and the FILs 113B ₁ and 113B ₂ as described above, and the signals PWB ₁ , PWB ₂ is generated.

  Thereafter, similarly to the case of the second embodiment, the processing by the phase calculation unit 120B, the upsampling unit 125, the reference signal generation unit 130A, and the signal processing unit 140 is sequentially performed, and the processing result signal MSA is generated. The machining result signal MSA generated in this way is output to the outside via the output terminal 192.

  As described above, according to the fourth embodiment, the same effects as those of the second embodiment described above can be obtained.

[Fifth Embodiment]
Next, a fifth embodiment of the present invention will be described with reference mainly to FIGS.

  FIG. 18 is a block diagram illustrating a schematic configuration of a signal processing device 100E according to the fifth embodiment. Note that the signal processing device 100E receives and processes the signal SIA output from the signal source 910A via the input terminal 191 and processes the result from the output terminal 192 in the same manner as the signal processing device 100B in the second embodiment. It is a device that outputs as signal MSA.

  As shown in FIG. 18, the signal processing device 100E includes an orthogonal signal generation unit 110E instead of the orthogonal signal generation unit 110B and a phase calculation instead of the phase calculation unit 120B, as compared with the signal processing device 100B described above. Only the point provided with the part 120E is different. Hereinafter, description will be made mainly focusing on these differences.

  In the fifth embodiment, the signal processing unit 140 functions as a second type signal processing means.

As shown in FIG. 19, the orthogonal signal generation unit 110E includes a band limitation filter 111E instead of the band limitation filter 111B as compared with the above-described orthogonal signal generation unit 110B, and the FILs 113B ₁ and 113B ₂ are replaced. The difference is that FILs 113E ₁ and 113E ₂ are provided as filter means. The only difference from the orthogonal signal generating unit 110B is that the band limited signal LSE output from the band limiting filter 111E is sent to the orthogonalizing unit 112B and also sent to the signal processing unit 140.

The band limiting filter 111E is configured as a digital filter such as a finite impulse response (FIR) filter. The band limiting filter 111E is a signal in a predetermined frequency band including the angular frequency ω _C in the signal SIA and the frequency band (angular frequencies ω _{c to} 3ω _c ) of the signal component SG2 in the signal SIA to be processed in the signal processing unit 140. The components are selectively passed and output to the orthogonalization unit 112B and the signal processing unit 140 as a band limited signal LSE with a limited frequency band.

In the fifth embodiment, the band limiting filter 111E has a component folded back at an angular frequency of 0 as a result of multiplication of a sine wave (∝sin (ω _SH · t)) performed at the time of angular frequency conversion in the orthogonalizing unit 112B. The frequency band of the signal to be passed so that it does not overlap (or is sufficiently suppressed) as a result of the angular frequency shift of the pilot signal PS (in this fifth embodiment, the angular frequency (ω _SH −ω _C )). Limit. Further, the band limiting filter 111E is configured so that the component folded at the angular frequency corresponding to 1/2 of the sample rate does not overlap (or is sufficiently suppressed) with the result of the angular frequency shift of the pilot signal PS. Limit the frequency band of the signal to pass through. Furthermore, the band limiting filter 111E has a characteristic that the signal transmittance is flat at the angular frequencies ω _{c to} 3ω _c . The band limiting filter 111E having such a function can be realized as, for example, an FIR low pass filter.

  In the fifth embodiment, it is assumed that a phase shift due to passing through the band limiting filter 111E does not occur.

The orthogonalizing unit 112B operates in the same manner as in the second embodiment. That is, the orthogonalizing unit 112B generates _two signals OSE ₁ and OSE ₂ that are orthogonal to each other and reflect the phase θ _C (t) of the pilot signal PS based on the band limited signal LSE. Then, the orthogonalizing unit 112B outputs the signals OSE ₁ and OSE ₂ to the FILs 113E ₁ and 113E ₂ .

The FIL 113E ₁ is configured as a digital filter such as an infinite impulse response (IIR) filter. The FIL 113E ₁ receives the signal OSE ₁ from the orthogonalizing unit 112B. The FIL 113E ₁ selectively passes _one of the signal components corresponding to the pilot signal PS in the signal OSE ₁ , and the angular frequency (ω _SH −ω _C ) component in the fifth embodiment, and is down-converted as the signal PSE _1. Output toward the sampling unit 115A. As the FIL 113E ₁ , for example, an IIR band-pass filter can be employed.

Note that a fixed phase shift Δθ ₂ associated with the filtering calculation may occur through the FIL 113E ₁ . In the fifth embodiment, it is assumed that such a phase shift Δθ ₂ occurs.

The signal PSE ₁ output from the FIL 113E ₁ is expressed by the following equation (12).
PSE ₁ (t) ∝A (t) · sin [θ _S (t)]
= A (t) · sin [ω _SH t−θ _C (t) −Δθ ₂ ]
= A (t) · sin [ω _SH t− (ω _C t + φ ₀ ) −Δθ ₂ ] (12)
Here, A (t) represents the amplitude value of the pilot signal PS.

The FIL 113E ₂ is configured in the same manner as the FIL 113E ₁ . The FIL 113E ₂ receives the signal OSE ₂ from the orthogonalizing unit 112B. The FIL 113E ₂ selectively passes one of the signal components corresponding to the pilot signal PS in the signal OSE ₂ while the angular frequency (ω _SH −ω _C ) component is passed in the fifth embodiment, thereby reducing the signal PSE ₂ . Output toward the sampling unit 115A.

The signal PSE ₂ output from the FIL 113E ₂ is expressed by the following equation (13).
PSE ₂ (t) ∝A (t) · cos [θ _S (t)]
= A (t) · cos [ω _SH t−θ _C (t) −Δθ ₂ ]
= A (t) · cos [ω _SH t− (ω _C t + φ ₀ ) −Δθ ₂ ] (13)

As in the case of the signals PSB ₁ and PSB ₂ in the second embodiment described above, the downsampling unit 115A reduces the respective sample rates f _SM1 of the signals PSE ₁ and PSE ₂ and samples them at the sample rate f _SM2. The signals PWE ₁ and PWE ₂ are generated. More specifically, the downsampling unit 115A, the data sequence of the signal _{PSE 1 (..., PSE 1 (} t 0), PSE 1 (t 1), ..., PSE 1 (t p), ...) from 3 Tsugoto , PSE ₁ (t ₀ ), PSE ₁ (t ₃ ),..., PSE ₁ (t _3p ,...) Are generated to generate the signal PWE ₁ . Further, the downsampling unit 115A selects data every three from the data sequence of the signal PSE ₂ (..., PSE ₂ (t ₀ ), PSE ₂ (t ₁ ),..., PSE ₂ (t _p ),. conducted, data sequence _{(..., PSE 2 (t 0} ), PSE 2 (t 3), ..., PSE 2 (t 3p), ...) by generating, for generating a signal PWE _2. The signals PWE ₁ and PWE ₂ thus generated are output toward the phase calculation unit 120E.

Returning to FIG. 18, the phase calculation unit 120E calculates the phase θ _C (t) of the pilot signal PS based on the signal PSE ₁ and the signal PSE ₂ from the orthogonal signal generation unit 110E. In such a calculation, the phase calculation section 120E, for example, the signal PSE ₁ and the signal PSE ₂ after performing operations such as arctan, by correcting the contribution of the phase shift [Delta] [theta] ₂ and the angular frequency omega _SH mentioned above, the phase Calculate θ _C.

The thus calculated θ _C (t) is output from the phase calculation unit 120E to the upsampling unit 125 as the signal PHW.

<Operation>
Next, a signal processing operation in the signal processing device 100E configured as described above will be described.

When the signal SIA from the signal source 910A is received by the signal processing device 100E via the input terminal 191, the signal SIA is supplied to the orthogonal signal generation unit 110E (see FIG. 19). In the orthogonal signal generation unit 110E that receives the supply of the signal SIA, first, the band limiting filter 111E has a predetermined frequency including the angular frequency ω _C in the signal SIA and the angular frequency band (angular frequencies ω _{C to} 3ω _C ) of the signal component SG2. The signal component of the band is selectively passed and sent to the orthogonalization unit 112B and the signal processing unit 140 as a band limited signal LSE whose frequency band is limited (see FIG. 19).

The orthogonalizing unit 112B that has received the band limited signal LSE generates _two signals OSE ₁ and OSE ₂ that are orthogonal to each other based on the band limited signal LSE, and sends them to the FILs 113E ₁ and 113E ₂ (see FIG. 19). Here, each of the signals OSE _j (j = 1, 2) is an angular frequency (ω _SH −ω) as a signal component corresponding to the pilot signal PS, similarly to the signal OSB _j in the case of the second embodiment. _C ) signal component PSM _j and angular frequency (ω _SH + ω _C ) signal component PSP _j (see FIG. 13).

In the fifth embodiment, the orthogonalizing unit 112B performs the angular frequency conversion of the band limited signal LSE. For this reason, as in the case of the second embodiment, noise is mixed into the signal component PSM _j and the signal component PSP _j that may be generated when the angular frequency conversion of the signal SIA having a wide signal component in the high angular frequency region is performed. Can be prevented (see FIGS. 11 to 13 referred to in the second embodiment).

Subsequently, down FIL113E ₁ which has received the signal OSE ₁ is the component of the angular frequency of the signal _{OSE 1 (ω SH -ω C)} , i.e., selectively pass a signal component corresponding to the pilot signal PS, as the signal PSE ₁ Output toward the sampling unit 115A. Also, down FIL113E ₂ which has received the signal OSE ₂ is the component of the angular frequency (ω _SH -ω _C) in the signal OSE _2, i.e., selectively pass a signal component corresponding to the pilot signal PS, as the signal PSE ₂ It outputs toward the sampling part 115A (refer FIG. 19).

Here, the signal PSE ₁ has a waveform represented by the above-described equation (12). On the other hand, the signal PSE ₂ has a waveform represented by the above-described equation (13).

Upon receiving the signals PSE ₁ and PSE ₂ , the downsampling unit 115A reduces the respective sample rates f _SM1 of the signals PSE ₁ and PSE ₂ as described above, and the signal PWE ₁ sampled at the sample rate f _SM2. , PWE ₂ is generated. The signals PWE ₁ and PWE ₂ thus generated are output toward the phase calculation unit 120E (see FIG. 19).

The phase calculation unit 120E that has received the signals PWE ₁ and PWE ₂ calculates the phase θ _C of the pilot signal PS based on the signals PWE ₁ and PWE ₂ , the phase shift Δθ ₂ and the angular frequency ω _SH as described above. calculate. The phase θ _C calculated in this way is sent to the upsampling unit 125 as a signal PHW (see FIG. 18).

  Thereafter, similarly to the case of the second embodiment, the processing by the upsampling unit 125, the reference signal generation unit 130A, and the signal processing unit 140 is sequentially performed, and the processing result signal MSA is generated. The machining result signal MSA generated in this way is output to the outside via the output terminal 192.

As described above, in the fifth embodiment, as in the case of the second embodiment, the signal SIA including the pilot signal PS having the angular frequency ω _{C is used} as a reference for the feedback loop and phase measurement in the PLL system. The phase θ _C of the pilot signal PS is derived without using the base signal. Therefore, it is possible to generate the reference signal BSA that is easily and quickly synchronized with the pilot signal PS.

  In the fifth embodiment, as in the case of the second embodiment, the signal component SG2 in the signal SIA is processed using the reference signal BSA, so that the signal component SG2 can be quickly and accurately processed. Desired processing can be performed.

In the fifth embodiment, similarly to the second embodiment, the signal SIA received at the input terminal 191 receives the sample rate f _SM2 of the signals PWE ₁ and PWE ₂ used for calculating the phase of the pilot signal PS. Since the sample rate is reduced from the sample rate f _SM1, the number of calculations with a large calculation amount such as arctan calculation in the phase calculation unit 120B can be reduced. Therefore, the total calculation amount can be reduced.

In the fifth embodiment, as in the second embodiment, the phase signal PHW obtained at the reduced sample rate f _SM2 is increased to the sample rate f _SM1 . Here, since the target for increasing the sample rate f _SM2 is the phase, it is possible to easily obtain the phase signal PHA increased to the sample rate f _SM1 without degrading the accuracy. Therefore, it is possible to obtain a reference signal BSA sample rate f _SM1 suitable for processing of the signal SIA sample rate f _SM1.

Further, in the fifth embodiment, as in the case of the first embodiment, the phase θ _C of the pilot signal PS is calculated in consideration of the phase shift Δθ ₂ in the FILs 113E ₁ and 113E ₂ , so that the phase is accurately determined. θ _C can be calculated. Therefore, it is possible to generate the reference signal BSA that is accurately synchronized with the pilot signal PS.

  Further, in the fifth embodiment, as in the case of the second embodiment, since orthogonalization is performed while performing frequency conversion in the orthogonalization unit 112D, the amount of calculation is larger than that in the case of orthogonalization using Hilbert transform or the like. Can be reduced.

  Further, in the fifth embodiment, as in the case of the second embodiment, the band limited signal LSE obtained by band limiting the signal SIA by the band limiting filter 111E is orthogonalized while performing angular frequency conversion, so that the pilot signal PS It is possible to reduce noise contamination in the angular frequency conversion result.

In the fifth embodiment, no phase shift occurs due to the band limiting filter 111E. On the other hand, when the phase shift Δθ ₁ occurs through the band limiting filter 111E, the phase calculation unit 120E performs phase compensation for the phase shift Δθ ₁ in the same manner as in the second embodiment. To do. Then, instead of the reference signal generation unit 130A, a reference signal generation unit 130E having the configuration shown in FIG. 20 may be provided.

  Based on the signal PHA from the upsampling unit 125, the reference signal generation unit 130E generates a reference signal BSE whose angular frequency is twice in synchronization with the pilot signal PS. The reference signal generation unit 130E having such a function is different from the reference signal generation unit 130A described above only in that a phase processing unit 131E is provided instead of the phase processing unit 131A.

The phase processing unit 131E receives the signal PHA from the upsampling unit 125 and processes the phase θ _C (t) indicated by the signal PHA. At the time of such processing, the phase processing unit 131E calculates the phase θ _M (t) by the following equation (14).
θ _M (t) = 2 (θ _C (t) −Δθ ₁ )
= 2 (ω _C t + φ ₀ −Δθ ₁ ) (14)

The phase θ _M (t) calculated in this way is output toward the signal generator 132 as the phase processing signal MPE. Upon receiving this phase processing signal MPE, the signal generator 132 generates a reference signal BSE. The signal generator 132 includes a sine value table in which amplitude values corresponding to the phase values are registered, and generates a sine wave signal having the phase θ _M (t) indicated by the phase processing signal MPE as the reference signal BSE. .

This reference signal BSE is expressed by the following equation (15).
BSE (t) = C ₀ · sin [θ _M (t)]
= C ₀ · sin [2 (θ _C (t) −Δθ ₁ )]
= C ₀ · sin [2 (ω _C t + φ ₀ −Δθ ₁ )] (15)

The reference signal BSE generated in this way is output from the reference signal generation unit 130E toward the signal processing unit 140. As a result, the signal processing unit 140 processes the signal component SG2 that is an amplitude-modulated signal using the reference signal BSE, and demodulates the signal component into a band of the angular frequency 0 to ω _c .

In the fifth embodiment, as the FIL 113E ₁ and FIL 113E ₂ , filters that do not cause a phase shift can be employed. In this case, the calculation of the phase compensation corresponding to FIL113E _1, FIL113E ₂ in the phase calculator 120E becomes unnecessary.

In the fifth embodiment, the output signal LSE from the band limiting filter 111E is supplied to the signal processing unit 140 as it is. On the other hand, after using the band limiting filter 111E as a decimation filter, the sample rate f _SM1 of the signal LSE is reduced, and the signal LWE of the sample rate f _SM3 (where f _SM2 <f _SM3 <f _SM1 ) is obtained. A downsampling unit to be generated may be newly provided, and the signal LWE may be supplied to the signal processing unit 140. In this case, the upsampling unit 125 may change the sample rate f _SM2 to the sample rate f _SM3 .

[Sixth Embodiment]
Next, a sixth embodiment of the present invention will be described with reference mainly to FIGS.

<Configuration>
FIG. 21 is a block diagram illustrating a schematic configuration of a signal processing device 100F according to the sixth embodiment. The signal processing device 100F is a device that receives and processes the signal SIF output from the signal source 910F via the input terminal 191, and outputs the processed result signals MSA and MSF from the output terminals 192 ₁ and 192 _2. .

In the sixth embodiment, as shown in FIG. 22, the signal SIF includes an angular frequency ω _{c in addition to} the pilot signal PS of the angular frequency ω _c and the signal component SG2 in the frequency band of the angular frequencies ω _{c to} 3ω _c. It is assumed that the signal component SG3 in the frequency band of 3ω _{c to} 5ω _c is further included. Here, the signal component SG3 is the band of the signal of the angular frequency 0～Omega _c, pilot signal signal (.alpha.sin of four times the angular frequency of the _{PS [4 (ω c t +} φ 0))]) the amplitude modulated signal Suppose that This in the sixth embodiment, the signal processing apparatus 100F, based on the pilot signal PS, the reference signal _{BSA (αsin [2 (ω c} t + φ 0)]) of angular frequency 2 [omega _c and the reference signal of the angular frequency 4Omega _c BSF (∝sin [4 (ω _c t + φ ₀₎ )]) is generated. The signal processing device 100F demodulates the amplitude modulation signal SG2 using the reference signal BSA, and demodulates the amplitude modulation signal SG3 using the reference signal BSF.

As shown in FIG. 21, the signal processing device 100F includes a reference signal generation unit 130F instead of the reference signal generation unit 130A as compared with the above-described signal processing device 100A, and two signal processing units as signal processing means. The only difference is that the portions 140 ₁ and 140 ₂ are provided. Hereinafter, description will be made mainly focusing on these differences.

  The reference signal generation unit 130F is synchronized with the pilot signal based on the signal PHA from the upsampling unit 125, and is a reference signal BSA whose angular frequency is twice and a reference whose frequency is four times that of the pilot signal. A signal BSF is generated. As shown in FIG. 23, the reference signal generation unit 130F having such a function is different from the reference signal generation unit 130A described above only in that it further includes a phase processing unit 131F and a signal generation unit 133. .

The phase processing unit 131F receives the signal PHA from the upsampling unit 125 and processes the phase θ _C (t) indicated by the signal PHA. In the sixth embodiment, the phase processing unit 131F calculates the phase θ _N (t) by the following equation (16).
θ _N (t) = 4θ _C (t) = 4 (ω _C t + φ ₀ ) (16)

That is, in the sixth embodiment, the phase processing unit 131F calculates the phase θ _N (t) that changes at the angular frequency 4ω _C. The phase θ _N (t) calculated in this way is output toward the signal generator 133 as the phase processing signal MPF.

  The signal generator 133 generates the reference signal BSF based on the phase processing signal MPF from the phase processing unit 131F. The signal generator 133 is configured in the same manner as the signal generator 132.

The reference signal BSF generated in this way is expressed by the following equation (17).
BSF (t) = C ₀ · sin [θ _N (t)]
= C ₀ · sin [4θ _C (t)]
= C ₀ · sin [4 (ω _C t + φ ₀ )] (17)

Reference signal BSF thus generated is outputted to the reference signal generator 130F to the signal processing unit 140 _2. The reference signal BSA generated by using the phase processing unit 131A and the signal generating unit 132 in the reference signal generator 130F is outputted to the reference signal generator 130F to the signal processing unit 140 _1.

Each of the signal processing units 140 ₁ and 140 ₂ is configured in the same manner as the signal processing unit 140 described above. Here, the signal processing unit 140 ₁ receives the signal SIF via the input terminal 191 and the reference signal BSA from the reference signal generation unit 130F. Then, the signal processing unit 140 ₁ processes the signal component SG2 that is an amplitude modulation signal using the reference signal BSA, and demodulates the signal component into a signal in the band of the angular frequency 0 to ω _c . The machining result signal MSA generated in this way is output to the outside via the output terminal 192 ₁ .

The signal processing unit 140 ₂ receives the signal SIF via the input terminal 191 and the reference signal BSF from the reference signal generation unit 130F. Then, the signal processing unit 140 _2, a signal component SG3 is an amplitude modulated signal is processed by using the reference signal BSF, demodulates the band of the signal of the angular frequency 0~ω _c. The machining result signal MSF generated in this way is output to the outside via the output terminal 192 ₂ .

<Operation>
Next, a signal processing operation in the signal processing device 100F configured as described above will be described.

When the signal SIF from the signal source 910F is received by the signal processing apparatus 100F via the input terminal 191, the signal SIF is supplied to the orthogonal signal generation unit 110A and the signal processing units 140 ₁ and 140 ₂ (FIG. 21). reference). The orthogonal signal generation unit 110A that has received the signal SIF uses the orthogonalization unit 112A, the FILs 113A ₁ and 113A _2, and the downsampling unit 115A in the same manner as in the first embodiment described above, and the signals PWA ₁ , PWA ₂ is generated. The signals PWA ₁ and PWA ₂ generated in this way are sent to the phase calculation unit 120A.

The phase calculation unit 120A that receives the signals PWA ₁ and PWA ₂ calculates the phase θ _C of the pilot signal PS in the same manner as in the first embodiment. The phase θ _C calculated in this way is sent to the upsampling unit 125 as a signal PHW (see FIG. 21). Subsequently, the upsampling unit 125 that has received the signal PHW increases the sample rate f _SM2 of the signal PHW, generates the signal PHA sampled at the sample rate f _SM1 , and sends it to the reference signal generation unit 130F (see FIG. 21). ).

The reference signal generator 130F that has received the signal PHA generates a reference signal BSA having an angular frequency of 2ω _C and a reference signal BSF having an angular frequency of 4ω _C based on the signal PHA. When generating the reference signal BSA, the reference signal generation unit 130F uses the phase processing unit 131A and the signal generation unit 132 in the same manner as the reference signal generation unit 130A described above. Is sent to the signal processing unit 140 ₁ .

When generating the reference signal BSF, in the reference signal generation unit 130F, the phase processing unit 131F calculates the phase θ _N (t) having the angular frequency 4ω _C by the above-described equation (16), and the signal generation unit 133 (see FIG. 23). Signal generating unit 133 which received the phase theta _N a (t) refers to the internal sine table based on the phase theta _N (t), it is a sine wave signal represented by the above-described (17) criteria A signal BSF is generated. The reference signal BSF generated in this way is sent to the signal processing unit 140 ₂ (see FIG. 23).

The signal processing unit 140 _{1 that} has received the reference signal BSA from the reference signal generation unit 130F and the signal SIF from the signal source 910F is a signal component that is an amplitude modulation signal in the same manner as the signal processing unit 140 described above. the SG2, and processed using a reference signal BSA, demodulates the band of the signal of the angular frequency 0~ω _c. Thus, corresponding to the signal component SG2, as shown in FIG. 24 (again shown Figure 7), processing result signal MSA band angular frequency 0～Omega _C is generated via the output terminal 192 _1, to the outside Is output.

The signal processing unit 140 ₂ that receives the reference signal BSF from the reference signal generation unit 130F and the signal SIF from the signal source 910F processes the signal component SG3, which is an amplitude modulation signal, using the reference signal BSF, demodulating the band of the signal of the angular frequency 0~ω _c. In such processing, in the signal processing unit 140 ₂ , the multiplication unit 141 in the signal processing unit 140 ₂ first multiplies the signal SIF and the reference signal BSF to generate a mixed signal MXF. Here, as described above, the signal component SG3 in the signal SIF has an angular frequency component of angular frequencies 3ω _{C to} 5ω _C , and the reference signal BSF has only an angular frequency component of 4ω _C. signal component corresponding to the signal component SG3 in the signal MXF is a component of the angular frequency 0～Omega _C, is the angular frequency converted into a component corresponding to the angular frequency 7ω _C ~9ω _C. Mixed signal MXF thus generated is sent towards the LPF142 signal processing unit 140 _2.

Receiving the mixed signal MXF, the LPF 142 selectively passes the signal component of the angular frequency 0 to ω _C. As a result, as shown in FIG. 28 corresponding to the signal component SG3, a processing result signal MSF in the frequency band of the angular frequency 0 to ω _C is generated and output to the outside via the output terminal 192 _2. .

As described above, in the sixth embodiment, as in the case of the first embodiment, a feedback loop in the PLL system and a reference for phase measurement are obtained from the signal SIF including the pilot signal PS of the angular frequency ω _C. The phase θ _C of the pilot signal PS is derived without using the base signal. Therefore, the reference signals BSA and BSF that are synchronized with the pilot signal PS can be generated easily and quickly.

  In the sixth embodiment, since the signal components SG2 and SG3 in the signal SIF are processed using the reference signals BSA and BSF, the signal components SG2 and SG3 can be processed quickly and accurately. It can be performed.

In the sixth embodiment, as in the first embodiment, the signal SIA received at the input terminal 191 receives the sample rate f _SM2 of the signals PWA ₁ and PWA ₂ used for calculating the phase of the pilot signal PS. Since the sample rate is reduced from the sample rate f _SM1, the number of calculations with a large calculation amount such as arctan calculation in the phase calculating unit 120A can be reduced. Therefore, the total calculation amount can be reduced.

In the sixth embodiment, as in the case of the first embodiment, the phase signal PHW obtained at the reduced sample rate f _SM2 is increased to the sample rate f _SM1 . Here, since the target for increasing the sample rate f _SM2 is the phase, it is possible to easily obtain the phase signal PHA increased to the sample rate f _SM1 without degrading the accuracy. Therefore, the reference signal BSA sample rate f _SM1 suitable for processing of the signal SIF sample rate f _SM1, it is possible to obtain a BSF.

In the sixth embodiment, similarly to the first embodiment, when there is a phase shift Δθ in the FILs 113A ₁ and 113A ₂ , the phase θ _C of the pilot signal PS is calculated in consideration of this. Therefore, the phase θ _C can be calculated with high accuracy. Therefore, it is possible to generate the reference signals BSA and BSF that are accurately synchronized with the pilot signal PS.

In the sixth embodiment, the reference signal BSA and the reference signal BSF have the same sample rate. On the other hand, an upsampling unit may be prepared for each of the reference signal BSA and the reference signal BSF so that the sample rate of the reference signal BSA is different from the sample rate of the reference signal BSF. In this case, it is possible to employ an optimum sample rate for processing in each of the signal processing unit 140 ₁ and the signal processing unit 140 ₂ .

[Seventh Embodiment]
Next, a seventh embodiment of the present invention will be described with reference mainly to FIGS.

<Configuration>
FIG. 26 is a block diagram illustrating a schematic configuration of a signal processing device 100G according to the seventh embodiment. The signal processing device 100G is a device that receives and processes the signal SIA output from the signal source 910A through the input terminal 191 and outputs the processed signal MSG from the output terminal 192.

  As shown in FIG. 26, the signal processing device 100G includes a reference signal generation unit 130G instead of the reference signal generation unit 130A and further includes a phase offset setting unit 150, as compared with the signal processing device 100A described above. Only is different. Hereinafter, description will be made mainly focusing on these differences.

The reference signal generation unit 130G generates a reference signal BSG having an angular frequency 2ω _C based on the signal PHA from the upsampling unit 125. As shown in FIG. 27, the reference signal generation unit 130G having such a function includes an adder 135 disposed between the phase processing unit 131A and the signal generation unit 132, as compared with the reference signal generation unit 130A described above. In addition.

The adding unit 135 receives the phase processing signal MPA from the phase processing unit 131A and the offset phase signal POS from the phase offset setting unit 150. Then, the adding unit 135 adds the phase θ _M (t) reported by the phase processing signal MPA and the offset phase Δφ designated by the offset phase signal POS, and calculates the phase θ _P.

Here, the phase θ _P is expressed by the following equation (18).
θ _P (t) = θ _M (t) + Δφ = 2θ _C (t) + Δφ
= 2 (ω _C t + φ ₀ ) + Δφ (18)

The phase θ _P (t) calculated in this way is output to the signal generator 132 as the phase processing signal MPG. As a result, the signal generator 132 generates the reference signal BSG represented by the following equation (19).
BSG (t) = C ₀ · sin [θ _P (t)]
= C ₀ · sin [2θ _C (t) + Δφ]
= C ₀ · sin [2 (ω _C t + φ ₀ ) + Δφ] (19)

  The reference signal BSG generated in this way is output from the reference signal generation unit 130G toward the signal processing unit 140.

Returning to FIG. 26, the phase offset setting unit 150 receives the signal SIA via the input terminal 191. Then, the phase offset setting unit 150 calculates the offset phase Δφ according to a predetermined algorithm based on the signal level of the signal SIA and the level of noise that is an angular frequency component higher than the angular frequency 3ω _C in the signal SIA. To do. The offset phase Δφ thus calculated is output to the reference signal generation unit 130G as the offset phase signal POS.

<Operation>
Next, a signal processing operation in the signal processing device 100G configured as described above will be described.

  When the signal SIA from the signal source 910A is received by the signal processing apparatus 100G via the input terminal 191, the signal SIA is supplied to the orthogonal signal generation unit 110A, the signal processing unit 140, and the phase offset setting unit 150 ( (See FIG. 26). The phase offset setting unit 150 that receives the supply of the signal SIA calculates the offset phase Δφ based on the signal level and noise level of the signal SIA as described above, and outputs the offset phase Δφ to the reference signal generation unit 130G as the offset phase signal POS. Send.

In addition, the orthogonal signal generation unit 110A that has received the signal SIA uses the orthogonalization unit 112A, the FILs 113A ₁ and 113A _2, and the downsampling unit 115A in the same manner as in the first embodiment described above, and the signal PWA ₁ and PWA ₂ are generated. The signals PWA ₁ and PWA ₂ generated in this way are sent to the phase calculation unit 120A.

The phase calculation unit 120A that receives the signals PWA ₁ and PWA ₂ calculates the phase θ _C of the pilot signal PS in the same manner as in the first embodiment. The phase θ _C calculated in this way is sent to the upsampling unit 125 as a signal PHW (see FIG. 26). Subsequently, the upsampling unit 125 that has received the signal PHW increases the sample rate f _SM2 of the signal PHW, generates the signal PHA sampled at the sample rate f _SM1 , and sends it to the reference signal generation unit 130G (see FIG. 26). ).

The reference signal generator 130G that has received the signal PHA generates a reference signal BSG having an angular frequency of 2ω _C based on the signal PHA and the offset phase signal POS from the phase offset setting unit 150. When generating the reference signal BSG, in the reference signal generation unit 130G, the phase processing unit 131A has the angular frequency 2ω _{C according} to the above-described equation (5), as in the case of the reference signal generation unit 130A. The phase θ _M (t) is calculated and sent to the adder 135 as the phase processing signal MPA (see FIG. 27).

The adding unit 135 that has received the phase θ _M adds the offset phase Δφ specified by the offset phase signal POS to the phase θ _M according to the above-described equation (18), and calculates the phase θ _P. The phase θ _P calculated in this way is sent to the signal generator 132 (see FIG. 27).

Signal generating unit 132 which receives the phase theta _P refers to the interior of the sine value table based on the phase theta _P, to generate a reference signal BSG is a sine wave signal represented by the above-described (19). The reference signal BSG generated in this way is sent to the signal processing unit 140 (see FIG. 27).

The signal processing unit 140 that receives the reference signal BSG from the reference signal generation unit 130G and the signal SIA from the signal source 910A uses the reference signal BSG as a reference signal for the signal component SG2 that is an amplitude modulation signal in the signal SIA. The signal is processed using BSG and demodulated into a signal in a band of angular frequency 0 to ω _c . The machining result signal MSG generated in this way is proportional to the value of cos (Δφ) as represented by the following equation (20).
MSG (t) ∝cos (Δφ) (20)

Therefore, the processing result signal MSG, as shown in FIG. 28, which has a signal component of the angular frequency band of the angular frequency 0～Omega _c, by changing the set value of the offset phase [Delta] [phi, changes the amplitude value Signal. The machining result signal MSG generated in this way is output to the outside via the output terminal 192.

As described above, in the seventh embodiment, as in the case of the first embodiment, a feedback loop in the PLL system and a reference for phase measurement are obtained from the signal SIA including the pilot signal PS of the angular frequency ω _C. The phase θ _C of the pilot signal PS is derived without using the base signal. Therefore, the reference signal BSG that is synchronized with the pilot signal PS can be generated easily and quickly.

In the seventh embodiment, as in the first embodiment, the signal SIA received at the input terminal 191 receives the sample rate f _SM2 of the signals PWA ₁ and PWA ₂ used for calculating the phase of the pilot signal PS. Since the sample rate is reduced from the sample rate f _SM1, the number of calculations with a large calculation amount such as arctan calculation in the phase calculating unit 120A can be reduced. Therefore, the total calculation amount can be reduced.

In the seventh embodiment, as in the case of the first embodiment, the phase signal PHW obtained at the reduced sample rate f _SM2 is increased to the sample rate f _SM1 . Here, since the target for increasing the sample rate f _SM2 is the phase, it is possible to easily obtain the phase signal PHA increased to the sample rate f _SM1 without degrading the accuracy. Therefore, it is possible to obtain a reference signal BSG sample rate f _SM1 suitable for processing of the signal SIA sample rate f _SM1.

  In the seventh embodiment, since the signal component SG2 in the signal SIA is processed using the reference signal BSG, desired processing can be performed on the signal component SG2 quickly and accurately.

In the seventh embodiment, as in the case of the first embodiment, when there is a phase shift Δθ in the FILs 113A ₁ and 113A ₂ , the phase θ _C of the pilot signal PS is calculated in consideration of this. Thus, the phase θ _C can be calculated with high accuracy. For this reason, it is possible to generate the reference signal BSG that is accurately synchronized with the pilot signal PS.

  In the seventh embodiment, since the offset phase Δφ corresponding to the signal level and noise level of the signal SIA from the signal source 910A is set and the reference signal BSG reflecting the offset phase Δφ is generated, the processing result signal MSG Can be changed in accordance with the signal level and noise level of the signal SIA. Note that the offset phase Δφ may be a predetermined fixed value.

[Modification of Embodiment]
The present invention is not limited to the first to seventh embodiments described above, and various modifications are possible.

For example, in the first to seventh embodiments, a signal having an angular frequency 2ω _C that is twice the angular frequency ω _C of the pilot signal PS is generated as the reference signal. On the other hand, a signal having an angular frequency that is an arbitrary multiple of the angular frequency ω _C can be generated as a reference signal in accordance with the mode of the signal component to be processed in the signal from the signal source. Furthermore, in the case of the second and third embodiments, a signal having an angular frequency obtained by an arbitrary linear combination of the angular frequency ω _C and the shift angular frequency ω _SH can be generated as the reference signal. .

In the first to seventh embodiments, the angular frequency band of the machining result signal is the angular frequency band of 0 to ω _C , but the angular frequency band of the machining result signal is changed by changing the angular frequency of the reference signal. Can be set to an arbitrary angular frequency band.

In the second to fifth embodiments, the signal having the angular frequency (ω _SH −ω _C ) is extracted and used from the signals corresponding to the pilot signal PS after the frequency conversion. An angular frequency (ω _SH + ω _C ) signal may be extracted and used.

  In the sixth embodiment, two reference signals are generated, and signals in two angular frequency bands included in the signal from the signal source are processed. However, three or more reference signals are generated. However, signals in three angular frequency bands included in the signal from the signal source can be processed.

  In the seventh embodiment, the offset phase Δφ is set according to the level of the signal from the signal source and the noise level. However, the offset phase Δφ may be set according to a user instruction. Good.

  Further, the modification of the first embodiment to the sixth or seventh embodiment can be applied to the second to fifth embodiments, or the modification of the first embodiment to the second to fifth embodiments. The sixth and seventh embodiments and the seventh and eighth embodiments can also be applied. Furthermore, the modification of the second embodiment to the third or fourth embodiment can be performed on the fifth embodiment.

  Further, if the decimation filter is arranged in front of the downsampling unit, a modification in which the downsampling unit is arranged at the same position as in the third or fourth embodiment is applied to the first, sixth, and seventh embodiments. It can also be done.

  Note that the signal processing devices of the first to seventh embodiments can also be realized by executing a program in a DSP (Digital Signal Processor). These programs may be acquired in the form recorded on a portable recording medium such as a CD-ROM or DVD, or may be acquired in the form of delivery via a network such as the Internet. Good.

1 is a block diagram illustrating a schematic configuration of a signal processing device according to a first embodiment of the present invention. It is a figure for demonstrating the frequency distribution of the signal from the signal source assumed in 1st Embodiment. It is a block diagram which shows the structure of the orthogonal signal generation part in the apparatus of FIG. It is a block diagram which shows the structure of the orthogonalization part in FIG. It is a block diagram which shows the structure of the reference signal production | generation part in the apparatus of FIG. It is a block diagram which shows the structure of the signal processing part in the apparatus of FIG. It is a figure for demonstrating the frequency distribution of the process result signal by the apparatus of FIG. It is a block diagram which shows the schematic structure of the signal processing apparatus which concerns on 2nd Embodiment of this invention. It is a block diagram which shows the structure of the orthogonal signal generation part in the apparatus of FIG. It is a block diagram which shows the structure of the orthogonalization part in FIG. It is a figure for demonstrating the frequency distribution of the signal from the signal source assumed in 2nd Embodiment. It is a figure for demonstrating the frequency distribution of the band limitation signal output from a band limitation filter. It is a figure for showing the frequency conversion result by the orthogonalization part of FIG. It is a block diagram which shows the schematic structure of the signal processing apparatus which concerns on 3rd Embodiment of this invention. It is a block diagram which shows the structure of the orthogonal signal generation part in the apparatus of FIG. It is a block diagram which shows schematic structure of the signal processing apparatus which concerns on 4th Embodiment of this invention. It is a block diagram which shows the structure of the orthogonal signal generation part in the apparatus of FIG. It is a block diagram which shows the schematic structure of the signal processing apparatus which concerns on 5th Embodiment of this invention. It is a block diagram which shows the structure of the orthogonal signal generation part in the apparatus of FIG. It is a block diagram which shows the structure of the modification of the reference signal production | generation part in the apparatus of FIG. It is a block diagram which shows the schematic structure of the signal processing apparatus which concerns on 6th Embodiment of this invention. It is a figure for demonstrating the frequency distribution of the signal from the signal source assumed in 6th Embodiment. It is a block diagram which shows the structure of the reference | standard signal production | generation part in the apparatus of FIG. It is FIG. (1) for demonstrating the frequency distribution of the process result signal by the apparatus of FIG. It is FIG. (2) for demonstrating the frequency distribution of the process result signal by the apparatus of FIG. It is a block diagram which shows schematic structure of the signal processing apparatus which concerns on 7th Embodiment of this invention. It is a block diagram which shows the structure of the reference signal production | generation part in the apparatus of FIG. It is a figure for demonstrating the frequency distribution and amplitude of a process result signal by the apparatus of FIG.

Explanation of symbols

100A to 100G ... Signal processing devices 110A to 110E ... Orthogonal signal generator (orthogonal signal generator)
111B, 111E ... Band limiting filter (band limiting means)
112A, 112B ... Orthogonalizing unit (orthogonalizing means)
113A ₁ , 113A ₂ ... Filter (filter means)
113B ₁ , 113B ₂ ... Filter (filter means)
113E ₁ , 113E ₂ ... Filter (filter means)
115A, 115D ... downsampling unit (downsampling means)
120A, 120B ... Phase calculation unit (phase calculation means)
120E ... Phase calculation unit (phase calculation means)
125 ... Upsampling section (upsampling means)
130A, 130E ... reference signal generator (reference signal generator)
130F, 130G... Reference signal generator (reference signal generator)
140... Signal processing section (first and second type signal processing means)
140 ₁ , 140 ₂ ... signal processing section (signal processing means)

Claims (14)

A signal processing apparatus that performs processing related to an input signal that is sampled at a first sample rate and includes a predetermined signal in a predetermined frequency band,
Orthogonal signal generation means for reflecting the phase of the predetermined signal and orthogonalizing each other and generating a first signal and a second signal sampled at a second sample rate lower than the first sample rate;
After calculating the ratio of the amplitude of the first signal and the amplitude of the second signal at the same time point, the calculated ratio is not fed back to the orthogonal signal generation means, and is calculated based on the time change of the calculated ratio. Phase calculating means for calculating the phase of the signal reflecting the predetermined signal;
Up-sampling means for calculating a phase equivalent to the case of sampling at a third sample rate higher than the second sample rate based on the calculation result by the phase calculation means;
Comprising a; the while reflecting the calculation result by the phase calculation means, a reference signal generating means for generating a reference signal at said predetermined signal and a predetermined relationship
The reference signal generating means generates the reference signal based on a calculation result by the upsampling means;
A signal processing apparatus.   The signal processing apparatus according to claim 1, wherein the reference signal is a signal having a phase that is a predetermined multiple of the predetermined signal.   2. The signal processing according to claim 1, wherein the reference signal is a signal having a phase represented by a sum of a predetermined multiple phase and a predetermined fixed phase value with respect to the phase of the predetermined signal. apparatus. The signal processing apparatus according to claim 1 , further comprising a first type processing unit that processes the input signal using the reference signal. The orthogonal signal generating means includes
Orthogonalizing means for generating a first orthogonalized signal and a second orthogonalized signal that reflect the phase of the predetermined signal and are orthogonalized with each other;
Filter means for selectively passing a signal corresponding to the first signal included in the first orthogonalized signal and a signal corresponding to the second signal included in the second orthogonalized signal;
A down-sampling unit that is arranged in at least one position before the orthogonalizing unit, between the orthogonalizing unit and the filter unit, and at a subsequent stage from the filter unit;
5. The signal processing apparatus according to claim 1 , comprising: The signal processing apparatus according to claim 5 , wherein the phase calculation unit calculates a phase of a signal reflecting the predetermined signal in consideration of a phase shift in the filter unit. The orthogonalizing means performs a frequency shift in addition to the orthogonalization,
The phase calculating means calculates a phase of a signal reflecting the predetermined signal in consideration of a frequency shift amount due to the frequency shift;
The signal processing apparatus according to claim 5 or 6 , wherein The signal processing apparatus according to claim 7 , wherein the reference signal has a phase calculated by linear combination of a phase of the signal reflecting the predetermined signal and a phase corresponding to the frequency shift amount. . The signal processing apparatus according to claim 8 , wherein the reference signal has a phase further shifted by a predetermined fixed phase value. A band limiting unit that generates a band limiting signal that limits the frequency band of the input signal and supplies the band limiting signal to the orthogonalizing unit;
The down-sampling means is arranged at a later stage than the band limiting means.
The signal processing device according to claim 7 , wherein the signal processing device is a signal processing device. The signal processing apparatus according to claim 10 , further comprising a second type processing unit that processes the band-limited signal using the reference signal. A signal processing method for performing processing related to an input signal sampled at a first sample rate and including a predetermined signal in a predetermined frequency band,
A quadrature signal generation step of generating a first signal and a second signal that are sampled at a second sample rate lower than the first sample rate while reflecting the phase of the predetermined signal and being orthogonalized with each other;
A ratio between the amplitude of the first signal and the amplitude of the second signal at the same time point is calculated, and the calculated ratio is not used in the orthogonal signal generation step, and is based on the time change of the calculated ratio. Calculating a phase of the signal reflecting the predetermined signal;
An up-sampling step of calculating a phase equivalent to the case of sampling at a third sample rate higher than the second sample rate based on the calculation result in the phase calculation step;
Comprising a; the reflecting the calculation result in the phase calculating step, the reference signal generating step of generating a reference signal at said predetermined signal and a predetermined relationship
In the reference signal generation step, the reference signal is generated based on the calculation result in the upsampling step.
And a signal processing method. A signal processing program for causing a calculation means to execute the signal processing method according to claim 12 . 14. A recording medium, wherein the signal processing program according to claim 13 is recorded so as to be readable by an arithmetic means.

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