JP4828044B2 - Power circuit - Google Patents

Description

【００４０】
また、磁心１１Ａ，１１Ｂの実効体積をＶｅ、比例定数をｋ、磁束密度幅を△Ｂ_ｍａｘとして上記式１を変形すれば、Ｐ_０＝（ｋ／２）×（△Ｂ_ｍａｘ）２×Ｖｅ×ｆとなるため、明らかに磁心の小型化をはかり、かっ高い△Ｂの設計値を許容することから、巻線の巻回数を下げた銅損低減効果によって、小型、高効率のアクティブフィルタを具備するスイッチング電源を提供することが可能となることは明白である。
【００４１】
（第２の実施の形態）
図２（ａ）は、本発明による磁心とそれを用いた線輪部品（トランス）及び電源回路をシングルエンドフライバック型のＤＣ−ＤＣコンバータに適用した第２の実施の形態であり、図２（ｂ）は、その主な回路動作波形を示し、図２（ｃ）は磁心とこの磁心を用いた線輪部品のＢ−Ｈ特性図である。
【００４２】
図２（ａ）において、Ｔ２１はトランス、Ｑ２１はスイッチング素子（主にトランジスタ）、Ｃｏｎｔ．２は制御回路、Ｄ２１はダイオード、Ｃ２１，Ｃ２２はコンデンサ、ＲＬは抵抗である。
【００４３】
トランスＴ２１は、第１の実施の形態で示した図１（ｂ）の磁心と同様の構成であり、巻線部は入力巻線１，２と出力巻線３，４とからなっている。従って、励磁巻線のインダクタンスは十分に高い値が確保出来るため、スイッチ素子Ｑ１が導通すると、図２（ｂ）に示したように励磁巻線に台形状の電流が流れるとともに磁気エネルギーを充電し、スイッチ素子Ｑ１が遮断すると同時に出力巻線３，４とダイオードＤ２１とを介して、やはり台形状の出力電流が流れて電力を伝達する動作を制御回路Ｃｏｎｔ．２の指令に従って繰り返す。
【００４４】
従って、この図２（ａ）に示したシングルエンドフライバックＤＣ−ＤＣコンバータの場合でも出力電力Ｐ_０［Ｗ］は、第１の実施の形態と同様に、Ｐ_０＝（１／２）×Ｌ×（（ｉｒ）^２−（ｉｒ）^２）×ｆ、Ｐ_０＝（ｋ／２）×（△Ｂ_ｍａｘ）２×Ｖｅ×ｆで示され、大出力のＤＣ―ＤＣコンバータに対しても、あえて回路構成の複雑な従来技術のシングルエンドフォワード方式やフルブリッジ構成を採らなくとも、トランスＴ２１の利用率を高め、かつ巻線の実効電流値も十分低減出来るシングルエンドフライバック型コンバータを、経済性と小型化を阻害することなく簡素な構成で小型高効率に提供できる。
【００４５】
（第３の実施の形態）
図３は、本発明の第３の実施の形態による磁心とそれを用いた線輪部品（トランス）及び電源回路を自励式のＲＣＣコンバータに適用した例を示す図である。
【００４６】
図３において、Ｔ３１はトランス、Ｑ３１はスイッチング素子（主にトランジスタ）、Ｄ３１，Ｄ３２，Ｄ３３はダイオード、Ｃ３１，Ｃ３２，Ｃ３３はコンデンサ、Ｒ３１，Ｒ３２，ＲＬは抵抗である。
【００４７】
トランスＴ３１は、第１の実施の形態で示した図１（ｂ）の磁心と同様の構成であり、巻線部は入力巻線１，２と出力巻線３，４、補助巻線５，６とからなっている。
【００４８】
出カトランスＴ３１が励磁巻線１−２に流れる電流によって形成される磁界と逆方向にバイアス磁界が印加するように永久磁石を配置しているため、出力巻線３，４からの出力は従来技術による磁心及び線輪部品と比較して大幅に負荷側に伝達され、小型、大容量、低損失に構成できる。
【００４９】
（第４の実施の形態）
図４（ａ）は、本発明の第４の実施の形態による磁心とそれを用いた線輪部品（リアクタ）、及び、電源回路を、第１の実施の形態に記述したアクティブフィルタ回路に対して適用して、スイッチ素子の損失を大幅に低減した例を示す図で、図４（ｂ）にはその主な波形を示している。
【００５０】
図４（ａ）において、Ｌ４１はリアクタ、Ｑ４１はスイッチング素子（主にトランジスタ）、Ｃｏｎｔ．３は制御回路、Ｄ４１，Ｄ４２，Ｄ４３はダイオード、Ｃｒ，Ｃｌはコンデンサ、ＲＬ，Ｒ４１，Ｒ４２は抵抗、Ｌ６は可飽和コイルである。
【００５１】
電源として高出力化するためにスイッチ素子に流れる電流を鋸歯状から台形状とすれば、リアクタの巻線電流の実効値は低減して損失も低減するものの、スイッチ素子自体のクロスカレント損失はターンオンを軸に増大するため、まず、スイッチ素子Ｑ４１のターンオン期間に対しては、遅延用として新たに設けた極めて小型の磁心にわずかの巻線ｘ−ｙを施し、トランスの励磁巻線とスイッチ素子Ｑ４１との間に直列に可飽和コイルＬｄとして接続する。そして、スイッチ素子Ｑ４１のターンオフ期間用としては、スイッチ素子Ｑ４１と並列に共振するようにコンデンサＣｒを設けている。
【００５２】
つまり、スイッチ素子Ｑ４１のターンオン期間は、上記可飽和コイルＬｄが未だ非飽和状態でスイッチ素子Ｑ４１には、その励磁電流分が流れ、飽和に達した時点でリアクタＬ４１への励磁電流を導通させるため、問題のクロスカレント損失は極めて小さくできる。
【００５３】
また、スイッチ素子Ｑ４１のターンオフ期間についても、リアクタＬ４１と可飽和コイルＬｄとの和のインダクタンスと、コンデンサＣｒとがダイオードＤ４３を介して並列共振を開始するため、スイッチ素子Ｑ４１の電圧は固有振動周波数１／（（Ｌ４１十Ｌｄ）×Ｃｒ）１／２に拘東されて上昇するため、同様にクロスカレント損失は極めて小さく出来る。
【００５４】
ダイオードＤ４３については、上記並列共振動作に介在させるとともに、スイッチ素子Ｑ４１がターンオンする際にコンデンサＣｒにチャージアップされた電荷を瞬時に放電してクロスカレント損失を増やさぬように抵抗Ｒ４３と並列に構成している。
【００５５】
尚、このような商用電源Ｖ_ＡＣＩＮ入力に供するアクティブフィルタの場合には、ダイオードＤ４２にファストリカバリダイオードを用いざるを得ないが、従来技術の構成の場合には、スイッチ素子Ｑ４１をターンオンすると同時にダイオードＤ４２のリカバリ期間と重なるため、大きな貫通電流が出力からスイッチ素子Ｑ４１に逆流して効率を低下させるとともに、大きなＥＭＩ障害をもたらしているが、上述した本発明の構成によれば可飽和リアクタＬｄが上記貫通電流をも阻止出来るため、更なる高効率と低ノイズのアクティブフィルタの提供が可能となる点も工業的に益するところ極めて大といえる。
【００５６】
（第５の実施の形態）
図５（ａ）は、本発明の第５の実施の形態による磁心とそれを用いた線輪部品（トランス）、及び電源回路を、第１の実施の形態に記述したシングルエンドフライバックコンバータ回路に対して適用して、スイッチ素子の損失を大幅に低減した例を示す図で、図５（ｂ）にはその主な波形を示している。
【００５７】
図５（ａ）において、Ｔ５１はトランス、Ｑ５１はスイッチング素子（主にトランジスタ）、Ｃｏｎｔ．５は制御回路、Ｄ５１，Ｄ５２はダイオード、Ｃｒ，Ｃ５１，Ｃ５２はコンデンサ、ＲＬ，Ｒ５１，Ｒ５２は抵抗、Ｌｄは可飽和コイルである。
【００５８】
上述した第４の実施の形態の場合と同様に、第５の実施の形態において、高出力化するためトランスの巻線電流を台形状としても、遅延用の可飽和コイルＬｄと、並列共振用のコンデンサＣｒを設けているためスイッチ素子Ｑ５１自体のクロスカレント損失を同様に大幅低減出来る。
【００５９】
続いて、上述した本発明による磁心、磁心を用いた線輪部品、及び電源回路に用いるバイアス用永久磁石に関する実施の形態を以下に記す。
【００６０】
（第６の実施の形態）
従来技術の問題に記述した熱減磁に対しては，以下の方策を施している。即ち、リフロー半田工程における熱に耐えるために、磁石粉末には高ＴＣであるＳｍＣｏ系磁石粉末を用いることで、熱減磁を生じさせない方策を施している。
【００６１】
第１の実施の形態に用いた構成に、第１の実施の形態で用いたＴｃが７７０℃の永久磁石を装着したものと、従来技術で用いられていたが４５０℃と低いＢａフェライト磁石を装着したものを、リフロー炉の条件である２７０℃の恒温槽で１時間保持し常温まで冷却後の直流重畳インダクタンス特性を測定した結果を下記表１に示す。
【００６２】
【表１】

【００６３】
本発明における高Ｔｃ材を使用したインダクタ部品はリフロー前後で、直流重畳インダクタンス特性の変化がないのに対して、Ｔｃが４５０℃と低いＢａフェライト磁石の場合、熱によって不可逆減磁が生じ、直流重畳インダクタンス特性の劣化が生じた。
【００６４】
従って、リフロー半田工程による加熱などに絶えるためには、Ｔｃが５００℃以上の磁石粉末を用いる必要がある。
【００６５】
加えてＳｍＣｏ系磁石粉末の中でも、組成がＳｍ（Ｃｏ_ｂａｌ．Ｆｅ_{０．１５−０．２５}Ｃｕ_{０．０５−０．０６}Ｚｒ_{０．０２−０．０３}）_{７．０−８．５}である俗に第３世代Ｓｍ_２Ｃｏ_１７磁石と呼ばれる組成の磁石粉末を用いることにより、熱による減磁はさらに抑えられる。
【００６６】
第１の実施の形態で用いた構成に、第１の実施の形態で用いた組成が、Ｓｍ（Ｃｏ_{０．７４２}Ｆｅ_０．２０Ｃｕ_{０．０５５}Ｚｒ_{０．０２９}）_７．７である永久磁石を装着したものと、組成がＳｍ（Ｃｏ_０．７８Ｆｅ_０．１１Ｃｕ_０．１０Ｚｒ_０．０１）_７．７である永久磁石を装着したものとをリフロー炉の条件である２７０℃の高温槽で１時間保持し、常温までの冷却後の直流重畳インダクタンス特性を測定した結果を下記表２に示す。
【００６７】
【表２】

【００６８】
本発明における組成が第３世代であるＳｍ（Ｃｏ_ｂａｌ．Ｆｅ_{０．１５−０．２５}Ｃｕ_{０．０５−０．０６}Ｚｒ_{０．０２−０．０３}）_{７．０−８．５}を用いたインダクタ部品はリフロー前後で、直流重畳インダクタンス特性の変化がないのに対して、俗に第２世代Ｓｍ_２Ｃｏ_１７と呼ばれるものの磁石粉末を用いたものは、直流重畳インダクタンス特性の劣化が起こった。
【００６９】
従って、リフロー半田工程による加熱などに耐えるためには、請求項３に示すような組成が第３世代であるＳｍ（Ｃｏ_ｂａｌ．Ｆｅ_{０．１５−０．２５}Ｃｕ_{０．０５−０．０６}Ｚｒ_{０．０２−０．０３}）_{７．０−８．５}を用いる必要がある。
【００７０】
（第７の実施の形態）
従来技術の問題に記載した過大電流による減磁に対しては以下の方策を施している。即ち、過大電流に伴う直流磁界によって、永久磁石の保磁力が消失しないように、高ｉＨｃであるＳｍＣｏ系磁石粉末を用いる方策を施している。
【００７１】
実施の形態１で用いた構成に、実施の形態１で用いた保磁力が２０ＫＯｅ（１．５８ＭＡ／ｍ）の永久磁石を装着したものと、従来技術で用いられていた保持力が２ＫＯｅ（１５８ｋＡ／ｍ）の磁石を装着したものに３００Ａ・５０μｓの過大電流を印加後、直流重畳インダクタンス特性を測定した結果を下記表３に示す。
【００７２】
【表３】

【００７３】
本発明における高ｉＨｃ材を用いたインダクタ部品は、過大電流印加前後で、直流重畳インダクタンス特性の変化がないのに対して保磁力が２ｋＯｅ（１５８ｋＡ／ｍ）しかないＢａフェライト磁石の場合、磁石に印加される逆向きの磁界による減磁が起こり、直流重畳インダクタンス特性が低下した。
【００７４】
従って、過大電流による直流磁界に耐えるためには、固有保磁力が１０ＫＯｅ（７９０ｋＡ／ｍ）以上の磁石粉末を用いる必要がある。
【００７５】
（第８の実施の形態）
従来技術の問題に記述した経時的な酸化が進むことによる永久減磁に対しては以下の実施を施している。即ち、磁石粉末が酸化を起こさないように金属や合金によるコーティングをする方策を施している。
【００７６】
第１の実施の形態で用いた構成に、第１の実施の形態１で用いたＺｎの被覆をした永久磁石を装着したものと、Ｚｎの被覆をしていない永久磁石を装着したものを、塩水に浸した後、２００時間自然放置し、直流重畳インダクタンス特性を測定した結果を下記表４に示す。
【００７７】
【表４】

【００７８】
本発明におけるコーティングを施したインダクタ部品はＰＣＴ前後で、直流重畳インダクタンス特性の変化がないのに対して、Ｚｎコートなしの磁石粉末は、経時的な酸化が進むことで減磁を生じ、直流重畳インダクタンス特性の劣化が生じた。
【００７９】
従って、酸化の進行による永久減磁を抑えるためには、Ｚｎ，Ａｌ，Ｂｉ，Ｇａ，Ｉｎ，Ｍｇ，Ｐｂ，Ｓｂ及びＳｎ等の磁石粉末をコーティングする必要がある。
【００８０】
加えて、粉末平均粒径を２．５〜２５μｍで最大粒径が５０μｍとすることで、作製工程中の酸化も抑えることが可能になる。
【００８１】
第１の実施の形態で用いた構成に、第１の実施の形態で用いた平均粒径が５μｍで最大粒径が４５μｍ磁石粉末を用いた永久磁石を装着したものと、平均粒径を２μｍにした永久磁石を装着したもので、直流重畳インダクタンス特性を測定した結果を下記表５に示す。
【００８２】
【表５】

【００８３】
本発明における粒径を用いたインダクタ部品は、磁気バイアスによる直流重畳インダクタンス特性の向上が５０％であるのに対して、平均粒径を２μｍにしたものは１５％しか仲びていない。
【００８４】
従って、作製工程中の酸化を抑えるには、粉末平均粒径を２．５〜２５μｍで最大粒径が５０μｍとする必要がある。
【００８５】
（第９の実施の形態）
従来技術の問題に記載した比抵抗が低いことによるコアロスの増加に対しては以下の実施を施している。即ち、高比抵抗にするために、樹脂量を体積比で３０％以上にする方策を施している。
【００８６】
第１の実施の形態で用いた構成に、第１の実施の形態で用いた樹脂量が４０Ｖｏｌ％とし、比抵抗を０．５Ωｃｍにした永久磁石を装着したものと、樹脂量を２０Ｖｏｌ％とし比抵抗を０．０５Ωｃｍにした永久磁石を装着したもの、および樹脂量を３０Ｖｏｌ％とし比抵抗を０．１Ωｃｍにした永久磁石を装着したもので、コアロスの測定を行った結果を下記表６に示す。
【００８７】
【表６】

【００８８】
本発明における樹脂量を３０Ｖｏｌ％以上としたインダクタ部品のコアロスに対して、樹脂量を体積比で２０％として比抵抗を０．０５Ωｃｍと低いものは、渦電流が流れることにより損失が生じ、コアロスが悪化している。また、樹脂量を体積比で３０％とし比抵抗を０．１Ωｃｍとしたものは、第１の実施の形態に用いた、樹脂量を体積比で４０％とし比抵抗を０．５Ωｃｍとしたものと同じ程度のコアロスを示している。
【００８９】
従って、比抵抗の低下に伴うコアロスの増加を抑えるためには、樹脂量は体積で３０％以上とし、比抵抗は０．１Ωｃｍ以上必要である。
【００９０】
【発明の効果】
以上、述べた通り、本発明による磁心、磁心を用いた線輪部品、及び電源回路を用いれば、スイッチング電源などに用いられる磁心やトランス、チョークコイル等の線輪部品の小型化・低損失化が図れるとともに、更に電源回路の簡素化、小型化、高効率化、省資源化、高信頼性に対して飛躍的に寄与することが出来るため、工業的に益するところ極めて大なるものである。
【図面の簡単な説明】
【図１】本発明の第１の実施の形態を示す図で、（ａ）はアクティブフィルタを具備したスイッチング電源の構成図、（ｂ）は磁心とそれを用いた線輪部品の断面構成図、（ｃ）は磁心とそれを用いた線輪部品の動作Ｂ−Ｈ特性図、（ｄ）は（ｃ）における主な動作波形を示す波形図、（ｅ）は磁心とそれを用いた線輪部品の直流重畳インダクタンスを説明した図である。
【図２】本発明の第２の実施の形態を示す図で、（ａ）はシングルエンドフライバック型ＤＣ−ＤＣコンバータ構成図、（ｂ）は（ａ）における主な動作波形を示す動作波形図、（ｃ）は磁心とそれを用いた線輪部品の動作Ｂ−Ｈ特性図である。
【図３】本発明の第３の実施の形態を示す図で、自励式のＲＣＣコンバータ構成図である。
【図４】本発明の第４の実施の形態を示す図で、（ａ）は磁心とそれを用いた線輪部品、及び電源回路構成図であり、（ｂ）は（ａ）における主な動作波形を示す波形図である。
【図５】本発明の第５の実施の形態を示す図で、（ａ）は磁心とそれを用いた線輪部品、及び電源回路構成図であり、（ｂ）は（ａ）における主な動作波形を示す波形図である。
【図６】（ａ）は従来技術によるアクティブフィルタを具備したスイッチング電源構成図、（ｂ）は従来技術によるアクティブフィルタに用いるトランスの断面構成図、（ｃ）は従来技術によるアクティブフィルタに用いるトランスの動作Ｂ−Ｈ特性図、（ｄ）は（ｃ）における主な動作波形を示す波形図である。
【図７】（ａ）は従来技術によるシングルエンドフォワード型ＤＣ−ＤＣコンバータ構成図、（ｂ）は動作Ｂ−Ｈ特性図である。
【図８】（ａ）は従来技術によるフルブリッジ型ＤＣ−ＤＣコンバータ構成図、（ｂ）線輪部品の動作Ｂ−Ｈ特性図である。
【符号の説明】
１１Ａ，１１Ｂ，１２，６２ 巻線
１３ 永久磁石
Ｑ１１，Ｑ２１，Ｑ３１，Ｑ４１，Ｑ５１，Ｑ６１，Ｑ７１，Ｑ８１，Ｑ８２，Ｑ８３，Ｑ８４ スイッチ素子
Ｌ１１，Ｌ４１，Ｌ６１ リアクタ
Ｌ７１，Ｌ８１ チョークコイル
Ｌｄ 可飽和コイル
Ｔ２１，Ｔ３１，Ｔ５１，Ｔ７１，Ｔ８１ トランス
Ｃ１１，Ｃ２１，Ｃ２２，Ｃ３１，Ｃ３２，Ｃ３３，Ｃｒ，Ｃ４１，Ｃ５１，Ｃ５２，Ｃ６１，Ｃ７１，Ｃ７２，Ｃ７３，Ｃ８１，Ｃ８２ コンデンサ
Ｒ１１，ＲＬ，Ｒ３１，Ｒ３２，Ｒ４１，Ｒ４２，Ｒ５１，Ｒ５２，Ｒ７１抵抗
Ｄ１１，Ｄ１２，Ｄ２１，Ｄ３１，Ｄ３２，Ｄ３３，Ｄ４１，Ｄ４２，Ｄ４３，Ｄ５１，Ｄ５２，Ｄ６２，Ｄ７１，Ｄ７２，Ｄ７３，Ｄ８１，Ｄ８２ ダイオード[0001]
BACKGROUND OF THE INVENTION
The present invention relates to a magnetic core frequently used in electronic equipment, a wire ring component using a magnetic core, and a power supply circuit, and in particular, a wire core component such as a magnetic core, choke coil, and transformer used in a switching power supply, and the like, and The present invention relates to a power circuit using wire ring parts.
[0002]
[Prior art]
Conventionally, a switching power supply shown in FIG. 6A is a configuration diagram of a switching power supply including an active filter according to the prior art, FIG. 6B is a cross-sectional view of a transformer used for the active filter according to the prior art, and FIG. 6C is used for the active filter according to the prior art. The operation | movement BH characteristic schematic diagram of a transformer is shown, FIG.6 (d) is a wave form diagram which shows the main operation | movement waveforms in (c).
[0003]
In the conventional switching power supply, the commercial power supply V_ACINIn order to comply with so-called harmonic regulation that eliminates problems such as an increase in power loss of the transmission line due to a decrease in power factor due to current and voltage distortion to the commercial power supply V_ACINAs shown in FIG. 6 (a), the reactor L61 is intentionally grounded via the switch element (mainly transistor) Q61 after rectification, and after being turned off. A so-called active filter configuration is employed in which the magnetic energy accumulated in the reactor L61 during the ON period is output to the output capacitor C61 and then output to the DC-DC converter 101, which is the main function.
[0004]
The control circuit Cont. 6 shows that the peak value of the current waveform flowing through the reactor L61 is the commercial power supply V._ACINBy adjusting the aging rate of on and off so as to have a value similar to the full-wave rectified waveform voltage VCE through the diode D61, power supply characteristics with an input power factor of approximately 1 are obtained.
[0005]
The E-type magnetic cores 61A and 61B used in the active filter circuit shown in FIG. 6B and the reactor L61 using the magnetic cores 61A and 61B are provided with a winding 62 on the magnetic core and the magnetic cores 61A and 61B are in contact with each other. A gap 63 is provided in the leg. Magnetic fields are formed in the magnetic cores 61A and 61B by the coil current iL flowing from the input side diode D65 in accordance with the conduction of the switch element Q61.
[0006]
The transformer used for the active filter exhibits an operation BH characteristic as shown in FIG. 6C, and its operation waveform is shown in FIG.
[0007]
Conventionally, the switching power supply has been promoted in order to reduce the size and increase the efficiency of the power supply. However, in order to obtain an output exceeding 100 W with an insulation type using a transformer, the conventional technique is shown in FIG. A single-end forward type DC-DC converter as shown in a) is frequently used, and FIG. 7B shows an operation BH characteristic of the wire ring component.
[0008]
7A, T71 is a transformer, L71 is a smoothing choke coil, Q71 is a switching element (mainly a transistor), Cont. 7 is a control circuit, D71, D72 and D73 are diodes, C71, C72 and C73 are capacitors, and RL and R71 are resistors.
[0009]
Further, in order to obtain a larger output, a full bridge type DC-DC converter as shown in FIG. 8A is frequently used in the prior art, and FIG. The characteristics are shown. The single-end forward type as shown in FIG. 7B is symmetrically expanded to half-wave excitation not only in the first quadrant of the BH characteristic but also in the third quadrant as shown in FIG. 8B. As a result, the utilization factor of the transformer T81 is increased to enable high output.
[0010]
8A, T81 is a transformer, L81 is a smoothing chiyoke coil, Q81, Q82, Q83, and Q84 are switching elements (mainly transistors), cont. 8 is a control circuit, D81 and D82 diodes, C81 and C82 are capacitors, and RL is a resistor.
[0011]
[Problems to be solved by the invention]
However, the above-described prior art has the following serious drawbacks. In other words, in the active filter circuit according to the prior art shown in FIG. 6A, the size of the reactor L61 is substantially the same as the output transformer of the DC-DC converter connected to the subsequent stage, and the winding 62 thereof. There was a serious drawback that the copper loss due to the decrease in the efficiency of the whole power supply.
[0012]
In addition, when a DC-DC converter having a large output to some extent is configured by the conventional technology, the excitation current becomes sawtooth and the effective value of the current waveform increases in the single-end flyback method, which is simple and economically desirable. Since an increase in size of the transformer is unavoidable, it is necessary to adopt a single-end forward converter configuration as shown in FIG. As a result, the circuit configuration is complicated and the economical efficiency is hindered.
[0013]
Further, in order to obtain a higher output, by adopting a full bridge converter configuration as shown in FIG. 8 (a), the forward method is different from the half-wave excitation as shown in FIG. As shown in b), the utilization of the transformer T81 is increased by expanding the target to the third quadrant as well as the first quadrant of the BH characteristic, and in this case, it is possible to cope with high output. In addition to requiring four switch elements (mainly transistors), the switching operation is also economical because the switch element Q81 and the switch element Q82 or the switch element Q83 and the switch element Q84 are turned on and off at the same time. In addition, there was a serious drawback in terms of loss that caused a large industrial disadvantage.
[0014]
  Therefore, the technical problem of the present invention is to solve the problems of the prior art, and to reduce the size and loss of the magnetic core and choke coil used in the switching power supply and the like, as well as to simplify and reduce the size of the circuit. , High efficiency, resource saving and economically excellent magnetic coreWithWire ring partsUsedIt is to provide a power supply circuit.
[0015]
[Means for Solving the Problems]
  According to the present invention, there is provided a power supply circuit having a half-wave excitation circuit using a wire ring component in which at least one winding and one or more turns are applied to a magnetic core of a closed magnetic circuit, wherein the magnetic core is A permanent magnet is disposed in at least one air gap of a magnetic material having a soft magnetic characteristic that forms a magnetic path in the closed magnetic path, and the wire ring component is excited when an input voltage is applied to an exciting winding. The polarity of the magnetic field generated by the current is opposite to the polarity of the magnetic field generated by the permanent magnet.Further, at least one or more other windings are applied to portions other than the windings of the magnetic core, and both ends of the other windings are respectively connected in series between the wire ring component and the switch element. A circuit including at least a capacitor is connected in parallel to the switch element.The power supply circuit characterized by this can be obtained.
[0018]
  According to the present invention, in any one of the power supply circuits, the permanent magnet is selected from polyamide imide resin, polyimide resin, epoxy resin, polyphenylene sulfide resin, silicon resin, polyester resin, aromatic polyamide resin, and liquid crystal polymer. The at least one kind of resin has an intrinsic coercive force of 790 kA / m or more, Tc of 500 ° C. or more, a powder average particle size of 2.5 to 25 μm, and Zn, Al, Bi, Ga, In, Mg, Pb , Sb and Sn, at least one metal or alloy thereofA first coating comprisingA power supply circuit is obtained in which the coated rare earth magnet powder is dispersed, the resin content is 30% or more by volume, and the specific resistance is 0.1 Ωcm or more.
[0019]
The composition of the rare earth magnet powder is Sm (Co_bal.Fe_0.15-0.25Cu_0.05-0.06Zr_0.02-0.03)_7.0-8.5It is preferable that
[0020]
  According to the present invention, in any one of the power supply circuits, the rare earth magnet powder coated with the first coating further includes a second non-metallic inorganic compound having a melting point of at least 300 ° C. A power supply circuit characterized by being covered with a covering is obtained.
[0021]
  Further, according to the present invention, the addition amount covered with the first, second and third coatings is the total amount,0.1-10% by volumeA power supply circuit characterized by the above can be obtained.
[0022]
Further, the magnet powder may be anisotropicized by orienting the rare earth magnet powder in the thickness direction with a magnetic field at the time of production.
[0023]
The permanent magnet preferably has a magnetization magnetic field of 2.5 T or more and a center line average roughness Ra of 10 μm or less.
[0024]
DETAILED DESCRIPTION OF THE INVENTION
Hereinafter, the present invention will be described in more detail. In the present invention, a magnetic core having a closed magnetic path, which has a soft magnetic characteristic and forms a magnetic path, and a magnetic core in which permanent magnets are arranged in at least one gap is linked to the magnetic path. The magnetic field generated by the permanent magnet and the polarity of the magnetic field generated by the excitation current that flows when the input voltage is applied to the winding applied to the wire ring component, which constitutes the wire ring component provided with at least one winding So that the polarities of these are opposite to each other.
[0025]
As a result, even in the case of a half-wave excitation coil or transformer, a magnetic material having soft magnetic characteristics can be preliminarily placed in the third quadrant by a permanent magnet even if the excitation direction is the first quadrant direction of the BH characteristic curve. Since the residual magnetic flux density Br substantially shifts to the third quadrant because it is biased in the direction, the available magnetic flux density width ΔB can be greatly expanded, so that the winding applied to the magnetic core can be greatly reduced. Thus, it is possible to contribute to downsizing and low loss of the wire ring parts.
[0026]
In addition, the magnetic core according to the present invention and the wire ring component using the magnetic core can be widely used not only in the first quadrant of the BH loop but also in the third quadrant for the half-wave excitation circuit due to the bias effect described above. Even if the single-end flyback method with the most cylindrical configuration is adopted, the current waveform can be designed in a trapezoidal shape without the sawtooth shape in the prior art described above, so the effective value of the winding current is Since the level can be reduced to the same level as the complicated forward converter and full-bridge converter having the above-described configuration, the output of the switching power source can be greatly simplified without complicating the circuit configuration.
[0027]
In addition, the power supply circuit of the present invention, that is, an extremely small magnetic core newly provided for delaying the turn-on current of the switch element, is provided with at least one or more windings. A power supply circuit is characterized in that both ends of the winding of the magnetic core are connected in series with each other, and a circuit including at least a parallel resonance capacitor at the time of turn-off is connected in parallel to the switch element. As a result, a power supply circuit that significantly reduces the switching loss due to the crossing of the current and voltage during the turn-on and turn-off periods of the switch element is realized with a simple configuration.
[0028]
In the present invention, the permanent magnet is made of SmCo-based magnet powder having a high Tc (Curie temperature) and a high iHc (coercive force), so that it can be thermally demagnetized even when placed in a heated state in the reflow soldering process. In addition, even when a DC magnetic field due to an excessive current is applied, the coercive force disappears and the initial characteristics can be maintained without demagnetization.
[0029]
In addition, by coating the surface of the magnet powder used for the permanent magnet with a metal such as Zn, permanent demagnetization due to the progress of oxidation over time does not occur.
[0030]
Further, by kneading the magnet powder with the resin at a volume ratio of 30% or more, it is possible to increase the specific resistance, and the eddy current loss of the permanent magnet can be greatly reduced.
[0031]
Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings.
[0032]
(First embodiment)
FIG. 1A shows a first embodiment in which a magnetic core and a wire ring component according to the present invention are configured as in the cross-sectional structure shown in FIG. It is a form.
[0033]
FIG. 1B is a cross-sectional configuration diagram of a magnetic core and a wire ring component using the magnetic core, and FIG. 1C is an operation BH characteristic diagram of the magnetic core and the wire ring component using the magnetic core. FIG. FIG. 1C is a waveform diagram showing main operation waveforms in FIG. 1C, and FIG. 1E is an explanatory diagram explaining the DC superimposed inductance characteristics of a magnetic core and a wire ring component using the magnetic core.
[0034]
In FIG. 1 (b), a winding 12 is applied to the core portions of a pair of EE type magnetic cores 11A and 11B having soft magnetic characteristics such as Mn—Zn ferrite so that the magnetic flux is linked to each other, and the opposing 1 Among the opposing surfaces of the pair of magnetic cores 11A and 11B, a gap is provided in the joint portion of the inner leg.
[0035]
In the active filter circuit shown in FIG. 1A, L11 is a wire ring component, Q11 is a switching element (mainly a transistor), Cont. Reference numeral 1 is a control circuit, D11 and D12 are diodes, C11 is a capacitor, and RL and R11 are resistors.
[0036]
The magnetic cores 11A and 11B used for the active filter circuit and the wire ring component L11 using the magnetic cores 11A and 11B have directions of magnetic fields formed in the magnetic cores 11A and 11B by the coil current iL flowing from the input side 1 according to the conduction of the switch element (mainly transistor) Q11. The magnetic cores 11A and 11B in which the permanent magnets 13 are arranged in the gaps between the magnetic cores 11A and 11B so as to have the opposite polarity are provided with at least one or more windings 12 of one turn or more.
[0037]
If the magnetic cores 11A and 11B and the wire ring part L11 according to the present invention are used, the magnetic cores 11A and 11B have a third direction opposite to the magnetic field direction formed in advance by the current flowing in the winding, as shown in FIG. Since a state in which a bias is applied by ΔH in advance is formed by the permanent magnet 13 on the quadrant side, the operation allowable width ΔB value of the magnetic flux density generated by the voltage and current applied to the winding can be increased as shown in the figure. In addition, the magnetic cores 11A and 11B having the same AL value can be compared with the characteristics of the related art ring-wheel component indicated by the broken line as in the DC superimposed inductance characteristic with respect to the exciting current flowing through the winding shown in FIG. If so, as shown by the arrow (1) shown in FIG. 1 (e), the current superposition value can be greatly increased by simply using the same inductance. Conversely, even when the AL value is increased by narrowing the gap as indicated by the arrow (2) shown in FIG. 1 (e), the inductance is increased in a flying manner and the superimposed current allowable value of the wire ring component according to the prior art is increased. It can also be secured.
[0038]
That is, by applying the magnetic core and the wire ring component shown in the first embodiment of the present invention to the wire ring component L11 of the active filter, the output power P that contributes to the boosting of the active filter.₀[W] is defined by the following equation 1 when the operating frequency is f and the peak value of the winding current is ip and ir shown in FIG. The output power can be increased up to four times the size and frequency of the same magnetic cores 11A and 11B by the square effect on the superposition value expansion.
[0039]
[Expression 1]

[0040]
Further, the effective volume of the magnetic cores 11A and 11B is Ve, the proportionality constant is k, and the magnetic flux density width is ΔB._maxIf the above equation 1 is modified, P₀= (K / 2) × (ΔB_max) Since 2 × Ve × f, it is obvious that the magnetic core is downsized and a high design value of ΔB is allowed, so the copper loss reduction effect by reducing the number of winding turns reduces the size and efficiency. It is obvious that it is possible to provide a switching power supply having a plurality of active filters.
[0041]
(Second Embodiment)
FIG. 2A shows a second embodiment in which a magnetic core according to the present invention, a wire ring component (transformer) using the magnetic core, and a power supply circuit are applied to a single-end flyback type DC-DC converter. (B) shows the main circuit operation waveforms, and FIG. 2 (c) is a BH characteristic diagram of a magnetic core and a wire ring component using the magnetic core.
[0042]
2A, T21 is a transformer, Q21 is a switching element (mainly a transistor), Cont. 2 is a control circuit, D21 is a diode, C21 and C22 are capacitors, and RL is a resistor.
[0043]
The transformer T21 has the same configuration as that of the magnetic core shown in FIG. 1B shown in the first embodiment, and the winding portion includes input windings 1 and 2 and output windings 3 and 4. Therefore, since the inductance of the exciting winding can secure a sufficiently high value, when the switch element Q1 is turned on, a trapezoidal current flows through the exciting winding and magnetic energy is charged as shown in FIG. , The switching element Q1 is cut off, and at the same time, the operation in which the trapezoidal output current flows through the output windings 3 and 4 and the diode D21 to transmit power is controlled by the control circuit Cont. Repeat in accordance with command 2.
[0044]
Therefore, even in the case of the single-end flyback DC-DC converter shown in FIG.₀[W] is P, as in the first embodiment.₀= (1/2) × L × ((ir)²-(Ir)²) × f, P₀= (K / 2) × (ΔB_max) Even with a large output DC-DC converter, the utilization rate of the transformer T21 can be reduced without adopting a conventional single-end forward method or full bridge configuration with a complicated circuit configuration. It is possible to provide a single-end flyback converter that can increase the effective current value of the winding sufficiently and can be reduced in size and efficiency with a simple configuration without impeding economy and downsizing.
[0045]
(Third embodiment)
FIG. 3 is a diagram illustrating an example in which a magnetic core according to a third embodiment of the present invention, a ring component (transformer) using the magnetic core, and a power supply circuit are applied to a self-excited RCC converter.
[0046]
In FIG. 3, T31 is a transformer, Q31 is a switching element (mainly a transistor), D31, D32, and D33 are diodes, C31, C32, and C33 are capacitors, and R31, R32, and RL are resistors.
[0047]
The transformer T31 has the same configuration as that of the magnetic core of FIG. 1B shown in the first embodiment, and the winding portion includes input windings 1 and 2, output windings 3 and 4, auxiliary windings 5, It consists of six.
[0048]
Since the permanent magnet is arranged so that the bias magnetic field is applied to the output transformer T31 in the direction opposite to the magnetic field formed by the current flowing in the exciting winding 1-2, the output from the output windings 3 and 4 is the conventional technique. Compared with the magnetic core and wire ring parts, it is transmitted to the load side greatly, and can be configured with small size, large capacity and low loss.
[0049]
(Fourth embodiment)
FIG. 4A shows a magnetic core according to a fourth embodiment of the present invention, a wire ring component (reactor) using the magnetic core, and a power supply circuit compared to the active filter circuit described in the first embodiment. FIG. 4B shows an example in which the loss of the switching element is greatly reduced by applying the above, and FIG.
[0050]
4A, L41 is a reactor, Q41 is a switching element (mainly a transistor), Cont. 3 is a control circuit, D41, D42 and D43 are diodes, Cr and Cl are capacitors, RL, R41 and R42 are resistors, and L6 is a saturable coil.
[0051]
If the current flowing through the switch element is changed from a sawtooth shape to a trapezoidal shape as a power source, the effective value of the winding current of the reactor is reduced and the loss is reduced, but the cross current loss of the switch element itself is turned on. First, for the turn-on period of the switch element Q41, a very small magnetic core newly provided for delay is provided with a small number of windings xy, and the exciting winding of the transformer and the switch element A saturable coil Ld is connected in series with Q41. For the turn-off period of the switch element Q41, a capacitor Cr is provided so as to resonate in parallel with the switch element Q41.
[0052]
That is, during the turn-on period of the switch element Q41, the saturable coil Ld is still in a non-saturated state, the excitation current flows through the switch element Q41, and the excitation current to the reactor L41 is made conductive when saturation is reached. The cross current loss in question can be made extremely small.
[0053]
Also, during the turn-off period of the switch element Q41, since the inductance of the sum of the reactor L41 and the saturable coil Ld and the capacitor Cr start parallel resonance via the diode D43, the voltage of the switch element Q41 has the natural vibration frequency. Similarly, the cross current loss can be made extremely small because it rises by being held at 1 / ((L41 + Ld) × Cr) 1/2.
[0054]
The diode D43 is arranged in parallel with the resistor R43 so as to intervene in the parallel resonance operation and not to increase the cross current loss by instantaneously discharging the charge charged up to the capacitor Cr when the switch element Q41 is turned on. is doing.
[0055]
In addition, such a commercial power supply V_ACINIn the case of an active filter used for input, a fast recovery diode must be used as the diode D42. However, in the case of the configuration of the conventional technique, the switching element Q41 is turned on and at the same time overlaps with the recovery period of the diode D42. While the through current flows back from the output to the switching element Q41 to reduce the efficiency and causes a large EMI failure, the saturable reactor Ld can also block the through current according to the configuration of the present invention described above. The fact that it is possible to provide an active filter with further high efficiency and low noise is extremely large from the industrial point of view.
[0056]
(Fifth embodiment)
FIG. 5A shows a single-end flyback converter circuit in which a magnetic core according to a fifth embodiment of the present invention, a wire ring component (transformer) using the magnetic core, and a power supply circuit are described in the first embodiment. FIG. 5B shows an example in which the loss of the switching element is significantly reduced by applying the above to FIG.
[0057]
5A, T51 is a transformer, Q51 is a switching element (mainly a transistor), Cont. Reference numeral 5 is a control circuit, D51 and D52 are diodes, Cr, C51 and C52 are capacitors, RL, R51 and R52 are resistors, and Ld is a saturable coil.
[0058]
As in the case of the fourth embodiment described above, in the fifth embodiment, even if the winding current of the transformer is trapezoidal in order to increase the output, the delay saturable coil Ld and the parallel resonance coil Since the capacitor Cr is provided, the cross current loss of the switch element Q51 itself can be greatly reduced as well.
[0059]
Subsequently, embodiments relating to the above-described magnetic core according to the present invention, a wire ring component using the magnetic core, and a permanent magnet for bias used in a power supply circuit will be described below.
[0060]
(Sixth embodiment)
The following measures are taken against the thermal demagnetization described in the problem of the prior art. That is, in order to withstand the heat in the reflow soldering process, a measure is taken to prevent thermal demagnetization by using a high TC SmCo magnet powder as the magnet powder.
[0061]
In the configuration used in the first embodiment, a permanent magnet having a Tc of 770 ° C. used in the first embodiment and a Ba ferrite magnet having a low 450 ° C. used in the prior art are used. Table 1 below shows the results of measuring the DC superimposed inductance characteristics after the mounted one was held in a constant temperature bath at 270 ° C., which is the condition of the reflow furnace, for 1 hour and cooled to room temperature.
[0062]
[Table 1]

[0063]
Inductor parts using a high Tc material in the present invention have no change in DC superimposed inductance characteristics before and after reflowing. On the other hand, in the case of a Ba ferrite magnet having a low Tc of 450 ° C., irreversible demagnetization occurs due to heat. Degradation of the superimposed inductance characteristics occurred.
[0064]
Therefore, it is necessary to use magnet powder having a Tc of 500 ° C. or higher in order to eliminate the heating by the reflow soldering process.
[0065]
In addition, among the SmCo-based magnet powders, the composition is Sm (Co_bal. Fe_0.15-0.25Cu_0.05-0.06Zr_0.02-0.03)_7.0-8.5The third generation Sm₂Co₁₇By using magnet powder having a composition called a magnet, demagnetization due to heat can be further suppressed.
[0066]
In the configuration used in the first embodiment, the composition used in the first embodiment is Sm (Co_0.742Fe_0.20Cu_0.055Zr_0.029)_7.7With a permanent magnet and a composition of Sm (Co_0.78Fe_0.11Cu_0.10Zr_0.01)_7.7Table 2 below shows the results of measuring the DC superimposed inductance characteristics after being cooled to room temperature for 1 hour in a high temperature bath of 270 ° C. which is the condition of the reflow furnace.
[0067]
[Table 2]

[0068]
The composition in the present invention is the third generation Sm (Co_bal.Fe_0.15-0.25Cu_0.05-0.06Zr_0.02-0.03)_7.0-8.5Inductor parts that use a dc have no change in the DC superimposed inductance characteristics before and after reflow, but second-generation Sm is commonly used.₂Co₁₇In the case of using the magnet powder, the DC superimposed inductance characteristics deteriorated.
[0069]
Therefore, in order to endure the heating by the reflow soldering process, the composition as shown in claim 3 is the third generation Sm (Co_bal.Fe_0.15-0.25Cu_0.05-0.06Zr_0.02-0.03)_7.0-8.5Must be used.
[0070]
(Seventh embodiment)
The following measures are taken against demagnetization due to excessive current described in the problem of the prior art. In other words, measures are taken to use high iHc SmCo-based magnet powder so that the coercive force of the permanent magnet does not disappear due to the DC magnetic field associated with the excessive current.
[0071]
The structure used in the first embodiment is mounted with the permanent magnet having the coercive force of 20 KOe (1.58 MA / m) used in the first embodiment, and the coercive force used in the prior art is 2 KOe (158 kA). Table 3 below shows the results of measuring the DC superimposed inductance characteristics after applying an excessive current of 300 A · 50 μs to a magnet equipped with a / m) magnet.
[0072]
[Table 3]

[0073]
  The inductor component using the high iHc material in the present invention isExcessiveIn the case of a Ba ferrite magnet having a coercive force of only 2 kOe (158 kA / m) while there is no change in the DC superimposed inductance characteristics before and after the current application, demagnetization occurs due to the reverse magnetic field applied to the magnet. Superimposed inductance characteristics deteriorated.
[0074]
Therefore, in order to withstand a DC magnetic field caused by an excessive current, it is necessary to use magnet powder having an intrinsic coercive force of 10 KOe (790 kA / m) or more.
[0075]
(Eighth embodiment)
For permanent demagnetization due to the progress of oxidation over time described in the problem of the prior art, the following implementation is performed. That is, measures are taken to coat the metal powder with a metal or alloy so that the magnet powder does not oxidize.
[0076]
In the configuration used in the first embodiment, the one equipped with the permanent magnet coated with Zn used in the first embodiment and the one equipped with the permanent magnet not coated with Zn, Table 4 below shows the results of measuring the direct current superimposed inductance characteristics after standing in salt water for 200 hours.
[0077]
[Table 4]

[0078]
Inductor parts with coating in the present invention have no change in DC superimposed inductance characteristics before and after PCT, whereas magnet powder without Zn coating causes demagnetization due to the progress of oxidation over time, and DC superposition. Inductance characteristics deteriorated.
[0079]
Therefore, in order to suppress permanent demagnetization due to the progress of oxidation, it is necessary to coat magnet powder such as Zn, Al, Bi, Ga, In, Mg, Pb, Sb and Sn.
[0080]
In addition, by setting the average particle size of the powder to 2.5 to 25 μm and the maximum particle size to 50 μm, oxidation during the manufacturing process can be suppressed.
[0081]
In the configuration used in the first embodiment, a permanent magnet using a magnet powder having an average particle size of 5 μm and a maximum particle size of 45 μm used in the first embodiment, and an average particle size of 2 μm are used. Table 5 below shows the results of measuring the DC superimposed inductance characteristics with the permanent magnets attached.
[0082]
[Table 5]

[0083]
The inductor component using the grain size according to the present invention has 50% improvement in the DC superimposed inductance characteristic due to the magnetic bias, whereas only 15% has an average grain size of 2 μm.
[0084]
Therefore, in order to suppress oxidation during the production process, it is necessary that the average particle size of the powder is 2.5 to 25 μm and the maximum particle size is 50 μm.
[0085]
(Ninth embodiment)
The following implementation is performed for the increase in the core loss due to the low specific resistance described in the problem of the prior art. That is, in order to obtain a high specific resistance, measures are taken to increase the resin amount to 30% or more by volume ratio.
[0086]
In the configuration used in the first embodiment, the amount of resin used in the first embodiment is 40 Vol%, and a permanent magnet with a specific resistance of 0.5 Ωcm is mounted, and the amount of resin is 20 Vol%. Table 6 below shows the results of measurement of core loss using a permanent magnet with a specific resistance of 0.05 Ωcm and a permanent magnet with a resin amount of 30 Vol% and a specific resistance of 0.1 Ωcm. Show.
[0087]
[Table 6]

[0088]
In contrast to the core loss of the inductor component in which the resin amount in the present invention is 30 vol% or more, the resin loss is 20% in volume ratio and the specific resistance is as low as 0.05 Ωcm. Is getting worse. In addition, the resin amount is 30% by volume and the specific resistance is 0.1 Ωcm, and the resin amount used in the first embodiment is 40% by volume and the specific resistance is 0.5 Ωcm. Shows the same level of core loss.
[0089]
Therefore, in order to suppress an increase in core loss accompanying a decrease in specific resistance, the amount of resin must be 30% or more by volume, and the specific resistance needs to be 0.1 Ωcm or more.
[0090]
【The invention's effect】
As described above, if the magnetic core according to the present invention, the wire ring component using the magnetic core, and the power supply circuit are used, miniaturization and low loss of the wire ring components such as the magnetic core used in the switching power supply, the transformer, the choke coil, etc. In addition, it can greatly contribute to the simplification, miniaturization, high efficiency, resource saving, and high reliability of the power supply circuit. .
[Brief description of the drawings]
1A and 1B are diagrams showing a first embodiment of the present invention, where FIG. 1A is a configuration diagram of a switching power supply including an active filter, and FIG. 1B is a cross-sectional configuration diagram of a magnetic core and a wire ring component using the magnetic core. , (C) is an operation BH characteristic diagram of a magnetic core and a wire ring part using the magnetic core, (d) is a waveform diagram showing main operation waveforms in (c), and (e) is a magnetic core and a line using the magnetic core. It is a figure explaining the direct current superposition inductance of a wheel component.
FIGS. 2A and 2B are diagrams showing a second embodiment of the present invention, in which FIG. 2A is a configuration diagram of a single-end flyback DC-DC converter, and FIG. 2B is an operation waveform showing main operation waveforms in FIG. FIG. 4C is an operation BH characteristic diagram of a magnetic core and a wire ring component using the magnetic core.
FIG. 3 is a diagram showing a third embodiment of the present invention and is a self-excited RCC converter configuration diagram.
4A and 4B are diagrams showing a fourth embodiment of the present invention, in which FIG. 4A is a configuration diagram of a magnetic core, a wire ring part using the magnetic core, and a power supply circuit, and FIG. 4B is a main diagram in FIG. It is a wave form diagram which shows an operation | movement waveform.
5A and 5B are diagrams showing a fifth embodiment of the present invention, in which FIG. 5A is a configuration diagram of a magnetic core, a wire ring component using the magnetic core, and a power supply circuit, and FIG. 5B is a main diagram in FIG. It is a wave form diagram which shows an operation | movement waveform.
6A is a configuration diagram of a switching power supply including an active filter according to the prior art, FIG. 6B is a cross-sectional configuration diagram of a transformer used for the active filter according to the prior art, and FIG. 6C is a transformer used for the active filter according to the prior art. (B) is a waveform diagram showing main operation waveforms in (c).
7A is a configuration diagram of a single-end forward DC-DC converter according to the prior art, and FIG. 7B is an operation BH characteristic diagram.
8A is a configuration diagram of a full-bridge DC-DC converter according to the prior art, and FIG. 8B is an operation BH characteristic diagram of a wire ring part.
[Explanation of symbols]
11A, 11B, 12, 62 Winding
13 Permanent magnet
Q11, Q21, Q31, Q41, Q51, Q61, Q71, Q81, Q82, Q83, Q84 switch element
L11, L41, L61 reactor
L71, L81 Choke coil
Ld saturable coil
T21, T31, T51, T71, T81 transformer
C11, C21, C22, C31, C32, C33, Cr, C41, C51, C52, C61, C71, C72, C73, C81, C82 capacitors
R11, RL, R31, R32, R41, R42, R51, R52, R71 resistors
D11, D12, D21, D31, D32, D33, D41, D42, D43, D51, D52, D62, D71, D72, D73, D81, D82 Diode

Claims (9)

A power supply circuit having a half-wave excitation circuit using a wire ring component in which at least one winding and one or more turns are applied to a magnetic core of a closed magnetic circuit, wherein the magnetic core is a magnetic circuit in the closed magnetic circuit. A permanent magnet is disposed in at least one gap of a magnetic material having a soft magnetic characteristic forming a path, and the wire ring component has a magnetic field polarity generated by an excitation current flowing when an input voltage is applied to the excitation winding. but the have a polarity opposite to each other with the polarity of the magnetic field generated by the permanent magnet in a part other than the winding of the magnetic core, further subjected to at least one other winding, the wire ring component and a switching element A power supply circuit , wherein both ends of the other windings are connected in series with each other, and a circuit including at least a capacitor is connected in parallel to the switch element . 2. The power supply circuit according to claim 1 , wherein the permanent magnet is at least one selected from a polyamideimide resin, a polyimide resin, an epoxy resin, a polyphenylene sulfide resin, a silicon resin, a polyester resin, an aromatic polyamide resin, and a liquid crystal polymer. The resin has an intrinsic coercive force of 790 kA / m or more, Tc of 500 ° C. or more, a powder average particle size of 2.5 to 25 μm, and Zn, Al, Bi, Ga, In, Mg, Pb, Sb and Sn. A rare earth magnet powder coated with a first coating comprising at least one kind of metal or an alloy thereof is dispersed, the resin content is 30% or more by volume, and the specific resistance is 0.1 Ωcm or more. A power supply circuit characterized by being. The power supply circuit according to claim 2, the composition of the rare earth magnet _{powder, Sm (Co bal, Fe 0.15-0.25} Cu 0.05-0.06 Zr 0.02-0.03) 7. A power supply circuit characterized by being _0-8.5 . The power supply circuit according to claim 2 or 3, rare earth magnet powder coated with the first coating further coated with a second coating comprising an inorganic compound of nonmetal having at least 300 ° C. or higher melting point Power supply circuit characterized by being made. 4. The power supply circuit according to claim 2 , wherein the rare earth magnet powder is coated with a third coating made of inorganic glass having a softening point of 220 ° C. or higher and 550 ° C. or lower. The power supply circuit according to any one of claims 2 to 5 , wherein the addition amount to be coated with the first, second, and third coatings is 0.1 in volume ratio in total amount. A power supply circuit characterized by being 10% to 10%. The power supply circuit according to any one of claims 2 to 6 , wherein the rare earth magnet powder is magnetically anisotropic when oriented in the thickness direction of the magnetic core during the production of the permanent magnet. Power supply circuit characterized by being made. The power supply circuit according to any one of claims 2 to 7 , wherein a magnetizing magnetic field of the permanent magnet is 2.5T or more. 9. The power supply circuit according to claim 2 , wherein the permanent magnet has a center line average roughness Ra of 10 μm or less.

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