JP4372812B2 - Drive control device for semiconductor switching element - Google Patents

Description

  The present invention relates to a drive control device for a semiconductor switching element, and more specifically to a technique for suppressing voltage fluctuation (surge voltage) generated when a semiconductor switching element constituting a semiconductor power conversion device such as an inverter or a converter is turned on. .

  As a countermeasure for increasing the output density of inverters and converters, active gate control applied when turning on / off semiconductor switching elements such as IGBTs (Insulated Gate Bipolar Transistors) and power MOS (Metal Oxide Semiconductor) transistors is known. Active gate control is a technique for optimizing the voltage drive speed (switching speed) of the gate (control electrode) by dynamically switching the gate resistance value during the turn-on or turn-off of the semiconductor switching element. By controlling the driving speed as described above, both switching loss reduction and surge voltage suppression, which are in a trade-off relationship with the switching speed, can be achieved.

  For example, Japanese Patent Application Laid-Open No. 2002-125363 (Patent Document 1), Japanese Patent Application Laid-Open No. 2006-222593 (Patent Document 2), or US Pat. No. 6,208,185 (Patent Document 3) discloses a semiconductor switching element. An active gate control is disclosed in which a gate resistance value is changed according to a current / voltage state at turn-on and / or turn-off.

  Further, as a configuration of a gate driving circuit for realizing active gate control, not only a configuration for switching a gate resistance value but also a circuit configuration to which a reactor is applied as in JP-A-2005-39988 (Patent Document 4) is proposed. Has been.

On the other hand, in an IGBT module or the like in which a freewheel diode is connected in parallel with the semiconductor switching element, it is reported that the operation behavior of the freewheel diode has a great influence on the surge voltage when the semiconductor switching element is turned on ( Non-patent document 1). In particular, it is described that when a turn-on is performed in a low current state, a reverse bias is applied from a state in which the number of carriers of the diode is small, so that a depletion layer develops in a very short time, resulting in an increase in surge voltage.
JP 2002-125363 A JP 2006-222593 A US Pat. No. 6,208,185 JP 2005-39988 A Fumio Nagahama and two others, “Analysis of diode reverse recovery phenomenon from transient on-state”, Fuji Jiho, Fuji Electric Co., Ltd., February 2001, Vol. 74 No. 2 2001, p. 149-152

  The general active gate control as disclosed in Patent Documents 1 to 4 has a control configuration in which the current or voltage of a semiconductor switching element that is turned on and off is monitored, and the switching speed is changed according to a detected value or a time differential value thereof. ing.

  For this reason, in applications where the switching frequency is high and the switching transition time to which the active gate control is applied is very short, very high control responsiveness is required for voltage / current monitoring and control of the switching speed in response thereto. . Furthermore, it is necessary to perform control in consideration of variations in characteristics of the gate drive circuit, the monitor circuit or the semiconductor switching element, and external fluctuation factors such as temperature.

  Because of these factors, it is practically difficult to ensure the control stability of active gate control based on feedback of monitoring results.If control fails due to the presence of a control delay, the semiconductor switching element will fail. There is also a possibility. In other words, in order to avoid such a control failure, the control design must have a margin, and the switching loss reduction effect is suppressed. Further, the addition of a voltage / current monitor circuit for the semiconductor switching element itself causes an increase in cost and circuit size.

  The present invention has been made to solve such a problem, and an object of the present invention is to provide a surge at turn-on without accompanying feedback of a monitoring result of voltage and / or current of a semiconductor switching element. It is to perform stable switching speed control for achieving both voltage suppression and switching loss reduction.

  A drive control device according to the present invention is a drive control device for a semiconductor switching element that constitutes a semiconductor power conversion device for controlling a supply current to a load in accordance with a current command value, and includes a current estimation unit, a switching speed control unit, Is provided. The semiconductor switching element is turned on or off in response to the voltage or current of the control electrode. The current estimation means estimates an energization current that flows when the semiconductor switching element is turned on based on the current command value when the turn-on command is generated. The switching speed control means is configured to obtain the semiconductor switching element based on the estimated energization current obtained by the current estimation means in accordance with a first current characteristic indicating a relationship between the energization current of the semiconductor switching element and the magnitude of the generated surge voltage. The drive speed of the voltage or current of the control electrode at the time of turn-on is changed.

  According to the drive control device for a semiconductor switching element, the energizing current flowing through the semiconductor switching element after turn-on is estimated based on the current command value to the semiconductor power conversion device, and the relationship between the energizing current and the surge voltage behavior is shown. The drive speed (that is, the switching speed) of the control electrode at the time of turn-on can be changed according to the characteristics obtained in advance. That is, by changing the switching speed based on the current command value at the time of turn-on command, the surge voltage can be reduced and reduced without providing a current and / or voltage monitor configuration and with a stable control configuration without feedback control. Surge voltage suppression can be achieved.

  Preferably, the switching speed control means includes a determination means and a switching speed instruction means. The determination means determines whether the estimated energization current belongs to the first current region where the surge voltage exceeds the allowable value or the second current region where the surge voltage is equal to or lower than the allowable value according to the first current characteristic. . The switching speed instruction means makes the drive speed at the start of turn-on slower when the estimated energization current is in the first current region than when the estimated energization current is in the second current region.

  With this configuration, the surge voltage can be suppressed by starting turn-on at a relatively low switching speed in the current region (first current region) where the surge voltage is expected to increase. . On the other hand, in the current region (second current region) where the surge voltage is expected to be small, the switching loss can be reduced by starting the turn-on at a relatively high switching speed.

  More preferably, the switching speed control means sets the driving speed at the start of turn-on to the first speed and starts driving the control voltage when the estimated energization current is in the first current region. The driving speed is increased from the first speed in accordance with the elapsed time. Particularly in such a configuration, the switching speed control means further includes a switching timing setting means. The switching timing setting means estimates the current in accordance with a second current characteristic that is obtained in advance and indicates the relationship between the energization current of the semiconductor switching element and the time from the start of driving the control voltage until the surge voltage is generated. A switching timing for increasing the driving speed is set based on the estimated energization current by the means. Furthermore, the switching speed instruction means increases the driving speed from the first speed to the second speed when the switching timing is reached based on the elapsed time.

  With such a configuration, in the current region where the surge voltage increases (first current region), the surge voltage is suppressed by reducing the switching speed at the start of turn-on, while the surge voltage is peaked. After the time exceeding, switching speed can be increased to reduce switching loss. In particular, the increase timing of the switching speed can be appropriately set without feedback according to a previously obtained characteristic indicating the relationship between the energization current and the surge voltage behavior (generation time delay).

  Alternatively, more preferably, the switching speed instruction means sets the driving speed at the start of turn-on to the first speed when the estimated energization current is in the first current region, while the estimated energization current is the second speed. In the current region, the drive speed at the start of turn-on is set to a second speed higher than the first speed, and the drive speed is fixed to the second speed throughout the turn-on period.

  By adopting such a configuration, in the current region (second current region) where the surge voltage is unlikely to be excessive, the switching speed is set to be relatively high throughout the on period from the start of turn-on. Loss can be reduced as much as possible.

  More preferably, the switching speed instruction means changes the driving speed from the first speed to the first speed according to the elapsed time from the start of driving the control voltage when the estimated energization current is in the first current region. Switch to 2 speed.

  By adopting such a configuration, the above-described switching speed control can be executed by a configuration in which the switching speed is changed in two stages, so that the circuit configuration of the drive control device can be simplified.

  Preferably, the drive control device further includes a capacitance value adjustment circuit for adjusting the capacitance value of the control electrode of the semiconductor switching element by trimming.

  By adopting such a configuration, it is possible to enhance the effect of the switching speed control described above corresponding to the manufacturing variation of the capacitance value of the control electrode of the semiconductor switching element.

  More preferably, the drive control device further includes learning control means. The learning control means sets a learning period in which a current command value is generated so that the load mainly consumes reactive power when the semiconductor power converter is started up, and sets the measured value of the surge voltage of the semiconductor switching element during the learning period. Based on this, the determination value indicating the boundary between the first and second current regions is corrected. Alternatively, the learning control means corrects the switching timing based on the measured value of the surge voltage of the semiconductor switching element during the learning period.

  With such a configuration, the first and second current regions are determined using the period until the load is actually operated as the power converter is started at the start of the load operation. It is possible to realize learning control in which the value or switching speed switching timing is updated based on the actual surge voltage during the learning period. Thereby, in actual operation after execution of learning control, switching speed control when the semiconductor switching element is turned on can be more appropriately executed, so that both suppression of surge voltage and reduction of switching loss can be achieved more effectively.

  Preferably, the load is an AC motor whose supply current is controlled by pulse width modulation control, and the current estimation means calculates an estimated energization current using a rotation angle of the AC motor and a current command value by vector control. .

  With such a configuration, in the drive control device for a semiconductor switching element constituting a semiconductor power device having an AC motor as a load, the current command value (d axis, q) of the AC motor according to the rotation angle and vector control of the AC motor. From the axial current command value), it becomes possible to easily and accurately estimate the on-current after the semiconductor switching element is turned on.

  According to the present invention, stable switching speed control for achieving both suppression of surge voltage and reduction of switching loss at the time of turn-on can be executed without accompanying feedback of the monitoring result of the voltage and / or current of the semiconductor switching element. .

  Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings. In the following description, the same or corresponding parts in the drawings are denoted by the same reference numerals, and the description thereof will not be repeated in principle.

  FIG. 1 is an overall configuration diagram of a motor drive system to which a drive control device for a semiconductor switching element according to the present invention is applied.

  Referring to FIG. 1, motor drive system 100 according to an embodiment of the present invention includes a DC power supply 10, a voltage sensor 13, a smoothing capacitor C0, an inverter 20, a control device 30, and an AC motor MG that is a load. With.

  AC motor MG is, for example, a drive motor for generating torque for driving drive wheels of a hybrid vehicle or an electric vehicle. Alternatively, AC motor MG may be configured to have a function of a generator driven by an engine, or may be configured to have both functions of an electric motor and a generator. Furthermore, AC motor MG may operate as an electric motor for the engine, and may be incorporated in a hybrid vehicle so that the engine can be started, for example.

  DC power supply 10 outputs a DC voltage between power supply line 7 and ground line 5. The DC power supply 10 can be charged by a DC voltage between the power supply line 7 and the ground line 5. The DC power supply 10 is typically composed of a secondary battery such as nickel metal hydride or lithium ion or a power storage device of an electric double layer capacitor. A configuration is provided in which a converter for changing the voltage level of the output voltage (DC) of the power storage device is provided to variably control the output voltage of DC power supply 10, that is, the voltage between power supply line 7 and ground line 5. Also good.

  Smoothing capacitor C 0 is connected between power supply line 7 and ground line 5. The voltage between the terminals of the smoothing capacitor C0, which corresponds to the DC side voltage of the inverter 20, is detected by the voltage sensor 13, and the detected value is sent to the control device 30 constituted by an electronic control unit (ECU).

  Inverter 20 includes a U-phase circuit 15, a V-phase circuit 16, and a W-phase circuit 17 provided in parallel between power supply line 7 and ground line 5. Each phase circuit includes a power semiconductor switching element connected in series between power supply line 7 and ground line 5. An IGBT (Insulated Gate Bipolar Transistor) is typically applied as a power semiconductor switching element (hereinafter simply referred to as “semiconductor switching element”).

  The U-phase circuit 15 includes a semiconductor switching element Q1 that is an upper arm element and a semiconductor switching element Q2 that is a lower arm element, and the V-phase circuit 16 includes a semiconductor switching element Q3 that is an upper arm element and a semiconductor that is a lower arm element. The W-phase circuit 17 includes a switching element Q4, and includes a semiconductor switching element Q5 that is an upper arm element and a semiconductor switching element Q6 that is a lower arm element. Further, free wheel diodes D1 to D6 are connected to the semiconductor switching elements Q1 to Q6, respectively, for causing a current to flow in a direction opposite to the semiconductor switching elements. On / off of the semiconductor switching elements Q1 to Q6 is controlled by switching control signals S1 to S6 from the control device 30.

  AC motor MG is typically a three-phase permanent magnet motor, and one end of three coils of U, V, and W phases is commonly connected to a neutral point. The other end of the U, V, W phase coil is connected to the connection point of the upper arm element and the lower arm element in the U, V, W phase of inverter 20.

  When the torque command value of AC motor MG is positive (Tqcom> 0), inverter 20 is a semiconductor switching element that responds to DC voltage from smoothing capacitor C0 in response to switching control signals S1 to S6 from control device 30. The AC motor MG is driven so as to convert it into an AC voltage and output a positive torque by the switching operation of Q1 to Q6.

  Further, when the torque command value of AC motor MG is zero (Tqcom = 0), inverter 20 converts the DC voltage to the AC voltage by the switching operation in response to switching control signals S1 to S6, and the torque is zero. The AC motor MG is driven so that Thereby, AC motor MG is driven to generate zero or positive torque designated by torque command value Tqcom.

  Further, during regenerative braking of a hybrid vehicle or electric vehicle equipped with motor drive system 100, torque command value Tqcom of AC motor MG is set to a negative value (Tqcom <0). In this case, the inverter 20 converts the AC voltage generated by the AC motor MG into a DC voltage by a switching operation in response to the switching control signals S1 to S6, and converts the converted DC voltage (system voltage) to the smoothing capacitor C0. Can be used to charge the DC power supply 10 via Note that regenerative braking here refers to braking with regenerative power generation when the driver driving a hybrid vehicle or electric vehicle performs foot braking, or turning off the accelerator pedal while driving, although the foot brake is not operated. This includes decelerating the vehicle (or stopping acceleration) while generating regenerative power.

  The current sensor 24 is configured by a hall sensor or the like, detects a motor current supplied from the inverter 20 to the AC motor MG, and outputs a detected value to the control device 30. Since the sum of instantaneous values of the three-phase currents iu, iv, and iw is zero, the current sensor 24 has a motor current for two phases (for example, a V-phase current iv and a W-phase current iw) as shown in FIG. It is sufficient to arrange it so as to detect.

  A rotation angle sensor (typically, a resolver) 25 detects the rotation angle θ of AC motor MG, and sends the detected rotation angle θ to control device 30. Control device 30 also calculates the rotational speed (rotational speed) of AC motor MG based on rotational angle θ.

  The control device 30 includes a torque command value Tqcom that indicates an output torque of the AC motor MG that is a load, a DC voltage Vdc detected by the voltage sensor 13, motor currents iv and iw from the current sensor 24, and a rotation angle sensor 25. Based on the rotation angle θ, the operation of the inverter 20 is controlled so that the AC motor MG outputs a torque according to the torque command value Tqcom. Typically, the semiconductor 20 determines the command value of the current (motor current) supplied from the inverter 20 to the AC motor MG in correspondence with the torque command value Tqcom, and generates a motor current according to the command value. Switching control signals S 1 to S 6 for turning on / off switching elements Q 1 to Q 6 are generated and output to inverter 20.

  Next, power conversion in the inverter 20 controlled by the control device 30 will be described in detail. Various systems can be applied to the motor control by the motor drive system shown in FIG.

  Typically, on / off control of the semiconductor switching elements Q1 to Q6 is performed according to sinusoidal pulse width modulation (PWM) control used as a general PWM control, thereby converting a DC voltage into an AC voltage and an AC motor current. Can be generated. Alternatively, rectangular wave voltage control is also applied to ensure the output of the AC motor in the high speed range.

  As is generally known, PWM control is a control method in which the average value of the output voltage for each period is changed by changing the pulse width of the square wave output voltage for each fixed period. In general, pulse width modulation control is performed by dividing a certain period into a plurality of switching periods corresponding to the period of the carrier wave and performing on / off control of the semiconductor switching element for each switching period.

  In the sine wave PWM control, current feedback control for controlling the motor current is performed according to the current deviation between the current command value set corresponding to the torque command value Tqcom and the detected value of the motor current. Specifically, a sinusoidal voltage command value for each phase of the inverter 20 is set so as to eliminate the current deviation, and a high frequency carrier wave (typically a triangular wave) and a voltage command value (sine wave). ) To control on / off of the upper arm element and the lower arm element of each phase.

  As a result, the alternating current cycle of the motor current is divided into a plurality of switching sections by the carrier wave cycle, and on / off of the semiconductor switching element is controlled for each switching period. That is, an AC voltage (modulation rate = 0..0) such that the fundamental wave component becomes a sine wave within the period of one rotation of the AC motor MG (electrical angle 720 °) by the set of pulsed voltages corresponding to the on / off control. 61) is generated. As described above, in the sine wave PWM control, each semiconductor switching element is controlled to be repeatedly turned on and off according to the carrier frequency.

Here, the motor control configuration by PWM control will be described in detail with reference to FIG.
Referring to FIG. 2, PWM control block 200 includes a current command generation unit 210, coordinate conversion units 220 and 250, a PI calculation unit 240, and a PWM signal generation unit 260. The PWM control block 200 is a functional block realized by executing a program stored in advance in the control device 30 at a predetermined cycle.

Current command generating unit 210, according to the map or the like which is prepared in advance, according to the torque command value Tqcom of AC motor M G, generates a current command value Idcom (d-axis) and Iqcom (q-axis).

  The coordinate conversion unit 220 converts the motor current (iv, iv) detected by the current sensor 24 by coordinate conversion (3 phase → 2 phase) using the rotation angle θ detected by the rotation angle sensor 25 provided in the AC motor MG. Based on iw, iu = − (iv + iw)), the d-axis current id and the q-axis current iq are calculated.

  A deviation ΔId (ΔId = Idcom-id) with respect to the command value of the d-axis current and a deviation ΔIq (ΔIq = Iqcom-iq) with respect to the command value of the q-axis current are input to the PI calculation unit 240. PI calculating section 240 performs PI calculation with a predetermined gain for each of d-axis current deviation ΔId and q-axis current deviation ΔIq to obtain a control deviation, and d-axis voltage command value Vd # and q-axis corresponding to this control deviation Voltage command value Vq # is generated.

  The coordinate conversion unit 250 converts the d-axis voltage command value Vd # and the q-axis voltage command value Vq # to the U-phase, V-phase, W-phase by coordinate conversion (2 phase → 3 phase) using the rotation angle θ of the AC motor MG. Each phase voltage command value Vu, Vv, Vw of the phase is converted.

  The PWM signal generation unit 260 generates the switching control signals S1 to S6 of the inverter 20 shown in FIG. 1 based on the comparison between the voltage command values Vu, Vv, Vw in each phase and a predetermined carrier wave.

  The inverter 20 is subjected to switching control according to the switching control signals S1 to S6 generated by the PWM control block 200, whereby a motor current for outputting torque according to the torque command value Tqcom is supplied to the AC motor MG. The

  On the other hand, in the rectangular wave control, a rectangular wave voltage having a frequency synchronized with the rotation speed of the AC motor MG is applied to increase the fundamental wave component. As a result, the modulation rate of the AC voltage is increased to 0.78. In the rectangular wave control, the voltage phase of the rectangular wave voltage is controlled according to the deviation between the torque command value and the actual torque value.

  Further, in the control configuration in which the sine wave PWM control and the rectangular wave control are switched in accordance with the operation state of the AC motor, in order to smoothly control the output in both switching regions, the intermediate between the sine wave PWM control and the rectangular wave control. In some cases, overmodulation PWM control is applied to obtain a typical modulation rate. In the overmodulation PWM control, the fundamental wave component can be distorted by distorting the voltage command amplitude to the increasing side under the current control similar to the sine wave PWM control, and the modulation factor is 0.61 to .0. It can be increased up to 78.

  As described above, the power conversion by the inverter 20 is roughly classified into rectangular wave control and PWM control (which generally represents sine wave PWM control and overmodulation PWM control). However, in the rectangular wave control, the number of on / off operations of the semiconductor switching element is extremely small, so that the switching loss is inherently low, and the switching speed control by the drive control device for the semiconductor switching element according to the present invention is generally unnecessary.

  On the other hand, in the PWM control, the carrier frequency is generally set to about several kHz to several tens of kHz, so that each semiconductor switching element is controlled to be repeatedly turned on and off at a relatively high frequency. For this reason, if the switching speed is lowered to suppress the surge voltage, an increase in switching loss becomes a problem. On the other hand, if the switching speed is increased to reduce the switching loss, the surge voltage becomes large. For this reason, in PWM control, it is necessary to set the switching speed from both aspects of switching loss reduction and surge voltage suppression.

[Switching speed control]
In the semiconductor switching element driving device according to the embodiment of the present invention, switching speed control as described below is executed in order to achieve both switching loss reduction and surge voltage suppression.

  FIG. 3 is a block diagram illustrating the configuration of the drive control device for the semiconductor switching element according to the embodiment of the present invention. FIG. 3 shows the configuration of one phase of the inverter 20.

  Referring to FIG. 3, two semiconductor switching elements Qa and Qb connected in series are connected between ground line 5 and power supply line 7 through an intermediate node Ni connected to a load. The semiconductor switching element Qa collectively represents the semiconductor switching elements Q2, Q4, and Q6 in FIG. 1, and the semiconductor switching element Qb generally represents the semiconductor switching elements Q1, Q3, and Q5 in FIG. It is to be described. A freewheeling diode Da as an antiparallel diode is connected in parallel to the semiconductor switching element Qa. Similarly, a free wheel diode Db is connected in parallel to the semiconductor switching element Qb.

  In the following, a configuration for voltage drive control of the gate (control electrode) of the semiconductor switching element Qa, which is typically the lower arm element, will be described, but the semiconductor switching element Qb, which is the upper arm element, has the same configuration. Assume that gate voltage drive is controlled.

  First, the voltage / current behavior when the semiconductor switching element Qa is turned on will be described with reference to FIG.

  Referring to FIG. 4, at time t0, a switching control signal is generated so as to switch on / off of semiconductor switching elements Qa, Qb, and turn-on of semiconductor switching element Qa is started. At this time, the transistor current Ic flowing through the semiconductor switching element Qa = 0. On the other hand, in the upper arm, as the semiconductor switching element Qb is turned off, the supply current (motor current) of the phase to the load L (corresponding to the L component of the AC motor MG) is circulated by the free wheel diode Db. (Diode current If ≠ 0). From this state, in response to the turn-on of the semiconductor switching element Qa, the transistor current Ic and the diode current If begin to change so that the motor current flows through the semiconductor switching element Qa.

  From time t0, the gate voltage Vge of the semiconductor switching element Qa is driven from a predetermined off voltage to a predetermined on voltage. As the gate voltage Vge increases, the transistor current Ic begins to flow, while the collector-emitter voltage Vce decreases accordingly.

  On the other hand, the diode current If of the freewheel diode Db gradually approaches 0 as the gate voltage Vge increases. Then, the diode voltage Vak starts increasing from the time when the diode current If = 0.

  Here, the time delay td from the time t0 when the turn-on command is generated to the diode current If = 0 and the diode voltage Vak rises, and the subsequent peak value of the diode voltage Vak, that is, the surge voltage, is the semiconductor switching. It depends on the motor current (corresponding phase) at the start of turn-on of the element Qa.

  As described above, in the PWM control, in response to the turn-on of the semiconductor switching element Qa, the motor current of the phase that was passed during the turn-on period of the semiconductor switching element Qb of the opposite arm is passed to the semiconductor switching element Qa. It becomes. Therefore, the motor current at the start of turn-on of the semiconductor switching element Qa is substantially equal to the energization current (transistor current Ic) after the semiconductor switching element Qa is turned on.

  Here, as disclosed in Non-Patent Document 1, when the semiconductor switching element Qa is turned on in a state where the energization current is low (hereinafter referred to as a low current state), the free wheel diode Db has a small number of carriers. Since the reverse bias voltage is applied, the depletion layer develops in a very short time. As a result, an excessive diode voltage Vak is easily generated in the freewheel diode Db, and the surge voltage rises.

  In the example of FIG. 4, an example of an operation waveform at a low surge voltage level is shown. However, the semiconductor switching element is in a lower current state, that is, a state where the absolute value of the diode current If at time t0 is smaller. Is turned on, the waveform of the diode voltage Vak oscillates and a large surge voltage is generated.

  FIG. 5 is a conceptual diagram showing the dependence of the surge voltage Vsg of the semiconductor switching element and the delay time td until the surge voltage is generated on the energization current.

  As described above, since the surge voltage Vsg depends on the diode voltage Vak, it is relatively large when the conduction current (transistor current) Ic of the semiconductor switching element is low, and decreases as the current increases. Has characteristics. Such a characteristic of Ic-Vsg can be grasped by executing an operation experiment of the semiconductor switching element in advance.

  On the other hand, as shown in FIG. 4, in the region where the energization current Ic is large, the absolute value of the diode current If at the start of turn-on is also large, so that the diode current If returns to 0 and the diode voltage Vak rises. The delay time (hereinafter, surge occurrence time delay) td also increases. For this reason, as shown in FIG. 5, the surge occurrence time delay td shows a characteristic that becomes longer as the energization current Ic becomes higher.

  When the allowable upper limit value Vsmax of the surge voltage Vsg determined according to the rated withstand voltage of the semiconductor switching element is determined based on the Ic-Vsg characteristic as shown in FIG. 5, the boundary current at which the surge voltage Vsg exceeds Vsmax It can be obtained. From the Ic-Vsg characteristic and the boundary current It, a current region 400 where the surge voltage Vsg exceeds the allowable upper limit value Vsmax and a current region 410 where the surge voltage Vsg falls within the allowable upper limit value Vsmax can be defined.

  In the Ic-Vsg characteristic as shown in FIG. 5, the surge voltage Vsg exceeding the allowable upper limit value Vsmax is expected to be generated in the current region 400 where Ic ≦ It, while in the current region 410 where Ic> It, Vsg is expected to be within the allowable upper limit value Vsmax. Hereinafter, the current region 400 is also referred to as a high surge region 400, and the current region 410 is also referred to as a low surge region 410. That is, the current region 400 corresponds to the “first current region” in the present invention, and the current region 410 corresponds to the “second current region” in the present invention.

  The Ic-Vsg characteristic and the Ic-td characteristic change depending on the element temperature. Therefore, the boundary current It is set to reflect the temperature dependence based on the detected value of the element temperature in actual operation by obtaining the change of the boundary current It with respect to the change of the element temperature in advance by experiments or the like. Can do.

  Similarly, in the configuration in which the output voltage Vdc of the DC power supply 10 is variably controlled, the applied voltage (Vce) to the semiconductor switching element changes, so that the Ic-Vsg characteristic and the Ic-td characteristic also change according to the voltage. Therefore, the change of the boundary current It with respect to the change of the applied voltage is obtained in advance by experiments or the like, so that in actual operation, the boundary reflecting the voltage dependence is reflected based on the detected value (DC voltage Vdc) of the voltage sensor 13. The current It can be set.

  As described in detail below, in the semiconductor switching element drive control apparatus according to the embodiment of the present invention, semiconductor switching is performed in a feed-forward manner in accordance with the characteristics of the surge voltage Vsg and the surge generation time delay td with respect to the energized current. The gate voltage driving speed of the element, that is, the switching speed is variably controlled.

  Referring to FIG. 3 again, a drive control circuit 50, drivers 120 and 125, and gate resistors 130 and 135 are provided for each semiconductor switching element.

The driver 120 drives the gate G from the off voltage VL to the on voltage VH via the gate resistor 130 (resistance value R1) when the control signal SD1 from the drive control circuit 50 is on. On the other hand, when the control signal SD2 from the drive control circuit 50 is turned on, the driver 125 drives the gate G from the off voltage VL to the on voltage VH via the gate resistor 135 (resistance value R2). The driver 120 and the gate resistor 130, and the driver 125 and the gate resistor 135 are provided in parallel to the gate G. Incidentally, the ON voltage VH corresponds to a gate voltage for turning on the semiconductor switching element, the off voltage VL corresponds to a gate voltage for turning turn-off the semiconductor switching element.

  Here, it is assumed that the resistance value R2 of the gate resistor 135 is smaller than the resistance value R1 of the gate resistor 130. Therefore, when the gate voltage is driven by the driver 120, relatively low speed switching is performed. On the other hand, when the gate voltage is driven by the driver 125 or by both the drivers 120 and 125, relatively high-speed switching is performed.

  The drive control circuit 50 includes drivers 120 and 125 based on the ON command signal SWon that instructs the ON period of the corresponding semiconductor switching element Qb and the motor current control command value Icom supplied from the inverter 20 to the AC motor MG. By generating the control signals SD1 and SD2, the switching speed when the semiconductor switching element Qb is turned on is controlled.

  Hereinafter, in the present embodiment, the drive control circuit 50 controls the switching speed at turn-on in two stages. Specifically, by turning on only the control signal SD1, only the driver 120 drives the gate voltage at a relatively low speed, and by turning on both the control signals SD1 and SD2, the driver 120 , 125 is used to selectively perform “high-speed switching” in which the gate voltage is driven at a relatively high speed.

  Note that the circuit configuration for realizing switching of the switching speed is not limited to the configuration illustrated in FIG. 3, and a point that any known circuit configuration can be applied will be described for confirmation. For example, instead of switching the gate resistance value, the switching speed may be changed by changing the impedance value including the L and C components. Alternatively, the gate voltage may be driven by a single driver, and the switching speed may be changed by switching the gate resistance value or the impedance value.

FIG. 6 is a block diagram for explaining in detail the configuration of the drive control circuit 50 shown in FIG.
Referring to FIG. 6, drive control circuit 50 includes a switching speed control unit 55 and a current level estimation unit 60. Switching speed control unit 55 includes a determination unit 70, a switching timing setting unit 80, a low speed switching instruction unit 90, and a high speed switching instruction unit 95.

  Based on the current command value Icom at the turn-on command of the semiconductor switching element, the current level estimation unit 60 calculates the energization current estimated value Ic # of the semiconductor switching device in the on period.

  The motor current supplied from the inverter 20 to the AC motor MG as a load is an AC current corresponding to the rotation cycle of the AC motor, but each semiconductor switching provided corresponding to each cycle of the carrier wave in the PWM control. The energization current during each ON period of the element corresponds to an instantaneous value at one point on the time axis of the alternating current.

  The current command value Icom corresponds to the d-axis current command value Idcom and the q-axis current command value Iqcom shown in FIG. 2 in PWM control. That is, the d-axis current command value Idcom and the q-axis current command value Iqcom, which are direct current values, are generally used in two phases (d and q phases) using the rotation angle θ detected by the rotation angle sensor 25 → By conducting the three-phase (U, V, W phase) conversion, it is possible to calculate the estimated current Ic # of the semiconductor switching element after the current turn-on.

  Based on the energized current estimated value Ic # estimated by the current level estimating unit 60, the determining unit 70 determines that the current turn-on is based on the high surge region (low current region) 400 and the Ic-Vsg characteristic shown in FIG. It is determined which of the low surge region (high current region) 410 is performed. Specifically, the above determination is performed based on a comparison between the energization current estimated value Ic # and the boundary current It (FIG. 5).

  The boundary current It is preferably set to reflect the temperature dependency and / or voltage dependency of the boundary current It as described above. For example, a setting map (not shown) of the boundary current It with respect to the element temperature and applied voltage is created in advance based on experimental data and the like, and in actual operation, each semiconductor switching element or the temperature provided in the arrangement region thereof The boundary current It can be appropriately set by referring to the map using the detection value of a sensor (not shown) and / or the detection value (DC voltage Vdc) of the voltage sensor 13.

  When energized current estimated value Ic # is in high surge region 400, determination unit 70 turns on control signal Isl for instructing low-speed switching at the start of turn-on. On the other hand, when energization current estimated value Ic # is in low surge region 410, determination unit 70 turns off control signal Isl at the start of turn-on. As described above, the control signal Isl is turned on when the low speed switching is selected, and is turned off when the high speed switching is selected.

  The low speed switching instruction unit 90 operates when the control signal Isl is turned on during the on period of the on command signal SWon. When operating, the low speed switching instruction unit 90 turns on the control signal SD1 while turning on the control signal SD1. On the other hand, the high-speed switching instruction unit 95 operates when the control signal Isl is OFF during the ON period of the ON command signal SWon. High-speed switching instruction unit 95 turns on both control signals SD1 and SD2 during operation. Note that both the control signals SD1 and SD2 are turned off during the off period of the on command signal SWon. That is, the low-speed switching instruction unit 90 and the high-speed switching instruction unit 95 correspond to the “switching speed instruction unit” according to the present invention.

  In this way, low speed switching and high speed switching are selectively performed based on the control signal Isl generated by the determination unit 70. Specifically, when energization current estimated value Ic # is in high surge region 400, turn-on is started by low-speed switching, and the gate voltage is gently driven from off voltage VL to on voltage VH. For this reason, the current / voltage change also becomes gradual, and the surge voltage is suppressed.

  On the other hand, when the energized current estimated value Ic # is in the low surge region 410, it is not necessary to worry about the surge voltage, so that high speed switching is selected from the start of turn-on. Thereby, a voltage and an electric current can be changed rapidly and reduction of switching loss can be aimed at.

  Further, when the energized current estimated value Ic # is in the high surge region 400, even when the turn-on is started by the low speed switching, the switching loss can be reduced by applying the high speed switching after the surge voltage exceeds the peak. desirable. The peak timing of such a surge voltage can be predicted from the surge occurrence time delay td shown in FIG.

  Therefore, switching timing setting unit 80 sets determination time tc for specifying switching timing based on estimated energization current value Ic #. For example, a setting map (not shown) of determination time tc for energization current estimated value Ic # can be created in advance based on the Ic-td characteristic shown in FIG.

  Then, the switching timing setting unit 80 measures the elapsed time from the turn-on start time by the built-in timer 85, and instructs the switching from the low speed switching to the high speed switching when the elapsed time exceeds the determination time tc. In response to this, the determination unit 70 turns off the control signal Isl.

  As a result, even when turn-on is started by low-speed switching to suppress surge voltage, switching from low-speed switching to high-speed switching is executed after the switching timing when the surge voltage is settled, and switching loss is reduced thereafter. Is planned.

  On the other hand, when energized current estimated value Ic # is in low surge region 410, by setting determination time tc = 0, high-speed switching is applied from the turn-on start point as described above. Accordingly, switching of the switching speed as described above is not executed, and the determination unit 70 maintains the control signal Isl off.

  As described above, since the Ic-td characteristic also has temperature dependency and voltage dependency, the determination time tc is preferably set to further reflect the element temperature and / or the applied voltage. For example, the detection value of the temperature sensor (not shown) and / or the detection value of the voltage sensor 13 (DC) is created by creating a setting map of the determination time tc for the energization current, the element temperature, and / or the applied voltage. The determination time tc can be set more appropriately by referring to the map using the voltage Vdc).

  FIG. 7 shows an operation example of switching speed control by the drive control device for a semiconductor switching element according to the embodiment of the present invention.

  As shown in FIG. 7A, the gate voltage Vge starts to rise in response to the turn-on command being generated at time t0. Accordingly, when the energization current Ic is low, the diode voltage Vak rises relatively quickly and the surge voltage peak increases.

  Therefore, as shown in FIG. 7B, when the energization current Ic is a low current, that is, the high surge region 400, the control signal SD1 is turned on at the start of turn-on, while the control signal SD2 is turned off. Thus, low speed switching is applied.

  Thereafter, at time t1 when the determination time tc has elapsed from the start of turn-on, the switching timing setting unit 80 (FIG. 6) instructs switching from low speed switching to high speed switching. As described above, the switching timing (time t1) is set corresponding to the timing at which the surge voltage exceeds the peak by setting the determination time tc based on the energization current estimated value Ic #.

  As a result, after the surge voltage peak occurs, both the control signals SD1 and SD2 are turned on, and the gate voltage is driven at high speed toward the on voltage VH, thereby reducing the switching loss.

  Referring again to FIG. 7A, when the energization current Ic is a high current, that is, the low surge region 410, the rising timing of the diode voltage Vak, that is, the generation timing of the surge voltage is delayed and its peak is reached. The value is also lowered. Therefore, in the low surge region 410, as shown in FIG. 7C, both the control signals SD1 and SD2 are turned on from the turn-on start time, and high-speed switching is continuously applied. Thereby, switching loss is reduced by rapidly changing the voltage / current.

  As described above, in the semiconductor switching element drive control device according to the embodiment of the present invention, focusing on the relationship between the energization current and the surge voltage behavior, the switching speed at the turn-on time based on the command value to the inverter 20 Can be controlled. Therefore, it is possible to achieve both reduction of surge voltage and suppression of surge voltage without providing a monitoring configuration of the current and / or voltage of the semiconductor switching element and stable switching speed control without feedback control. .

[Modification 1]
FIG. 8 shows a configuration of gate capacitance value adjusting circuit 300 according to the modification of the embodiment of the present invention.

  Referring to FIG. 8, gate capacitance value adjustment circuit 300 includes a plurality of capacitance value adjustment units 310 provided in parallel to gate G of semiconductor switching element Q (which comprehensively represents Q1 to Q6). Have.

  The capacitance value adjustment unit 310 includes an adjustment capacitor 320 and a link element 325 that are connected in series to the gate G. In each capacitance value adjustment unit 310, when the link element 325 is in a non-disconnected state and electrically conductive, the capacitance value of the adjustment capacitor 320 is added to the gate capacitance value. On the other hand, when the link element 325 is cut off by laser input from the outside or the like to be in a non-conductive state, the capacitance value of the adjustment capacitor 320 is not added to the gate capacitance value.

  Therefore, by appropriately cutting the link element 325 from the outside with a laser, so-called trimming can be executed to adjust the gate capacitance value corresponding to the manufacturing variation of the semiconductor switching element.

  By providing such a gate capacitance value adjusting circuit 300, the semiconductor switching element can be adjusted so that the characteristics of Ic-Vsg, td shown in FIG.

  In particular, since the surge generation time delay td from the generation of the turn-on command to the generation of the surge voltage greatly depends on the gate capacitance value, by providing the gate capacitance value adjusting mechanism according to this modification, the switching speed control described above can be performed. It becomes possible to suppress the variation and improve the control accuracy.

[Modification 2]
FIG. 9 is a block diagram showing a learning control configuration in the drive control apparatus for a semiconductor switching element according to the embodiment of the present invention.

  Referring to FIG. 9, drive control circuit 50 further includes a learning control unit 110 in addition to the configuration shown in FIG. 6. Further, a voltage dividing circuit 27 for dividing the diode voltage Vak is provided for each freewheel diode. The divided voltage Vak # generated by the voltage dividing circuit 27 is detected by the learning control unit 110.

  The learning control unit 110 executes the switching operation of the semiconductor switching elements Qa and Qb during a predetermined learning period, and grasps the actual surge voltage based on the divided voltage Vak # at that time. Then, learning control for correcting (updating) the boundary current It and the determination time tc described above is executed based on the results of the surge voltage during the learning period.

  For example, during the learning period, when the motor drive system 100 is activated, the current command command value (for example, Idcom, FIG. 2) is set such that the AC motor MG mainly consumes reactive power and does not generate torque. Iqcom) can be realized. In this way, it is possible to sample the surge voltage behavior at the time of turn-on by turning on / off each semiconductor switching element constituting the inverter 20 without causing output to the AC motor MG as a load. As an example, when the switching frequency of each semiconductor switching element is 10 kHz, sampling of 10,000 turn-on timings can be executed by securing a learning period of 1 second. That is, it is sufficiently possible to provide a learning period when the motor drive system 100 is activated.

FIG. 10 is a flowchart for explaining the learning control operation by the learning control unit 110.
Referring to FIG. 10, learning control unit 110 determines in step S100 whether or not an operation start command for motor drive system 100 including inverter 20 has been generated. For example, in the motor drive system 100 mounted on a hybrid vehicle, the operation start instruction corresponds to the start instruction of the hybrid system. If the operation start command is not generated (NO at S100), the learning is not started and the process is terminated as it is.

  When the operation start command is generated (YES in S100), learning control unit 110 further determines whether or not the learning start condition is satisfied in step S110. The establishment of the learning start condition is determined based on, for example, whether each sensor is normal or whether the semiconductor switching element is abnormal. When an output instruction is issued to AC motor MG as a load, S100 is determined as NO and learning is not executed. That is, learning is performed using a period during which no output instruction is issued to AC motor MG.

  The learning control unit 110 starts learning in step S120 when the learning start condition is satisfied (when YES is determined in S110). Then, in step S130, the learning control unit 110 uses the current command Iqcom (Idcom, Iqcom) according to the learning current command pattern so that the AC motor MG consumes reactive power without generating torque as described above. Is generated.

  Then, according to the current command value generated in this manner, each semiconductor switching element is switched by applying switching control based on the currently set boundary current It and the determination time tc. Then, the learning control unit 110 samples the diode voltage Vak at the time of turn-on that has occurred at the time of switching in step S140 (step S150).

In step S160, learning control unit 110 evaluates the surge voltage based on diode voltage Vak sampled in step S150. For example, in the current region near the current boundary current It, when there is a margin in the margin to the allowable upper limit value of the surge voltage maximum value, the boundary current It is updated to the lower side, while the margin is insufficient The boundary current It is updated to the rising side. Thus, it modified according to the learning result boundaries current It of the high surge region 4 00 and the low surge region 4 10.

Similarly, in the high surge region, the surge voltage at the switching timing from the low speed switching to the high speed switching according to the determination time tc is evaluated, and when there is a margin in the allowable upper limit value, the determination time tc is shortened. On the other hand, if the margin is insufficient, the determination time tc is updated to the extension side. Thereby, the switching timing from the low speed switching to the high speed switching at the time of turn-on in the high surge region 400 can be corrected according to the learning result.

  The learning control unit 110 continues the learning operation in steps S130 to S160 until a predetermined learning end condition is satisfied in step S170. For example, the learning end condition is satisfied by elapse of a predetermined time, completion of a desired learning item, generation of an output instruction to the load (AC motor MG), or the like.

  According to the embodiment of the present invention, by configuring the map values of the setting map of the boundary current It and the determination time tc described above by the learning operation as described above, it is possible to cope with the manufacturing variation and change with time of each semiconductor element. Switching control can be executed more effectively.

  Note that it is difficult to perform learning including temperature dependency and voltage dependency in the learning period, but for the boundary current It and the determination time tc, a basic value based on the energized current and a correction value based on temperature and voltage changes. If the configuration is set according to the sum, the map value can be appropriately updated by the learning operation for the map for setting the basic value.

  In the semiconductor switching element drive control device according to the present embodiment, the switching control is changed by switching between two switching speeds of low speed and high speed, thereby suppressing surge voltage and reducing switching loss with a simpler circuit configuration. However, it is possible to subdivide the switching speed into three or more stages.

  In the present embodiment described above, a three-phase inverter using a motor (or motor generator) mounted on a hybrid vehicle or an electric vehicle as an example of an application example of a drive control device for a semiconductor switching element has been illustrated. The application of is not limited to this. That is, a semiconductor switching element that performs so-called hard switching, such as a single-phase inverter, a DC / DC converter, a step-up chopper, a switching mode amplifier, etc., as long as it constitutes a semiconductor power conversion device that controls a supply current to a load according to a command value The present invention can be applied to the switching control. Moreover, the point which is not specifically limited also about a load is described definitely.

Furthermore, although the IGBT is exemplified in the present embodiment for the semiconductor switching element, the switching speed control according to the present invention is applied to other voltage-driven switching elements such as a power MOS (Metal Oxide Semiconductor) transistor. it can. Further, similarly to the gate voltage driving speed in the present embodiment, the semiconductor power conversion device configured by a current driving semiconductor switching element such as a power bipolar transistor by controlling the current driving speed of the control electrode (base). The switching control according to the present invention can also be applied to.

  The embodiment disclosed this time should be considered as illustrative in all points and not restrictive. The scope of the present invention is defined by the terms of the claims, rather than the description above, and is intended to include any modifications within the scope and meaning equivalent to the terms of the claims.

1 is a block diagram illustrating an overall configuration of a motor drive system to which a drive control device for a semiconductor switching element according to the present invention is applied. It is a control block diagram explaining the motor control structure by PWM control. It is a block diagram explaining the structure of the drive control apparatus of the switching element by embodiment of this invention. It is an operation | movement waveform diagram which shows the voltage and current behavior at the time of turn-on of a switching element. It is a conceptual diagram which shows the dependence with respect to the electric current of the delay time until the surge voltage of a switching element and surge voltage generation | occurrence | production. FIG. 4 is a block diagram for explaining in detail the configuration of a drive control circuit shown in FIG. 3. It is a conceptual diagram which shows the operation example of the switching speed control by the drive control apparatus of the semiconductor switching element by embodiment of this invention. It is a conceptual diagram which shows the structure of the gate capacity | capacitance adjustment circuit according to the modification of embodiment of this invention. It is a block diagram which shows the learning control structure in the drive control apparatus of the semiconductor switching element according to embodiment of this invention. It is a flowchart explaining the learning control operation | movement by the learning control part of FIG.

Explanation of symbols

  5 Ground wire, 7 Power line, 10 DC power supply, 13 Voltage sensor, 15 U phase circuit, 16 V phase circuit, 17 W phase circuit, 20 Inverter, 24 Current sensor, 25 Rotation angle sensor, 27 Voltage divider circuit, 30 Control Equipment (ECU), 50 drive control circuit, 55 switching speed control section, 60 current level estimation section, 70 determination section, 80 switching timing setting section, 85 timer, 90 low speed switching instruction section, 95 high speed switching instruction section, 100 motor drive System, 110 Learning control unit, 120, 125 driver, 130, 135 Gate resistance, 200 PWM control block, 210 Current command generation unit, 220, 250 Coordinate conversion unit, 240 PI operation unit, 260 PWM signal generation unit, 300 Gate capacity Value adjustment circuit, 310 capacitance value adjustment unit, 32 Adjustment capacity, 325 link element, 400 High surge region (low current region), 410 Low surge region (high current region), C0 smoothing capacitor, D1-D6, Da, Db Freewheel diode, G gate, Ic transistor current (energization) Current), Ic # estimated current value, Icom current command value, Idcom d-axis current command value, Idcom q current command value, If diode current, Isl control signal (high speed / low speed switching), It boundary current, iu, iv, iw Three-phase current (motor current), MG AC motor (load), Q, Q1 to Q6, Qa, Qb semiconductor switching element, R1, R2 resistance value (gate resistance), S1 to S6 switching control signal, SD1, SD2 control Signal (driver), SWon ON command signal 310 Capacity adjustment unit 320 adjustment capacity, 325 link element, tc determination time (low speed to high speed switching timing), td surge generation time delay, Tqcom torque command value, Vak diode voltage, Vak # divided voltage (diode voltage), Vce collector-emitter voltage , Vdc DC voltage (inverter DC side voltage), Vge gate voltage, VH on voltage, VL off voltage, Vsg surge voltage, Vsmax allowable upper limit (surge voltage), Vu, Vv, Vw phase voltage command value, ΔId Current deviation (D-axis), ΔIq Current deviation (q-axis), θ rotation angle (AC motor).

Claims (8)

A drive control device for a semiconductor switching element constituting a semiconductor power conversion device for controlling a supply current to a load according to a current command value,
The semiconductor switching element is turned on or off in response to the voltage or current of the control electrode,
The drive control device includes:
Current estimation means for estimating an energization current that flows when the semiconductor switching element is turned on based on the current command value when a turn-on command is generated;
In accordance with a first current characteristic indicating a relationship between the energizing current of the semiconductor switching element and the magnitude of a surge voltage generated in advance, the turn-on of the semiconductor switching element is based on the estimated energizing current by the current estimating means. Switching speed control means for changing the driving speed of the voltage or current of the control electrode at the time,
The switching speed control means includes
According to the first current characteristic, it is determined whether the estimated energized current belongs to a first current region where the surge voltage exceeds an allowable value or a second current region where the surge voltage is equal to or less than the allowable value. Determination means to perform,
Switching speed indicating means for lowering the driving speed at the start of turn-on when the estimated energized current is in the first current region than when the estimated energized current is in the second current region; seen including,
The drive control device includes:
Setting a learning period in which the current command value is generated so that the load mainly consumes reactive power when the semiconductor power converter is started up, and an actual measured value of the surge voltage of the semiconductor switching element in the learning period A drive control device for a semiconductor switching element , further comprising learning control means for correcting a determination value indicating a boundary between the first and second current regions based on the above . The switching speed instruction means sets the drive speed at the start of the turn-on to the first speed when the estimated energization current is in the first current region, and sets the voltage or current of the control electrode. 2. The drive control device for a semiconductor switching element according to claim 1, wherein the drive speed is increased from the first speed after the surge voltage is settled after a predetermined time has elapsed since the drive was started. The switching speed control means includes
According to a second current characteristic indicating a relationship between the energization current of the semiconductor switching element and the voltage of the control electrode or the time from when the drive of the current is started until the surge voltage is generated, obtained in advance. Switching timing setting means for setting switching timing for increasing the drive speed based on the estimated energization current by the current estimation means;
3. The drive control of a semiconductor switching element according to claim 2, wherein the switching speed instruction means increases the drive speed from the first speed to the second speed when the switching timing is reached based on the elapsed time. apparatus.   The switching speed instruction means sets the driving speed at the start of the turn-on to the first speed when the estimated energization current is in the first current region, while the estimated energization current is the first energization current. When the current is within a current range of 2, the drive speed at the start of turn-on is set to a second speed higher than the first speed, and the drive speed is fixed to the second speed throughout the turn-on period. The drive control apparatus of the semiconductor switching element of Claim 1.   The switching speed instructing means is configured such that when the estimated energization current is in the first current region, the surge voltage is settled after a predetermined time has elapsed from the start of driving the voltage or current of the control electrode. 5. The drive control device for a semiconductor switching element according to claim 4, wherein the drive speed is switched from the first speed to the second speed.   6. The drive control device for a semiconductor switching element according to claim 1, further comprising a capacitance value adjustment circuit for adjusting a capacitance value of the control electrode of the semiconductor switching element by trimming. A drive control device for a semiconductor switching element constituting a semiconductor power conversion device for controlling a supply current to a load according to a current command value,
The semiconductor switching element is turned on or off in response to the voltage or current of the control electrode,
The drive control device includes:
Current estimation means for estimating an energization current that flows when the semiconductor switching element is turned on based on the current command value when a turn-on command is generated;
In accordance with a first current characteristic indicating a relationship between the energizing current of the semiconductor switching element and the magnitude of a surge voltage generated in advance, the turn-on of the semiconductor switching element is based on the estimated energizing current by the current estimating means. Switching speed control means for changing the driving speed of the voltage or current of the control electrode at the time,
The switching speed control means includes
According to the first current characteristic, it is determined whether the estimated energized current belongs to a first current region where the surge voltage exceeds an allowable value or a second current region where the surge voltage is equal to or less than the allowable value. Determination means to perform,
Switching speed indicating means for lowering the driving speed at the start of turn-on when the estimated energized current is in the first current region than when the estimated energized current is in the second current region; Including
The switching speed instruction means sets the drive speed at the start of the turn-on to the first speed when the estimated energization current is in the first current region, and sets the voltage or current of the control electrode. After the surge voltage has settled after a predetermined time has elapsed since the start of driving, the driving speed is increased above the first speed,
The switching speed control means includes
According to a second current characteristic indicating a relationship between the energization current of the semiconductor switching element and the voltage of the control electrode or the time from when the drive of the current is started until the surge voltage is generated, obtained in advance. Switching timing setting means for setting switching timing for increasing the drive speed based on the estimated energization current by the current estimation means;
The switching speed instruction means increases the driving speed from the first speed to the second speed when the switching timing is reached based on the elapsed time,
The drive control device includes:
Setting a learning period in which the current command value is generated so that the load mainly consumes reactive power when the semiconductor power converter is started up, and an actual measured value of the surge voltage of the semiconductor switching element in the learning period A drive control device for a semiconductor switching element, further comprising learning control means for correcting the switching timing based on the above. The load is an AC motor whose supply current is controlled by pulse width modulation control,
2. The drive control device for a semiconductor switching element according to claim 1, wherein the current estimation means calculates the estimated energization current using a rotation angle of the AC motor and a current command value by vector control.

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