WO2009014245A1

WO2009014245A1 - Device for controlling drive of semiconductor switching element

Info

Publication number: WO2009014245A1
Application number: PCT/JP2008/063488
Authority: WO
Inventors: Satoshi Hirose; Hiroo Fuma
Original assignee: Toyota Jidosha Kabushiki Kaisha
Priority date: 2007-07-23
Filing date: 2008-07-22
Publication date: 2009-01-29
Also published as: JP2009027881A; JP4372812B2

Abstract

A semiconductor switching element has a characteristic that the diode voltage (Vak) in the low current region rises relatively sharp and has a high peak value, and that in the high current region rises relatively slowly and has a low peak value. When receiving a command to turn on the semiconductor switching element, a control device estimates the applied current (Ic) on the basis of the current command value. When the operation in the low-current region is inferred, turn-on is started by low-speed switching. When the operation in the high-current region is inferred, turn-on is carried out by high-speed switching. When it is inferred that the peak of the surge voltage passes in the low-current region after a predetermined time from the start of the turn-on, the low-speed switching is changed to the high-speed switching. This change timing is determined according to the applied current estimated on the basis of the current command value. Consequently, without causing feedback of the voltage and current, the switching speed control of the semiconductor switching element can be carried out when turning on the semiconductor switching element.

Description

Description: Semiconductor switching element drive control device Technical Field

The present invention relates to a drive control device for a semiconductor switching element, and more specifically, a technique for suppressing voltage fluctuation (surge voltage) generated when a semiconductor switching element constituting a semiconductor power conversion device such as an inverter converter is turned on. Concerning. Background art

As a countermeasure to increase the output density of inverters and converters, active gate control applied when turning on and off semiconductor switching elements such as IGBT (Insulated Gate Bipolar Transistor J and NOS MONO (Metal Oxide Semiconductor) transistor) is known. Active gate control is a technology that optimizes the voltage drive speed (switching speed) of the gate (control electrode) by switching the gate resistance directly to dynamic during the turn-on or turn-off of the semiconductor switching element. By controlling the driving speed, it is possible to achieve both switching loss reduction and surge voltage suppression, which are in a trade-off relationship with the switching speed.

For example, Japanese Patent Laid-Open No. 2 0 2-1 2 5 3 6 3 (hereinafter referred to as Patent Document 1), Japanese Patent Laid-Open No. 2 0 06-2 2 2 5 93 (hereinafter referred to as Patent Document 2), or US Pat. No. 6,208,185 (hereinafter referred to as Patent Document 3) discloses an active gate that changes a gate resistance value according to a current / voltage state when a semiconductor switching element is turned on and / or turned off. Controls are disclosed.

Further, the configuration of the gate drive circuit for realizing the active gate control is not limited to the configuration for switching the gate resistance value, but as disclosed in Japanese Patent Application Laid-Open No. 2 0 5-3 9 98 8 (hereinafter referred to as Patent Document 4). A circuit configuration using a rear tuttle has also been proposed. On the other hand, in IGBT modules with a free wheel diode connected in parallel with the semiconductor switching element, the operation behavior of the free diode has a great influence on the surge voltage when the semiconductor switching insulator is turned on. Fumio Nagahama, 2 others, “Analysis of reverse recovery phenomenon of diode from transient on state j, Fuji Times, Fuji Electric Co., Ltd., February 2000, Vol. 7 4 No. 2 2 0 0 1 1 4 9-1 5 2 (hereinafter referred to as Non-Patent Document 1) In particular, in Non-Patent Document 1, a reverse bias is applied from a state where there are few diode carriers at the time of turn-on in a low current state. Therefore, it is described that the surge voltage increases as a result of the depletion layer developing in a very short time.

The general active control controls disclosed in Patent Documents 1 to 4 monitor the current or voltage of a semiconductor switching element that is turned on and off, and control the switching speed according to the detected value or its time differential value. The control configuration is changed.

For this reason, in applications where switching frequency is high and switching time with active gate control is very short, very high control responsiveness is required to monitor voltage and current and control the switching speed in response. Is done. Furthermore, it is necessary to control the gate drive circuit, monitor circuit or semiconductor switching element characteristics, and external fluctuation factors such as temperature.

Because of these factors, it is practically difficult to ensure the control stability of the active gate control based on the feedback of the monitoring results.If control fails due to the presence of control delay, etc., semiconductor switching There is a possibility that the device will fail. In other words, in order to avoid such a control failure, the control design must have a margin, and the switching loss reduction effect is suppressed. In addition, the addition of a voltage / current monitor circuit for the semiconductor switching element itself increases costs and circuit dimensions.

Disclosure of the invention

The present invention has been made in order to solve the above-described problems. The object of the present invention is to provide a semiconductor switching element voltage and Z or current monitoring result without feedback and at the time of turn-on. Surge voltage suppression and switch It is to perform stable switching speed control to achieve both reduction in switching loss.

A drive control device according to the present invention is a drive control device for a semiconductor switching element that constitutes a semiconductor power conversion device for controlling a supply current to a load according to a current command value, and includes a current estimation unit, a switching speed control unit With. Then, the semiconductor switching element is turned on or off in response to the voltage or current of the control electrode. The current estimation unit estimates an energization current that flows when the semiconductor switching element is turned on based on the current command value when the turn-on command is generated. The switching speed control unit determines the semiconductor switching based on the estimated conduction current by the current estimation unit according to the first current characteristic indicating the relationship between the conduction current of the semiconductor switching element and the generated surge voltage. Change the drive speed of the voltage or current of the control electrode when the device is turned on.

According to the drive control device of the semiconductor switching element, the energization current flowing through the semiconductor switching element after the turn-on is estimated based on the current command value to the semiconductor power converter, and the relationship between the energization current and the surge voltage behavior is shown. The drive speed of the control electrode at the time of turn-on (ie, switching speed) can be changed according to the characteristics obtained in advance. In other words, by changing the switching speed based on the current command value at the time of turn-on command, the current control Z or voltage monitor configuration is not provided, and the stable control configuration without feedback control is used. Surge voltage reduction and surge voltage suppression can be achieved. Preferably, the switching speed control unit includes a determination unit and a switching speed instruction unit. The determination unit determines whether the estimated energization current belongs to the first current region where the surge voltage exceeds the allowable value or the second current region where the surge voltage is less than the allowable value according to the first current characteristic. . The switching speed instruction unit makes the driving speed at the start of turn-on slower when the estimated energization current is in the first current region than when the estimated energization current is in the second current region.

With this configuration, the surge voltage can be suppressed by starting the turn-on at a relatively low switching speed in the current region (first current region) where the surge voltage is expected to increase. it can. 'On the other hand, Sir In the current region (second current region) where the voltage is expected to be small, switching loss can be reduced by starting the turn-on at a relatively high switching speed.

More preferably, the switching speed control unit sets the drive speed at the start of turn-on to the first speed and drives the control electrode voltage or current when the estimated energization current is within the first current region. The drive speed is increased from the first speed according to the elapsed time from the start. Particularly in such a configuration, the switching speed control unit further includes a switching timing setting unit. The switching timing setting unit is a second current characteristic showing a relationship between a conduction current of the semiconductor switching element obtained in advance and a time from when the drive of the control electrode voltage or current is started until the surge voltage is generated. Accordingly, the switching timing for increasing the drive speed is set based on the estimated energization current by the current estimation unit. Further, the switching speed instruction unit increases the driving speed from the first speed to the second speed when the switching timing is reached based on the elapsed time.

With this configuration, in the current region where the surge voltage increases (first current region), the surge voltage is suppressed by reducing the switching speed at the start of turn-on, while the surge voltage is generated. After the time exceeds the peak, the switching speed can be increased to reduce the switching loss. In particular, the switching speed rise timing can be set appropriately without feedback according to a pre-determined characteristic that shows the relationship between the energization current and the surge voltage behavior (delay time). .

Alternatively, more preferably, the switching speed indicator sets the driving speed at the start of turn-on to the first speed when the estimated energizing current is in the first current region, while the estimated energizing current is the second In the current region, the drive speed when the turn-on is open is set to a second speed higher than the first speed, and the drive speed is fixed to the second speed throughout the turn-on period.

With this configuration, the surge voltage can be excessive. In the low current region (second current region), the switching speed is set to a relatively high speed from the start of turn-on to the on-period. Maximum reduction of switching loss Can be aimed at.

More preferably, when the estimated energization current is within the first current region, the switching speed instruction unit sets the drive speed according to the elapsed time since the start of driving of the control electrode voltage or current. Switch from the first speed to the second speed.

By adopting such a configuration, the above switching speed control can be executed by changing the switching speed in two stages, so that the circuit configuration of the drive control device can be simplified.

More preferably, the drive control device further includes a learning control unit. The learning control unit sets a learning period in which a current command value is generated so that the load mainly consumes reactive power when the semiconductor power converter is started up, and based on the measured value of the surge voltage during the learning period, Correct the judgment value indicating the boundary between the 1st and 2nd current regions. Alternatively, the learning control unit corrects the switching timing based on the measured value of the surge voltage during the learning period.

With this configuration, the first and second current regions are determined using the period until the load is actually operated as the power converter is started at the start of load operation. It is possible to realize learning control in which the value or switching speed switching timing is updated based on the actual surge voltage during the learning period. As a result, the switching speed control when the semiconductor switching element is turned on can be more appropriately executed in actual operation after execution of learning control, so that both suppression of surge voltage and reduction of switching loss can be achieved more effectively. .

Preferably, the drive control device further includes a capacitance value adjustment circuit for adjusting the capacitance value of the control electrode of the semiconductor switching element by trimming.

By adopting such a configuration, the effect of the switching speed control described above can be enhanced in response to manufacturing variations in the capacitance value of the control electrode of the semiconductor switching element.

Preferably, the load is an AC motor whose supply current is controlled by pulse width modulation control, and the current estimation unit uses the rotation angle of the AC motor and a current command value by vector control to estimate the energization current. Is calculated. By adopting such a configuration, in the drive control device for the semiconductor switching element constituting the semiconductor cover device having the AC motor as a load, the current command value of the AC motor (d-axis) according to the rotation angle and vector control of the AC motor. , Q-axis current command value), it is possible to easily and accurately estimate the on-current after the semiconductor switching element is turned on.

Therefore, the main advantage of the present invention is that a stable switching for achieving both suppression of surge voltage and reduction of switching loss at the time of turn-on without involving the feed pack of the voltage and z or current monitoring result of the semiconductor switching element. It is in the point which can execute the etching speed control. Brief Description of Drawings

FIG. 1 is a block diagram illustrating an overall configuration of a motor drive system to which a drive control device for a semiconductor switching element according to the present invention is applied.

FIG. 2 is a control block diagram for explaining a motor control configuration by PWM control. Figure 3 shows a book! FIG. 5 is a block diagram illustrating a configuration of a drive control device for a switching element according to a clear embodiment.

Fig. 4 is an operation waveform diagram showing the voltage / current behavior when the switching element is turned on.

Fig. 5 is a conceptual diagram showing the dependence of the switching element's surge voltage and the delay time until the surge voltage is generated on the energizing current.

FIG. 6 is a block diagram illustrating in detail the configuration of the drive control circuit shown in FIG. FIGS. 7A to 7C are conceptual diagrams showing an operation example of switching speed control by the drive control device of the semiconductor switching element according to the embodiment of the present invention.

FIG. 8 is a conceptual diagram showing a configuration of a gate capacitance adjustment circuit according to a modification of the embodiment of the present invention.

FIG. 9 is a block diagram showing a learning control configuration in the drive control apparatus for a semiconductor switching element according to the embodiment of the present invention.

FIG. 10 is a flowchart for explaining the learning control operation by the learning control unit of FIG. BEST MODE FOR CARRYING OUT THE INVENTION

Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings. In the following, the same or corresponding parts in the drawings are denoted by the same reference numerals, and the description thereof will not be repeated in principle.

FIG. 1 is an overall configuration diagram of a motor drive system to which a drive control device for a semiconductor switching element according to the present invention is applied.

Referring to FIG. 1, motor drive system 100 according to the embodiment of the present invention includes direct current power supply 10, voltage sensor 13, smoothing capacitor CO, inverter 20, and control device 30. And an AC motor MG as a load.

AC motor MG is, for example, a drive motor for generating torque for driving drive wheels of a hybrid vehicle or an electric vehicle. Alternatively, the AC motor MG may be configured to have a function of a generator driven by an engine, or may be configured to have a function of an electric motor and an electric machine. Furthermore, AC motor MG operates as an electric motor for the engine, and may be incorporated into a hybrid vehicle, for example, so that it can start an engine.

The DC power supply 10 outputs a DC voltage between the power supply line 7 and the ground line 5. The DC power supply 10 can be charged by a DC voltage between the power supply line 7 and the ground line 5. The DC power supply 10 is typically composed of a secondary battery such as nickel metal hydride or lithium ion or a power storage device of an electric double layer capacitor. A converter for changing the voltage level of the output voltage (DC) of the power storage device is provided to variably control the output voltage of DC power supply 10, that is, the voltage between power supply line 7 and ground line 5. It is good also as a structure to be.

The smoothing capacitor C O is connected between the power line 7 and the ground line 5. The voltage across the smoothing capacitor CO corresponding to the DC voltage of the inverter 20 is detected by the voltage sensor 13, and the detected value is the control device 3 configured by the electronic control unit (ECU) 3. Sent to 0.

Inverter 20 is provided in parallel between power line 7 and ground line 5, Circuit 15, phase V circuit 16, and phase W circuit 17. Each phase circuit includes a power semiconductor switching element connected in series between a power line 7 and a ground line 5. IGBT (Insulated Gate Bipolar Transistor) is typically applied as a power semiconductor switching element (hereinafter simply referred to as “semiconductor switching element”).

The U-phase circuit 15 is composed of a semiconductor switching element Q 1 which is an upper arm element and a semiconductor switching element Q 2 which is a lower arm element. The V-phase circuit 16 is a semiconductor switching element Q 3 which is an upper arm element. The W-phase circuit 17 is composed of a semiconductor switching element Q5, which is an upper arm element, and a semiconductor switching element Q6, which is a lower arm element. In addition, free wheel diodes D1 to D6 are connected to the semiconductor switching elements Q1 to Q6, respectively, for flowing a current in the direction opposite to the semiconductor switching elements. On / off of the semiconductor switching elements Q 1 to Q 6 is controlled by switching control signals S 1 to S 6 from the control device 30.

The AC motor MG is typically a three-phase permanent magnet motor, with U, V, and W phases.

One end of the three coils is commonly connected to the neutral point. The other end of the U, V, W phase coil is connected to the connection point of the upper arm element and the lower arm element in the U, V, W phase of inverter 20.

When the torque command value of the AC motor MG is positive (Tq com> 0), the inverter 20 applies the DC voltage from the smoothing capacitor C 0 to the switching control signals S 1 to S 6 from the control device 30. The AC motor MG is driven to output positive torque by converting to AC voltage by the switching operation of the semiconductor switching elements Q1 to Q6 that responded.

Further, when the torque command value of the AC motor MG is zero (Tq com = 0), the inverter 20 converts the DC voltage into the AC voltage by the switching operation in response to the switching control signal S1-S6. Then, drive AC motor MG so that the torque becomes zero. As a result, AC motor MG is driven to generate zero or positive torque designated by torque command value T q c om.

In addition, a hybrid vehicle equipped with the motor drive system 100 or an electric vehicle. During regenerative braking of an electric vehicle, the torque command value T qcom of AC motor MG is set negative (T ci com <0). In this case, the inverter 20 converts the AC voltage generated by the AC motor MG into a DC voltage by a switching operation in response to the switching control signals S1 to S6, and converts the converted DC voltage (system voltage). It can be used to charge a DC power supply 10 through a smoothing capacitor C0. In this context, regenerative braking refers to braking with regenerative power generation when a driver operating a hybrid vehicle or an electric vehicle is operated with regenerative power generation, or without operating the foot brake, but the accelerator pedal is This includes decelerating (or stopping acceleration) the vehicle while regenerating power by turning it off.

The current sensor 24 is composed of a hall sensor or the like, detects the motor current supplied from the inverter 20 to the AC motor MG, and outputs the detected value to the control device 30. Since the sum of the instantaneous values of the three-phase currents iu, i V and iw is zero, as shown in Fig. 1, the current sensor 24 has a motor current for two phases (for example, V-phase current i V and It suffices to arrange it to detect the W-phase current iw).

A rotation angle sensor (typically, a resolver) 25 detects the rotation angle 0 of AC motor MG and sends the detected rotation angle 0 to control device 30. The control device 30 also calculates the rotational speed (rotational speed) of the AC motor MG based on the rotational angle 0. Controller 30 is a Tonlek command value that indicates the output torque of AC motor MG, which is a load. T qcom DC voltage V dc detected by voltage sensor 1 3, motor current i V, iw from current sensor 2 4 Based on the rotation angle 0 from the rotation angle sensor 25, the operation of the inverter 20 is controlled so that the AC motor MG outputs a torque according to the torque command value Tqcom. Typically, the command value of the current (motor current) supplied from the inverter 20 to the AC motor MG is determined in accordance with the torque command value T qcom, and the motor current is generated according to the command value. Thus, switching control signals S 1 to S 6 for turning on / off the semiconductor switching elements Q 1 to Q 6 are generated and output to the inverter 20.

Next, power conversion in the inverter 20 controlled by the control device 30 will be described in detail. Various methods can be applied to motor control by the motor drive system shown in Fig. 1. Typically, the semiconductor switching elements Q1 to Q6 are turned on and off according to sinusoidal pulse width modulation (PWM) control, which is used as general PWM control, to convert DC voltage to AC voltage, AC motor current can be generated. Alternatively, rectangular wave voltage control is also applied to ensure output in the high speed range of the AC motor.

As is generally known, PWM control is a control method in which the average value of the output voltage for each period can be changed by changing the pulse width of the square wave output voltage for each fixed period. In general, the pulse width modulation control is performed by dividing a fixed period into a plurality of switching periods corresponding to the carrier period and performing on / off control of the semiconductor switching element for each switching period.

In sine wave PWM control, current feedback control is performed to control the motor current in accordance with the current deviation between the current command value set corresponding to the torque command value T q co m and the detected motor current value. Specifically, a sinusoidal voltage command value is set for each phase of the inverter 20 so as to eliminate the current deviation, and a high-frequency carrier wave (typically a triangular wave) and a voltage command Based on the voltage comparison with the value (sine wave), on / off of the upper and lower arm elements of each phase is controlled.

As a result, one AC period of the motor current is divided into a plurality of switching sections by the carrier wave period, and the on / off state of the semiconductor switching element is controlled for each switching period. In other words, an AC voltage (modulation rate) whose fundamental wave component becomes a sine wave within the period of one rotation of the AC motor MG (electrical angle 720 °) is obtained by a collection of pulse voltages corresponding to this on / off control. = 0. 6 1) is generated. Thus, in the sine wave PWM control, each semiconductor switching element is controlled to be turned on and off according to the carrier frequency.

Here, the motor control configuration by PWM control will be described in detail with reference to FIG.

Referring to FIG. 2, PWM control block 200 includes a current command generator 2 1 0, a coordinate converter 2 2 0, 2 5 0, a PI calculator 2 4 0, and a P WM signal generator. 2 6 0 included. The PWM control block 200 indicates a function block realized by executing a program stored in the control device 30 in advance at a predetermined cycle. The

Current command generation unit 210 generates current command values I d com (d-axis) and I q com (q-axis) according to torque command value Tq com of AC motor MG according to a map created in advance.

The coordinate conversion unit 220 is detected by the current sensor 24 by coordinate conversion (3 phase → 2 phase) using the rotation angle 0 detected by the rotation angle sensor 25 provided in the AC motor MG.

Calculate the d-axis current i d and the q-axis current i q based on the motor current (iv, iw, iu = — (iv + iw)).

The PI calculation unit 240 includes a deviation Δ I d (Δ I d = I dc om—id) for the d-axis current command value and a deviation Δ I q (Δ I q = I qco m- iq) is entered. The PI calculation unit 240 obtains a control deviation by performing PI calculation with a predetermined gain for each of the d-axis current deviation Δ I d and the q-axis current deviation Δ I q, and the d-axis voltage command value V corresponding to the control deviation is calculated. d # and q axis voltage command value Vq # is generated.

The coordinate converter 250 converts the d-axis voltage command value Vd # and q-axis voltage command value (1 # into 1; phase, V-phase) by coordinate conversion using the rotation angle Θ of the AC motor MG (2 phase → 3 phase). , W phase phase voltage command value Vu, V v. Convert to Vw.

The PWM signal generation unit 260 generates the switching control signals S1 to S6 of the inverter 20 shown in FIG. 1 based on the comparison of the voltage command values Vu, VV, Vw in each phase and a predetermined carrier wave. .

The inverter 20 is switching-controlled according to the switching control signals S 1 to S 6 generated by the PWM control block 200, so that the AC motor MG can output torque according to the torque command value Tq com. Motor current is supplied.

On the other hand, in the rectangular wave control, a rectangular wave voltage having a frequency synchronized with the rotational speed of the AC motor MG is applied to increase the fundamental wave component. This increases the modulation rate of the AC voltage to 0.78. In the rectangular wave control, the voltage phase of the rectangular wave voltage is controlled according to the deviation between the torque command value and the actual torque value. In addition, in the control configuration that switches between sine wave PWM control and rectangular wave control according to the operating state of the AC motor, the sine wave P WM control and rectangular wave control are used to smoothly control the output in the switching area of both. Overmodulation P WM control may be applied to obtain an intermediate modulation rate. In overmodulation P WM control, the fundamental wave component can be distorted by distorting the voltage command amplitude to the increasing side under current control similar to sine wave P WM control, and the modulation factor is set to 0.6. It can be increased to a range of 1 to 0.78. As mentioned above, the power conversion by the inverter 20 is the rectangular wave control and? WM control (sine wave P WM control and over-modulation P WM control are generally described). However, in the rectangular wave control, the number of times the semiconductor switching element is turned on and off is extremely small compared to the PWM control, so the switching loss is inherently low, and the switching speed control by the drive control device of the semiconductor switching element according to the present invention is common. Is unnecessary. '

In contrast, in PWM control, the carrier frequency is generally set to several kilohertz to several tens of kilohertz, so each semiconductor switching element is controlled to be turned on and off at a relatively high frequency. . For this reason, if the switching speed is lowered to suppress the surge voltage, an increase in switching loss becomes a problem. On the other hand, if the switching speed is increased to reduce the switching loss, the surge voltage increases. For this reason, in PWM control, it is necessary to set the switching speed in terms of both switching loss reduction and surge voltage suppression.

(Switching speed control)

In the semiconductor switching element driving apparatus according to the embodiment of the present invention, switching speed control as described below is executed in order to achieve both switching loss reduction and surge voltage suppression.

FIG. 3 is a block diagram illustrating the configuration of the drive control device for the semiconductor switching element according to the embodiment of the present invention. FIG. 3 shows the configuration of one phase of inverter 20.

Referring to FIG. 3, two semiconductor switching elements Q a and Q b connected in series via an intermediate node N i connected to the load are connected between the ground wire 5 and the power supply wire 7. It is connected. The semiconductor switching element Q a is the semiconductor switching element shown in Fig. 1. The switching elements Q 2, Q 4, and Q 6 are collectively described, and the semiconductor switching element Q b is a comprehensive description of the semiconductor switching elements Q l, Q 3, and Q 5 in FIG. It is. A freewheeling diode Da as an antiparallel diode is connected in parallel to the semiconductor switching element Qa. Similarly, a freewheeling diode Db is connected in parallel to the semiconductor switching element Qb.

In the following, the configuration for voltage drive control of the gate (control electrode) of the semiconductor switching element Qa, which is typically the lower arm element, will be described, but the same applies to the semiconductor switching element Qb, which is the upper arm element. The Gout voltage drive shall be controlled by the configuration.

First, the voltage and current behavior when the semiconductor switching element Qa is turned on will be described with reference to FIG.

Referring to FIG. 4, at time t O, a switching control signal is generated so as to switch on / off of semiconductor switching elements Q a and Q b, and turn-on of semiconductor switching element Q a is started. At this time, the transistor current I c flowing through the semiconductor switching element Q a is 0. On the other hand, in the upper arm, as the semiconductor switching element Q b is turned off, the current supplied to the load (corresponding to the L component of the AC motor MG) by the free wheel diode D b (motor) Current) is recirculated (diode current I f ≠ 0). From this state, in response to the turn-on of the semiconductor switching element Q a, the transistor current I c and the diode current I ί start to change so that the motor current is caused to flow by the semiconductor switching element Q a.

At time t 0, the gate voltage Vge of the semiconductor switching element Qa is driven from a predetermined off voltage toward a predetermined on voltage. As the gate voltage V g e increases, the transistor current I c begins to flow, while the collector-emitter voltage V c e decreases accordingly.

On the other hand, the diode current If of the free wheel diode Db gradually approaches 0 as the gate voltage Vge increases. Then, when the diode current I ί = 0, the diode voltage V ak starts to rise. Here, the time delay td from the time t 0 when the turn-on command is generated to the diode current If f becomes 0 and the diode voltage V ak rises, and the subsequent peak value of the diode voltage V ak, that is, the surge voltage Depends on the motor current (corresponding phase) at the start of turn-on of the semiconductor switching element Qa.

As described above, in the PWM control, in response to the turn-on of the semiconductor switching element Q a, the motor current of the relevant phase that was passed during the turn-on period of the semiconductor switching element Q b of the opposite arm is transferred to the semiconductor switching element Q a. Will be washed away. Therefore, the motor current at the start of turn-on of the semiconductor switching element Qa is substantially equal to the energization current (transistor current Ic) after the turn-on of the semiconductor switching element Qa.

Here, as disclosed in Non-Patent Document 1, when the semiconductor switching element Q a is turned on in a state where the energization current is low (hereinafter referred to as a low current state), the free wheel diode D b carries the carrier. Since the reverse bias voltage is applied from a state with a small amount of depletion, the depletion layer develops in a very short time. As a result, in the freewheel diode Db, an excessive diode voltage Vak is likely to occur, and the surge voltage rises.

In the example of Fig. 4, the 1S semiconductor switching element, which shows an example of the operation waveform at a low surge voltage level, is in a lower current state, that is, the absolute value of the diode current If at time t O is When turned on in a smaller state, the waveform of the diode voltage V ak oscillates and a large surge voltage is generated.

FIG. 5 is a conceptual diagram showing the dependence of the surge voltage V s g and the delay time t d until the surge voltage is generated on the energizing current of the semiconductor switching element.

As described above, since the surge voltage V sg depends on the diode voltage V ak, when the conduction current (transistor current) I c of the semiconductor switching element is low, it becomes relatively large and high. It has the characteristic that it decreases according to. Such a characteristic of I c –V s g can be grasped by performing an operation experiment of the semiconductor switching element in advance.

On the other hand, as shown in Fig. 4, the turn-on Since the absolute value of the diode current I ί at the start is also large, the delay time (hereinafter referred to as surge generation time delay) td until the diode current V ak rises after the diode current I f returns to 0 also increases. For this reason, as shown in FIG. 5, the surge occurrence time delay td shows a characteristic that becomes longer as the conduction current I c becomes higher. When the allowable upper limit V smax of the surge voltage V sg determined according to the rated breakdown voltage of the semiconductor switching element is determined based on the I c – V sg characteristics as shown in Fig. 5, the surge voltage V sg The boundary current It that exceeds V sma X can be obtained. Then, from the I c-V sg characteristics and the boundary current It, the surge voltage V sg force S is the current region exceeding the allowable upper limit value V sma X, and the surge voltage V sg is below the allowable upper limit value V smax. The current region 4 1 0 that can be defined can be defined.

In the I c _ V sg characteristic as shown in Fig. 5, in the current region 4 0 0 of I c ≤ I t, surge voltage V sg exceeding the allowable upper limit V sma X is expected to occur, but I c > In the current region 4 1 0, the surge voltage V sg is expected to be within the allowable upper limit V sma X. Hereinafter, the current region 4 0 0 is also referred to as a high surge region 4 0 0, and the current region 4 1 0 is also referred to as a low surge region 4 1 0. That is, the current region 400 corresponds to the “first current region” in the present invention, and the current region 41 0 corresponds to the “second current region” in the present invention.

The I c vs V s g characteristics and I c – t d characteristics vary depending on the element temperature. Therefore, by determining the change in the boundary current I t with respect to the change in the element temperature through experiments and the like in advance, the boundary current I reflects the temperature dependency based on the detected element temperature during actual operation. t can be set.

Similarly, in a configuration in which the output voltage V dc of the DC power supply 10 is variably controlled, the applied voltage (V ce) to the semiconductor switching element changes, so I c-V sg characteristics and I c The characteristics also change depending on the voltage. Therefore, the change in the boundary current It with respect to the change in the applied voltage is obtained in advance through experiments, etc., so that in actual operation, the voltage dependence is determined based on the detection value (DC voltage V dc) of the voltage sensor 13. The boundary current It can be set reflecting the characteristics.

As will be described in detail below, in the semiconductor switching element drive control device according to the embodiment of the present invention, the surge voltage V sg and According to the characteristics of surge generation time delay td, the gate voltage drive speed of the semiconductor switching element, that is, the switching speed is variably controlled in a feed forward manner. Referring again to FIG. 3, a drive control circuit 50, drivers 120 and 125, and gate resistors 130 and 135 are provided for each semiconductor switching element.

The driver 120 drives the gate G from the off voltage VL to the on voltage V H through the gate resistor 1 30 (resistance value R1) when the control signal S D 1 of 50 drive control circuits is turned on. On the other hand, when the control signal SD2 from the drive control circuit 50 is turned on, the driver 125 drives the gate G from the off voltage VL to the on voltage VH via the gate resistor 135 (resistance R2). The driver 120 and the gate resistor 130, and the driver 125 and the gate resistor 135 are provided in parallel to the gate G. The on-voltage VH corresponds to the gate voltage for turning on the semiconductor switching element, and the off-voltage VL corresponds to the gate voltage for turning off the semiconductor switching element.

Here, it is assumed that the resistance value R 2 of the gate resistor 135 is smaller than the resistance value R 1 of the gout resistor 130. Therefore, when the gate voltage is driven by the driver 120, relatively low speed switching is performed. On the other hand, when the gate voltage is driven by the driver 125 or by both the drivers 120 and 125, relatively high-speed switching is performed.

The drive control circuit 50 generates an ON command signal S Wo indicating the ON period of the corresponding semiconductor switching element Q b and a motor current control command value I com supplied from the inverter 20 to the AC motor MG. Based on this, the control signals SD 1 and SD 2 of the drivers 120 and 125 are generated to control the switching speed when the semiconductor switching element Qb is turned on.

Hereinafter, in the present embodiment, it is assumed that the drive control circuit 50 controls the switching speed at turn-on in two stages. Specifically, by turning on only the control signal SD1, turning on both the control signals SD1 and SD2 as well as "low speed switching" where the gate voltage is driven at a relatively low speed only by the driver 120. Therefore, “high-speed switching” in which the gate voltage is driven at a relatively high speed by the drivers 120 and 125 is selectively performed. Note that the circuit configuration for realizing switching of the switching speed is not limited to the configuration illustrated in FIG. 3, but a point that any known circuit configuration can be applied will be described. For example, instead of switching the gate resistance value, the switching speed may be changed by changing the impedance value including the L and C components. Alternatively, the gate voltage may be driven by a single driver, and the switching speed may be changed by switching the gate resistance value or impedance value.

FIG. 6 is a block diagram for explaining in detail the configuration of the drive control circuit 50 shown in FIG.

Referring to FIG. 6, drive control circuit 50 includes a switching speed control unit 55 and a current level estimation unit 60. Switching speed control unit 55 includes a determination unit 70, a switching timing setting unit 80, a low speed switching instruction unit 90, and a high speed switching instruction unit 95.

Based on the current command value I com at the turn-on command of the semiconductor switching element, the current level estimation unit 60 calculates the energization current estimated value I _c # of the semiconductor switching device in the on period.

The motor current supplied from the inverter 20 to the AC motor MG as a load is an AC current corresponding to the rotation cycle of the AC motor, but each semiconductor provided corresponding to each cycle of the carrier wave in PWM control. The energization current during each ON period of the switching element corresponds to the instantaneous value at one point on the time axis of the AC current. The current command value I com is the d-axis current command value shown in Fig. 2 in PWM control.

Corresponds to 1 d c o m and q-axis current command value I q c o m. That is, the d-axis current command value I d com and the q-axis current command value I q com that are direct current values are

2 Semiconductors after the turn-on this time by using the rotation angle 0 detected by 5 and converting the commonly used 2 phase (d, q phase) → 3 phase (U, V, W phase) The estimated conduction current I c # of the switching element can be calculated.

Based on the estimated current I c # estimated by the current level estimation unit 60, the judgment unit 70 determines that the current turn-on is in the high surge region (low) according to the I c – V sg characteristics shown in Fig. 5. Current region) 4 0 0 and low surge region (high current region) 4 1 0 It is determined in which way it is performed. Specifically, the above determination is performed based on a comparison between the energized current estimated value I c # and the boundary current It (FIG. 5).

The boundary current I t is preferably set reflecting the temperature dependency and / or voltage dependency of the boundary current It as described above. For example, a setting map (not shown) of the boundary current I t with respect to the element temperature and applied voltage is created in advance based on experimental data, etc., and in actual operation, it is provided in each semiconductor switching element or in its arrangement area. The boundary current It can be appropriately set by referring to the map using the detection value of the temperature sensor (not shown) and / or the detection value of the voltage sensor 13 (DC voltage V dc).

When the estimated energization current value I c # is in the high surge region 400, the determination unit 70 turns on the control signal I s 1 for instructing low-speed switching at the start of turn-on. On the other hand, when the energized current estimated value I c # is in the low surge region 4 10, the determination unit 70 turns off the control signal Is 1 at the start of turn-on. Thus, the control signal Is 1. is turned on when low speed switching is selected, and is turned off when high speed switching is selected.

The low-speed switching instruction unit 90 operates when the control signal I s 1 is on during the on-period of the on-command signal SWon. During operation, the low-speed switching instruction unit 90 turns on the control signal S D 1 while turning off the control signal S D 2. On the other hand, the high-speed switching instruction unit 95 operates when the control signal I s 1 is off during the on period of the on command signal SWon. The high-speed switching instruction unit 95 turns on both the control signals S D 1 and S D 2 during operation. Note that both the control signals S D 1 and S D 2 are turned off during the off period of the on command signal S W on. That is, the low-speed switching instruction unit 90 and the high-speed switching instruction unit 95 correspond to the “switching speed instruction unit” according to the present invention.

In this manner, the low speed switching and the high speed switching are selectively executed based on the control signal I s 1 generated by the determination unit 70. Specifically, when the estimated current I c # is in the high surge region 4 0 0, turn-on is started by slow switching, and the gate voltage gradually decreases from the off voltage VL to the on voltage VH. Driven. For this reason, the current 'voltage change also becomes gradual, Surge voltage is suppressed.

On the other hand, when the estimated current I c # is in the low surge region 4 10, there is no need to worry about the surge voltage, so fast switching is selected from the start of turn-on. As a result, voltage loss and current can be changed quickly to reduce switching loss.

Furthermore, when the energized current estimate I c # is in the high surge region 40 °, even if the turn-on starts with low-speed switching, switching loss is applied by applying high-speed switching after the surge voltage exceeds the peak. It is desirable to reduce Such surge voltage peak timing can be predicted from the surge occurrence time delay td shown in FIG.

Therefore, the switching timing setting unit 80 sets the determination time t c for designating the switching timing based on the energized current estimated value I c #. For example, a setting map (not shown) of the determination time t c for the estimated energization current value I c # can be created in advance based on the I c -t d characteristic shown in FIG.

The switching timing setting unit 80 measures the elapsed time from the turn-on start time by the built-in timer 85, and instructs the switching from the low speed switching to the high speed switching when the elapsed time exceeds the determination time tc. To do. In response to this, the determination unit 70 turns off the control signal Is1.

As a result, even when turn-on is started by low-speed switching to suppress the surge voltage, switching from low-speed switching to high-speed switching is executed after the switching timing when the surge voltage has settled, and switching loss thereafter. Can be reduced.

On the other hand, when the energized current estimated value I c # is in the low surge region 4 10, the high-speed switching is applied from the turn-on start point as described above by setting the determination time t c = 0. Therefore, the switching speed switching as described above is not executed, and the determination unit 70 keeps the control signal Is 1 off.

As described above, since the I c -td characteristic also has temperature dependency and voltage dependency, the determination time t (for the first time can also be set to further reflect the element temperature and / or the applied voltage). For example, current flow, element temperature and By using the detection value of the temperature sensor (not shown) and / or the detection value of the voltage sensor 13 (DC voltage V dc) By referring to the map, the judgment time tc can be set more appropriately.

7A to 7C show operation examples of switching speed control by the drive control device for a semiconductor switching element according to the embodiment of the present invention.

As shown in Fig. 7A, the gate voltage Vge begins to rise in response to the turn-on command being generated at time t0. Along with this, when the current I c is low, the diode voltage V ak rises relatively quickly and the surge voltage peak increases.

Therefore, as shown in FIG. 7B, when the conduction current I c is a low current, that is, a high surge region 400, the control signal SD 1 is turned on at the start of turn-on, while the control signal SD 2 Is turned off and slow switching is applied.

Thereafter, at time t 1 when the determination time t c has elapsed from the start of turn-on, switching from the low speed switching to the high speed switching is instructed by the switching timing setting unit 80 (FIG. 6). As described above, the switching timing (time t 1) is determined based on the estimated current I c #.

By setting the constant time t c, it is set according to the timing when the surge voltage exceeds the peak.

As a result, after the surge voltage peak occurs, both the control signals S D 1 and S D 2 are turned on, and the gate voltage is driven toward the on voltage V H at a high speed, thereby reducing the switching loss.

Referring again to FIG. 7A, when the conduction current I c is high, that is, in the low surge region 4 10, the rise timing of the diode voltage V ak, that is, the surge voltage generation timing is delayed. And the peak value is also lowered. Therefore, in the low surge region 4 1 0, as shown in Figure 7C, both control signals SD 1 and SD 2 are turned on from the start of turn-on, and high-speed switching is applied continuously. The As a result, switching loss is reduced by quickly changing the voltage and current. As described above, in the drive control device for the semiconductor switching element according to the embodiment of the present invention, paying attention to the relationship between the energizing current and the surge voltage behavior, the turn-on time is determined based on the command value to the inverter 20. The switching speed can be controlled. Therefore, it is possible to achieve both reduction of surge voltage and suppression of surge voltage by providing stable switching speed control without feedback control without providing a monitoring structure for the current and / or voltage of the semiconductor switching element. It becomes possible.

(Modification 1)

FIG. 8 shows a configuration of gate capacitance value adjusting circuit 300 according to the modification of the embodiment of the present invention.

Referring to FIG. 8, the gate capacitance value adjustment circuit 300 is composed of a plurality of capacitors provided in parallel to the gate G of the semiconductor switching element Q (which comprehensively represents Q 1 to Q 6). It has a value adjustment unit 3 1 0.

The capacitance value adjustment unit 3 1 0 has an adjustment capacitance 3 2 0 and a link element 3 2 5 connected in series to the gate G. In each capacitance value adjustment unit 3 1 0, when the link element 3 2 5 is in a non-disconnected state and is electrically conductive, the capacitance value of the adjustment capacitance 3 2 0 is added to the gate capacitance. On the other hand, when the link element 3 25 is cut off by a laser input from the outside or the like and brought into a non-conductive state, the capacitance value of the adjustment capacitor 3 20 is not added to the gate capacitance value.

Therefore, by appropriately cutting the link element 3 2 5 with a laser from the outside, so-called trimming can be executed to adjust the gate capacitance value corresponding to the manufacturing variation of the semiconductor switching element.

By providing such a gate capacitance value adjusting circuit 300, the semiconductor switching device can be adjusted so that the characteristics of I c -V s g, t d shown in FIG.

In particular, the surge generation time delay td from the generation of the turn-on command to the generation of the surge voltage greatly depends directly on the gate capacitance, so by providing the gate capacitance value adjusting mechanism according to this modification, the switching speed described above Control accuracy can be improved by suppressing control variations. (Modification 2)

FIG. 9 is a block diagram showing a learning control configuration in the drive control device for the semiconductor switching element according to the embodiment of the present invention.

Referring to FIG. 9, drive control circuit 50 further includes a learning control unit 110 in addition to the configuration shown in FIG. Further, a voltage dividing circuit 27 for dividing the diode voltage V ak is provided for each freewheeling diode. The divided voltage V a k # generated by the voltage dividing circuit 2 7 is detected by the learning control unit 110.

The learning control unit 1 1 0 performs the switching operation of the semiconductor switching elements Q a and Q b during a predetermined learning period, and grasps the actual surge voltage based on the divided voltage V ak # at that time. . Then, based on the actual surge voltage during the learning period, learning control for correcting (updating) the boundary current It and the determination time tc is executed.

For example, during the learning period, when the motor drive system 100 is started, the current command command value (for example, FIG. 2) is set so that the AC motor MG mainly consumes reactive power and does not generate torque. (Idcom, Iqcom). In this way, it is possible to sample the surge voltage behavior at turn-on by turning on / off each semiconductor switching element constituting the inverter 20 without causing output to the AC motor MG as a load. As an example, when the switching frequency of each semiconductor switching element is 1 kHz, sampling at a turn-on timing of 100 0 times can be executed by securing a learning period of 1 second. In other words, it is sufficiently possible to provide a learning period when the motor drive system 100 is started.

FIG. 10 is a flowchart for explaining the learning control operation by the learning control unit 110.

Referring to FIG. 10, the learning control unit 1 1 0 is the inverter in step S 1 0 0.

Motor drive system including 2 0 Determine whether an operation start command for 1 0 0 has been generated. For example, in the motor drive system 100 installed in a hybrid vehicle, the operation start instruction corresponds to the start instruction of the hybrid system. If the operation start command is not generated (when S 1 0 0 is NO), learning is not activated. The process is terminated as it is.

The learning control unit 110 determines whether or not the learning start condition is satisfied at step s i 1 0 when the operation start command is generated (when YES is determined in S1100). The establishment of the learning start condition is determined based on, for example, whether each sensor is normal or whether there is an abnormality in the semiconductor switching element. Further, when an output instruction is issued to the AC motor MG as a load, S 1 0 0 is determined as NO, and learning is not executed. That is, learning is performed using a period during which no output instruction is issued to AC motor MG.

The learning control unit 1 1 0 starts learning in step S 1 2 0 when the learning start condition is satisfied (when Y ES is determined in S 1 1 0). Then, in step S 1 30, the learning control unit 1 1 Q determines the current command according to the current command pattern for learning so that the AC motor MG consumes reactive power without generating torque as described above. Generate qcom (I dcom, I qcom).

Then, according to the current command value generated in this way, switching control based on the currently set boundary current It and the determination time tc is applied to switch each semiconductor switching element. Then, the learning control unit 110 samples the diode voltage V ak at the time of turn-on, which is generated at the time of switching by the step S 14 40 (step S 15 50).

In step S 1 60, learning control unit 110 evaluates the surge voltage based on diode voltage V ak sampled in step S 15 50. For example, in the current region in the vicinity of the current boundary current It, if there is a margin for the maximum surge voltage allowable upper limit, the boundary current It is updated to the lower side, but the margin is insufficient. If it is, update the boundary current It to the rising side. Thereby, the boundary current It of the high surge region 400 and the low surge region 41 can be corrected according to the learning result.

Similarly, in the high surge region, the surge voltage at the switching timing from low speed switching to high speed switching according to the judgment time tc is evaluated, and if there is a margin in the allowable upper limit value, the judgment time tc is shortened. If the margin is insufficient, the judgment time tc is updated to the extension side. This The switching timing from low-speed switching to high-speed switching at turn-on in the high surge region 400 can be corrected according to the learning results.

The learning control unit 1 1 0 continues the learning operation of steps S 1 3 0 to S 1 60 until a predetermined learning end condition is satisfied in step S 1 70. For example, the learning end condition is satisfied by elapse of a predetermined time, completion of a desired learning item, generation of an output instruction to a load (AC motor MG), or the like.

By implementing the above-described learning operation and configuring the map values of the setting map of the boundary current It and the determination time tc described above, the present invention can be implemented in response to manufacturing variations and changes with time of each semiconductor element. The switching control according to the form can be executed more effectively.

Although it is difficult to perform learning including temperature dependency and voltage dependency during the learning period, the boundary current It and the judgment time tc are based on the basic value based on the conduction current and the temperature and voltage changes. If the configuration is set by the sum with the correction value, the map value can be appropriately updated by the learning operation for the map for setting the basic value.

In the semiconductor switching element drive control device according to the present embodiment, the switching control is changed by switching between two switching speeds, low speed and high speed, thereby suppressing surge voltage and switching loss with a simpler circuit configuration. Although the configuration is designed to achieve both reduction in loss, the switching speed can be subdivided into three or more stages.

In the present embodiment described above, a three-phase inverter using a motor (or a motor generator) mounted on a hybrid vehicle or an electric vehicle as a load is illustrated as an application example of a drive control device for a semiconductor switching element. The application of the present invention is not limited to this. In other words, so-called hard switching such as a single-phase inverter, D CZD C converter, step-up chopper, switching mode amplifier, etc. is performed if it constitutes a semiconductor power conversion device that controls the supply current to the load according to the command value. The present invention can be applied to switching control of a semiconductor switching element. Also, note that the load is not particularly limited. Furthermore, with regard to semiconductor switching elements, IGBTs have been exemplified in the present embodiment, but switching speed control according to the present invention is also applicable to other voltage-driven switching elements such as power MOS S (Metal Oxide Semiconductor) transistors. Can be applied. Similarly to the gate voltage drive speed in the present embodiment, by controlling the current drive speed of the control electrode (base), a semiconductor power configured by a current drive semiconductor switching element such as a power bipolar transistor is provided. The switching control according to the present invention can also be applied to the conversion device.

The embodiment disclosed this time should be considered as illustrative in all points and not restrictive. The scope of the present invention is defined by the terms of the claims, rather than the description above, and is intended to include any modifications within the scope and meaning equivalent to the terms of the claims. Industrial applicability

The present invention can be applied to switching control of a semiconductor switching element constituting a semiconductor power conversion device that controls a supply current to a load according to a command value.

Claims

The scope of the claims

1. Drive control device for the semiconductor switching elements (Q1-Q6) that constitute the semiconductor power conversion device ( ₂₀ ) for controlling the current supplied to the load (MG) according to the current command value (Icom) (50 ) And

The semiconductor switching element is turned on or off in response to the voltage or current of the control electrode (G).

The drive control device includes:

A current estimation unit (60) for estimating an energization current (I c) flowing when the semiconductor switching element is turned on based on the current command value when the turn-on command is generated, and the semiconductor switching element of the semiconductor switching element previously obtained In accordance with the first current characteristic indicating the relationship between the conduction current and the generated surge voltage (V sg), the semiconductor switching element is turned on based on the estimated conduction current (I c #) by the current estimation unit. And a switching speed control unit (55) for changing the driving speed of the voltage or current of the control electrode at the time.

2. The switching speed controller (55)

According to the first current characteristic, the estimated energization current (I c #) 1 the first current region (400) in which the surge voltage (V sg) exceeds an allowable value (V smax), and the surge voltage is A determination unit (70) for determining which of the second current region (410) is less than or equal to the allowable value;

The drive speed at the start of turn-on is made slower when the estimated energization current (I c #) is in the first current region than when the estimated energization current is in the second current region. The drive control device for a semiconductor switching element according to claim 1, further comprising a switching speed instruction section (90, 95). '

3. The switching speed control unit (55) sets the driving speed at the start of turn-on as the first speed when the estimated energization current (I c #) is in the first current region (400). And the driving speed is set to the first speed according to an elapsed time from the start of driving of the voltage or current of the control electrode. The drive control device for a semiconductor switching element according to claim 2, wherein the drive control device is further raised.

4. The switching speed controller (5 5)

The energization current (Ί c) of the semiconductor switching element (Q 1− Q 6) and the surge voltage (V sg) obtained after the start of driving the voltage or current of the control electrode are calculated in advance. In accordance with a second current characteristic indicating the relationship with the time until occurrence (td), a switching timing (in which the drive speed is increased based on the estimated energization current (I c #) by the current estimation unit (60)) ( It further includes a switching timing setting section (8 0) for setting t 1) ',

The switching speed instruction section (90, 95) increases the driving speed from the first speed to the second speed when the switching timing is reached based on the elapsed time. The drive control apparatus of the semiconductor switching element of description.

5. A learning period is set in which the current command value (I com) is generated so that the load (MG) mainly consumes reactive power when the semiconductor power converter (20) is started up, and the learning 5. The learning apparatus according to claim 4, further comprising a learning control unit (1 1 0) for correcting the switching timing (t 1) based on an actual measurement value of the surge voltage (V sg) during a period. Drive control device for semiconductor switching element.

6. The switching speed instruction unit (5 5) sets the driving speed at the start of turn-on when the estimated turtle current (I c #) is in the first current region (400). On the other hand, when the estimated energization current is within the second current region (4 10), the driving speed at the start of the turn-on is higher than the first speed. 3. The drive control device for a semiconductor switching element according to claim 2, wherein the drive speed is set to a speed of 2 and the drive speed is fixed to the second speed throughout a turn-on period.

7. The switching speed instruction section (5 5) drives the voltage or current of the control electrode when the estimated energization current (I c #) is within the first current region (40 0). 7. The drive control device for a semiconductor switching element according to claim 6, wherein the drive speed is switched from the first speed to the second speed in accordance with an elapsed time since the start.

8. A learning period is set during which the current command value (I com) is generated so that the load (MG) mainly consumes reactive power when the semiconductor power converter (20) is started up, and the learning Based on the measured value of the surge voltage (V sg) during the period, the judgment value (It) indicating the boundary between the first current region (400) and the second current region (410) is corrected. The drive control apparatus for a semiconductor switching element according to claim 2, further comprising a learning control unit (1 10) for performing the above operation. .

9. The capacitance value adjusting circuit (300) for adjusting the capacitance value of the control electrode (G) of the semiconductor switching element (Q 1 to Q6) by trimming, further comprising: The drive control device for a semiconductor switching element according to any one of the above.

10. The load is an AC motor (MG) whose supply current is controlled by pulse width modulation control.

The current estimation unit (60) calculates the estimated energization current (I c #) using the rotation angle (0) of the AC motor and a current command value (I dc om, I qc om) by vector control. 9. The drive control device for a semiconductor switching element according to any one of claims 1 to 8, which is calculated.