JP3927046B2 - Resonant type DC voltage converter - Google Patents

Description

【００１３】
請求項２に記載の発明によれば、前記所定の閾値に対応する前記スイッチング信号の周期Ｔlimは、上記計算式を演算することによって求められる。この場合、ＬＵＴ（Ｌｏｏｋ Ｕｐ Ｔａｂｌｅ）メモリに前記求められた周期Ｔlimを予め記憶しておく態様としてもよい。
【００１４】
【発明の実施の形態】
図１は本発明の一実施形態に係る共振型直流電圧変換器を示す回路図である。本変換器は、コンバータ回路１と、制御部３とを備え、コンバータ回路１は、トランジスタ（スイッチング手段）Ｑ１と、共振回路２とを備えている。
【００１５】
コンバータ回路１は、入力端子Ｐｉ１，Ｐｉ２間に印加される直流入力電圧Ｖｉより低い直流出力電圧Ｖｏを生成して出力端子Ｐｏ１，Ｐｏ２間に接続される負荷８に印加するもので、公知の全波形ゼロ電流スイッチング方式の降圧形コンバータを構成している。
【００１６】
すなわち、このコンバータ回路１は、入力電圧ＶｉをチョッピングするトランジスタＱ１と、このトランジスタＱ１と逆並列に接続され、電流を逆方向に流すためのダイオードＤ１とを備えると共に、トランジスタＱ１に直列に接続された共振用リアクトルＬ１および共振用コンデンサＣ１とからなる共振回路２と、リアクトルＬ２およびコンデンサＣ２とからなり出力電圧Ｖｏの脈動を抑制するための低域通過フィルタ（平滑回路）と、トランジスタＱ１がオフしたときにリアクトルＬ２に蓄積されたエネルギーを放出するための還流用ダイオードＤ２とから構成されている。
【００１７】
制御部３は、演算部３０とドライブ回路３１とから構成されている。ドライブ回路３１は、演算部３０からの制御信号３２に従ってトランジスタＱ１をオンオフさせるものである。演算部３０は、マイコン、メモリ、Ａ／Ｄ変換器等からなり、ドライブ回路３１に制御信号３２を出力してトランジスタＱ１のオンオフを制御するもので、以下の機能▲１▼〜▲４▼を有する。
【００１８】
▲１▼出力電圧Ｖｏと入力電圧Ｖｉを検出する機能
▲２▼トランジスタＱ１をオンにした後、ｉ＜０、すなわち共振電流ｉが反転してダイオードＤ１に流れている間に、トランジスタＱ１をオンからオフに切り替えるゼロ電流スイッチングを行う機能
▲３▼検出した出力電圧Ｖｏが予め設定された値に一致するように、トランジスタＱ１のスイッチング周波数を制御する機能
▲４▼出力電流Ｉｏが、前記所定の閾値Ｉlim以下の値を維持するように、トランジスタＱ１のスイッチング周波数を制御する機能
上記▲１▼〜▲３▼の機能は、公知の手段により実現されている。
【００１９】
次に、▲４▼の機能の実現手段について説明する。
図１のコンバータ回路１において、入力端子Ｐｉ１、Ｐｉ２に入力されるエネルギーと、出力端子Ｐｏ１、Ｐｏ２から出力されるエネルギーはロスがないとすると等しくなる。
従って、下記式の関係が成立する。
【００２０】
【数３】

【００２１】
また、スイッチング周波数ｆｓは、ｆｓ＝１／Ｔｓ の関係があるため、式（１）から、
Ｖｉ・Ｉｉ・Ｔｏ・ｆｓ＝Ｖｏ・Ｉｏ ・・・（２）
となる。
【００２２】
上記式（２）から、
ｆｓ＝Ｖｏ・Ｉｏ／（Ｖｉ・Ｉｉ・Ｔｏ）・・・（３）
上記式（３）で、トランジスタＱ１がオンしている状態の時に入力端子Ｐｉ１、Ｐｉ２から見たコンバータ回路１の入力インピーダンスをＺｎとすると、
ｆｓ＝Ｖｏ・Ｉｏ・Ｚｎ／（Ｖｉ・Ｖｉ・Ｔｏ）
＝ｋ・Ｖｏ・Ｉｏ／（Ｖｉ・Ｖｉ） ・・・（４）
ただし、 ｋ＝Ｚｎ／Ｔｏ ・・・・（５）
が得られる。
【００２３】
また、共振用リアクトルＬ１に流れる共振電流ｉは、下記式（６）で示される。
【００２４】
【数４】

【００２５】
ここで、ｆｎは共振用リアクトルＬ１および共振用コンデンサＣ１からなる共振回路の共振周波数、Ｉｐは共振電流ｉの交流成分の振幅である。この共振周波数ｆｎ、および共振周期Ｔｎは、下記式（７）で示される。
１／ｆｎ＝Ｔｎ＝２π√（Ｌｒ・Ｃｒ） ・・・（７）
但し、共振用リアクトルＬ１のインダクタンスをＬｒ、共振用コンデンサＣ１のキャパシタンスをＣｒとする。
【００２６】
図２は、式（６）で示される共振電流ｉの電流波形を示したものである。ここで、出力電流Ｉｏ＝０のときは、Ｔｏ＝Ｔｎ／２となり（図２(ｂ)参照）、Ｉｏ＝Ｉｐのときは、Ｔｏ＝Ｔｎになる（図２(ｃ)参照）。また、出力電流Ｉｏ＞Ｉｐになるとゼロ電流スイッチングができなくなってしまうため、ＩｏはＩｐ以下の電流値である必要がある。そこで、出力電流値が越えないように抑制する所定の閾値を、例えばＩlim＝Ｉｐと定める。そうすると、出力電流Ｉｏ＝Ｉlimになるとき、トランジスタＱ１の導通時間Ｔｏ＝Ｔｎになるので、Ｉｏが零からＩlimまで変化すると共に、ＴｏはＴｎ／２からＴｎまで変化することになる。
【００２７】
出力電流Ｉｏが零からＩlimまで変化した場合、前記式（４）により、スイッチング周波数ｆｓは、零からｆｓ＝ｋ・Ｖｏ・Ｉlim／（Vｉ・Vｉ）まで変化する。このとき前記式（５）により、ｋはＺｎ・２／ＴｎからＺｎ／Ｔｎまで変化し、ｆｓを小さくする方向に働くが、ｋの変化率はＩｏの変化率より小さいため、この区間の前記式（４）の傾きは正となり、ｆｓは常に増加方向となる。
【００２８】
したがって、出力電流Ｉｏ＝Ｉlimのときのｆｓの周波数をｆlimとすると、スイッチング周波数ｆｓが出力電流Ｉｏの増加に伴って増加し、ｆｓ＝ｆlim、すなわちＴｓ＝Ｔlimとなったとき、同時にＩｏ＝Ｉlimおよび、ｋ＝Ｚｎ／Ｔｎが成立する。
【００２９】
ここで、ＺｎはトランジスタＱ１がオンしている状態の時に入力端子Ｐｉ１、Ｐｉ２から見た共振回路側の入力インピーダンスであり、コンバータ回路１に固有の値となる。また、Ｔｎは、前記式（７）で示されるとおり、共振用リアクトルＬ１のインダクタンスＬｒと共振用コンデンサＣ１のキャパシタンスＣｒで決定される値である。したがって、ｋはコンバータ回路１に固有の固定値となる。以上の関係により制限周波数ｆlimおよび制限周期Ｔlimは、
【００３０】
【数５】

【００３１】
となる。
【００３２】
また、演算部３０のメモリにｋの値とＩlimの値、および前記式（８）を予め格納しておく。演算部３０は、出力電圧Ｖｏを検出し、Ｖｏが出力電圧の目標値よりも低い場合には現在のスイッチング周期Ｔｓを短くして新たなスイッチング周期Ｔｓ２を算出し、Ｖｏが出力電圧の目標値よりも高い場合には現在のスイッチング周期Ｔｓを長くして新たなスイッチング周期Ｔｓ２を算出する。そして、演算部３０はスイッチング周期Ｔｓを前記算出されたＴｓ２の時間幅に変更し、前記変更されたスイッチング周期Ｔｓで制御信号３２を出力する。ドライブ回路３１は、前記出力された制御信号３２に従ってスイッチング信号３３を出力する。トランジスタＱ１は、前記出力されたスイッチング信号３３によってオンオフされる。したがって、負荷状態が重くなって出力電圧が目標値よりも低くなると周期Ｔｓがより短い時間に設定され、出力電流Ｉｏが増加させられる。また、負荷状態が軽くなって出力電圧が目標値よりも高くなると周期Ｔｓがより長い時間に設定され、出力電流Ｉｏが減少させられる。その結果、出力電圧Ｖｏは、ほぼ前記出力電圧の目標値になるように制御される。
【００３３】
しかしながら、負荷８が例えば異常等を含めて必要以上に重くなった場合、出力電流が増加し過ぎてＩlimを越える虞れがある。そこで、演算部３０は、Ｖｏが出力電圧の目標値よりも低い場合にはスイッチング周期Ｔｓをより短い時間に設定して新たなスイッチング周期Ｔｓ２を算出するとともに、入力電圧Ｖｉを検出し、前記検出した出力電圧Ｖｏ、入力電圧Ｖｉ、前記メモリに格納したｋの値とＩlimの値、および前記式（８）とからＩlimに対応するスイッチング信号の周期Ｔlimを算出する。そして、前記算出したＴｓ２とＴlimとを比較演算し、その結果がＴｓ２＜Ｔlimになった場合には、スイッチング周期ＴｓをＴlimの時間幅に変更し、スイッチング周期ＴｓをＴlim以上の時間幅に維持する。
【００３４】
スイッチング周期ＴｓがＴlim以上の時間幅に維持されると、前記式（８）の関係によって、出力電流Ｉｏは、ほぼＩlim以下の電流値に維持される。
【００３５】
なお、本実施形態では、前記式（８）の関係から、Ｔlimを演算算出する例を説明したが、あらかじめ入力電圧値と出力電圧値との関係とからＩlimに対応するスイッチング信号の周期Ｔlimを演算算出、あるいは実験により測定する等の手段により求め、演算部３０のＬＵＴメモリに予め記憶しておき、このＬＵＴメモリを参照してＴlimを算出する構成としてもよい。
【００３６】
また、本実施形態では、平滑回路がリアクトルおよびコンデンサからなる構成例を示したが、出力電圧Ｖｏの脈動を抑制するものであれば、コンデンサ等による他の構成としてもよい。
【００３７】
【発明の効果】
以上のように、請求項１に記載の発明によれば、出力電流検出回路を設けることなく出力電流を定格値等、所定の閾値以下に維持制御し得る共振型直流電圧変換器が提供できる。
【００３８】
請求項２に記載の発明によれば、前記所定の閾値に対応するスイッチング信号の周期Ｔlimは、下記式により計算することができる。
【００３９】
【数６】

【図面の簡単な説明】
【図１】本発明の一実施形態を説明するための回路図である。
【図２】本発明の一実施形態を説明するための電流波形図である。
【図３】従来例を説明するための回路図である。
【図４】従来例を説明するための信号波形図である。
【符号の説明】
１ コンバータ回路
２ 共振回路
３ 制御部（制御手段）
３０ 演算部（入力電圧検知手段、出力電圧検知手段、出力電流制限手段）
３１ ドライブ回路
３２ 制御信号
３３ スイッチング信号
３４ 出力電流検出回路
８ 負荷
Ｃ１ 共振用コンデンサ
Ｃ２ コンデンサ
Ｄ１ ダイオード
Ｄ２ 還流用ダイオード
Ｌ１ 共振用リアクトル
Ｌ２ リアクトル
Ｐｉ１，Ｐｉ２ 入力端子
Ｐｏ１，Ｐｏ２ 出力端子
Ｑ１ トランジスタ（スイッチング手段）[0001]
BACKGROUND OF THE INVENTION
The present invention relates to a resonance type DC voltage converter, and more particularly to means for limiting the output current thereof.
[0002]
[Prior art]
Conventionally, as a DC voltage converter, a switch mode converter in which switching elements are periodically turned on and off is known. In this switch mode converter, the switching loss increases as the frequency of the switching period signal, that is, the switching frequency fs increases. For this reason, conventional switch mode converters use a resonant circuit consisting of a resonant reactor and a resonant capacitor to switch at zero voltage using voltage resonance, or switch at zero current using current resonance. The switching loss is reduced by adopting a method to perform.
[0003]
FIG. 3 is a circuit diagram of a conventional resonant DC voltage converter. In this converter, the switching element Q1, the resonance circuit 2 and the smoothing circuit connected to the input terminals Pi1 and Pi2 are connected in series, and the input voltage is converted and supplied to the load 8 connected to the output terminals Po1 and Po2. On the other hand, it is provided with a calculation unit 30 that takes in the input voltage Vi and the output voltage Vo and sets the feedback amount. The control unit 3 sets a switching cycle Ts for turning on / off the switching element from the input voltage Vi and the output voltage Vo, and controls so as to obtain a predetermined output voltage Vo.
[0004]
FIG. 4 is a signal waveform diagram for explaining the operation of the zero current switching method in the circuit of FIG. In general, the DC voltage converter of the zero current switching type is configured so that no current flows through the switching element when the resonance current i <0 by a diode. It is controlled to be turned off after a lapse of time. When the time of resonance current i ≧ 0 is To, the ON time T100 of the switching element is set to a constant value of T100> To.
[0005]
By the way, the control unit 3 sets the output voltage so as to change the switching cycle Ts according to the fluctuation state of the load 8 (for example, when the load state becomes heavy, the cycle Ts is shortened, that is, the switching frequency fs is increased). In order to stabilize, the output current is increased.
[0006]
However, if the load 8 becomes heavier than necessary including, for example, an abnormality, the output current may increase excessively and exceed the rated current value. Therefore, conventionally, an output current detection circuit 34 is provided, and the control unit 3 should maintain the output current at the rated current value when the output current detected by the output current detection circuit 34 has reached the rated current value. It is configured to perform current control.
[0007]
[Problems to be solved by the invention]
However, in the conventional configuration as described above, in order to maintain and control the output current below the rated current value, it is necessary to separately provide an output current detection circuit composed of, for example, a Hall element or a low resistance and wiring. The circuit size is increased and the cost is increased.
[0008]
The present invention has been made in view of the above circumstances, and an object thereof is to provide a resonance type DC voltage converter capable of maintaining and controlling an output current below a predetermined threshold value such as a rated value without providing an output current detection circuit. And
[0009]
[Means for Solving the Problems]
The invention according to claim 1 is a switching means for turning on / off an input voltage, a resonance circuit connected to the switching means and generating a resonance current at a predetermined cycle, and smoothing the generated resonance current to a load. A resonance type DC voltage converter comprising: a smoothing circuit that outputs current and voltage; and a control unit that outputs a switching signal at a switching period set to turn on and off the switching unit. Input voltage detection means for detecting voltage, output voltage detection means for detecting the output voltage of the smoothing circuit, and output current limiting means for suppressing the output current value of the smoothing circuit to a predetermined threshold value or less, and The current limiting means includes an input voltage value detected by the input voltage detection means and an output voltage value detected by the output voltage detection means. Serial calculates the cycle of the switching signal corresponding to a predetermined threshold value, is characterized in that so as to output the switching signal at the calculated cycle or more.
[0010]
According to the first aspect of the present invention, the switching signal is output by the control unit at a set switching period, the switching unit is turned on / off by the output switching signal, and the input voltage is It is turned on and off by the switching means. The resonance current is generated by a resonance circuit that resonates at a predetermined period, and is smoothed by a smoothing circuit. Furthermore, the switching signal is output by the control means at a period equal to or greater than a period corresponding to the predetermined threshold value calculated from the detected input voltage value and output voltage value. In this case, the output current value is suppressed substantially below the predetermined threshold value.
[0011]
According to a second aspect of the present invention, in the resonant DC voltage converter according to the first aspect, the cycle Tlim of the switching signal corresponding to the predetermined threshold is given by the following equation.
[0012]
[Expression 2]

[0013]
According to the second aspect of the present invention, the period Tlim of the switching signal corresponding to the predetermined threshold value is obtained by calculating the calculation formula. In this case, the obtained cycle Tlim may be stored in advance in a LUT (Look Up Table) memory.
[0014]
DETAILED DESCRIPTION OF THE INVENTION
FIG. 1 is a circuit diagram showing a resonant DC voltage converter according to an embodiment of the present invention. The converter includes a converter circuit 1 and a control unit 3, and the converter circuit 1 includes a transistor (switching means) Q1 and a resonance circuit 2.
[0015]
The converter circuit 1 generates a DC output voltage Vo lower than the DC input voltage Vi applied between the input terminals Pi1 and Pi2 and applies it to the load 8 connected between the output terminals Po1 and Po2. It constitutes a step-down converter with waveform zero current switching.
[0016]
That is, the converter circuit 1 includes a transistor Q1 that chops the input voltage Vi, a diode D1 that is connected in antiparallel with the transistor Q1 and allows current to flow in the reverse direction, and is connected in series to the transistor Q1. The resonance circuit 2 including the resonance reactor L1 and the resonance capacitor C1, the low-pass filter (smoothing circuit) including the reactor L2 and the capacitor C2 for suppressing the pulsation of the output voltage Vo, and the transistor Q1 are turned off. And a recirculation diode D2 for releasing the energy accumulated in the reactor L2.
[0017]
The control unit 3 includes a calculation unit 30 and a drive circuit 31. The drive circuit 31 turns on and off the transistor Q1 in accordance with a control signal 32 from the arithmetic unit 30. The arithmetic unit 30 is composed of a microcomputer, a memory, an A / D converter, etc., and outputs a control signal 32 to the drive circuit 31 to control on / off of the transistor Q1, and has the following functions (1) to (4). Have.
[0018]
(1) Function for detecting output voltage Vo and input voltage Vi (2) After turning on transistor Q1, i <0, that is, turning on transistor Q1 while resonance current i is inverted and flowing to diode D1 Function for performing zero current switching for switching from OFF to OFF (3) Function for controlling the switching frequency of the transistor Q1 so that the detected output voltage Vo matches a preset value (4) The output current Io is the predetermined current Functions for controlling the switching frequency of the transistor Q1 so as to maintain a value equal to or less than the threshold value Ilim The above functions (1) to (3) are realized by known means.
[0019]
Next, means for realizing the function (4) will be described.
In the converter circuit 1 of FIG. 1, the energy input to the input terminals Pi1 and Pi2 and the energy output from the output terminals Po1 and Po2 are equal if there is no loss.
Accordingly, the following relationship is established.
[0020]
[Equation 3]

[0021]
Further, since the switching frequency fs has a relationship of fs = 1 / Ts, from the equation (1),
Vi · Ii · To · fs = Vo · Io (2)
It becomes.
[0022]
From the above equation (2),
fs = Vo · Io / (Vi · Ii · To) (3)
In the above equation (3), when the input impedance of the converter circuit 1 viewed from the input terminals Pi1 and Pi2 when the transistor Q1 is on is Zn,
fs = Vo · Io · Zn / (Vi · Vi · To)
= K · Vo · Io / (Vi · Vi) (4)
However, k = Zn / To (5)
Is obtained.
[0023]
The resonance current i flowing through the resonance reactor L1 is expressed by the following formula (6).
[0024]
[Expression 4]

[0025]
Here, fn is the resonance frequency of the resonance circuit composed of the resonance reactor L1 and the resonance capacitor C1, and Ip is the amplitude of the AC component of the resonance current i. The resonance frequency fn and the resonance period Tn are expressed by the following formula (7).
1 / fn = Tn = 2π√ (Lr · Cr) (7)
However, the inductance of the resonance reactor L1 is Lr, and the capacitance of the resonance capacitor C1 is Cr.
[0026]
FIG. 2 shows a current waveform of the resonance current i represented by the equation (6). Here, when the output current Io = 0, To = Tn / 2 (see FIG. 2B), and when Io = Ip, To = Tn (see FIG. 2C). Further, since zero current switching cannot be performed when the output current Io> Ip, Io needs to have a current value equal to or less than Ip. Therefore, a predetermined threshold value for suppressing the output current value from exceeding is determined, for example, as Ilim = Ip. Then, when the output current Io = Ilim, the conduction time To of the transistor Q1 becomes To = Tn, so that Io changes from zero to Ilim and To changes from Tn / 2 to Tn.
[0027]
When the output current Io changes from zero to Ilim, the switching frequency fs changes from zero to fs = k · Vo · Ilim / (Vi · Vi) according to the equation (4). At this time, according to the equation (5), k changes from Zn · 2 / Tn to Zn / Tn and works in the direction of decreasing fs. However, since the rate of change of k is smaller than the rate of change of Io, The slope of equation (4) is positive and fs is always increasing.
[0028]
Accordingly, if the frequency of fs when the output current Io = Ilim is assumed to be flim, the switching frequency fs increases as the output current Io increases. When fs = flim, that is, when Ts = Tlim, Io = Ilim. And k = Zn / Tn is established.
[0029]
Here, Zn is an input impedance on the resonance circuit side as viewed from the input terminals Pi1 and Pi2 when the transistor Q1 is on, and is a value inherent to the converter circuit 1. Further, Tn is a value determined by the inductance Lr of the resonance reactor L1 and the capacitance Cr of the resonance capacitor C1 as shown in the equation (7). Therefore, k is a fixed value unique to the converter circuit 1. Due to the above relationship, the limiting frequency flim and the limiting period Tlim are
[0030]
[Equation 5]

[0031]
It becomes.
[0032]
Further, the value of k, the value of Ilim, and the formula (8) are stored in advance in the memory of the arithmetic unit 30. The arithmetic unit 30 detects the output voltage Vo, and when Vo is lower than the target value of the output voltage, calculates the new switching period Ts2 by shortening the current switching period Ts, and Vo is the target value of the output voltage. If higher, the current switching period Ts is lengthened and a new switching period Ts2 is calculated. Then, the arithmetic unit 30 changes the switching cycle Ts to the calculated time width of Ts2, and outputs the control signal 32 at the changed switching cycle Ts. The drive circuit 31 outputs a switching signal 33 in accordance with the output control signal 32. The transistor Q1 is turned on / off by the output switching signal 33. Therefore, when the load state becomes heavy and the output voltage becomes lower than the target value, the cycle Ts is set to a shorter time, and the output current Io is increased. Further, when the load state becomes light and the output voltage becomes higher than the target value, the period Ts is set to a longer time, and the output current Io is reduced. As a result, the output voltage Vo is controlled to be approximately the target value of the output voltage.
[0033]
However, if the load 8 becomes heavier than necessary including, for example, an abnormality, the output current may increase excessively and exceed Ilim. Therefore, when Vo is lower than the target value of the output voltage, the calculation unit 30 calculates the new switching cycle Ts2 by setting the switching cycle Ts to a shorter time, detects the input voltage Vi, and detects the detection. The period Tlim of the switching signal corresponding to Ilim is calculated from the output voltage Vo, the input voltage Vi, the k value and the Ilim value stored in the memory, and the equation (8). Then, the calculated Ts2 and Tlim are compared, and if the result is Ts2 <Tlim, the switching period Ts is changed to the time width of Tlim, and the switching period Ts is maintained at the time width equal to or larger than Tlim. To do.
[0034]
When the switching period Ts is maintained at a time width equal to or greater than Tlim, the output current Io is maintained at a current value approximately equal to or less than Ilim due to the relationship of the above equation (8).
[0035]
In the present embodiment, the example in which Tlim is calculated and calculated from the relationship of the above equation (8) has been described. However, the cycle Tlim of the switching signal corresponding to Ilim is previously determined from the relationship between the input voltage value and the output voltage value. A configuration may be used in which calculation is performed or calculation is performed by means such as experimentation, which is stored in advance in the LUT memory of the calculation unit 30 and Tlim is calculated with reference to the LUT memory.
[0036]
Further, in the present embodiment, the configuration example in which the smoothing circuit includes the reactor and the capacitor has been described. However, as long as the pulsation of the output voltage Vo is suppressed, another configuration using a capacitor or the like may be used.
[0037]
【The invention's effect】
As described above, according to the first aspect of the present invention, it is possible to provide a resonant DC voltage converter that can maintain and control the output current below a predetermined threshold such as a rated value without providing an output current detection circuit.
[0038]
According to the second aspect of the present invention, the period Tlim of the switching signal corresponding to the predetermined threshold can be calculated by the following equation.
[0039]
[Formula 6]

[Brief description of the drawings]
FIG. 1 is a circuit diagram for explaining an embodiment of the present invention.
FIG. 2 is a current waveform diagram for explaining an embodiment of the present invention.
FIG. 3 is a circuit diagram for explaining a conventional example.
FIG. 4 is a signal waveform diagram for explaining a conventional example.
[Explanation of symbols]
DESCRIPTION OF SYMBOLS 1 Converter circuit 2 Resonant circuit 3 Control part (control means)
30 arithmetic unit (input voltage detecting means, output voltage detecting means, output current limiting means)
31 drive circuit 32 control signal 33 switching signal 34 output current detection circuit 8 load C1 resonance capacitor C2 capacitor D1 diode D2 recirculation diode L1 resonance reactor L2 reactor Pi1, Pi2 input terminal Po1, Po2 output terminal Q1 transistor (switching means)

Claims (2)

Switching means for turning on / off an input voltage; a resonance circuit connected to the switching means for generating a resonance current at a predetermined period; and a smoothing circuit for smoothing the generated resonance current and outputting a current and a voltage to a load; And a control means for outputting a switching signal at a switching cycle set to turn on and off the switching means, wherein the control means comprises an input voltage detection means for detecting the input voltage. Output voltage detecting means for detecting the output voltage of the smoothing circuit; and output current limiting means for suppressing the output current value of the smoothing circuit to a predetermined threshold value or less, wherein the output current limiting means is configured to detect the input voltage. Corresponding to the predetermined threshold value from the input voltage value detected by the means and the output voltage value detected by the output voltage detection means Serial to calculate the period of the switching signal, the resonant DC voltage converter, characterized in that to output said switching signal at the calculated cycle or more. 2. The resonant DC voltage converter according to claim 1, wherein the period Tlim of the switching signal corresponding to the predetermined threshold is given by the following equation.

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