JP2008535456A - Control of resonant converter - Google Patents

Control of resonant converter Download PDF

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Publication number
JP2008535456A
JP2008535456A JP2008503652A JP2008503652A JP2008535456A JP 2008535456 A JP2008535456 A JP 2008535456A JP 2008503652 A JP2008503652 A JP 2008503652A JP 2008503652 A JP2008503652 A JP 2008503652A JP 2008535456 A JP2008535456 A JP 2008535456A
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Prior art keywords
resonant
control
interval
switch
voltage
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JP2008503652A
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ハルバースタッツ ハンス
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エヌエックスピー ビー ヴィ
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Priority to EP05104271 priority
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Priority to PCT/IB2006/050910 priority patent/WO2006103609A2/en
Publication of JP2008535456A publication Critical patent/JP2008535456A/en
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    • HELECTRICITY
    • H02GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02MAPPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00Conversion of dc power input into dc power output
    • H02M3/22Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/337Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only in push-pull configuration
    • H02M3/3376Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only in push-pull configuration with automatic control of output voltage or current
    • HELECTRICITY
    • H02GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02MAPPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00Details of apparatus for conversion
    • H02M2001/0048Circuits or arrangements for reducing losses
    • H02M2001/0054Transistor switching losses
    • H02M2001/0058Transistor switching losses by employing soft switching techniques, i.e. commutation of transistor when voltage applied to it is zero and/or when current flowing through it is zero
    • YGENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02BCLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion
    • Y02B70/14Reduction of losses in power supplies
    • Y02B70/1416Converters benefiting from a resonance, e.g. resonant or quasi-resonant converters
    • Y02B70/1433Converters benefiting from a resonance, e.g. resonant or quasi-resonant converters in galvanically isolated DC/DC converters
    • YGENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02BCLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion
    • Y02B70/14Reduction of losses in power supplies
    • Y02B70/1491Other technologies for reduction of losses, e.g. non-dissipative snubbers, diode reverse recovery losses minimisation, zero voltage switching [ZVS], zero current switching [ZCS] or soft switching converters

Abstract

The present invention deals with the control of a resonant LLC converter through the use of control parameters. The primary current flowing through the resonant tank and the voltage at a predetermined point of the resonant tank are monitored, the control parameter is set for the high side conduction interval, the control parameter is set for the low side conduction interval, The control parameters for these two conduction intervals are the peak current of the interval and the predetermined voltage of the interval. The resonant converter comprises a series-controllable switch connected to a power source. The resonant converter is operated by setting a criterion to switch off according to a criterion that includes four control parameters.

Description

  The present invention relates to control of resonant converters, and more particularly to control of resonant converters through the use of control parameters.

  In a conventional resonant LLC converter, a resonant capacitor and two inductors are connected to additional elements when possible to form a resonant circuit. The converter further includes a transformer and a rectifier circuit that are used to generate a DC output voltage. An output current can be continuously generated by adding a series inductance to the rectifier circuit. Many variations exist in this type of converter, including rectification by bridge rectifiers or no rectification as used in lighting devices. Transformers can be excluded in some cases.

  In order for the resonator to perform satisfactorily, the switch that generates the alternating current is switched on and off in an exact instant. The frequency at which the switch is operated defines the mode of operation of the converter.

  Conventional converters are typically controlled by a control logic circuit at a frequency of 50% duty cycle, where the output power of the converter is controlled by changing the operating frequency. Duty cycles other than 50% can also be used. In this case, the duty cycle determines the output power.

  This control principle by frequency gives serious disadvantages.

  When controlled by frequency, the voltage and current of the resonator at the start of each time interval (high-side or low-side conduction interval) depends not only on the frequency but also on the history of previous conduction strokes. This history results in a larger transition response, especially when the converter operates near resonance, and further complicates the problem of control loop stabilization.

  For recent designs, low load efficiency and input power during standby (with no output power extracted) are major issues. The conventional 50% duty cycle control circulates a large amount of energy at a low load, so it is difficult to solve this problem. The solution to this is burst mode, but burst mode is not always allowed due to large ripples in the output. A sudden takeover from normal mode to burst mode or vice versa can give an unacceptable transition at the output.

  Low duty cycle operation is a solution that maintains high efficiency at low power. However, changing the duty cycle at a fixed frequency can cause unacceptable changes in the sign of the loop gain.

  US Pat. No. 6,711,034 describes a resonant converter that controls the frequency of switch conduction time by a timer while compensating for the asymmetry of the secondary diode current by measuring the electrical magnitude, eg, primary current. Is disclosed.

  The present invention seeks to provide an improved resonant converter. The invention is defined by the independent claims. The dependent claims define preferred embodiments.

  Embodiments of the present invention provide an improved means of controlling a converter so that a simpler, more robust and less expensive resonant converter can be provided. Preferably, the present invention reduces or eliminates one or more of the above or other disadvantages, alone or in any combination.

  In an embodiment of the present invention, a resonant converter is provided that monitors a change in state and implements a control algorithm to operate the resonant converter according to a criterion set from control parameters. The current flowing through the resonant tank is also referred to as the primary current.

  Operating the resonant converter in this manner results in a number of advantages. By operating the switch directly from monitoring the primary current and the voltage at a predetermined point, faster control and higher stability of the system can be provided. Further, it is possible to directly have a risk prevention form such as short circuit output protection from the setting of the reference and / or the setting of the control parameter. In addition, a smooth combination of 50% duty cycle high output power can be combined into a high efficiency low output power / standby mode by setting criteria where different criteria consider different aspects, if desired Allows operation with a duty cycle different from 50%. A further advantage is that a symmetrically controlled converter can be obtained if necessary. The scope of the present invention covers embodiments in which the voltage is replaced by indirect measurements as will be described later in connection with FIG.

  The optional features of claim 2 are advantageous. The reason is that in addition to setting a specific reference for the primary current and the voltage at a predetermined point in relation to the control parameters, the reference can be ignored even if the reference is not satisfied at the beginning of the conduction period. Because. Satisfying the criterion that the minimum time has passed is not necessarily obtained by the timer means. This can be obtained, for example, from the known progression of the primary current and / or the voltage at a predetermined point associated with a particular criterion or other equivalent criterion.

  The optional features of claim 3 are advantageous. The reason is that a dedicated converter circuit is not required to operate the resonant converter in a desired operation mode, so that a versatile resonant converter can be provided.

  The optional features of claim 4 are advantageous. The reason is that stabilization is not performed directly against state fluctuations such as transition response, ripple, burst, etc., so that a more stable resonant converter can be provided.

  The optional features of claim 5 are advantageous. The reason is that since the resonant converter is operated in modes having various output powers, a dedicated converter circuit is not required, so that a versatile resonant converter can be provided.

  The optional features of claims 6 and 7 are advantageous. This is because the relationship between the control parameters can be considered by a single operating parameter, thereby avoiding specifying a plurality of control parameters to operate the resonant converter in a predetermined operating mode.

  The optional features of claim 8 are advantageous. The reason is that a direct relationship between the control parameter and the output power can be obtained by taking the power supply voltage and operating frequency of the converter into the flourishing of the resonant converter.

  According to another aspect of the invention, there is provided control logic for controlling a resonant converter, a method for controlling a resonant converter, and computer readable code for performing the steps of the method. In general, the various aspects of the invention can be combined in possible ways within the scope of the invention. These and other aspects, features and / or advantages of the present invention will become apparent with respect to the embodiments described later.

  Embodiments of the present invention will be described with reference to the accompanying drawings by way of example.

  An embodiment of a resonant converter is shown in FIG. This circuit is a resonant LLC converter, and includes a resonant capacitor Cr, an inductor L1, a magnetic inductor L2, and an element that forms part of a resonant circuit or a resonant tank. The transformer and the rectifier circuit are used here to generate the DC output voltage Vo. The output current can be generated continuously by adding a series inductance L3. The circuit has three parts. The first part is a control unit comprising a control logic CL for generating a control signal for opening and closing the switches 6 and 7 by the drivers HSD and LSD. The second part is a primary circuit and the third part is a secondary circuit. The resonant converter is connected, for example, to a power supply Vs that supplies electrical energy to a load that can be connected to the output terminal 4 on the secondary side. The resonant converter comprises a controllable first switch 6 and a second switch 7 arranged in series connected to a power source, the first switch being a high side switch (HSS), And the second switch is a low side switch (LSS), and the low side switch is grounded at one end. Embodiments having a full bridge configuration can also be envisaged. An embodiment of the full bridge configuration is shown in FIG.

  The converter is usually controlled by a frequency of 50% duty cycle by the block CL. The output power of the converter can be controlled by changing the operating frequency. Duty cycles other than 50% can also be used. Again, the duty cycle determines the output power. The present invention relates to a new and inventive method of operating a resonant converter.

  In the converter according to the invention, the duty cycle and the frequency variation can be combined so that a smooth adjustment of the output power can be realized. However, the converter is not directly controlled by frequency and duty cycle, but by current and voltage on the primary side of the transformer. Therefore, variables Iprim and Vcap1 as shown in FIG. 1 are used. These two variables are compared with two control values in each conduction period, and thus the converter is cyclically controlled. The current Iprim is defined as a current that flows in the resonant tank according to the opening / closing of the switch. The current is measured by other equivalent methods based on, for example, the voltage across Rs, the switch current, and the like. The current Iprim is also referred to as a primary current. Let Vcap1 be the voltage at a predetermined point. In the present embodiment, a predetermined point is a point given a reference number 9.

  The first conduction period occurs while the first switch is turned on, and the second conduction period occurs while the second switch is turned on. The two control parameters of the first conduction period and the second conduction period are the peak current of the conduction interval and a predetermined voltage existing at the Vcap1 point.

  The first and second switches are turned off according to criteria including four control parameters: IpeakH, VcapH, which are control parameters during the conduction interval of the high side switch, and IpeakL, VcapL, which are control parameters during the conduction interval of the low side switch. Can be switched.

The high side switch and low side switch are switched off according to the following criteria or control algorithm. The control algorithm is calculated by control logic (CL).
-HSS turn-off 1) [Primary current> IpeakH] OR
2) [Reaching the peak of Vcap1] AND [Vcap1 <VcapH]
-LSS turn-off 3) [Primary current <IpeakL] OR
4) [Reach Vcap1 valley] AND [Vcap1> VcapL]

The detection of primary current peaks or valleys creates a practical problem for identification with noise and disturbances. The peaks or valleys of Vcap correspond to primary current> 0 or primary current <0, respectively. Therefore, the equivalent control algorithm is
HSS turn-off 1A) [Primary current> IpeakH] OR
2A) [Primary current> 0] AND [Vcap1 <VcapH]
-LSS turn-off 3A) [Primary current <IpeakL] OR
4A) [Primary current <0] AND [Vcap1> VcapL]
It becomes. In the present embodiment, the four control parameters result in IpeakH, IpeakL, VcapH, and VcapL.

  Depending on the control algorithm, the initial state at the beginning of each conduction interval is more closely related to the control variable. Therefore, the influence of the history of the previous cycle is reduced.

  Hereinafter, various embodiments in which the control parameters are specified in more detail and complicated will be described.

  The high side switch (HSS) and the low side switch (LSS) can be switched on using various modes of operation, for example, HSS / LSS turn-on can be adapted non-overlap, fixed non-overlap, Or it can be determined by other criteria according to the prior art. This means that the reverse switch is turned on after a fixed time after the conduction switch is turned off. The reverse switch can also be turned on after detecting that the half bridge has rectified. This is referred to as adapted non-overlap, which can be achieved, for example, by detecting dV / dt at the half-bridge point. The scope of the present invention deals with the criteria for turning off a particular switch, but as already explained, the switch is turned on again to drive the converter.

The resonant converter can be operated in the desired operating mode by setting the control parameter to a specific value, for example the standby mode can be realized by giving the following values to the control variables.
IpeakH = fixed value. The fixed value is determined according to the desired output power and rectified energy.
IpeakL = maximum negative value. This ensures that the control algorithm is not determined by this parameter.
VcapH = a value lower than Vcap1 at the end of the high-side switch conduction stroke. This ensures that the control algorithm is not determined by this parameter.
A value close to VcapL = 0. This ensures that the low-side switch is switched off at the moment when Iprim is the negative maximum value.

  As IpeakH increases, the output power increases. At a given output power, VcapH takes control, and IpeakH no longer determines the high-side switch conduction interval. In this way, smooth control of output power is possible without sudden changes in output power or changes in the sign of loop gain.

  The fact that the output power can be controlled by setting appropriate boundary conditions for the four control parameters will be described in more detail.

  It can be calculated that the converted energy is approximately equal to the voltage difference between Vcap1 at the beginning of the conduction interval and Vcap1 at the end of the conduction interval. This is because the operating frequency of the converter is almost constant. Thus, Vcap1 is selected as a state parameter starting at the peak or valley of Vcap1. The reason is that peak / valley energy is supplied to Cr and after the peak this energy is supplied from Cr to the rest of the resonant tank and load. Thus, the conversion is given a nearly linear conversion from Vcap1 to output power. Advantageously, the transformation is a linear function, which also represents that Vcap1 is a good parameter for use in the control algorithm.

The following equation describing the output power of the converter can be obtained:
Pout = [Vsupply- (VcapH-VcapL)] x (CrxFswitchxVsupply) xeff
In this case, eff is the efficiency of the converter.
From this equation:
・ Vsupply- (VcapH-VcapL) represents output power.
A 50% duty cycle occurs when VcapL = VcapH.
<> 50% duty cycle occurs when VcapL <>-VcapH.
-While [Vsupply- (VcapH-VcapL)] x (CrxFswitchxVsupply) decreases, the power decreases with a varying duty cycle.

Other control protocols can be envisaged.
For high / medium loads, use VcapH = −VcapL to determine the end of the switch conduction interval, giving a 50% duty cycle.
For low loads, use Ipeak (IpeakH) in HSS conduction and VcapL = 0 (giving dI / dt at maximum negative current) to get the desired low duty cycle mode.
Increase VcapL to 0 while decreasing Vsuply- (VcapH-VcapL) to take over the region between the two duty cycle modes.

As a result, the following occurs.
One operating parameter Poutrel for controlling the output power
・ Pout = effxVsupply 2 x (1- [VcapH-VcapL] / Vsupply) xCrxFswitch
・ Pout = effxVsupply 2 xPoutrelxCrxFswitch
Poutrel = 1- [VcapH-VcapL] / Vsupply

Further examination of Poutrel reveals that there are three regions of parameters.
Region 1 (R1): Poutrel> Prelborder:
VcapH = Vsupply / 2 [1-Poutrel]
VcapL = -Vsupply / 2 [1-Poutrel]
Region 2 (R2): Poutrel <Prelborder and VcapL <0
VcapL = -Vsupply / 2 [1-Poutrel] + VsupplyxK2x (Poutrelborder-Poutrel)
VcapH = Vsupply / 2 [1-Poutrel] + VsupplyxK2x (Poutrelborder-Poutrel)
Region 3 (R3): Other than that
VcapL = 0
VcapH = Vsupply / [1-Poutrel]

  The constant K2 defines the width of the region 2. K2 is selected to be greater than a predetermined minimum value, so that takeover from VcapH control to IpeakH control preferably occurs in any of the K2 regions, during which time the amplitude of IpeakH is HSS conduction The maximum value is not exceeded during the interval. K2 is also selected to be less than a predetermined maximum value, so that takeover from VcapH control to IpeakH control preferably occurs while the amplitude of IpeakH is greater than a predetermined minimum value at the takeover point.

  FIG. 2 shows the behavior of Vcap1 as a function of Poutrel in the region 0-2. In region 3, VcapL is equal to zero, whereas VcapH decreases from a large positive value. In region 2, both VcapL and VcapH decrease, whereas in region 1, VcapL and VcapH are symmetric around zero on the Y axis.

In region 2, the common mode term VsupplyxK2x (Poutrelorder-Poutrel) is selected to be proportional to Vsupply to obtain the same value of Poutrel that does not depend on Vsupply and that takes over from region 2 to region 1 occurs. It should be understood that Vsuply can be taken as both a variable parameter and a constant that cannot be used to control the resonant circuit. If Vsupply is not considered, the three regions are
R1) Poutrel> Prelorder:
VcapH = Vsupply / 2 [1-Poutrel]
VcapL = -Vsupply / 2 [1-Poutrel]
R2) Poutrel <Prelorder and VcapL <0
VcapL = -Vsupply / 2 [1-Poutrel] + K2x (Poutrelborder-Poutrel)
VcapH = Vsupply / 2 [1-Poutrel] + K2x (Poutrelborder-Poutrel)
R3) Otherwise VcapL = 0
VcapH = Vsupply / [1-Poutrel]
It is prescribed.

  Since VcapH and VcapL depend on a single operating parameter Poutrel, it is possible to know how to close the switch. The insertions labeled 20 and 21 show the time dependence of Poutrel and the corresponding Vcap21 and half-bridge voltage Vhb20. First, the low side switch is switched off and the high side switch is switched on, which can be obtained from the high bridge voltage shifting from a low value to a high value. Vcap1 rises rapidly, as shown at 24, in the first state where Vcap1 is less than VcapH, but does not reach the top of Vcap1, so the switch remains on (according to criterion 1). After a predetermined time, Vcap1 increases above VcapH as shown at 25, but just before reaching the top as shown at 26, the first part of the reference is satisfied, and VcapL becomes VcapH as shown at 28. Just before descending below, the second part of the reference is met and the switch is turned off. This is seen with the half-bridge voltage 27 dropping to a low value 27. In this situation, turn-off is controlled by the course of VcapL.

  FIG. 3 shows the corresponding situation of the primary current as a function of Poutrel. In this case, IpeakH is given a value IpeakH = K3 × VsupplyxPoutrel having the maximum value of Ipeak_max. IpeakL is given the value IpeakL_max. K3 can be selected according to a predetermined mode of operation, in which situation K3 is selected such that IpeakH takes control from VcapH in region 2. Iprim is shown as an amplitude, and in some regions 1 and 2, Iprim is always present within the boundary set by IpeakH and IpeakL, but in region 3, Iprim is greater than or equal to IpeakH. An important reason for taking over Ipeak control in the middle of Region 2 and in Region 3 is to make a smooth transition from a 50% duty cycle to a low duty cycle standby method (standby mode as determined by IpeakH and VcapL). Because. In region 1, the converter is protected from very large currents due to IpeakH_max and IpeakL (eg when a short circuit load occurs). In this case, Iprim is significantly greater than that shown in FIG. 3 at a given Poutrel, so that the primary current is limited by IpeakH and IpeakL. Thus, it can be seen that in region 3 the converter is controlled by the parameters IpeakH and VcapL and in regions 1 and 2 the converter is controlled only by VcapH and VcapL.

  When a very low voltage is supplied, the converter operates in mode R3. In practical use, Poutrel does not decrease to 0 in this case, but is maintained at a predetermined minimum value. This minimum value gives exactly a predetermined minimum value for the HSS and thus gives a predetermined minimum energy to the resonant inductor. This minimum energy circulates during the next LSS conduction period, so that Cp can be charged to the positive supply rail after the LSS is switched off and at the beginning of the next period the HSS Can be switched on smoothly. Since Poutrel is held constant, another control mechanism for controlling the output power is required. In the control mechanism, the LSS turn-off instant is not the first instant when the primary current reaches the maximum negative value, but after one or more complete resonance cycles, and is therefore the n-th valley of Iprim. . This control method can be realized by the present invention by setting IpeakH to a desired minimum value and setting VcapL to 0V. The skip of the resonance cycle can be realized as follows.

  Reference 4 or 4a ([Vcap1 valley reached] AND [Vcap1> VcapL] or equivalent reference [primary current <0] AND [Vcap1> VcapL]) is satisfied and the second is satisfied during the short time window Tw1 The LSS is switched off when the criteria are met. A time interval begins that instantly starts when the LSS is switched on.

  4 and 5 show the time variation of various parameters in the simulation of the situation of use. The top graphs 40 and 50 show the primary current. The next graphs 41 and 51 show the voltage at the half bridge point 8. A high voltage indicates that the high side switch is on and thereby connects the circuit to the feed rail, while a low voltage indicates that the low side switch is switched on. The next graph represents the voltage Vcap1, and the last graphs 43 and 53 represent the current flowing through the inductor L3 on the secondary side of the resonator.

  FIG. 4 shows a situation where a small duty cycle is used, i.e. a situation where the low switch is open longer than the high switch. In the situation represented by reference numeral 44, the high-side switch is on until the primary current is greater than IpeakH (48) (reference 1), and when the primary current is greater than IpeakH, the high-side switch is turned off. The half-bridge voltage drops to a low level. IpeakL is not specified in this situation (eg, by setting IpeakL to a large negative value), so the low switch off is not determined by the course of the primary current (45) and is determined by criterion 4. Is done. The reason is that a valley is reached in Vcap1 (402), and Vcap1 becomes larger than VcapL (47). Thus, after setting VcapL to a predefined value (49), the switch opens when Vcap1 reaches the valley and Vcap1 is greater than VcapL, which occurs instantaneously at the time indicated by reference numeral 47. The primary current rises instantaneously at this time (45), and the half-bridge voltage rises to a high value indicating that the high-side switch is turned on.

  The control algorithm shown in FIG. 5 shows a high power state with a 50% duty cycle. In FIG. 5, the focus is on VcapH and VcapL as control parameters. The high side switch is turned on at 58. First, it is detected that the peak of Vcap1 has been reached (54), and while it is detected that Vcap1 is greater than VcapH (55), the high-side switch remains on. However, once Vcap1 <VcapH, criterion 2 is satisfied and the high side switch is turned off. The corresponding algorithm follows 56, 57 which determines when the low side switch should be switched off.

The control algorithm may have the feature that the operating parameters are compensated for Vsupply and / or Fswitch. In this case, a new operation parameter Poutrel comp is defined as follows.
Poutrel = Poutrel comp x1 / [Vsupply 2 xFswitch]
In this case, Vsupply is the actual power supply voltage of the converter, and Fswitch is the actual operating voltage of the converter. Thus, the output power equation is as follows:
Pout = effxVsupply 2 xPoutrelxCrxFswitch
Therefore,
Pout = effxVsupply 2 xPoutrel comp x1 / [Vsupply 2 xFswitch] xCrxFswitch
= effxPoutrel comp xCr
It becomes.

This operating parameter Poutrel comp creates a direct relationship between the control parameter and the output power.

  The present invention can also be used in combination with a full bridge converter. This is also within the scope of the present invention. An example of a circuit diagram of the full bridge converter is shown in FIG.

In the context of a full bridge converter, a reference is set for all four switches. For example, HSS1 and HSS2 are conducted or LSS1 and LSS2 are conducted. However, an equivalent mode associated with a half-bridge converter can be realized, in which case the switch combination is controlled as follows.
State 1: HSS1 and LSS2 conduction State 2: HSS2 and LSS1 conduction

  The main difference from the half-bridge converter is that the voltage of the resonant tank is doubled. In this mode, when state 1 ends, HSS1 and LSS2 are switched off. Both Cp2 and Cp1 are charged to the opposite feed rails by the primary current as in the half-bridge converter. In this case, HSS2 and LSS1 can be switched on similarly to the half-bridge converter.

Control parameter criteria or switch control algorithms can be extended in this situation as follows.
-HSS2 and LSS1 turn-off:
-[Primary current> IpeakH] OR
-[Vtrafo1 mountain reached] AND [Vtrafo1 <VcapH]
HSS1 and LSS2 turn-off:
− [Primary current <IpeakL] OR
-[Vtrafo1 valley reached] AND [Vtrafo1> VcapL]

  FIG. 7 shows an embodiment of the invention showing the coupling of the state parameter circuit to the control logic CL, which is connected to the analog control function ACF or has an analog control function AFC. Further, the position of the resonance capacitor Cr is changed as compared with FIG. Such changes in the circuit, for example the position of the resonant capacitor Cr, are within the scope of the present invention. Changing the position of the resonant capacitor also changes the predetermined point at which the resonant tank voltage is monitored. In the embodiment of FIG. 7, the capacitance Vcap1 is monitored at a point with a reference number 60 different from the point 9 shown in FIG. These two points 9 and 60 are the most measurable points, and point 9 is preferable. This is because the voltage of 9 is a direct display of Vhb (8) + Vcr (9-8). The voltage across the resonant capacitor (9-8) can also be measured, for example, while placed between L1 and L2 or on the other side of Rs (see FIG. 1). The desired parameter (voltage of 9) can then be constructed by measuring Vhb (8) and Vcr and adding these two together. The scope of the present invention is that Cr is placed between L1 and L2 or Rs and ground or excludes Rs (if the current is measured otherwise) and Vhb + Vcr is 9 or 60 Embodiments used as an alternative to voltage are also covered. Vcr is actually the integral of the resonant tank current (capacitor Cr acts as an integrator), so that the current is integrated and the integrated current instead of the voltage measured at 9 or 60 is added to Vhb Is used. This embodiment is also covered by the present invention. Modifications of the control algorithm, such as the exclusion of Vcap1 peak or valley detection, or replacement of peaks or valleys with (fixed) time after the start of the conduction interval, or other adaptation criteria are also within the scope of the present invention.

  The resonance tank current Iprim is monitored as a current flowing through the current detection resistor Rs. In general, all current sensing methods can be used here, for example, Hall elements, current measurement transformers, and the like. The resistor Rs is placed at the point of the circuit between the capacitor Cr and ground. The monitored voltage signal, Vcap1, and the monitoring signal representing the monitored current are fed back to control logic CL through monitoring lines 71 and 72, respectively.

  In FIG. 8, the implementation of the control logic and analog control functions CL, ACF will be described in more detail. The control logic and analog control function blocks represent functions that implement embodiments of the proposed control algorithm.

  A monitoring voltage signal 71 representing Vcap1 is input to the control logic CL. Even if the coupling of state parameters from the circuit to the control logic is shown for other embodiments of the resonant circuit other than the resonant circuit shown in FIG. 1, Iprim and Vcap1 may be represented by the circuit shown in FIG. The same can be obtained for all the other resonance circuits in the circuit. With the circuit of FIG. 7, a DC voltage component is present in Vcap1, complicating Vcap1 control. The control logic comprises equivalent means for determining when the peak / valley point of the peak and valley detector VT or Vcap1 is reached. The output of the peak and valley detector VT is input to the output logic OL.

  The monitored current signal 71 representing Vcap1 is further connected to two comparators 82 and 83. In the comparators 82 and 83, the Vcap1 signal is compared with the values of the control parameters VcapH and VcapL.

  The monitored current signal representing the voltage Iprim is input to the control logic CL and the two comparators 84 and 85. In the comparators 84 and 85, the Iprim signal 72 is compared with the values of the control parameters IpeakL and IpeakH.

  In this embodiment, the control parameters IpeakH, IpeakL, VcapL, and VcapH are given by the control parameter determination block CPD86. In this case, a desired Poutrel is input to the control parameter determination block CPD86, and the control parameters have been described above. Thus, it is determined based on the value of Poutrel. However, block 86 exists only in embodiments where the mode of operation is a single parameter, ie, a mode that controls output power with Poutrel. Control parameters can also be supplied to the comparator by other means.

  The outputs of the comparators 82 to 85 are supplied to the output logic OL. The output logic is manipulated according to inputs from comparators 82-85 and peak / valley detector VT. Based on these inputs, the output logic outputs HSS and LSS status outputs 80 and 81 to the HS and LS drivers HSD and LSD.

  In an embodiment that includes Vsupply and frequency compensation, a Vsupply-frequency compensation block VFC 90 can be added for Poutrel 86 compensation.

  FIG. 9 shows an embodiment where Poutrel is compensated for power supply voltage and switching frequency at 90. In this embodiment, the operating frequency of the switch is supplied to the Vsupply-Fswitch compensation block VFS90 through 91, and Poutrel is output according to the algorithm described so far.

  In this embodiment, the control logic CL can be provided by general-purpose computer means or dedicated programmable computer means, in which case a monitoring signal can be input and the computer means, for example, the method of the present invention. It can be implemented to manipulate the control logic by executing computer code that implements it.

  Although the present invention has been described in connection with preferred embodiments, these embodiments are not intended to be limited to the specific forms described herein. The scope of the invention is limited only by the appended claims. Here, reference to two switches does not exclude embodiments of more than two switches.

  In the present specification, specific details of the disclosed embodiments, such as specific implementations, circuit diagrams, etc., have been set forth for purposes of explanation and not limitation in order to provide a clear and complete understanding of the invention. However, it will be readily appreciated by those skilled in the art that the present invention may be practiced in other embodiments that do not follow the details described herein without departing from the scope of the invention as defined by the claims. Can do. Furthermore, in this connection, for the sake of brevity and clarity, detailed descriptions of well-known devices, circuits and procedures have been omitted to avoid unnecessary details and possible confusion.

  Although reference signs are included in the claims, inclusion of reference signs is only for clarity reasons and should not be construed as limiting the scope of the claims. The term “comprising” does not exclude the presence of elements or steps other than those listed in a claim. An element does not exclude the presence of a plurality of elements. The present invention can be implemented by hardware comprising a plurality of individual elements and / or appropriately programmed processors. In the device claim enumerating several means, several of these means can be embodied by one and the same item of hardware. Certain measures recited in mutually different dependent claims do not imply that a combination of these measures cannot be used to advantage.

FIG. 1 shows a first embodiment of a resonant converter. FIG. 2 shows Vcap1 as a function of Poutrel in the range 0-2. FIG. 3 shows Iprim as a function of Poutrel in the range 0-2. FIG. 4 shows the time variation of the various parameters in the simulation of the situation used for the first set of parameters. FIG. 5 shows the time variation of the various parameters in the simulation of the situation used for the second set of parameters. FIG. 6 shows an embodiment of a full bridge configuration. FIG. 7 shows the coupling from the state parameter circuit to the control logic and analog control functions. FIG. 8 shows a first implementation of control logic and analog control functions. FIG. 9 shows a second implementation of the control logic and analog control functions.

Claims (11)

  1. A resonant converter for supplying electrical energy from a power source to a load,
    A controllable first and second switch arranged in series connected between power supply terminals;
    A control device for generating a control signal for opening and closing the first and second switches;
    A resonant tank electrically connected to the first and second switches;
    The resonant tank comprises a resonant capacitor;
    One of the first and second switches is switched on, the current flowing through the resonant tank is monitored, the voltage at a predetermined point of the resonant tank is monitored, and the first switch is switched on. A first conduction interval occurs while the second switch is turned on, a second conduction interval occurs, and two control parameters are set for the first conduction interval. , Two control parameters are set for the second conduction interval, the control parameters for the two conduction intervals are the peak current of the interval and the predetermined voltage of the interval, and the first and second The resonance converter is turned off according to a criterion including the four control parameters.
  2.   2. The resonant converter according to claim 1, wherein the first switch is turned off when the first criterion or the second criterion is satisfied, and the current exceeds the peak current based on the first criterion. The second reference is that the minimum time has elapsed after the switch is turned on and the voltage is below a level set by a predetermined voltage of the interval; When the criterion or the second criterion is satisfied, the second switch is turned off, the first criterion is that the current is less than the peak current, and the second criterion is the switch The voltage is above the level set by the predetermined voltage of the interval as the minimum time elapses after the is switched on. Resonant converter.
  3.   2. The resonant converter according to claim 1, wherein the four control parameters are set to operate the resonant circuit according to a desired operation mode.
  4.   The resonant converter of claim 1, wherein the four control parameters are set to stabilize a desired operating mode of the resonant converter, and a stabilization loop incorporates the four control parameters into the stabilization procedure. A characteristic resonant converter.
  5.   4. The resonant converter according to claim 3, wherein the desired operation mode is a mode in which output power is controlled.
  6.   6. The resonant converter according to claim 5, wherein the output power is controlled by setting the control parameter in accordance with a value of a single operating parameter.
  7.   7. The resonant converter according to claim 6, wherein a region of the single operating parameter is defined, and the output power is controlled by setting the control parameter according to a value of the single operating parameter of the region. Resonant converter.
  8.   7. The resonant converter according to claim 6, wherein the single operating parameter is adjusted according to a power supply voltage and an operating frequency of the converter.
  9. Control logic for controlling the resonant circuit,
    Output logic for controlling the first and second switches arranged in series connected between the power supply terminals;
    While the current of the resonant tank of the resonant converter and the voltage of the predetermined point of the resonant tank are input, the current and the voltage are compared with control parameters, and the first switch is turned on. Two control parameters are set for the first conduction interval that occurs, two parameters are set for the second conduction interval that occurs while the second switch is turned on, The control parameter for the conduction interval is a peak current of the interval and a predetermined voltage of the interval, and at least the first and second switches are turned off according to a criterion including the four control parameters. Control logic characterized by comprising a comparator.
  10. A method of controlling a resonant converter that supplies electrical energy from a power source to a load,
    The resonant circuit is
    First and second switches arranged in series connected between power terminals;
    A control device for generating a control signal for opening and closing the first and second switches;
    A resonant tank electrically connected to the first and second switches;
    The resonant tank comprises a resonant capacitor;
    Monitoring the current flowing through the resonant tank and a predetermined point of the resonant tank;
    Comparing the resonant tank current of the resonant converter and the voltage at a predetermined point of the resonant tank with control parameters;
    The current and the voltage are compared with a control parameter, two control parameters are set for a first conduction interval that occurs while the first switch is turned on, and the second switch is turned on Setting two parameters for a second conduction interval that occurs while being switched to, and setting the control parameters for the two conduction intervals to a peak current of the interval and a predetermined voltage of the interval When,
    Switching off said first and second switches according to a criterion comprising four said control parameters.
  11.   A computer readable code capable of causing a programmable device to perform the method of claim 10.
JP2008503652A 2005-04-01 2006-03-24 Control of resonant converter Withdrawn JP2008535456A (en)

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US20080266908A1 (en) 2008-10-30
EP1869759B1 (en) 2019-08-07
WO2006103609A2 (en) 2006-10-05
EP1869759A2 (en) 2007-12-26
US7944716B2 (en) 2011-05-17

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