JP3266389B2 - Series resonant converter - Google Patents

Series resonant converter

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Publication number
JP3266389B2
JP3266389B2 JP30336493A JP30336493A JP3266389B2 JP 3266389 B2 JP3266389 B2 JP 3266389B2 JP 30336493 A JP30336493 A JP 30336493A JP 30336493 A JP30336493 A JP 30336493A JP 3266389 B2 JP3266389 B2 JP 3266389B2
Authority
JP
Japan
Prior art keywords
series
resonance
current
sets
period
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Expired - Fee Related
Application number
JP30336493A
Other languages
Japanese (ja)
Other versions
JPH07135769A (en
Inventor
清嗣 小津
克彦 山本
Current Assignee (The listed assignees may be inaccurate. Google has not performed a legal analysis and makes no representation or warranty as to the accuracy of the list.)
Shindengen Electric Manufacturing Co Ltd
Nippon Telegraph and Telephone Corp
Original Assignee
Shindengen Electric Manufacturing Co Ltd
Nippon Telegraph and Telephone Corp
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Shindengen Electric Manufacturing Co Ltd, Nippon Telegraph and Telephone Corp filed Critical Shindengen Electric Manufacturing Co Ltd
Priority to JP30336493A priority Critical patent/JP3266389B2/en
Publication of JPH07135769A publication Critical patent/JPH07135769A/en
Application granted granted Critical
Publication of JP3266389B2 publication Critical patent/JP3266389B2/en
Anticipated expiration legal-status Critical
Expired - Fee Related legal-status Critical Current

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Description

【発明の詳細な説明】DETAILED DESCRIPTION OF THE INVENTION

【0001】[0001]

【産業上の利用分野】本発明は、スイッチング電源等に
適用される直列共振コンバータに関するものである。
BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a series resonance converter applied to a switching power supply or the like.

【0002】[0002]

【従来の技術】従来の直列共振コンバータの回路構成の
一例を図3に示す。例えばMOSFETなどの2個の主
スイッチング素子11,12が直列接続され1組のブリ
ッジ回路が構成され、別に2個の主スイッチング素子1
3,14が直列接続され、もう1組のブリッジ回路が構
成され、各々のブリッジ回路が直流電源10に接続され
て、いわゆるフルブリッジ回路が構成されている。
2. Description of the Related Art FIG. 3 shows an example of a circuit configuration of a conventional series resonance converter. For example, two main switching elements 11 and 12 such as MOSFETs are connected in series to form a set of bridge circuits.
3 and 14 are connected in series to form another set of bridge circuits, and each bridge circuit is connected to the DC power supply 10 to form a so-called full bridge circuit.

【0003】主スイッチング素子11,12の接続点
と、主スイッチング素子13,14の接続点間にそれぞ
れのトランス19,20の一次巻線の一端が接続され、
トランス19,20の他端間に共振インダクタンス21
と共振コンデンサ22が接続されている。
One end of a primary winding of each of transformers 19 and 20 is connected between a connection point of main switching elements 11 and 12 and a connection point of main switching elements 13 and 14,
A resonance inductance 21 between the other ends of the transformers 19 and 20;
And the resonance capacitor 22 are connected.

【0004】共振インダクタンス21と共振コンデンサ
22の両端には、2組のブリッジ回路に対応して、主ス
イッチング素子11,12及び13,14がOFFの期
間エネルギーを直流電源10に帰還する方向に、帰還ダ
イオード15,16の直列回路と帰還ダイオード17,
18の直列回路が直流電源10に接続されている。
[0004] Both ends of the resonance inductance 21 and the resonance capacitor 22 correspond to the two sets of bridge circuits, and return the energy to the DC power supply 10 while the main switching elements 11, 12, 13 and 14 are OFF. The series circuit of the feedback diodes 15, 16 and the feedback diode 17,
18 series circuits are connected to the DC power supply 10.

【0005】又、トランス19,20の二次巻線にはそ
れぞれ整流用ダイオード23,24及び25,26が全
波整流回路として構成され、各々の出力が並列接続さ
れ、平滑コンデンサ27及び負荷抵抗28が接続されて
いる。
The secondary windings of the transformers 19 and 20 are provided with rectifying diodes 23 and 24 and 25 and 26, respectively, as full-wave rectifier circuits, the outputs of which are connected in parallel, a smoothing capacitor 27 and a load resistor. 28 are connected.

【0006】従来技術の直列共振コンバータは次の様な
動作をする。主スイッチング素子11と主スイッチング
素子14をターンオンさせると、直流電源10−主スイ
ッチング素子11−トランス19の一次巻線−共振イン
ダクタンス21−共振コンデンサ22−トランス20の
一次巻線−主スイッチング素子14−直流電源10のル
ープで図3の点線に示す様な共振電流I1 が流れる。
The prior art series resonant converter operates as follows. When the main switching element 11 and the main switching element 14 are turned on, the DC power supply 10-the main switching element 11-the primary winding of the transformer 19-the resonance inductance 21-the resonance capacitor 22-the primary winding of the transformer 20-the main switching element 14- loop resonance current I 1 such as shown in dotted line in FIG. 3 flows in the DC power supply 10.

【0007】この共振電流I1 は図4の様にON幅期間
ONの期間流れ続ける。共振電流I1 は、トランス1
9,20のそれぞれの一次巻線と二次巻線との比をNと
すると、それぞれのトランスの二次巻線に1/N倍の共
振電流となって流れるので、この共振電流を整流用ダイ
オード23,24及び25,26で整流し、平滑コンデ
ンサ27を充電して直流電圧を負荷抵抗28に供給す
る。
The resonance current I 1 continues to flow during the ON width period T ON as shown in FIG. The resonance current I 1 is
Assuming that the ratio between the primary and secondary windings 9 and 20 is N, a resonance current of 1 / N times flows through the secondary windings of the respective transformers. The current is rectified by the diodes 23, 24 and 25, 26, and the smoothing capacitor 27 is charged to supply a DC voltage to the load resistor 28.

【0008】又、主スイッチング素子11,14のOF
Fの期間に於いては、共振コンデンサ22に図示の極性
で充電された電荷が、図3の点線で示す帰還電流I2
なって電源に帰還され、その電流は図4の帰還電流期間
1 の期間流れる。
The OF of the main switching elements 11 and 14
Is In the period F, the electric charge charged in the shown polarity to the resonant capacitor 22, is fed back to the power supply becomes the feedback current I 2 indicated by the dotted line in FIG. 3, the feedback current period T of the current 4 Flows for one period.

【0009】この方式では、主スイッチング素子11,
14のON幅は一定であり、OFF幅、すなわち図4の
休止期間T2 の期間を可変させて出力の定電圧制御を行
う様に動作する。従って、軽負荷時は、OFF期間T
OFF の内、休止期間T2 が長くなり、動作周波数が下が
り、やがて20KHZ前後の可聴周波数領域となる。
In this system, the main switching element 11,
ON width of 14 is constant, operated as to perform OFF width, ie the constant voltage control of the output by varying the duration of rest period T 2 of the FIG. Therefore, at light load, the OFF period T
Among OFF, the longer the idle period T 2, lower the operating frequency and eventually 20KHZ around audible frequency range.

【0010】尚、以上は半サイクル動作についての説明
であるが、次の半サイクルは主スイッチング素子12,
13がONする事によって上に述べた事と全く同じ動作
を行う。
The above description is for a half cycle operation, but the next half cycle is for the main switching element 12,
When 13 is turned on, the same operation as described above is performed.

【0011】[0011]

【発明が解決しようとする課題】しかしながら、従来の
技術には次の様な問題点が生ずる。すなわち、負荷の重
い時は、ON幅期間すなわちTONは一定であるので、O
FF幅期間すなわちTOF F の内、休止期間T2 の期間が
短く、従って、動作周波数が30〜60KHZで安定動
作を行っている。しかし、負荷が軽くなると、TONは一
定である為にTOF F の内、休止期間T2 の期間が長くな
り、従って、動作周波数が20KHZ程度まで下がり可
聴周波数領域で動作を行うようになる。
However, the prior art has the following problems. That is, when the load is heavy, the ON width period, that is, T ON is constant.
Of FF width period or T OF F, duration of rest period T 2 is short, therefore, the operating frequency is performing stability with 30~60KHZ. However, when the load decreases, among T OF F for T ON is constant, the period of rest period T 2 becomes longer, therefore, the operating frequency is to perform the operation at audio frequency range down to about 20KHZ .

【0012】この周波数領域になると動作時の発振音が
耳ざわりになり、騒音問題も引き起こす。又、無負荷に
なるに従ってOFF幅期間(TOFF )が限りなく長くな
り、動作が不安定になるという欠点も生ずる。
In this frequency range, an oscillating sound during operation becomes harsh and causes a noise problem. In addition, there is a disadvantage that the OFF width period (T OFF ) becomes infinitely longer as the load is reduced, and the operation becomes unstable.

【0013】従って、上記の如く従来技術の有する問題
点を克服すべく、本発明の目的は次の通りである。すな
わち、直列共振コンバータのサージ電圧、電流によるノ
イズが小さいという特徴を備えたまま、軽負荷時には、
ON幅期間制御、すなわちPWM(PULSE WIDTH MODULA
TION)制御を行う動作を併用して、OFF幅期間が必要
以上に長くならない様な動作を行う。この様にして、可
聴周波数領域にまで動作周波数が下がらない様にした直
列共振コンバータを提供することを目的とする。
Accordingly, in order to overcome the problems of the prior art as described above, the objects of the present invention are as follows. In other words, at the time of light load, with the characteristic that the noise due to the surge voltage and current of the series resonance converter is small,
ON width period control, that is, PWM (PULSE WIDTH MODULA)
TION) An operation is performed so that the OFF width period does not become unnecessarily long together with the operation for performing the control. Thus, an object of the present invention is to provide a series resonant converter in which the operating frequency is not reduced to an audible frequency range.

【0014】[0014]

【課題を解決するための手段】すなわち本発明は2個の
主スイッチング素子が直列接続された、2組のブリッジ
回路が直流電源に接続され、この2組のブリッジ回路の
2個直列主スイッチング素子の接続点には、各々トラン
スの一次巻線の一端が接続され、各々のトランスの一次
巻線の他端間に共振インダクタンスと共振コンデンサの
直列回路が接続されている。さらに共振インダクタンス
と共振コンデンサの直列回路の両端より、2組のブリッ
ジ回路毎の2個の直列主スイッチング素子に対応して、
電源へエネルギーを帰還する方向に、2個の帰還ダイオ
ードの直列回路が、2組直流電源に接続されている。
又、各々のトランスの二次巻線に全波整流回路と平滑コ
ンデンサが接続され、負荷に電力を供給する様に直列共
振コンバータが構成されている。
That is, according to the present invention, two sets of bridge circuits in which two main switching elements are connected in series are connected to a DC power supply, and two series main switching elements of the two sets of bridge circuits are connected. Is connected to one end of a primary winding of each transformer, and a series circuit of a resonance inductance and a resonance capacitor is connected between the other ends of the primary windings of each transformer. Further, from both ends of the series circuit of the resonance inductance and the resonance capacitor, corresponding to two series main switching elements for each of two sets of bridge circuits,
A series circuit of two feedback diodes is connected to two sets of DC power supplies in a direction of returning energy to the power supply.
Further, a full-wave rectifier circuit and a smoothing capacitor are connected to the secondary winding of each transformer, and a series resonant converter is configured to supply power to a load.

【0015】そしてこの直列共振コンバータの2組の2
個直列帰還ダイオードの各々に、電源に順方向の補助ス
イッチング素子を並列接続して、主スイッチング素子の
動作周波数が可聴周波数領域より低い周波数の時は、2
組2個の補助スイッチング素子が、それぞれの主スイッ
チング素子の導通期間内で導通幅制御(PWM)を行う
如く動作する様に構成された事を特徴としている。
The two series 2 of this series resonance converter
When a forward auxiliary switching element is connected in parallel to the power supply to each of the series feedback diodes, and the operating frequency of the main switching element is lower than the audible frequency range, 2
The two auxiliary switching elements are characterized in that they are configured to operate so as to perform conduction width control (PWM) within the conduction period of each main switching element.

【0016】[0016]

【作用】この様な手段を有する直列共振コンバータは、
負荷の重い時は従来の回路と全く同じ様にOFF幅期間
制御を行うが、負荷が軽くなって動作周波数が可聴周波
数に近くなるまで下がると、ON幅期間制御(PWM制
御)に切替わり可聴周波数まで下らない様な制御を行
う。
The series resonant converter having such means is
When the load is heavy, the OFF width period control is performed in exactly the same way as with the conventional circuit. However, when the load becomes lighter and the operating frequency falls close to the audio frequency, it is switched to the ON width period control (PWM control), Control so that it does not fall to the frequency.

【0017】すなわち軽負荷になると、主スイッチング
素子の導通期間内で、それぞれの主スイッチング素子に
対応する補助スイッチング素子が導通して、トランス1
次電流として流れていた共振電流を補助スイッチング素
子に転流させ、トランス1次電流を流さない様にする事
によってON幅期間制御を行うものである。
That is, when the load becomes light, the auxiliary switching elements corresponding to the respective main switching elements conduct during the conduction period of the main switching elements, and the transformer 1
The ON width period control is performed by diverting the resonance current flowing as the secondary current to the auxiliary switching element so that the primary current of the transformer does not flow.

【0018】[0018]

【実施例】図1はこの発明の一実施例を示し、従来技術
の図3と対応する部分は同一符号がつけてある。又、図
2はこの発明による主要部分の電流波形を示している。
この発明では、従来技術に対して補助スイッチング素子
29,30,31,32が追加され、直流電源10に順
方向となる様にそれぞれの帰還ダイオード15,16,
17,18に並列接続されている。
FIG. 1 shows an embodiment of the present invention, and portions corresponding to those of FIG. 3 of the prior art are denoted by the same reference numerals. FIG. 2 shows a current waveform of a main part according to the present invention.
According to the present invention, auxiliary switching elements 29, 30, 31, and 32 are added to the prior art, and respective feedback diodes 15, 16, and
17 and 18 are connected in parallel.

【0019】この発明の直列共振コンバータの動作は次
の様である。負荷が重い時、すなわち負荷抵抗28が小
さい時は、従来技術で述べた通り、主スイッチング素子
11,14のONにより、図1の点線の様に共振電流I
1 を流す。この共振電流I1は、従来技術の図4と同様
半サイクル期間トランス19,20に流れ、トランス2
次巻線に変換されて負荷抵抗28に負荷電流として流れ
る。
The operation of the series resonant converter of the present invention is as follows. When the load is heavy, that is, when the load resistance 28 is small, as described in the related art, when the main switching elements 11 and 14 are turned on, the resonance current I as shown by the dotted line in FIG.
Pour 1 The resonance current I 1 flows in the same half cycle transformer 19, 20 and 4 of the prior art, trans-2
It is converted to the next winding and flows as a load current through the load resistor 28.

【0020】そして半サイクル後、共振コンデンサ22
にたくわえられた電荷が帰還電流I2 となって流れ、そ
の後休止期間T2 が継続されることは従来技術の図4の
波形と全く同じである。
After a half cycle, the resonance capacitor 22
That the stored charge flows as the feedback current I 2 and then the rest period T 2 is continued is exactly the same as the waveform in FIG. 4 of the prior art.

【0021】次に軽負荷となり負荷抵抗28が大きくな
った時、定電圧制御により、OFF幅期間TOFF の内、
休止期間T2 期間が長くなり動作周波数が下がる。そし
て20KHZ程度の可聴周波数領域に近ずくと、ON幅
期間制御すなわちPWM(PULSE WIDTH MODULATION)制
御による定電圧制御に切替わる。
Next, when the load becomes lighter and the load resistance 28 becomes larger, the constant voltage control is performed for the OFF width period T OFF .
Rest period period T 2 is the operating frequency is lowered long. Then, when approaching the audible frequency range of about 20 KHZ, the control is switched to the ON voltage period control, that is, the constant voltage control by the PWM (PULSE WIDTH MODULATION) control.

【0022】このPWM制御の動作以後が本発明に係わ
る部分である。すなわち、可聴周波数領域に近づくと、
次の様な動作によりPWM制御が行われる。主スイッチ
ング素子11,14がONしていて、共振電流であるト
ランス1次電流I1 が図1の点線の様に流れている期間
に、補助スイッチング素子29,32をターンオンさせ
ると、トランス19及びトランス20の一次巻線にはト
ランス1次電流I1 による一次巻線電圧が発生している
為、共振電流(トランス1次電流)I1 は主スイッチン
グ素子から補助スイッチング素子側に転流する。
The portion after the operation of the PWM control is a portion related to the present invention. That is, when approaching the audible frequency range,
PWM control is performed by the following operation. The main switching element 11, 14 have turned ON, the transformer primary current I 1 is resonant current during flowing like a dotted line in FIG. 1, to turn on the auxiliary switching element 29 and 32, transformer 19 and since the primary winding of the transformer 20, which primary winding voltage by the transformer primary current I 1 is generated, the resonant current (transformer primary current) I 1 is commutated from the main switching element in the auxiliary switching element side.

【0023】すなわち図2の様にトランス1次電流I1
がPWM ON幅期間TA の期間流れている時に補助ス
イッチング素子29,32をONさせると、トランス1
次電流I1 は直流電源10−補助スイッチング素子29
−共振インダクタンス21−共振コンデンサ22−補助
スイッチング素子32−直流電源10のループに転流さ
れ、図1の一点鎖線の様にバイパス共振電流I1 ’に切
替わり共振電流を流し続ける。
That is, as shown in FIG. 2, the transformer primary current I 1
When There is ON the auxiliary switching element 29 and 32 when the flowing period of the PWM ON width period T A, the transformer 1
The secondary current I 1 is supplied from the DC power supply 10
- commutated loop of the resonant inductance 21 resonance capacitor 22-auxiliary switching element 32 DC power supply 10, continues to flow switching instead resonance current to bypass the resonance current I 1 'as a one-dot chain line in FIG. 1.

【0024】このバイパス共振電流I1 ’はトランス1
9,20の一次巻線には流れない為、図2の様にPWM
OFF幅期間TB の期間トランス1次電流I1 は零と
なり、バイパス共振電流I1 ’がバイパスループに流れ
るので、共振電流としては半サイクルの期間連続的に流
し続けることになる。
This bypass resonance current I 1 ′ is
Since the current does not flow through the primary windings 9 and 20, as shown in FIG.
OFF width period T period transformer primary current I 1 becomes zero in B, since the bypass resonance current I 1 'flows through the bypass loop, it will continue to flow in continuous of the half-cycle period as the resonance current.

【0025】従って、共振電流は連続的に流し続けなが
ら、トランス19,トランス20の一次巻線には、補助
スイッチング素子をONさせる事によって、必要な期間
のみ共振電流(トランス1次電流)I1 を流すことが出
来るので、結果的にON幅期間TONが制御されて定電圧
制御が行われることになる。
Accordingly, the resonance current (transformer primary current) I 1 is only provided for a necessary period by turning on the auxiliary switching element in the primary winding of the transformer 19 and the transformer 20 while the resonance current is continuously flowing. As a result, the ON width period T ON is controlled and constant voltage control is performed.

【0026】共振電流が流れ終わった後、共振コンデン
サ22にたくわえられた電荷が帰還電流I2 となって、
帰還電流期間T1 の期間電源に帰還されるモード、及び
休止期間T2 のモードについては従来技術と全く同じで
ある。
After the end of the resonance current, the charge stored in the resonance capacitor 22 becomes the feedback current I 2 ,
The mode in which the power is fed back to the power supply during the feedback current period T 1 and the mode during the idle period T 2 are exactly the same as those in the related art.

【0027】従って、図2に於いて、負荷の重い時のO
N幅期間はTA +TB であるので、動作責務(DUT
Y)は(TA +TB )/Tであり、軽負荷時のON幅期
間はTA であるので動作責務(DUTY)はTA /Tと
なる。この様にして重負荷時は周波数制御を行い、軽負
荷時はPWM制御を行う事によって、常に安定的に定電
圧制御を行う事が出来るものである。
Accordingly, in FIG. 2, when the load is heavy, O
Since the N width period is T A + T B , the operation duty (DUT
Y) is (T A + T B) / T, operating duty since ON width period of the light load is a TA (DUTY) becomes T A / T. In this way, the frequency control is performed at the time of heavy load, and the PWM control is performed at the time of light load, so that the constant voltage control can always be stably performed.

【0028】又、図示しないがトランス19及びトラン
ス20に固有するリーケージインダクタンス、又は別に
トランスと直列に設けた共振インダクタンス21、ある
いは共振インダクタンス21から帰還回路に直列に新た
な共振インダクタンス21を設ける事により、転流時の
トラブルを防ぐことが出来る。例えば、共振電流I1
ら帰還電流I2 に転流する時、帰還ダイオードのリカバ
リータイムの間、短絡電流の発生を防止したり、各転流
時に一定時間かけて切替わる為に発生するスイッチング
ロスやスイッチングノイズを上に述べたインダクタンス
を設ける事により防止する事が出来る。
Although not shown, a leakage inductance inherent to the transformer 19 and the transformer 20, a resonance inductance 21 separately provided in series with the transformer, or a new resonance inductance 21 provided in series from the resonance inductance 21 to a feedback circuit is provided. Therefore, trouble during commutation can be prevented. For example, when commutating from the resonance current I 1 to the feedback current I 2 , during the recovery time of the feedback diode, a short-circuit current is prevented from occurring, or a switching loss occurs due to switching over a certain time at each commutation. And switching noise can be prevented by providing the above-described inductance.

【0029】[0029]

【発明の効果】本発明は以上説明した様な構成である
為、以下に示すような効果を有する。すなわち、重負荷
時の周波数制御、軽負荷時のPWM制御を併用する事に
よって、全負荷から無負荷までの全領域に於いて、動作
周波数が低くなる事による不安定動作を防止したり、
又、可聴周波数での動作領域をさけている為、騒音防止
の効果も呈する。さらに転流電流の急峻な切替わりによ
るスイッチングロスやスイッチングノイズの発生を防止
出来る従来の直列共振コンバータの特徴を失う事もな
く、本発明回路は実現出来るので、より有用なスイッチ
ング電源を提供し得るものである。
Since the present invention has the configuration as described above, it has the following effects. That is, by using the frequency control at the time of heavy load and the PWM control at the time of light load together, it is possible to prevent the unstable operation due to the lowering of the operating frequency in the entire region from full load to no load,
In addition, since the operating region at the audible frequency is avoided, the effect of preventing noise is exhibited. Further, since the circuit of the present invention can be realized without losing the features of the conventional series resonant converter that can prevent the occurrence of switching loss and switching noise due to abrupt switching of the commutation current, a more useful switching power supply can be provided. Things.

【図面の簡単な説明】[Brief description of the drawings]

【図1】本発明の直列共振コンバータ。FIG. 1 is a series resonance converter of the present invention.

【図2】本発明の主要部電流波形。FIG. 2 is a main part current waveform of the present invention.

【図3】従来の直列共振コンバータ。FIG. 3 shows a conventional series resonance converter.

【図4】従来型の主要部電流波形。FIG. 4 is a conventional main part current waveform.

【符号の説明】[Explanation of symbols]

10 直流電源 11,12,13,14 主スイッチング素子 15,16,17,18 帰還ダイオード 19,20 トランス 21 共振インダクタンス 22 共振コンデンサ 23,24,25,26 整流用ダイオード 27 平滑コンデンサ 28 負荷抵抗 29,30,31,32 補助スイッチング素子 I1 トランス1次電流(共振電流) I1 ’バイパス共振電流 I2 帰還電流 T 1サイクル期間 TON ON幅期間 TOFF OFF幅期間 TA PWM ON幅期間 TB PWM OFF幅期間 T1 帰還電流期間 T2 休止期間Reference Signs List 10 DC power supply 11, 12, 13, 14 Main switching element 15, 16, 17, 18 Feedback diode 19, 20 Transformer 21 Resonant inductance 22 Resonant capacitor 23, 24, 25, 26 Rectifier diode 27 Smoothing capacitor 28 Load resistance 29, 30, 31 and 32 the auxiliary switching element I 1 transformer primary current (resonance current) I 1 'bypass resonance current I 2 feedback current T 1 cycle period T ON ON width period T OFF OFF width period T A PWM ON width period T B PWM OFF width period T 1 feedback current period T 2 idle period

───────────────────────────────────────────────────── フロントページの続き (56)参考文献 特開 平6−70543(JP,A) 特開 昭62−64259(JP,A) 特開 平1−202162(JP,A) 特開 平2−155467(JP,A) (58)調査した分野(Int.Cl.7,DB名) H02M 3/00 - 3/44 ──────────────────────────────────────────────────続 き Continuation of front page (56) References JP-A-6-70543 (JP, A) JP-A-62-64259 (JP, A) JP-A-1-202162 (JP, A) JP-A-2- 155467 (JP, A) (58) Field surveyed (Int. Cl. 7 , DB name) H02M 3/00-3/44

Claims (1)

(57)【特許請求の範囲】(57) [Claims] 【請求項1】 2個の主スイッチング素子が直列接続さ
れた、2組のブリッジ回路が直流電源に接続され、前記
2組のブリッジ回路の2個直列主スイッチング素子の接
続点には、各々トランスの一次巻線の一端が接続され、
前記各々のトランスの一次巻線の他端間に共振インダク
タンスと共振コンデンサの直列回路が接続され、前記共
振インダクタンスと共振コンデンサの直列回路の両端よ
り、前記2組のブリッジ回路毎の2個直列主スイッチン
グ素子に対応して、電源へエネルギーを帰還する方向
に、2個の帰還ダイオードの直列回路が、2組直流電源
に接続され、かつ前記各々のトランスの二次巻線に全波
整流回路と平滑コンデンサが接続され、負荷に電力を供
給する直列共振コンバータに於いて、前記2組の2個直
列帰還ダイオードの各々に、電源に順方向の補助スイッ
チング素子を並列接続し、前記主スイッチング素子の動
作周波数が可聴周波数領域より低い周波数の時は、前記
2組2個の補助スイッチング素子が、それぞれの主スイ
ッチング素子の導通期間内で導通幅制御(PWM)を行
う如く動作する様に構成された事を特徴とする直列共振
コンバータ。
1. Two sets of bridge circuits in which two main switching elements are connected in series are connected to a DC power supply, and a connection point of each of the two series main switching elements of the two sets of bridge circuits is connected to a transformer. One end of the primary winding is connected,
A series circuit of a resonance inductance and a resonance capacitor is connected between the other ends of the primary windings of the respective transformers, and two series main circuits for each of the two sets of bridge circuits are connected from both ends of the series circuit of the resonance inductance and the resonance capacitor. A series circuit of two feedback diodes is connected to two sets of DC power supplies in a direction of returning energy to the power supply corresponding to the switching element, and a full-wave rectifier circuit is connected to the secondary winding of each of the transformers. In a series resonance converter to which a smoothing capacitor is connected and supplies power to a load, a forward auxiliary switching element is connected in parallel to a power supply to each of the two sets of two series feedback diodes, When the operating frequency is lower than the audible frequency range, the two sets of two auxiliary switching elements are turned on by the respective main switching elements. Series resonant converter, characterized in that configured so as to operate as performing the conduction width control (PWM) in between.
JP30336493A 1993-11-09 1993-11-09 Series resonant converter Expired - Fee Related JP3266389B2 (en)

Priority Applications (1)

Application Number Priority Date Filing Date Title
JP30336493A JP3266389B2 (en) 1993-11-09 1993-11-09 Series resonant converter

Applications Claiming Priority (1)

Application Number Priority Date Filing Date Title
JP30336493A JP3266389B2 (en) 1993-11-09 1993-11-09 Series resonant converter

Publications (2)

Publication Number Publication Date
JPH07135769A JPH07135769A (en) 1995-05-23
JP3266389B2 true JP3266389B2 (en) 2002-03-18

Family

ID=17920106

Family Applications (1)

Application Number Title Priority Date Filing Date
JP30336493A Expired - Fee Related JP3266389B2 (en) 1993-11-09 1993-11-09 Series resonant converter

Country Status (1)

Country Link
JP (1) JP3266389B2 (en)

Families Citing this family (6)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN1992493B (en) * 2005-12-30 2011-05-18 艾默生网络能源系统北美公司 Resonance DC/DC converter and control method thereof
KR101288230B1 (en) * 2011-12-23 2013-07-24 명지대학교 산학협력단 Battery charging device for electric vehicle
JP5867476B2 (en) 2013-09-30 2016-02-24 サンケン電気株式会社 Current resonance type power supply
JP6007931B2 (en) 2014-03-06 2016-10-19 サンケン電気株式会社 Current resonance type power supply
JP6007935B2 (en) 2014-03-26 2016-10-19 サンケン電気株式会社 Current resonance type power supply
CN111786568A (en) * 2020-08-06 2020-10-16 石家庄通合电子科技股份有限公司 Bidirectional power converter, circuit and system

Also Published As

Publication number Publication date
JPH07135769A (en) 1995-05-23

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