JP3266389B2 - Series resonant converter - Google Patents
Series resonant converterInfo
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- JP3266389B2 JP3266389B2 JP30336493A JP30336493A JP3266389B2 JP 3266389 B2 JP3266389 B2 JP 3266389B2 JP 30336493 A JP30336493 A JP 30336493A JP 30336493 A JP30336493 A JP 30336493A JP 3266389 B2 JP3266389 B2 JP 3266389B2
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- series
- resonance
- current
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Description
【０００１】[0001]
【産業上の利用分野】本発明は、スイッチング電源等に
適用される直列共振コンバータに関するものである。BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a series resonance converter applied to a switching power supply or the like.
【０００２】[0002]
【従来の技術】従来の直列共振コンバータの回路構成の
一例を図３に示す。例えばＭＯＳＦＥＴなどの２個の主
スイッチング素子１１，１２が直列接続され１組のブリ
ッジ回路が構成され、別に２個の主スイッチング素子１
３，１４が直列接続され、もう１組のブリッジ回路が構
成され、各々のブリッジ回路が直流電源１０に接続され
て、いわゆるフルブリッジ回路が構成されている。2. Description of the Related Art FIG. 3 shows an example of a circuit configuration of a conventional series resonance converter. For example, two main switching elements 11 and 12 such as MOSFETs are connected in series to form a set of bridge circuits.
3 and 14 are connected in series to form another set of bridge circuits, and each bridge circuit is connected to the DC power supply 10 to form a so-called full bridge circuit.
【０００３】主スイッチング素子１１，１２の接続点
と、主スイッチング素子１３，１４の接続点間にそれぞ
れのトランス１９，２０の一次巻線の一端が接続され、
トランス１９，２０の他端間に共振インダクタンス２１
と共振コンデンサ２２が接続されている。One end of a primary winding of each of transformers 19 and 20 is connected between a connection point of main switching elements 11 and 12 and a connection point of main switching elements 13 and 14,
A resonance inductance 21 between the other ends of the transformers 19 and 20;
And the resonance capacitor 22 are connected.
【０００４】共振インダクタンス２１と共振コンデンサ
２２の両端には、２組のブリッジ回路に対応して、主ス
イッチング素子１１，１２及び１３，１４がＯＦＦの期
間エネルギーを直流電源１０に帰還する方向に、帰還ダ
イオード１５，１６の直列回路と帰還ダイオード１７，
１８の直列回路が直流電源１０に接続されている。[0004] Both ends of the resonance inductance 21 and the resonance capacitor 22 correspond to the two sets of bridge circuits, and return the energy to the DC power supply 10 while the main switching elements 11, 12, 13 and 14 are OFF. The series circuit of the feedback diodes 15, 16 and the feedback diode 17,
18 series circuits are connected to the DC power supply 10.
【０００５】又、トランス１９，２０の二次巻線にはそ
れぞれ整流用ダイオード２３，２４及び２５，２６が全
波整流回路として構成され、各々の出力が並列接続さ
れ、平滑コンデンサ２７及び負荷抵抗２８が接続されて
いる。The secondary windings of the transformers 19 and 20 are provided with rectifying diodes 23 and 24 and 25 and 26, respectively, as full-wave rectifier circuits, the outputs of which are connected in parallel, a smoothing capacitor 27 and a load resistor. 28 are connected.
【０００６】従来技術の直列共振コンバータは次の様な
動作をする。主スイッチング素子１１と主スイッチング
素子１４をターンオンさせると、直流電源１０−主スイ
ッチング素子１１−トランス１９の一次巻線−共振イン
ダクタンス２１−共振コンデンサ２２−トランス２０の
一次巻線−主スイッチング素子１４−直流電源１０のル
ープで図３の点線に示す様な共振電流Ｉ1 が流れる。The prior art series resonant converter operates as follows. When the main switching element 11 and the main switching element 14 are turned on, the DC power supply 10-the main switching element 11-the primary winding of the transformer 19-the resonance inductance 21-the resonance capacitor 22-the primary winding of the transformer 20-the main switching element 14- loop resonance current I 1 such as shown in dotted line in FIG. 3 flows in the DC power supply 10.
【０００７】この共振電流Ｉ1 は図４の様にＯＮ幅期間
ＴONの期間流れ続ける。共振電流Ｉ1 は、トランス１
９，２０のそれぞれの一次巻線と二次巻線との比をＮと
すると、それぞれのトランスの二次巻線に１／Ｎ倍の共
振電流となって流れるので、この共振電流を整流用ダイ
オード２３，２４及び２５，２６で整流し、平滑コンデ
ンサ２７を充電して直流電圧を負荷抵抗２８に供給す
る。The resonance current I 1 continues to flow during the ON width period T ON as shown in FIG. The resonance current I 1 is
Assuming that the ratio between the primary and secondary windings 9 and 20 is N, a resonance current of 1 / N times flows through the secondary windings of the respective transformers. The current is rectified by the diodes 23, 24 and 25, 26, and the smoothing capacitor 27 is charged to supply a DC voltage to the load resistor 28.
【０００８】又、主スイッチング素子１１，１４のＯＦ
Ｆの期間に於いては、共振コンデンサ２２に図示の極性
で充電された電荷が、図３の点線で示す帰還電流Ｉ2 と
なって電源に帰還され、その電流は図４の帰還電流期間
Ｔ1 の期間流れる。The OF of the main switching elements 11 and 14
Is In the period F, the electric charge charged in the shown polarity to the resonant capacitor 22, is fed back to the power supply becomes the feedback current I 2 indicated by the dotted line in FIG. 3, the feedback current period T of the current 4 Flows for one period.
【０００９】この方式では、主スイッチング素子１１，
１４のＯＮ幅は一定であり、ＯＦＦ幅、すなわち図４の
休止期間Ｔ2 の期間を可変させて出力の定電圧制御を行
う様に動作する。従って、軽負荷時は、ＯＦＦ期間Ｔ
OFF の内、休止期間Ｔ2 が長くなり、動作周波数が下が
り、やがて２０ＫＨＺ前後の可聴周波数領域となる。In this system, the main switching element 11,
ON width of 14 is constant, operated as to perform OFF width, ie the constant voltage control of the output by varying the duration of rest period T 2 of the FIG. Therefore, at light load, the OFF period T
Among OFF, the longer the idle period T 2, lower the operating frequency and eventually 20KHZ around audible frequency range.
【００１０】尚、以上は半サイクル動作についての説明
であるが、次の半サイクルは主スイッチング素子１２，
１３がＯＮする事によって上に述べた事と全く同じ動作
を行う。The above description is for a half cycle operation, but the next half cycle is for the main switching element 12,
When 13 is turned on, the same operation as described above is performed.
【００１１】[0011]
【発明が解決しようとする課題】しかしながら、従来の
技術には次の様な問題点が生ずる。すなわち、負荷の重
い時は、ＯＮ幅期間すなわちＴONは一定であるので、Ｏ
ＦＦ幅期間すなわちＴOF F の内、休止期間Ｔ2 の期間が
短く、従って、動作周波数が３０〜６０ＫＨＺで安定動
作を行っている。しかし、負荷が軽くなると、ＴONは一
定である為にＴOF F の内、休止期間Ｔ2 の期間が長くな
り、従って、動作周波数が２０ＫＨＺ程度まで下がり可
聴周波数領域で動作を行うようになる。However, the prior art has the following problems. That is, when the load is heavy, the ON width period, that is, T ON is constant.
Of FF width period or T OF F, duration of rest period T 2 is short, therefore, the operating frequency is performing stability with 30～60KHZ. However, when the load decreases, among T OF F for T ON is constant, the period of rest period T 2 becomes longer, therefore, the operating frequency is to perform the operation at audio frequency range down to about 20KHZ .
【００１２】この周波数領域になると動作時の発振音が
耳ざわりになり、騒音問題も引き起こす。又、無負荷に
なるに従ってＯＦＦ幅期間（ＴOFF ）が限りなく長くな
り、動作が不安定になるという欠点も生ずる。In this frequency range, an oscillating sound during operation becomes harsh and causes a noise problem. In addition, there is a disadvantage that the OFF width period (T OFF ) becomes infinitely longer as the load is reduced, and the operation becomes unstable.
【００１３】従って、上記の如く従来技術の有する問題
点を克服すべく、本発明の目的は次の通りである。すな
わち、直列共振コンバータのサージ電圧、電流によるノ
イズが小さいという特徴を備えたまま、軽負荷時には、
ＯＮ幅期間制御、すなわちＰＷＭ（PULSE WIDTH MODULA
TION）制御を行う動作を併用して、ＯＦＦ幅期間が必要
以上に長くならない様な動作を行う。この様にして、可
聴周波数領域にまで動作周波数が下がらない様にした直
列共振コンバータを提供することを目的とする。Accordingly, in order to overcome the problems of the prior art as described above, the objects of the present invention are as follows. In other words, at the time of light load, with the characteristic that the noise due to the surge voltage and current of the series resonance converter is small,
ON width period control, that is, PWM (PULSE WIDTH MODULA)
TION) An operation is performed so that the OFF width period does not become unnecessarily long together with the operation for performing the control. Thus, an object of the present invention is to provide a series resonant converter in which the operating frequency is not reduced to an audible frequency range.
【００１４】[0014]
【課題を解決するための手段】すなわち本発明は２個の
主スイッチング素子が直列接続された、２組のブリッジ
回路が直流電源に接続され、この２組のブリッジ回路の
２個直列主スイッチング素子の接続点には、各々トラン
スの一次巻線の一端が接続され、各々のトランスの一次
巻線の他端間に共振インダクタンスと共振コンデンサの
直列回路が接続されている。さらに共振インダクタンス
と共振コンデンサの直列回路の両端より、２組のブリッ
ジ回路毎の２個の直列主スイッチング素子に対応して、
電源へエネルギーを帰還する方向に、２個の帰還ダイオ
ードの直列回路が、２組直流電源に接続されている。
又、各々のトランスの二次巻線に全波整流回路と平滑コ
ンデンサが接続され、負荷に電力を供給する様に直列共
振コンバータが構成されている。That is, according to the present invention, two sets of bridge circuits in which two main switching elements are connected in series are connected to a DC power supply, and two series main switching elements of the two sets of bridge circuits are connected. Is connected to one end of a primary winding of each transformer, and a series circuit of a resonance inductance and a resonance capacitor is connected between the other ends of the primary windings of each transformer. Further, from both ends of the series circuit of the resonance inductance and the resonance capacitor, corresponding to two series main switching elements for each of two sets of bridge circuits,
A series circuit of two feedback diodes is connected to two sets of DC power supplies in a direction of returning energy to the power supply.
Further, a full-wave rectifier circuit and a smoothing capacitor are connected to the secondary winding of each transformer, and a series resonant converter is configured to supply power to a load.
【００１５】そしてこの直列共振コンバータの２組の２
個直列帰還ダイオードの各々に、電源に順方向の補助ス
イッチング素子を並列接続して、主スイッチング素子の
動作周波数が可聴周波数領域より低い周波数の時は、２
組２個の補助スイッチング素子が、それぞれの主スイッ
チング素子の導通期間内で導通幅制御（ＰＷＭ）を行う
如く動作する様に構成された事を特徴としている。The two series 2 of this series resonance converter
When a forward auxiliary switching element is connected in parallel to the power supply to each of the series feedback diodes, and the operating frequency of the main switching element is lower than the audible frequency range, 2
The two auxiliary switching elements are characterized in that they are configured to operate so as to perform conduction width control (PWM) within the conduction period of each main switching element.
【００１６】[0016]
【作用】この様な手段を有する直列共振コンバータは、
負荷の重い時は従来の回路と全く同じ様にＯＦＦ幅期間
制御を行うが、負荷が軽くなって動作周波数が可聴周波
数に近くなるまで下がると、ＯＮ幅期間制御（ＰＷＭ制
御）に切替わり可聴周波数まで下らない様な制御を行
う。The series resonant converter having such means is
When the load is heavy, the OFF width period control is performed in exactly the same way as with the conventional circuit. However, when the load becomes lighter and the operating frequency falls close to the audio frequency, it is switched to the ON width period control (PWM control), Control so that it does not fall to the frequency.
【００１７】すなわち軽負荷になると、主スイッチング
素子の導通期間内で、それぞれの主スイッチング素子に
対応する補助スイッチング素子が導通して、トランス１
次電流として流れていた共振電流を補助スイッチング素
子に転流させ、トランス１次電流を流さない様にする事
によってＯＮ幅期間制御を行うものである。That is, when the load becomes light, the auxiliary switching elements corresponding to the respective main switching elements conduct during the conduction period of the main switching elements, and the transformer 1
The ON width period control is performed by diverting the resonance current flowing as the secondary current to the auxiliary switching element so that the primary current of the transformer does not flow.
【００１８】[0018]
【実施例】図１はこの発明の一実施例を示し、従来技術
の図３と対応する部分は同一符号がつけてある。又、図
２はこの発明による主要部分の電流波形を示している。
この発明では、従来技術に対して補助スイッチング素子
２９，３０，３１，３２が追加され、直流電源１０に順
方向となる様にそれぞれの帰還ダイオード１５，１６，
１７，１８に並列接続されている。FIG. 1 shows an embodiment of the present invention, and portions corresponding to those of FIG. 3 of the prior art are denoted by the same reference numerals. FIG. 2 shows a current waveform of a main part according to the present invention.
According to the present invention, auxiliary switching elements 29, 30, 31, and 32 are added to the prior art, and respective feedback diodes 15, 16, and
17 and 18 are connected in parallel.
【００１９】この発明の直列共振コンバータの動作は次
の様である。負荷が重い時、すなわち負荷抵抗２８が小
さい時は、従来技術で述べた通り、主スイッチング素子
１１，１４のＯＮにより、図１の点線の様に共振電流Ｉ
1 を流す。この共振電流Ｉ1は、従来技術の図４と同様
半サイクル期間トランス１９，２０に流れ、トランス２
次巻線に変換されて負荷抵抗２８に負荷電流として流れ
る。The operation of the series resonant converter of the present invention is as follows. When the load is heavy, that is, when the load resistance 28 is small, as described in the related art, when the main switching elements 11 and 14 are turned on, the resonance current I as shown by the dotted line in FIG.
Pour 1 The resonance current I 1 flows in the same half cycle transformer 19, 20 and 4 of the prior art, trans-2
It is converted to the next winding and flows as a load current through the load resistor 28.
【００２０】そして半サイクル後、共振コンデンサ２２
にたくわえられた電荷が帰還電流Ｉ2 となって流れ、そ
の後休止期間Ｔ2 が継続されることは従来技術の図４の
波形と全く同じである。After a half cycle, the resonance capacitor 22
That the stored charge flows as the feedback current I 2 and then the rest period T 2 is continued is exactly the same as the waveform in FIG. 4 of the prior art.
【００２１】次に軽負荷となり負荷抵抗２８が大きくな
った時、定電圧制御により、ＯＦＦ幅期間ＴOFF の内、
休止期間Ｔ2 期間が長くなり動作周波数が下がる。そし
て２０ＫＨＺ程度の可聴周波数領域に近ずくと、ＯＮ幅
期間制御すなわちＰＷＭ（PULSE WIDTH MODULATION）制
御による定電圧制御に切替わる。Next, when the load becomes lighter and the load resistance 28 becomes larger, the constant voltage control is performed for the OFF width period T OFF .
Rest period period T 2 is the operating frequency is lowered long. Then, when approaching the audible frequency range of about 20 KHZ, the control is switched to the ON voltage period control, that is, the constant voltage control by the PWM (PULSE WIDTH MODULATION) control.
【００２２】このＰＷＭ制御の動作以後が本発明に係わ
る部分である。すなわち、可聴周波数領域に近づくと、
次の様な動作によりＰＷＭ制御が行われる。主スイッチ
ング素子１１，１４がＯＮしていて、共振電流であるト
ランス１次電流Ｉ1 が図１の点線の様に流れている期間
に、補助スイッチング素子２９，３２をターンオンさせ
ると、トランス１９及びトランス２０の一次巻線にはト
ランス１次電流Ｉ1 による一次巻線電圧が発生している
為、共振電流（トランス１次電流）Ｉ1 は主スイッチン
グ素子から補助スイッチング素子側に転流する。The portion after the operation of the PWM control is a portion related to the present invention. That is, when approaching the audible frequency range,
PWM control is performed by the following operation. The main switching element 11, 14 have turned ON, the transformer primary current I 1 is resonant current during flowing like a dotted line in FIG. 1, to turn on the auxiliary switching element 29 and 32, transformer 19 and since the primary winding of the transformer 20, which primary winding voltage by the transformer primary current I 1 is generated, the resonant current (transformer primary current) I 1 is commutated from the main switching element in the auxiliary switching element side.
【００２３】すなわち図２の様にトランス１次電流Ｉ1
がＰＷＭ ＯＮ幅期間ＴA の期間流れている時に補助ス
イッチング素子２９，３２をＯＮさせると、トランス１
次電流Ｉ1 は直流電源１０−補助スイッチング素子２９
−共振インダクタンス２１−共振コンデンサ２２−補助
スイッチング素子３２−直流電源１０のループに転流さ
れ、図１の一点鎖線の様にバイパス共振電流Ｉ1 ’に切
替わり共振電流を流し続ける。That is, as shown in FIG. 2, the transformer primary current I 1
When There is ON the auxiliary switching element 29 and 32 when the flowing period of the PWM ON width period T A, the transformer 1
The secondary current I 1 is supplied from the DC power supply 10
- commutated loop of the resonant inductance 21 resonance capacitor 22-auxiliary switching element 32 DC power supply 10, continues to flow switching instead resonance current to bypass the resonance current I 1 'as a one-dot chain line in FIG. 1.
【００２４】このバイパス共振電流Ｉ1 ’はトランス１
９，２０の一次巻線には流れない為、図２の様にＰＷＭ
ＯＦＦ幅期間ＴB の期間トランス１次電流Ｉ1 は零と
なり、バイパス共振電流Ｉ1 ’がバイパスループに流れ
るので、共振電流としては半サイクルの期間連続的に流
し続けることになる。This bypass resonance current I 1 ′ is
Since the current does not flow through the primary windings 9 and 20, as shown in FIG.
OFF width period T period transformer primary current I 1 becomes zero in B, since the bypass resonance current I 1 'flows through the bypass loop, it will continue to flow in continuous of the half-cycle period as the resonance current.
【００２５】従って、共振電流は連続的に流し続けなが
ら、トランス１９，トランス２０の一次巻線には、補助
スイッチング素子をＯＮさせる事によって、必要な期間
のみ共振電流（トランス１次電流）Ｉ1 を流すことが出
来るので、結果的にＯＮ幅期間ＴONが制御されて定電圧
制御が行われることになる。Accordingly, the resonance current (transformer primary current) I 1 is only provided for a necessary period by turning on the auxiliary switching element in the primary winding of the transformer 19 and the transformer 20 while the resonance current is continuously flowing. As a result, the ON width period T ON is controlled and constant voltage control is performed.
【００２６】共振電流が流れ終わった後、共振コンデン
サ２２にたくわえられた電荷が帰還電流Ｉ2 となって、
帰還電流期間Ｔ1 の期間電源に帰還されるモード、及び
休止期間Ｔ2 のモードについては従来技術と全く同じで
ある。After the end of the resonance current, the charge stored in the resonance capacitor 22 becomes the feedback current I 2 ,
The mode in which the power is fed back to the power supply during the feedback current period T 1 and the mode during the idle period T 2 are exactly the same as those in the related art.
【００２７】従って、図２に於いて、負荷の重い時のＯ
Ｎ幅期間はＴA ＋ＴB であるので、動作責務（ＤＵＴ
Ｙ）は（ＴA ＋ＴB ）／Ｔであり、軽負荷時のＯＮ幅期
間はＴA であるので動作責務（ＤＵＴＹ）はＴA ／Ｔと
なる。この様にして重負荷時は周波数制御を行い、軽負
荷時はＰＷＭ制御を行う事によって、常に安定的に定電
圧制御を行う事が出来るものである。Accordingly, in FIG. 2, when the load is heavy, O
Since the N width period is T A + T B , the operation duty (DUT
Y) is (T A + T B) / T, operating duty since ON width period of the light load is a TA (DUTY) becomes T A / T. In this way, the frequency control is performed at the time of heavy load, and the PWM control is performed at the time of light load, so that the constant voltage control can always be stably performed.
【００２８】又、図示しないがトランス１９及びトラン
ス２０に固有するリーケージインダクタンス、又は別に
トランスと直列に設けた共振インダクタンス２１、ある
いは共振インダクタンス２１から帰還回路に直列に新た
な共振インダクタンス２１を設ける事により、転流時の
トラブルを防ぐことが出来る。例えば、共振電流Ｉ1か
ら帰還電流Ｉ2 に転流する時、帰還ダイオードのリカバ
リータイムの間、短絡電流の発生を防止したり、各転流
時に一定時間かけて切替わる為に発生するスイッチング
ロスやスイッチングノイズを上に述べたインダクタンス
を設ける事により防止する事が出来る。Although not shown, a leakage inductance inherent to the transformer 19 and the transformer 20, a resonance inductance 21 separately provided in series with the transformer, or a new resonance inductance 21 provided in series from the resonance inductance 21 to a feedback circuit is provided. Therefore, trouble during commutation can be prevented. For example, when commutating from the resonance current I 1 to the feedback current I 2 , during the recovery time of the feedback diode, a short-circuit current is prevented from occurring, or a switching loss occurs due to switching over a certain time at each commutation. And switching noise can be prevented by providing the above-described inductance.
【００２９】[0029]
【発明の効果】本発明は以上説明した様な構成である
為、以下に示すような効果を有する。すなわち、重負荷
時の周波数制御、軽負荷時のＰＷＭ制御を併用する事に
よって、全負荷から無負荷までの全領域に於いて、動作
周波数が低くなる事による不安定動作を防止したり、
又、可聴周波数での動作領域をさけている為、騒音防止
の効果も呈する。さらに転流電流の急峻な切替わりによ
るスイッチングロスやスイッチングノイズの発生を防止
出来る従来の直列共振コンバータの特徴を失う事もな
く、本発明回路は実現出来るので、より有用なスイッチ
ング電源を提供し得るものである。Since the present invention has the configuration as described above, it has the following effects. That is, by using the frequency control at the time of heavy load and the PWM control at the time of light load together, it is possible to prevent the unstable operation due to the lowering of the operating frequency in the entire region from full load to no load,
In addition, since the operating region at the audible frequency is avoided, the effect of preventing noise is exhibited. Further, since the circuit of the present invention can be realized without losing the features of the conventional series resonant converter that can prevent the occurrence of switching loss and switching noise due to abrupt switching of the commutation current, a more useful switching power supply can be provided. Things.
【図１】本発明の直列共振コンバータ。FIG. 1 is a series resonance converter of the present invention.
【図２】本発明の主要部電流波形。FIG. 2 is a main part current waveform of the present invention.
【図３】従来の直列共振コンバータ。FIG. 3 shows a conventional series resonance converter.
【図４】従来型の主要部電流波形。FIG. 4 is a conventional main part current waveform.
１０ 直流電源 １１，１２，１３，１４ 主スイッチング素子 １５，１６，１７，１８ 帰還ダイオード １９，２０ トランス ２１ 共振インダクタンス ２２ 共振コンデンサ ２３，２４，２５，２６ 整流用ダイオード ２７ 平滑コンデンサ ２８ 負荷抵抗 ２９，３０，３１，３２ 補助スイッチング素子 Ｉ1 トランス１次電流（共振電流） Ｉ1 ’バイパス共振電流 Ｉ2 帰還電流 Ｔ １サイクル期間 ＴON ＯＮ幅期間 ＴOFF ＯＦＦ幅期間 ＴA ＰＷＭ ＯＮ幅期間 ＴB ＰＷＭ ＯＦＦ幅期間 Ｔ1 帰還電流期間 Ｔ2 休止期間Reference Signs List 10 DC power supply 11, 12, 13, 14 Main switching element 15, 16, 17, 18 Feedback diode 19, 20 Transformer 21 Resonant inductance 22 Resonant capacitor 23, 24, 25, 26 Rectifier diode 27 Smoothing capacitor 28 Load resistance 29, 30, 31 and 32 the auxiliary switching element I 1 transformer primary current (resonance current) I 1 'bypass resonance current I 2 feedback current T 1 cycle period T ON ON width period T OFF OFF width period T A PWM ON width period T B PWM OFF width period T 1 feedback current period T 2 idle period
───────────────────────────────────────────────────── フロントページの続き (56)参考文献 特開 平６−70543（ＪＰ，Ａ) 特開 昭62−64259（ＪＰ，Ａ) 特開 平１−202162（ＪＰ，Ａ) 特開 平２−155467（ＪＰ，Ａ) (58)調査した分野(Int.Cl.7，ＤＢ名) H02M 3/00 - 3/44 ──────────────────────────────────────────────────続 き Continuation of front page (56) References JP-A-6-70543 (JP, A) JP-A-62-64259 (JP, A) JP-A-1-202162 (JP, A) JP-A-2- 155467 (JP, A) (58) Field surveyed (Int. Cl. 7 , DB name) H02M 3/00-3/44
Claims (1)
れた、２組のブリッジ回路が直流電源に接続され、前記
２組のブリッジ回路の２個直列主スイッチング素子の接
続点には、各々トランスの一次巻線の一端が接続され、
前記各々のトランスの一次巻線の他端間に共振インダク
タンスと共振コンデンサの直列回路が接続され、前記共
振インダクタンスと共振コンデンサの直列回路の両端よ
り、前記２組のブリッジ回路毎の２個直列主スイッチン
グ素子に対応して、電源へエネルギーを帰還する方向
に、２個の帰還ダイオードの直列回路が、２組直流電源
に接続され、かつ前記各々のトランスの二次巻線に全波
整流回路と平滑コンデンサが接続され、負荷に電力を供
給する直列共振コンバータに於いて、前記２組の２個直
列帰還ダイオードの各々に、電源に順方向の補助スイッ
チング素子を並列接続し、前記主スイッチング素子の動
作周波数が可聴周波数領域より低い周波数の時は、前記
２組２個の補助スイッチング素子が、それぞれの主スイ
ッチング素子の導通期間内で導通幅制御（ＰＷＭ）を行
う如く動作する様に構成された事を特徴とする直列共振
コンバータ。1. Two sets of bridge circuits in which two main switching elements are connected in series are connected to a DC power supply, and a connection point of each of the two series main switching elements of the two sets of bridge circuits is connected to a transformer. One end of the primary winding is connected,
A series circuit of a resonance inductance and a resonance capacitor is connected between the other ends of the primary windings of the respective transformers, and two series main circuits for each of the two sets of bridge circuits are connected from both ends of the series circuit of the resonance inductance and the resonance capacitor. A series circuit of two feedback diodes is connected to two sets of DC power supplies in a direction of returning energy to the power supply corresponding to the switching element, and a full-wave rectifier circuit is connected to the secondary winding of each of the transformers. In a series resonance converter to which a smoothing capacitor is connected and supplies power to a load, a forward auxiliary switching element is connected in parallel to a power supply to each of the two sets of two series feedback diodes, When the operating frequency is lower than the audible frequency range, the two sets of two auxiliary switching elements are turned on by the respective main switching elements. Series resonant converter, characterized in that configured so as to operate as performing the conduction width control (PWM) in between.
Priority Applications (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| JP30336493A JP3266389B2 (en) | 1993-11-09 | 1993-11-09 | Series resonant converter |
Applications Claiming Priority (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| JP30336493A JP3266389B2 (en) | 1993-11-09 | 1993-11-09 | Series resonant converter |
Publications (2)
| Publication Number | Publication Date |
|---|---|
| JPH07135769A JPH07135769A (en) | 1995-05-23 |
| JP3266389B2 true JP3266389B2 (en) | 2002-03-18 |
Family
ID=17920106
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| JP30336493A Expired - Fee Related JP3266389B2 (en) | 1993-11-09 | 1993-11-09 | Series resonant converter |
Country Status (1)
| Country | Link |
|---|---|
| JP (1) | JP3266389B2 (en) |
Families Citing this family (6)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN1992493B (en) * | 2005-12-30 | 2011-05-18 | 艾默生网络能源系统北美公司 | Resonance DC/DC converter and control method thereof |
| KR101288230B1 (en) * | 2011-12-23 | 2013-07-24 | 명지대학교 산학협력단 | Battery charging device for electric vehicle |
| JP5867476B2 (en) | 2013-09-30 | 2016-02-24 | サンケン電気株式会社 | Current resonance type power supply |
| JP6007931B2 (en) | 2014-03-06 | 2016-10-19 | サンケン電気株式会社 | Current resonance type power supply |
| JP6007935B2 (en) | 2014-03-26 | 2016-10-19 | サンケン電気株式会社 | Current resonance type power supply |
| CN111786568A (en) * | 2020-08-06 | 2020-10-16 | 石家庄通合电子科技股份有限公司 | Bidirectional power converter, circuit and system |
-
1993
- 1993-11-09 JP JP30336493A patent/JP3266389B2/en not_active Expired - Fee Related
Also Published As
| Publication number | Publication date |
|---|---|
| JPH07135769A (en) | 1995-05-23 |
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