JP3480283B2 - Power supply - Google Patents

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a power supply device for controlling load power by a high frequency inverter.

[0002]

2. Description of the Related Art FIG. 31 is a circuit diagram of a conventional example (Japanese Patent Application No. 7-35740). In this circuit, a series circuit of a DC power supply 1 and a capacitor Co and a series circuit of switching elements Qa and Qb are connected in parallel, and the DC power supply 1 and the capacitor Co are connected.
Is connected to one end of the primary winding of the transformer Tf, and the connection point of the switching elements Qa and Qb is 1 of the transformer Tf.
It is formed by connecting the other end of the secondary winding and the load circuit 7a to the secondary winding of the transformer Tf.
By alternately turning on and off a and Qb, the DC power supply 1 is converted into high frequency power and supplied to the load circuit 7a. Diodes Da and Db are connected in antiparallel to the switching elements Qa and Qb, respectively. FIG. 32 shows an example of the load circuit 7a.

The operation of the high frequency inverter 2 of FIG. 31 will be described. First, when the switching element Qb is on, a current flows through the path of the DC power supply 1, the primary winding Lo of the transformer Tf, and the switching element Qb, and the transformer is turned on. Power is supplied to the secondary side of Tf. When the switching element Qb is off, a portion of the magnetic energy stored in the transformer Tf through the path of the primary winding Lo of the transformer Tf, the diode Da, and the capacitor Co and the magnetic energy on the secondary side is stored in the capacitor Co.
Move to. Next, when the switching element Qa is turned on, a current flows through the path of the capacitor Co, the switching element Qa, and the primary winding Lo of the transformer Tf, and the current of the transformer Tf becomes 2
Supply power to the secondary side. When the switching element Qa is off, a regenerative current flows in the path of the primary winding Lo of the transformer Tf, the DC power supply 1, and the diode Db to regenerate a part of the magnetic energy of the circuit in the DC power supply 1. In this way, a high-frequency alternating current flows through the primary winding Lo of the transformer Tf, the voltage is boosted by the transformer Tf, and the necessary power is supplied to the load circuit 7a on the secondary side. The switching elements Qa and Qb are alternately turned on and off, and the power is adjusted by the on duty. The power supplied to the load circuit 7a increases as the ON duty of the switching element Qb increases.

FIG. 33 shows a configuration in which a discharge lamp such as a metal halide lamp is used as the load. A diode bridge DB1 is connected to the secondary side of the transformer Tf via a current limiting inductor L2, and the rectified output is connected to a capacitor C.
After smoothing with 2 to obtain a DC voltage, it is converted into a low frequency rectangular wave voltage by the inverter circuit 4 of the full bridge configuration,
Electric power is supplied to the discharge lamp 5 which is a load. The same operation is performed in a circuit having a high frequency inverter circuit 2a as shown in FIG. 34 (Japanese Patent Application No. 7-35736).

In the above circuit, when the load impedance is low and the output is large, when the switching element Qb changes from on to off, the inductor L2 and the transformer T are turned on.
Before the leakage inductance and the excitation energy of f are completely discharged to the load and the capacitor Co, the switching element Q
b may turn on again. As a result, the DC bias of the transformer Tf becomes excessive and may be saturated.

To solve this problem, the charging capacitor Cx in the rectifying / smoothing circuit loop of at least one of the polarities of the voltage generated on the secondary side of the transformer Tf.
The capacitance of is smaller than the smoothing capacitor, especially
The present inventors have proposed to prevent saturation by setting the off time Tboff of the switching element Qb to a predetermined constant such that Tboff ≧ π√ (CxL2) / 2. A specific example of this configuration is shown in FIG.

In the circuit of FIG. 35, with one polarity on the secondary side of the transformer Tf, the secondary side output of the transformer Tf is charged in the capacitor Cx having a small capacitance via the current limiting inductor L2 and the diode D2b, and the other side. Transformer T in polarity
When the charge of the capacitor Cx exists, the secondary side output of f is the inductor L2, the capacitor Cx, and the diode D.
Power is sent to the smoothing capacitor C2 and the load while discharging the capacitor Cx via 2a, and when the capacitor Cx is completely discharged, it is directly output to the load side via the inductor L2. That is, since the clamping diode D2c is connected to the capacitor Cx, when the voltage of the capacitor Cx becomes zero, the secondary side output is the diode D2c,
The smoothing capacitor C2 is charged through D2a, and the load 5
Power is supplied to a. The capacitor C2 is a smoothing capacitor that removes ripples.

As a result, with the secondary side output polarity that charges the capacitor Cx, the discharge time of the energy stored in the transformer Tf and the inductor L2 can be shortened and the saturation of the core due to excessive DC bias of the transformer Tf. Can be prevented. Further, when the on-duty of the switching element Qb in which the power supply voltage is directly applied to the primary side is large, saturation of the core of the transformer Tf easily occurs, so that the voltage of the capacitor Co is applied to the primary side in the switching operation. Setting the polarity of the transformer Tf so that the capacitor Cx is charged when the element Qa is on is more effective in suppressing saturation of the transformer Tf.

Further, in this circuit, the capacitor Cx is first discharged, and in the polarity in which power is supplied to the smoothing capacitor C2, the capacitor Cx is first discharged. This is because the current rises due to the resonance action of the capacitor Cx and the inductor L2, so under the condition that the capacitor Cx is set small, the current rises quickly, the current peak can be reduced, and the efficiency can be improved. Further, since it is basically in the form of a voltage doubler rectifier circuit, the step-up ratio of the transformer Tf can be reduced and the current peak on the primary side can also be lowered, so that efficiency can be further improved.

[0010]

As described above, FIG.
In the above circuit, in the rectifying / smoothing circuit provided on the secondary side of the transformer Tf, at least one of the voltage polarities generated on the secondary side of the transformer Tf has a closed circuit that only charges the small capacity capacitor Cx. 2 of the transformer Tf when the charge of the capacitor Cx is discharged to the load side.
If the current polarity when discharging the secondary side and discharging the charge is opposite to the polarity of charging the capacitor Cx from the transformer Tf, the secondary side rectifying / smoothing capacitor Cx is charged. When the primary side switching element Qa that generates a voltage having a polarity
Since the capacitance of x is small, after the switching element Qa is turned on and charging of the capacitor Cx is completed, the charging voltage to the capacitor Cx becomes higher than the voltage obtained by adding the secondary voltage and the load voltage due to the resonance action with the inductor L2. When it becomes, the electric charge of the capacitor Cx starts discharging to the load side,
A current flows in the secondary winding n2 of the transformer Tf in a direction opposite to the generated voltage polarity, and at the same time, a reverse current flows in the primary winding n1 and the switching element Qa. Even if the switching element Qa is turned off in this state, the switching element Qa
The current continues to flow in the diode Da connected in anti-parallel to the diode Da, and if the other switching element Qb is turned on at this point, the capacitors Da and Da acting as voltage sources during the reverse recovery time of the diode Da to the diode Da, A switching phenomenon occurs in a phase advancing state in which a surge current flows through the switching element Qb.

The present invention has been made in view of the above circumstances, and an object of the present invention is to provide a DC-DC conversion circuit including a high frequency inverter, a step-up transformer, a current limiting element, and a rectifying / smoothing circuit. In the power supply device including the transformer, even if only a small-capacity capacitor is charged in at least one of the secondary side polarities of the transformer, the switching timing of the primary side switching element is not advanced. To prevent surge current.

[0012]

According to the present invention, in order to solve the above-mentioned problems, as shown in FIG. 1, there are provided first and second switching elements each having reverse conduction elements in parallel. a circuit connected in series so the forward direction coincides, a DC power supply and the first capacitor connecting the circuit connected in series to the parallel connection point between the first and second direct-current power supply and the first capacitor connect the transformer primary winding between the connection point of the second switching element, the second co <br/> capacitor for smoothing through the inductor and the rectifying circuit current limiting on the transformer secondary windings A rectifier circuit and a rectifier circuit, which are connected to each other and have a power circuit configuration for supplying the power obtained by the second capacitor to a load .
In a power supply device in which a third capacitor having a smaller capacity than the second capacitor is connected between the second capacitor and
At least the polarity of the voltage generated by the secondary winding of the transformer
Also, with one polarity, only charging the third capacitor
Has a closed circuit and discharges the charge of the third capacitor to the load side
Is discharged through the secondary winding of the transformer when
The current polarity when discharging the
Configuration that is opposite to the polarity of charging the capacitor of No. 3
And at least immediately before the switching element on the primary side of the transformer is turned off, a positive current is superimposed on the switching element. Specifically, the transformer Tf has a tertiary winding n3.
Is connected to the circuit 20 for supplying the lagging current. Further, as in the embodiment of FIGS. 11 to 16, a power source or current limiting element for canceling the reverse current is provided on the primary side or the secondary side of the transformer immediately before the switching element on the primary side of the transformer is turned off. You may connect in parallel.

Alternatively, in the embodiment of FIGS. 17-19,
While the primary side switching element that charges the capacitor Cx is turned on, the capacitor Cx is prevented from discharging, and therefore, a switching device is connected on the discharge path from the capacitor Cx to the load side. .

Further, as in the embodiment of FIG. 20, the primary side switching element Qa having the charging polarity to the capacitor Cx.
There is also a means to make the ON period of the capacitor shorter than the maximum charging time of the capacitor Cx. Alternatively, in the charging polarity of the capacitor Cx, when discharging is started from the capacitor Cx, the ON period is set such that the reverse current of the secondary winding of the transformer Tf due to the discharging current becomes zero.
That is, the primary side switching element Qa that charges the capacitor Cx is operated at a predetermined value that satisfies the above condition during the ON state.

Alternatively, since the voltage at the completion of charging of the capacitor Cx is high and discharge with the reverse polarity voltage as described above occurs, the voltage at the completion of charging of the capacitor Cx may be reduced. Alternatively, by inserting an inductance element that lowers the resonance frequency in the charging path of the capacitor Cx, even if the ON time of the switching element is an arbitrary value, the capacitor Cx is prevented from discharging within the charging polarity period. But good.

Furthermore, the charging time to the capacitor Cx is shortened by reducing the current limiting element in the charging path, or the charging / discharging time of the capacitor Cx is shortened by reducing the current limiting element in the charging / discharging path. Transformer Tf
The time during which the current flows in the opposite direction to the secondary side output is shortened, and the primary side switching element is operated for a longer ON time.

[0017]

DESCRIPTION OF THE PREFERRED EMBODIMENTS (Embodiment 1) FIG. 1 shows a first embodiment of the present invention.
An example of is shown. In this embodiment, in the conventional circuit configuration shown in FIG. 35, a transformer Tf is provided with a tertiary winding n3 and a circuit 20 for supplying a lag current is connected. In the circuit configuration of FIG. 1, since the capacitor Cx is charged when the switching element Qa is on, the current continues to flow in the opposite direction while the switching element Qa is on.
There is a possibility of switching in a phase advance state in which the switching element Qb is turned on. Therefore, at least immediately before the switching element Qa is turned off, the transformer T is set so that the current flowing through the switching element Qa is in the positive direction.
A circuit 20 for supplying a lagging phase current that cancels the leading phase current of the primary winding is provided in the tertiary winding n3 of f.

FIG. 2 shows an example of the lagging current circuit 20 connected to the tertiary winding n3 of the transformer Tf. The configuration of FIG. 2 includes a DC power supply 1 ′, an inductor L3, a diode Dc, and a switching element Qc, and the switching element Qc is switched in conjunction with the switching element Qa. As a result, when the switching element Qa is turned on, the switching element Qc is turned on and a lagging current flows through the tertiary winding n3. The number of turns of the tertiary winding n3 and the voltage of the DC power supply 1 ′ are set so that the lagging current has a current value that cancels the reverse current flowing in the switching element Qa (or the diode Da) immediately before the switching element Qa is turned off. The inductance value of the inductor L3 is designed to be a predetermined value.

Another example of the lagging current circuit 20 is shown in FIG. In the circuit of FIG. 3, in order to prevent a phase-advanced state caused by the regeneration of power from the secondary side of the transformer Tf to the primary side due to the charging voltage of the capacitor Cx, regeneration from the secondary side of the transformer Tf is performed. By sending the generated energy to the tertiary winding n3 instead of the primary winding n1, the reverse current flowing in the primary winding n1 is prevented. That is, with the switching element Qa turned on, the switching element Qa
When the current flowing in the opposite direction is reversed, the switching element Qc
Is turned on, a regenerative current is caused to flow to the side of the tertiary winding n3, and energy is stored in the inductor L3. Then, when the switching element Qa is turned off and the switching element Qc is also turned off at the same time, a lag current flows through the primary side of the transformer Tf, and the energy of the inductor L3 is a DC power supply 1 ′.
Sent to. The DC power supply 1 ′ may be the same as the DC power supply 1.

Another example of the lagging phase current circuit 20 is shown in FIG.
Shown in. This is because the inductor L3 in the circuit of FIG.
Instead of sending the energy stored in the DC power supply 1 ',
Energy is stored in the capacitor Cc, and the energy stored in the capacitor Cc is used as a power source for the control circuit and the drive circuit.

(Embodiment 2) FIG. 5 shows a second embodiment of the present invention. In this embodiment, in the conventional circuit configuration shown in FIG. 35, a transformer Tf is provided with a tertiary winding n3 and is connected so as to form a closed loop via a diode Dc and a switching element Qa. Explaining the switching phenomenon in the phase advance state, when the switching element Qa is on, the voltage charged in the small capacity capacitor Cx is equal to the voltage of the secondary winding n2 of the transformer Tf and the smoothing capacitor C2.
When it becomes higher than the superposed voltage with the voltage of
current flows in the direction opposite to the voltage polarity of the secondary winding n2 of f,
Current also flows in the reverse direction in the primary winding n1 of the transformer Tf,
As a result, a reverse current flows through the switching element Qa. In this state, even if the switching element Qa is turned off, the current continues to flow in the diode Da, and if the switching element Qb is turned on, the reverse recovery time of the diode Da
A short circuit current surge occurs in the loop of the capacitor Cs, the capacitor Co, the diode Da, and the switching element Qb. That is, the fact that the current does not always flow in the positive direction while the switching element Qa is on leads to the switching phenomenon in the phase advance state.

In the configuration of FIG. 5, when the switching element Qa is turned on and a voltage is applied to the primary winding n1 of the transformer Tf, a voltage is also generated in the tertiary winding n3 by electromagnetic coupling. If the winding direction is set so that this voltage is in the forward direction of the diode Dc, the tertiary winding n3, the diode Dc,
A current flows through the switching element Qa. If this current value is set to be equal to or more than the reverse current value when the switching element Qa is turned off, a current always flows in the forward direction in the switching element Qa while the switching element Qa is on.

In the configuration of FIG. 5, the tertiary winding n
3, a current limiting inductor L3 may be connected in the closed loop of the diode Dc and the switching element Qa, as shown in FIG. Further, the leakage inductance of the tertiary winding n3 itself may be used as the current limiting inductor L3.

(Embodiment 3) FIG. 7 shows a third embodiment.
In this embodiment, similar to the circuit shown in FIG. 5, the transformer Tf is provided with the tertiary winding n3 and one end thereof is connected to one end of the switching element Qa through the diode Dc. Compared with the third winding n of the transformer Tf
The connection point of the other end of 3 is different. That is, in the present embodiment, in the conventional circuit configuration shown in FIG. 35, the transformer Tf is provided with the tertiary winding n3 and is connected so as to form a closed loop with the capacitor Co via the diode Dc. The capacitor Co is used as a voltage source of the primary winding n1 of the transformer Tf when the switching element Qa is turned on. The number of turns of the tertiary winding n3 is 1
If the number of turns of the secondary winding n1 is set to a value larger than the number of turns of the secondary winding n1, when the switching element Qa is on and energy is regenerated from the secondary side to the primary side, a current flows from the primary winding n1 to the tertiary winding n3. Since energy flows from the tertiary winding n3 to the capacitor Co, no reverse current flows in the switching element Qa. Therefore, switching does not occur in the advanced phase.

In the configuration of FIG. 7, the tertiary winding n
3, in the closed loop of the diode Dc, the capacitor Co,
As shown in FIG. 8, a current limiting inductor L3 may be connected. This is because the number of turns of the tertiary winding n3 is larger than the number of turns of the primary winding n1 when the switching element Qa is on, so that a current constantly flows, so that this current is limited. Inductor L3 and tertiary winding n3
Is a predetermined value that prevents the reverse current of the switching element Qa immediately before the switching element Qa is turned off.

(Embodiment 4) FIG. 9 shows a fourth embodiment.
Also in this embodiment, similarly to the circuit shown in FIG. 5, the transformer Tf is provided with the tertiary winding n3 and one end thereof is connected to one end of the switching element Qa via the diode Dc. Compared with the third winding n of the transformer Tf
The connection point of the other end of 3 is different. That is, in this embodiment, in the conventional circuit configuration shown in FIG. 35, the transformer Tf is provided with the tertiary winding n3, and a closed loop with the capacitor Co and the capacitor Cs (and the DC power supply 1) is formed via the diode Dc. Are connected as described. Tertiary winding n
The winding direction of 3 is a direction in which a forward voltage is generated with respect to the diode Dc when the switching element Qa that generates a voltage of the polarity that charges the capacitor Cx is turned on.

In the present embodiment, the number of turns of the tertiary winding n3 is made larger than that of the primary winding n1 so that the charging voltage of the capacitor Cx is equal to the voltage of the secondary winding n2 when the switching element Qa is in the ON state. The reverse direction that flows from the primary winding n1 to the switching element Qa by flowing the regenerative current that is higher than the superimposed voltage with the voltage of C2 and returns to the primary side of the transformer Tf to the side of the tertiary winding n3 Prevent current,
Tertiary winding n to avoid switching in the advanced phase
3 is a predetermined number of turns.

Further, the charging current of the capacitor Cx increases and the reverse current flows in the switching element Qa because the switching element Qb for applying the voltage of the DC power supply 1 to the primary winding n1 has a certain on-duty range. When it exceeds. The voltage Vco of the capacitor Co is related to the on-duty D of the switching element Qb,
In, if the voltage of the DC power supply 1 is Vin, Vco = V
in.D / (1-D). If the on-duty D of the switching element Qb is not more than a certain value, the reverse current flowing through the switching element Qa does not occur. Therefore, when the on-duty D of the switching element Qb is small,
It is not necessary to pass a current through the tertiary winding n3.

Therefore, in the present embodiment, the number of turns of the tertiary winding n3 is made larger than that of the primary winding n1, and the voltage generated in the tertiary winding n3 is equal to or higher than a predetermined on-duty and the voltage is generated in the capacitor. The predetermined number of turns is higher than the superimposed voltage of Co and the voltage of the capacitor Cs (DC power supply 1). This makes it possible to automatically stop the current by the commutation action of the diode Dc when it is not necessary to pass the current through the tertiary winding n3.

In the configuration of FIG. 9, the tertiary winding n
As shown in FIG. 10, a current limiting inductor L3 may be connected in the closed loop of the diode 3, the diode Dc, and the capacitors Co and Cs. This limits the current value of the three-character winding n3. The current limiting inductor L3 and the tertiary winding n3 have predetermined values so as to prevent the reverse current of the switching element Qa immediately before the switching element Qa is turned off. It is efficient to set the current limiting inductor L3 to a predetermined inductance value such that the minimum current required to prevent the reverse current of the switching element Qa flows.

(Embodiment 5) FIG. 11 shows a fifth embodiment. In this embodiment, in the conventional circuit configuration shown in FIG. 35, the power supply 1b is connected in parallel with the primary winding n1 of the transformer Tf.
Is connected. In the power supply 1b, when the switching element Qa is turned on, the charging voltage to the capacitor Cx is high and a current flows in the direction opposite to the voltage polarity of the secondary winding n2, so that a reverse current flows in the switching element Qa. It is a power source for supplying a predetermined current that cancels the reverse current in order to prevent the current from flowing, and is a current source in the example of FIG. 11. This current source may be a constant current source that flows the above-described predetermined current, or may flow a current that cancels the reverse current of the switching element Qa only immediately before the switching element Qa is turned off. Further, as shown in FIG. 12, the power supply 1b may be configured as a series circuit of a voltage source and a resistor.

(Embodiment 6) FIG. 13 shows a sixth embodiment. In this embodiment, the inductor L4 is connected in parallel with the primary winding n1 of the transformer Tf in the conventional circuit configuration shown in FIG. This is a circuit in which a lagging current for canceling a reverse current flowing for the same reason as described above is provided in parallel with the primary winding n1 when the switching element Qa is turned on. Then, the inductor L4 constitutes a circuit for obtaining the delayed phase current. The inductor L4 has a predetermined value that cancels the reverse current of the switching element Qa.

As shown in FIG. 14, the inductor L
You may connect the diode Dc in series with 4. this is,
The lagging current due to the inductor L4 is the switching element Q.
Since it is not necessary when b is on, the switching element Q
Only when a is on, a current is passed from the capacitor Co to the inductor L4 to obtain a lagging current.

(Embodiment 7) FIG. 15 shows a seventh embodiment. In this embodiment, the inductor L5 is connected in parallel with the secondary winding n2 of the transformer Tf in the conventional circuit configuration shown in FIG. This is a circuit in which a lagging current for canceling a reverse current flowing for the same reason as described above is provided in parallel with the primary winding n2 when the switching element Qa is turned on. Then, the inductor L5 constitutes a circuit for obtaining the delayed phase current. The inductor L5 has a predetermined value that cancels the reverse current of the switching element Qa. That is,
In this embodiment, a circuit for obtaining a lagging current for canceling the leading current as shown in FIG. 13 (Embodiment 6) is provided with a transformer Tf.
In contrast to the configuration provided on the primary side, the same circuit is provided on the secondary side to prevent the phase advance current from flowing on the secondary side and prevent it from flowing on the primary side.

As shown in FIG. 16, the inductor L
The diode Dc may be connected in series with 5. this is,
The lagging current due to the inductor L5 is the switching element Q.
Since it is not necessary when b is on, the switching element Q
Only when a is on, a current is passed through the inductor L5 to obtain a lagging current, and the series circuit of the inductor L4 and the diode Dc shown in FIG. 14 (a modification of the sixth embodiment) is connected to the transformer Tf. It is provided on the secondary side.

(Embodiment 8) FIG. 17 shows an eighth embodiment. In this embodiment, the switching element Qd is provided in the discharge path from the capacitor Cx to the load side. The switching element Qd turns on when the switching element Qb turns on, and turns off when the current becomes substantially zero. Further, when the switching element Qa is on, the switching element Qd does not turn on, and performs a switching operation to keep the off state. As a result, the capacitor Cx
When is the charging polarity, the discharging phenomenon to the load does not occur, so that the reverse current does not flow to the primary side.

(Embodiment 9) FIG. 18 shows a ninth embodiment. This embodiment is the diode D in the embodiment of FIG.
2c and D2a, the connection of the diode D2c is changed in the path for sending the power to the load.
The configuration is changed to send power to the load only through 2c. The switching element Qd is provided in series with the diode D2a. Since the switching element Qd performs the same switching operation as the switching element Qb, unnecessary discharge of the capacitor Cx can be prevented.

(Embodiment 10) FIG. 19 shows a tenth embodiment. In this embodiment, charging of the small-capacity capacitor Cx is 2
The discharge path is connected to the load side through the auxiliary winding n4. Even if the voltage polarity of the secondary winding n2 is in the direction of charging the capacitor Cx, the charging voltage of the capacitor Cx becomes high due to the resonance action, and the voltage of the secondary winding n2 and the load voltage (voltage of the smoothing capacitor C2) are increased. When the voltage becomes higher than the superposed voltage, the secondary winding n
The auxiliary winding n4 has a predetermined number of turns so that the voltage of 2 + the load voltage + the voltage of the auxiliary winding n4 becomes higher than the voltage of the capacitor Cx. As a result, when the switching element Qa is in the ON state and the charging polarity of the capacitor Cx is present, no discharging current flows due to the switching action of the diode D2a, and no reverse current flows through the switching element Qa.

(Embodiment 11) FIG. 20 is an explanatory view of an eleventh embodiment of the present invention. In this embodiment, the capacitor Cx
The change width of the ON period Ta-on of the switching element Qa having the polarity to be charged to is suppressed within a predetermined range.
Alternatively, the output may be adjusted by fixing Ta-on to a predetermined value and mainly changing the ON period Tb-on of the switching element Qb. If the ON period Ta-on of the switching element Qa is fixed as shown in FIG. 20, the ON period Tb- of the switching element Qb is as shown in FIG.
The output is larger as on is larger, and the output is smaller as ON period Tb-on of the switching element Qb is shorter as shown in FIG.

ON period Ta-o of switching element Qa
During the n period, the current flowing through the switching element Qa is
As shown in (a), the current peak due to the resonance of the capacitor Cx passes, becomes negative once, then becomes positive again, and then the switching element Qa is turned off.
turn it on. Alternatively, a predetermined ON period Ta-on is set such that the switching element Qa is turned off before the current of the switching element Qa reverses from the positive polarity to the negative polarity as shown in FIG.

(Embodiment 12) In Embodiment 11 described above, since the ON period Ta-on of the switching element Qa operates within a certain predetermined range, it is necessary to change the switching frequency in order to control the output in a wide range. is there. Therefore, in the twelfth embodiment, as shown in FIG. 22, the switching element Qa
After turning off, the switching element Qb is turned on, and after a predetermined time, a period Tdead in which both the switching elements Qa and Qb are in the off state is provided for a predetermined time to keep the switching frequency constant. As a result, it is possible to fix the switching frequency at which the filter or the like can effectively operate.

(Embodiment 13) FIG. 23 shows a thirteenth embodiment. The structure of the main circuit is the same as that of the conventional example shown in FIG. 35. In this example, the control circuit 10b shown in FIG. 23 creates drive signals for the switching elements Qa and Qb. The load voltage Vla is detected from both ends of the capacitor C2 in the circuit shown in FIG. 35, and the load current is detected as a current flowing from the capacitor C2 to the load side. Control circuit 1 shown in FIG.
0b indicates the ON period Ta-on of the switching element Qa.
Is short, the ON period Tb-on of the switching element Qb is long, the output is large, and the ON period Ta-on of the switching element Qa is long, the ON period Tb-on of the switching element Qb is short, and the output is small. Switching pattern B is included.

In both switching patterns A and B,
During the on period Ta-on period of the switching element Qa, both have a predetermined value that does not cause a phase advance operation. The output state is detected, and when the output is excessive, the switching operation is performed with the switching pattern B. When the output is small,
The switching operation is performed according to the switching pattern A.

In FIG. 23, the load voltage Vload and the load current Iload are detected, the load power calculator calculates the power, and the comparator compares it with the power command value Wref to switch the switching pattern. The switching pattern A is based on the output of the reference signal generator, and the PWM signal generator A
The switching pattern B is created by the PWM signal generation circuit B based on the output of the reference signal generation unit. This can prevent the switching element Qa from operating in a switching period in which a reverse current flows.

(Embodiment 14) FIG. 24 shows a fourteenth embodiment. In the present embodiment, on the secondary side of the transformer Tf,
A tap serving as a charging path for the capacitor Cx is provided, and the capacitor Cx is charged from this tap via the diode D2b and the inductor L2.
As a result, the charging voltage for the capacitor Cx can be reduced. By setting the position of the tap to a predetermined number of turns, the superimposed voltage of the voltage across the secondary side of the transformer Tf and the load voltage becomes higher than the charging voltage of the capacitor Cx. This can suppress an unnecessary discharge phenomenon from the capacitor Cx in the polarity of charging the capacitor Cx when the switching element Qa is on. Also, FIG.
Reference numeral 5 is a modified example of this embodiment, which is another configuration example in which the charging voltage of the capacitor Cx is reduced by the secondary side tap of the transformer Tf.

(Fifteenth Embodiment) FIG. 26 shows a fifteenth embodiment. In this embodiment, by connecting an impedance element in the charging path to the capacitor Cx, the charging speed to the capacitor Cx is reduced, or the capacitor Cx is reduced.
By reducing the charging voltage of, the switching in the advanced phase is avoided. In the circuit of FIG. 26, an example in which impedance elements are connected to the portions Z2a and Z2b is shown, but as shown in FIG.
It may be configured to connect to either one of 2b.

FIG. 27 shows a configuration in which the inductor L2a is connected to the portion Z2a in FIG. 26, and FIG. 28 shows a configuration in which the inductor L2b is connected to the portion Z2b in FIG. 27 or 28
Then, the resonance frequency at the time of charging the capacitor Cx is lowered by the inductors L2a and L2b, and the capacitor Cx
The charging is prevented from ending within the ON period of the primary side switching element with the polarity for charging.

(Embodiment 16) FIG. 29 shows a sixteenth embodiment. In this embodiment, the inductor L2 connected to the secondary side of the transformer Tf and the inductor L2b in the charging path of the capacitor Cx are magnetically coupled to each other so that the combined inductance becomes small at least when the capacitor Cx is charged. Then, the charging period to the capacitor Cx is shortened, and as shown in the current waveform of FIG. 21 (a), after discharging from the capacitor Cx, the range of the ON period is set so that the current of the primary side switching element is again in the forward direction. To widen In addition, as shown in FIG.
Inductor L2 connected to the next side and capacitor C
The inductor L2a in the charging / discharging path of x may be magnetically coupled to reduce the combined inductance of the charging / discharging path.

[0049]

According to the present invention, a circuit in which first and second switching elements each having a reverse current-carrying element in parallel are connected in series so that their forward directions coincide with each other, and a first circuit
In parallel with the circuit in which the capacitor and the DC power supply are connected in series.
Connect the connection point of the first capacitor and the DC power supply to the first and
Between the connection point of the second switching element and the
The secondary winding is connected to the secondary winding of the transformer,
The second smoothing capacitor through the rectifier circuit
Then, supply the electric power obtained by the second capacitor to the load.
A rectifier circuit and a second capacitor.
A third capacitor with a smaller capacity than the second capacitor between
In a power supply device with a capacitor connected,
At least one of the voltage polarities generated by the secondary winding
Then, with a closed circuit that only charges the third capacitor,
However, when discharging the charge of the third capacitor to the load side,
Emits through the secondary winding of the lance and releases its charge
When the current polarity is from the secondary winding of the transformer to the third capacitor
Has a configuration that is opposite to the polarity of charging the sensor, and
At least the switching element on the primary side of the transformer
Immediately before turning off, the voltage in the opposite direction to the switching element
Since the flow does not flow, it is possible to prevent the switching operation in a phase advance state where a current flows in the opposite direction just before the switching element on the primary side is turned off, and to prevent stress on the switching element. Further, there is an effect that it becomes unnecessary to make the withstand capacity of the element excessive.

[Brief description of drawings]

FIG. 1 is a circuit diagram of a first embodiment of the present invention.

FIG. 2 is a circuit diagram showing an example of a lagging phase current circuit according to the first embodiment of the present invention.

FIG. 3 is a circuit diagram showing another example of the lagging current circuit according to the first embodiment of the present invention.

FIG. 4 is a circuit diagram showing another example of the lagging current circuit according to the first embodiment of the present invention.

FIG. 5 is a circuit diagram of a second embodiment of the present invention.

FIG. 6 is a circuit diagram of a modification of the second embodiment of the present invention.

FIG. 7 is a circuit diagram of a third embodiment of the present invention.

FIG. 8 is a circuit diagram of a modified example of the third embodiment of the present invention.

FIG. 9 is a circuit diagram of a fourth embodiment of the present invention.

FIG. 10 is a circuit diagram of a modification of the fourth embodiment of the present invention.

FIG. 11 is a circuit diagram of a fifth embodiment of the present invention.

FIG. 12 is an equivalent circuit diagram of a current source according to a fifth embodiment of the present invention.

FIG. 13 is a circuit diagram of a sixth embodiment of the present invention.

FIG. 14 is a circuit diagram of a modification of the sixth embodiment of the present invention.

FIG. 15 is a circuit diagram of a seventh embodiment of the present invention.

FIG. 16 is a circuit diagram of a modification of the seventh embodiment of the present invention.

FIG. 17 is a circuit diagram of Embodiment 8 of the present invention.

FIG. 18 is a circuit diagram of Embodiment 9 of the present invention.

FIG. 19 is a circuit diagram of Embodiment 10 of the present invention.

FIG. 20 is a waveform diagram showing drive signals for a pair of switching elements according to Example 11 of the present invention.

FIG. 21 is a waveform diagram showing the relationship between the ON period of one switching element and the current flowing through the element according to Example 11 of the present invention.

FIG. 22 is a waveform diagram showing drive signals for a pair of switching elements according to Embodiment 12 of the present invention.

FIG. 23 is a block circuit diagram showing a configuration of a control circuit according to a thirteenth embodiment of the present invention.

FIG. 24 is a circuit diagram of Embodiment 14 of the present invention.

FIG. 25 is a circuit diagram of a modification of the fourteenth embodiment of the present invention.

FIG. 26 is a circuit diagram of Embodiment 15 of the present invention.

FIG. 27 is a circuit diagram of a modification of the fifteenth embodiment of the present invention.

FIG. 28 is a circuit diagram of another modification of the fifteenth embodiment of the present invention.

FIG. 29 is a circuit diagram of Embodiment 16 of the present invention.

FIG. 30 is a circuit diagram of a modification of the sixteenth embodiment of the present invention.

FIG. 31 is a circuit diagram of Conventional Example 1.

FIG. 32 is an equivalent circuit diagram of a load circuit in Conventional Example 1.

FIG. 33 is a circuit diagram of Conventional Example 2.

FIG. 34 is a circuit diagram of Conventional Example 3.

FIG. 35 is a circuit diagram of Conventional Example 4.

[Explanation of symbols]

1 DC power supply 2 High frequency inverter circuit 3 Rectification / smoothing circuit 4 Low frequency inverter circuit 5a load (discharge lamp) 6 power supply Q1 switching element Q2 switching element Cs capacitor Co capacitor Tf transformer n3 tertiary winding 20 Late phase current circuit Cx small capacity capacitor C2 smoothing capacitor

─────────────────────────────────────────────────── ─── Continuation of front page (58) Fields surveyed (Int.Cl. ⁷ , DB name) H02M 3/28 H02M 7/48 H02M 7/5387 H05B 41/282

Claims (10)

(57) [Claims] 1. A circuit in which first and second switching elements each having a reverse current-carrying element in parallel are connected in series so that their forward directions match, and a first capacitor and a DC power supply are connected in series. connecting the circuit in parallel, connecting the primary winding of the transformer between the connection point of the first co <br/> capacitor and the DC power supply connection point and the first and second switching elements,
A power supply circuit configuration in which a second smoothing capacitor is connected to a secondary winding of a transformer via a current limiting inductor and a rectifying circuit, and the power obtained by the second capacitor is supplied to a load. And a third capacitor having a smaller capacity than the second capacitor is connected between the rectifier circuit and the second capacitor, the secondary winding of the transformer is generated.
At least one of the voltage polarities has the third polarity.
It has a closed circuit that only charges the capacitor,
The secondary winding of the transformer is used to discharge the electric charge from the transformer to the load side.
The current polarity when discharging the electric charge is
Polarity for charging the third capacitor from the secondary winding of the lance
A power supply having a configuration that is the reverse of the above, and preventing a reverse current from flowing to the switching element at least immediately before the switching element on the primary side of the transformer is turned off. apparatus. 2. The transformer according to claim 1, at least immediately before the switching element on the primary side of the transformer is turned off, a positive current is superimposed on the switching element, and the superimposed current is supplied from the tertiary winding of the transformer. A power supply device characterized by supplying. 3. The switching device according to claim 2, wherein both ends of the tertiary winding of the transformer are connected to at least one switching element, and when the switching element is turned on, the voltage generated in the tertiary winding causes the switching to occur. Through the element 3
A power supply device having a configuration in which a current flows in the secondary winding, wherein the current flowing from the tertiary winding is connected to the switching element so as to have a positive polarity. 4. A power supply device according to claim 2, wherein both ends of the tertiary winding of the transformer are connected to both ends of a capacitor forming a series circuit with the power supply in the primary side circuit. 5. The power supply device according to claim 2, wherein both ends of the tertiary winding of the transformer are connected to both ends of the power supply in the primary side circuit. 6. The power supply device according to claim 2, wherein both ends of the tertiary winding of the transformer are connected to both ends of a series circuit of a power supply and a capacitor in the primary side circuit. 7. The method according to any one of claims 3 to 6,
A power supply device, to which a current limiting element for limiting a current flowing through a tertiary winding of a transformer is connected. 8. The power supply device according to claim 1, wherein a current source is connected in parallel to the primary winding of the transformer. 9. The power supply device according to claim 1, wherein a current limiting element is connected in parallel to the primary winding of the transformer. 10. The power supply device according to claim 1, wherein a current limiting element is connected in parallel to the secondary winding of the transformer.