JP2020085574A - Detection circuit - Google Patents

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JP2020085574A
JP2020085574A JP2018217791A JP2018217791A JP2020085574A JP 2020085574 A JP2020085574 A JP 2020085574A JP 2018217791 A JP2018217791 A JP 2018217791A JP 2018217791 A JP2018217791 A JP 2018217791A JP 2020085574 A JP2020085574 A JP 2020085574A
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detection circuit
voltage
magnetic field
output voltage
differential amplifier
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圭 田邊
Kei Tanabe
圭 田邊
晶裕 海野
Akihiro Unno
晶裕 海野
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TDK Corp
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TDK Corp
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Abstract

To provide a detection circuit which can be reduced in size and in cost as compared with an existing detection circuit.SOLUTION: A detection circuit 1A includes: a reception unit 13 in which GMR elements 15a and 15c are half-bridge connected; a signal generator 18b; and a low-pass filter 18a. The directions of magnetization of the pin layers of the GMR elements 15a and 15c are parallel to each other and face opposite directions. The reception unit 13 receives a magnetic field Hpropagating in a space. The signal generation unit 18b inputs, in the reception unit 13, an alternation voltage Vof the frequency for which amplitude information is to be taken out. A high-frequency component is removed from the output voltage of the reception unit 13 through the low-pass filter 18a. The output voltage Vout of the low-path filter 18a is a synchronous detected magnetic field H.SELECTED DRAWING: Figure 1

Description

本発明は、磁界信号を検波する検波回路に関する。 The present invention relates to a detection circuit that detects a magnetic field signal.

従来より、軟磁性体歯車等の移動体の位置検出（回転検出）に、磁気検出装置が用いられている。下記特許文献１の磁気検出装置は、移動体に交番磁界を印加し、移動体の相対移動による磁界変化を磁気センサで検出する構成である。 Conventionally, a magnetic detection device has been used for position detection (rotation detection) of a moving body such as a soft magnetic gear. The magnetic detection device of Patent Document 1 below has a configuration in which an alternating magnetic field is applied to a moving body and a magnetic sensor detects a magnetic field change due to relative movement of the moving body.

再公表特許WO2017/073280号公報Republished Patent WO2017/073280

特許文献１の磁気検出装置は、磁気センサの出力信号を同期検波する同期検波部を有する。同期検波では、一般に、検波用の信号と検波対象信号とを乗算器で乗算し、ローパスフィルタで高周波成分を除去する。乗算器は回路規模が大きいため、検波回路の小型化や低コスト化が困難であった。 The magnetic detection device of Patent Document 1 has a synchronous detection unit that synchronously detects the output signal of the magnetic sensor. In synchronous detection, generally, a detection signal and a detection target signal are multiplied by a multiplier, and a high frequency component is removed by a low pass filter. Since the multiplier has a large circuit scale, it has been difficult to reduce the size and cost of the detection circuit.

本発明はこうした状況を認識してなされたものであり、その目的は、従来と比較して小型化、低コスト化が可能な検波回路を提供することにある。 The present invention has been made in recognition of such a situation, and an object thereof is to provide a detection circuit which can be downsized and reduced in cost as compared with a conventional one.

本発明のある態様は、検波回路である。この検波回路は、
少なくとも１つの磁気感応素子を含む受信部と、
前記受信部を構成する磁気感応素子に所定周波数の交番電圧を印加する電圧印加部と、
前記受信部の出力信号を通すローパスフィルタと、を備える。 One aspect of the present invention is a detection circuit. This detection circuit
A receiver including at least one magnetically sensitive element,
A voltage applying section for applying an alternating voltage of a predetermined frequency to the magnetically sensitive element forming the receiving section,
A low-pass filter that passes the output signal of the receiving unit.

前記受信部は、ブリッジ接続された複数の磁気感応素子からなるブリッジ回路を含んでもよい。 The receiving unit may include a bridge circuit including a plurality of magnetically sensitive elements connected in a bridge.

前記受信部は、前記ブリッジ回路の出力電圧が入力される差動増幅器を有してもよい。 The receiver may include a differential amplifier to which the output voltage of the bridge circuit is input.

前記差動増幅器から電流を供給され、前記ブリッジ回路を磁気平衡状態にする負帰還磁界を発生する磁界発生導体と、
前記差動増幅器から前記磁界発生導体に供給される電流を電圧に変換して前記ローパスフィルタに出力する電流電圧変換手段と、を備えてもよい。 A magnetic field generating conductor that is supplied with a current from the differential amplifier and generates a negative feedback magnetic field that brings the bridge circuit into a magnetic equilibrium state,
Current-voltage converting means for converting a current supplied from the differential amplifier to the magnetic field generating conductor into a voltage and outputting the voltage to the low-pass filter.

なお、以上の構成要素の任意の組合せ、本発明の表現を方法やシステムなどの間で変換したものもまた、本発明の態様として有効である。 It should be noted that any combination of the above constituent elements and one obtained by converting the expression of the present invention between methods and systems are also effective as an aspect of the present invention.

本発明によれば、従来と比較して小型化、低コスト化が可能な検波回路を提供することができる。 According to the present invention, it is possible to provide a detection circuit that can be downsized and reduced in cost as compared with the related art.

本発明の実施の形態１に係る検波回路１Ａの回路図。1 is a circuit diagram of a detection circuit 1A according to Embodiment 1 of the present invention. 検波回路１Ａの受信部１３に１ｋＨｚの矩形波（方形波）の磁界が印加された場合において、信号生成部１８ｂから１ｋＨｚ、２ｋＨｚ、３ｋＨｚ、４ｋＨｚ、５ｋＨｚ、６ｋＨｚの正弦波電圧を受信部１３に印加した各場合における、センサ出力電圧Ｖoutのシミュレーションによる波形図。When a 1 kHz rectangular wave (square wave) magnetic field is applied to the receiving unit 13 of the detection circuit 1A, a sine wave voltage of 1 kHz, 2 kHz, 3 kHz, 4 kHz, 5 kHz, 6 kHz is applied to the receiving unit 13 from the signal generating unit 18b. The waveform diagram by simulation of the sensor output voltage Vout in each case applied. 信号生成部１８ｂから１ｋＨｚの正弦波電圧を受信部１３に印加した場合のセンサ出力電圧Ｖoutの強度を基準（１００％）として、図２に示す各場合のセンサ出力電圧Ｖoutの強度を示した棒グラフ。A bar graph showing the intensity of the sensor output voltage Vout in each case shown in FIG. 2 with the intensity of the sensor output voltage Vout when a 1 kHz sine wave voltage is applied to the receiving unit 13 from the signal generation unit 18b as a reference (100%). . 比較例１に係る検波回路の回路図。6 is a circuit diagram of a detection circuit according to Comparative Example 1. FIG. 本発明の実施の形態２に係る検波回路１Ｂの回路図。The circuit diagram of the detection circuit 1B which concerns on Embodiment 2 of this invention. 比較例２に係る検波回路の回路図。6 is a circuit diagram of a detection circuit according to Comparative Example 2. FIG. 本発明の実施の形態３に係る検波回路１Ｃの回路図。The circuit diagram of the detection circuit 1C which concerns on Embodiment 3 of this invention. 比較例３に係る検波回路の回路図。7 is a circuit diagram of a detection circuit according to Comparative Example 3. FIG.

以下、図面を参照しながら本発明の好適な実施の形態を詳述する。なお、各図面に示される同一または同等の構成要素、部材等には同一の符号を付し、適宜重複した説明は省略する。また、実施の形態は発明を限定するものではなく例示であり、実施の形態に記述されるすべての特徴やその組み合わせは必ずしも発明の本質的なものであるとは限らない。 Hereinafter, preferred embodiments of the present invention will be described in detail with reference to the drawings. Note that the same or equivalent constituent elements, members, and the like shown in each drawing are denoted by the same reference numerals, and duplicated description will be omitted as appropriate. In addition, the embodiments do not limit the invention, but are exemplifications, and all features and combinations described in the embodiments are not necessarily essential to the invention.

（実施の形態１）
図１は、本発明の実施の形態１に係る検波回路１Ａの回路図である。検波回路１Ａは、同期検波を行う回路であって、受信部１３と、ローパスフィルタ１８ａと、電圧印加部としての信号生成部１８ｂと、を備える。受信部１３は、磁気感応素子としてのＧＭＲ素子１５ａ、１５ｃ（ＧＭＲ：Giant Magneto Resistive effect）をハーフブリッジ接続したＧＭＲ素子ハーフブリッジ回路を有する。ＧＭＲ素子１５ａの一端は、信号生成部１８ｂの出力端子に接続される。ＧＭＲ素子１５ａの他端は、ＧＭＲ素子１５ｃの一端に接続される。ＧＭＲ素子１５ｃの他端は、固定電圧端子としてのグランドに接続される。 (Embodiment 1)
FIG. 1 is a circuit diagram of a detection circuit 1A according to the first embodiment of the present invention. The detection circuit 1A is a circuit that performs synchronous detection, and includes a reception unit 13, a low-pass filter 18a, and a signal generation unit 18b as a voltage application unit. The receiving unit 13 has a GMR element half-bridge circuit in which GMR elements 15a and 15c (GMR: Giant Magneto Resistive effect) as magnetic sensitive elements are half-bridge connected. One end of the GMR element 15a is connected to the output terminal of the signal generator 18b. The other end of the GMR element 15a is connected to one end of the GMR element 15c. The other end of the GMR element 15c is connected to the ground as a fixed voltage terminal.

ＧＭＲ素子１５ａ、１５ｃの相互接続点は、ローパスフィルタ１８ａの入力端子に接続される。ＧＭＲ素子１５ａ、１５ｃのピン層（固定層）の磁化方向は、互いに平行かつ反対向きである。図示は省略したが、信号生成部１８ｂの出力する交番電圧の位相を調整してＧＭＲ素子ハーフブリッジ回路に入力する位相調整手段を設け、ＧＭＲ素子ハーフブリッジ回路への入力磁界ＨINの位相（すなわちＧＭＲ素子１５ａ、１５ｃの抵抗値変化の位相）と、ＧＭＲ素子ハーフブリッジ回路に印加される電圧の位相と、を合わせるようにしてもよい。 The interconnection point of the GMR elements 15a and 15c is connected to the input terminal of the low pass filter 18a. The magnetization directions of the pinned layers (fixed layers) of the GMR elements 15a and 15c are parallel and opposite to each other. Although illustration is omitted, the phase adjusting means for adjusting the phase of the alternating voltage output from the signal generating unit 18b and inputting it to the GMR element half bridge circuit is provided, and the phase of the input magnetic field H IN to the GMR element half bridge circuit (that is, The phase of the resistance value change of the GMR elements 15a and 15c) may be matched with the phase of the voltage applied to the GMR element half bridge circuit.

受信部１３は、空間を伝搬する磁界ＨINを受信する。磁界ＨINは、矩形波であってもよいし、振幅変調波（ＡＭ波）でもよい。信号生成部１８ｂは、振幅情報を取り出したい周波数の交番電圧ＶEXTを受信部１３に入力する（動作電圧として供給する）。受信部１３の出力電圧（ＧＭＲ素子１５ａ、１５ｃの相互接続点の電圧）は、ローパスフィルタ１８ａに通されて高周波成分が除去される。ローパスフィルタ１８ａの出力端子の電圧が、センサ出力電圧Ｖoutとなる。センサ出力電圧Ｖoutは、後述のように、受信部１３に印加される磁界信号を同期検波したものとなる。センサ出力電圧Ｖoutを２値に変換する増幅器を設けてもよい。 The receiver 13 receives the magnetic field H IN propagating in space. The magnetic field H IN may be a rectangular wave or an amplitude modulation wave (AM wave). The signal generation unit 18b inputs the alternating voltage V EXT of the frequency at which the amplitude information is desired to be extracted to the reception unit 13 (supplies it as an operating voltage). The output voltage of the receiver 13 (the voltage at the interconnection point of the GMR elements 15a and 15c) is passed through the low-pass filter 18a to remove high frequency components. The voltage at the output terminal of the low-pass filter 18a becomes the sensor output voltage Vout. The sensor output voltage Vout is obtained by synchronously detecting the magnetic field signal applied to the receiving unit 13, as described later. An amplifier for converting the sensor output voltage Vout into a binary value may be provided.

図２は、検波回路１Ａの受信部１３（ＧＭＲ素子ハーフブリッジ回路）に１ｋＨｚの矩形波（方形波）の磁界が印加された場合において、信号生成部１８ｂから１ｋＨｚ、２ｋＨｚ、３ｋＨｚ、４ｋＨｚ、５ｋＨｚ、６ｋＨｚの正弦波電圧を参照信号として受信部１３に印加した各場合における、センサ出力電圧Ｖoutのシミュレーションによる波形図である。図３は、信号生成部１８ｂから１ｋＨｚの正弦波電圧を受信部１３に印加した場合のセンサ出力電圧Ｖoutの強度を基準（１００％）として、図２に示す各場合のセンサ出力電圧Ｖoutの強度を示した棒グラフである。 FIG. 2 shows that when a 1 kHz rectangular wave (square wave) magnetic field is applied to the receiving unit 13 (GMR element half bridge circuit) of the detection circuit 1A, the signal generating unit 18b outputs 1 kHz, 2 kHz, 3 kHz, 4 kHz, 5 kHz. , 6 kHz is a waveform diagram by simulation of the sensor output voltage Vout in each case where a sine wave voltage of 6 kHz is applied to the receiving unit 13 as a reference signal. FIG. 3 shows the intensity of the sensor output voltage Vout in each case shown in FIG. 2 with the intensity of the sensor output voltage Vout when the sine wave voltage of 1 kHz is applied to the receiving unit 13 from the signal generation unit 18b as a reference (100%). It is a bar graph showing.

１ｋＨｚの矩形波は、基本波となる１ｋＨｚの正弦波の他に、３次、５次、・・・の奇数次の高調波成分（３ｋＨｚ、５ｋＨｚ、・・・の成分）を含み、２次、４次、・・・の偶数次の高調波成分（２ｋＨｚ、４ｋＨｚ、・・・の成分）は含まない。３次成分は基本波成分の１／３であり、５次成分は基本波成分の１／５である。Ｎを正の奇数とすれば、Ｎ次成分は基本波成分の１／Ｎである。 The rectangular wave of 1 kHz includes a sine wave of 1 kHz which is a fundamental wave and an odd harmonic component (3 kHz, 5 kHz,...) Of the third, fifth,... The even harmonic components (2 kHz, 4 kHz,...) of the fourth order... Are not included. The third-order component is 1/3 of the fundamental wave component, and the fifth-order component is 1/5 of the fundamental wave component. If N is a positive odd number, the Nth order component is 1/N of the fundamental wave component.

図２及び図３に示す結果は、２次、４次、６次の高調波成分（２ｋＨｚ、４ｋＨｚ、６ｋＨｚの成分）が若干あるものの、３次の高調波成分（３ｋＨｚの成分）が基本波成分である１ｋＨｚの成分の３２％（≒１／３）の強度、５次の高調波成分（５ｋＨｚの成分）が基本波成分である１ｋＨｚの成分の１９％（≒１／５）の強度という結果であった。この結果から、検波回路１Ａにより、１ｋＨｚの矩形波から１ｋＨｚ、３ｋＨｚ、５ｋＨｚの各成分（振幅情報）をセンサ出力電圧Ｖoutとして得られること、すなわち同期検波が可能であることが確認できた。 The results shown in FIGS. 2 and 3 show that there are some second, fourth, and sixth harmonic components (2 kHz, 4 kHz, and 6 kHz components), but the third harmonic component (3 kHz component) is the fundamental wave. The intensity of 32% (≅1/3) of the 1 kHz component, which is the component, the intensity of the fifth harmonic component (5 kHz component) is 19% (≅1/5) of the 1 kHz component, which is the fundamental wave component. It was a result. From this result, it was confirmed that the detection circuit 1A can obtain each component (amplitude information) of 1 kHz, 3 kHz, 5 kHz from the rectangular wave of 1 kHz as the sensor output voltage Vout, that is, the synchronous detection is possible.

図４は、比較例１に係る検波回路の回路図である。図４の回路は、図１に示した実施の形態１のものと比較して、ＧＭＲ素子ハーフブリッジ回路への入力電圧が信号生成部１８ｂの出力電圧ＶEXTから電源電圧Ｖccに替わった点と、ＧＭＲ素子１５ａ、１５ｃの相互接続点とローパスフィルタ１８ａの入力端子との間に乗算器１８ｃが追加された点と、信号生成部１８ｂの出力電圧ＶEXTが乗算器１８ｃに入力される点で相違し、その他の点で一致する。 FIG. 4 is a circuit diagram of a detection circuit according to Comparative Example 1. The circuit of FIG. 4 is different from that of the first embodiment shown in FIG. 1 in that the input voltage to the GMR element half bridge circuit is changed from the output voltage V EXT of the signal generator 18b to the power supply voltage Vcc. , A point where a multiplier 18c is added between the interconnection point of the GMR elements 15a and 15c and the input terminal of the low-pass filter 18a, and a point where the output voltage V EXT of the signal generator 18b is input to the multiplier 18c. They differ and are otherwise identical.

以下、図１及び図４の各回路において、ＧＭＲ素子ハーフブリッジ回路への入力磁界をＨIN、ＧＭＲ素子１５ａの抵抗値をＲMR+、ＧＭＲ素子１５ｃの抵抗値をＲMR-、ＧＭＲ素子１５ａ、１５ｃの相互接続点の電圧をＶa、信号生成部１８ｂの出力電圧をＶEXT、乗算器１８ｃの出力電圧をＶMULTI（図４のみ）、ローパスフィルタ１８ａの出力電圧をＶoutとする。 1 and 4, the input magnetic field to the GMR element half bridge circuit is H IN , the resistance value of the GMR element 15a is R MR+ , the resistance value of the GMR element 15c is R MR− , the GMR element 15a, The voltage at the interconnection point of 15c is Va, the output voltage of the signal generator 18b is V EXT , the output voltage of the multiplier 18c is V MULTI (FIG. 4 only), and the output voltage of the low-pass filter 18a is Vout.

図１に示す実施の形態１の検波回路１Ａでは、図４に示す比較例１の検波回路における乗算器１８ｃが存在しない。しかし、図１に示す検波回路１ＡにおけるＧＭＲ素子ハーフブリッジ回路の出力電圧（ＧＭＲ素子１５ａ、１５ｃの相互接続点の電圧）Ｖaは、ＨIN×ＶEXTに比例する演算結果、すなわち図４に示す比較例１の検波回路における乗算器１８ｃの出力電圧と同等の電圧信号（同期検波の過程における乗算済みの電圧信号）となる。これは、ＧＭＲ素子ハーフブリッジ回路への入力電圧を、信号生成部１８ｂの出力電圧ＶEXTとしたことによる。 The detection circuit 1A of the first embodiment shown in FIG. 1 does not include the multiplier 18c in the detection circuit of the first comparative example shown in FIG. However, the output voltage (voltage at the interconnection point of the GMR elements 15a and 15c) Va of the GMR element half-bridge circuit in the detection circuit 1A shown in FIG. 1 is proportional to H IN ×V EXT , that is, shown in FIG. The voltage signal is equivalent to the output voltage of the multiplier 18c in the detection circuit of Comparative Example 1 (voltage signal that has been multiplied in the process of synchronous detection). This is because the input voltage to the GMR element half bridge circuit is the output voltage V EXT of the signal generator 18b.

ＧＭＲ素子１５ａ、１５ｃの無磁界時の抵抗値をＲ0、磁界による抵抗値の変化量をΔｒとすると、
ＲMR+＝Ｒ0＋Δｒ 式１
ＲMR-＝Ｒ0−Δｒ 式２
と表される。Δｒは、ＧＭＲ素子１５ａ、１５ｃに印加される磁界ＨINによって変化し、
Δｒ＝ＡMRＨIN 式３
と表される。ＡMRは、ＧＭＲ素子１５ａ、１５ｃの抵抗変化率によって決まる定数である。 If the resistance value of the GMR elements 15a and 15c in the absence of a magnetic field is R 0 , and the change amount of the resistance value due to the magnetic field is Δr,
R MR+ =R 0 +Δr Formula 1
R MR- =R 0 −Δr Equation 2
Is expressed as Δr changes depending on the magnetic field H IN applied to the GMR elements 15a and 15c,
Δr=A MR H IN formula 3
Is expressed as A MR is a constant determined by the rate of resistance change of the GMR elements 15a and 15c.

図４に示す比較例１の検波回路では、ＧＭＲ素子１５ａ、１５ｃの相互接続点の電圧Ｖaは、

Figure 2020085574
となる。乗算器１８ｃの出力電圧ＶMULTIは、
ＶMULTI＝Ｖa×ＶEXT 式５
で表される。式４より、
Figure 2020085574
 となる。さらに、式３より、
Figure 2020085574
 となる。 In the detection circuit of Comparative Example 1 shown in FIG. 4, the voltage Va at the interconnection point of the GMR elements 15a and 15c is
Figure 2020085574
 Becomes The output voltage V MULTI of the multiplier 18c is
V MULTI =Va×V EXT formula 5
It is represented by. From Equation 4,
Figure 2020085574
 Becomes Furthermore, from Equation 3,
Figure 2020085574
 Becomes

図１に示す実施の形態１の回路では、ＧＭＲ素子１５ａ、１５ｃの相互接続点の電圧Ｖaは、

Figure 2020085574
となる。式３及び式８より、
Figure 2020085574
 となる。 In the circuit of the first embodiment shown in FIG. 1, the voltage Va at the interconnection point of the GMR elements 15a and 15c is
Figure 2020085574
 Becomes From Equation 3 and Equation 8,
Figure 2020085574
 Becomes

このように、図１に示す検波回路１ＡにおけるＧＭＲ素子１５ａ、１５ｃの相互接続点の電圧Ｖa（式９）は、図４に示す比較例１の検波回路における乗算器１８ｃの出力電圧ＶMULTI（式７）と比例する。よって、図１に示す検波回路１Ａでは、ＧＭＲ素子１５ａ、１５ｃの相互接続点の電圧Ｖaをローパスフィルタ１８ａに通した後の信号（センサ出力電圧Ｖout）は、ＧＭＲ素子ハーフブリッジ回路に印加される磁界信号ＨINを同期検波した結果の信号となる。すなわち、図１に示す検波回路１Ａは、図４に示す比較例１と異なり乗算器１８ｃを有さないにもかかわらず、ＧＭＲ素子１５ａ、１５ｃの相互接続点の電圧Ｖaとして乗算済みの信号が得られることから、乗算器１８ｃを有さずに同期検波が可能である。 Thus, the voltage Va (Equation 9) at the interconnection point of the GMR elements 15a and 15c in the detection circuit 1A shown in FIG. 1 is equal to the output voltage V MULTI (of the multiplier 18c in the detection circuit of Comparative Example 1 shown in FIG. It is proportional to Equation 7). Therefore, in the detection circuit 1A shown in FIG. 1, the signal (sensor output voltage Vout) after the voltage Va at the interconnection point of the GMR elements 15a and 15c is passed through the low pass filter 18a is applied to the GMR element half bridge circuit. It becomes a signal as a result of synchronous detection of the magnetic field signal H IN . That is, unlike the comparative example 1 shown in FIG. 4, the detection circuit 1A shown in FIG. 1 does not have the multiplier 18c, but the signal that has been multiplied as the voltage Va at the interconnection point of the GMR elements 15a and 15c is Since it is obtained, synchronous detection is possible without the multiplier 18c.

以下、ＨIN及びＶEXTの時間変化を考慮して、
ＨIN＝ＨIN・sin（ωt＋θ） 式１０
ＶEXT＝ＶEXT・sin（ωt） 式１１
とする。位相調整によりθ＝０とすると、図１に示す検波回路１Ａにおいて、式９〜式１１より、

Figure 2020085574
となる。式１２の導出過程において、下記式１３に示す三角関数の公式、
Figure 2020085574
 を利用した。ローパスフィルタ１８ａの出力電圧Ｖoutは、式１２のＶaから低周波成分を取り出したものであり、
Figure 2020085574
 となる。このＶoutが検波後の信号強度出力となる。 Hereinafter, considering the time changes of H IN and V EXT ,
H IN =H IN ·sin(ωt+θ) Equation 10
V EXT =V EXT ·sin(ωt) Equation 11
And When θ=0 is set by the phase adjustment, in the detection circuit 1A shown in FIG.
Figure 2020085574
 Becomes In the process of deriving Expression 12, the trigonometric function formula shown in Expression 13 below,
Figure 2020085574
 Was used. The output voltage Vout of the low-pass filter 18a is obtained by extracting the low frequency component from Va of the equation 12,
Figure 2020085574
 Becomes This Vout becomes the signal strength output after detection.

本実施の形態によれば、信号生成部１８ｂの出力する交番電圧ＶEXTをＧＭＲ素子ハーフブリッジ回路に動作電圧として供給するため、ＧＭＲ素子ハーフブリッジ回路の出力電圧Ｖaは乗算済みの信号（ＨIN×ＶEXTに比例する電圧）となる。このため、乗算のための専用回路（例えば図４に示す比較例１の検波回路の乗算器１８ｃ）を設けずに、ＧＭＲ素子ハーフブリッジ回路に印加される磁界信号の同期検波が可能となる。よって、本実施の形態の検波回路１Ａは、乗算のための専用回路が不要な分、小型かつ低コストなものとなる。 According to the present embodiment, since the alternating voltage V EXT output from the signal generator 18b is supplied to the GMR element half bridge circuit as the operating voltage, the output voltage Va of the GMR element half bridge circuit is the multiplied signal (H IN ×V EXT ). Therefore, the synchronous detection of the magnetic field signal applied to the GMR element half bridge circuit can be performed without providing a dedicated circuit for multiplication (for example, the multiplier 18c of the detection circuit of Comparative Example 1 shown in FIG. 4). Therefore, the detection circuit 1A of the present embodiment is small in size and low in cost because a dedicated circuit for multiplication is unnecessary.

（実施の形態２）
図５は、本発明の実施の形態２に係る検波回路１Ｂの回路図である。以下、図１に示した実施の形態１の検波回路１Ａとの相違点を中心に説明する。検波回路１Ｂにおいて、受信部１３は、ＧＭＲ素子１５ａ〜１５ｄをフルブリッジ接続したＧＭＲ素子フルブリッジ回路を有する。検波回路１Ｂは、ＧＭＲ素子フルブリッジ回路とローパスフィルタ１８ａとの間に、オペアンプ等の差動増幅器１７を有する。 (Embodiment 2)
FIG. 5 is a circuit diagram of the detection circuit 1B according to the second embodiment of the present invention. Hereinafter, differences from the detection circuit 1A of the first embodiment shown in FIG. 1 will be mainly described. In the detection circuit 1B, the reception unit 13 has a GMR element full bridge circuit in which the GMR elements 15a to 15d are full bridge connected. The detection circuit 1B has a differential amplifier 17 such as an operational amplifier between the GMR element full bridge circuit and the low pass filter 18a.

ＧＭＲ素子１５ａ、１５ｂのピン層の磁化方向は、互いに平行かつ反対向きである。ＧＭＲ素子１５ｂ、１５ｄのピン層の磁化方向は、互いに平行かつ反対向きである。ＧＭＲ素子１５ａ、１５ｂの相互接続点は、信号生成部１８ｂの出力端子に接続される。ＧＭＲ素子１５ａ、１５ｃの相互接続点は、差動増幅器１７の反転入力端子に接続される。ＧＭＲ素子１５ｂ、１５ｄの相互接続点は、差動増幅器１７の非反転入力端子に接続される。ＧＭＲ素子１５ｃ、１５ｄの相互接続点はグランドに接続される。 The magnetization directions of the pinned layers of the GMR elements 15a and 15b are parallel and opposite to each other. The magnetization directions of the pinned layers of the GMR elements 15b and 15d are parallel and opposite to each other. The interconnection point of the GMR elements 15a and 15b is connected to the output terminal of the signal generator 18b. The interconnection point of the GMR elements 15a and 15c is connected to the inverting input terminal of the differential amplifier 17. The interconnection point of the GMR elements 15b and 15d is connected to the non-inverting input terminal of the differential amplifier 17. The interconnection point of the GMR elements 15c and 15d is connected to the ground.

差動増幅器１７は、電源電圧Ｖcc、−Ｖccの供給を受けて動作する。差動増幅器１７の出力端子は、ローパスフィルタ１８ａの入力端子に接続される。差動増幅器１７の出力端子とグランドとの間に、抵抗１９が接続される。抵抗１９は、グランドに対する差動増幅器１７の出力端子の電圧を確実に決めるために設けられるが、不要であれば省略してもよい。ＧＭＲ素子１５ａ〜１５ｄの出力電圧は、差動増幅器１７によって増幅され、ローパスフィルタ１８ａに入力される。ローパスフィルタ１８ａは、差動増幅器１７の出力信号の高周波成分を除去する。検波回路１Ｂの回路構成のその他の点は、実施の形態１の検波回路１Ａと同様である。 The differential amplifier 17 operates by being supplied with the power supply voltages Vcc and -Vcc. The output terminal of the differential amplifier 17 is connected to the input terminal of the low pass filter 18a. The resistor 19 is connected between the output terminal of the differential amplifier 17 and the ground. The resistor 19 is provided to surely determine the voltage of the output terminal of the differential amplifier 17 with respect to the ground, but may be omitted if unnecessary. The output voltages of the GMR elements 15a to 15d are amplified by the differential amplifier 17 and input to the low pass filter 18a. The low pass filter 18a removes high frequency components of the output signal of the differential amplifier 17. The other points of the circuit configuration of the detection circuit 1B are the same as those of the detection circuit 1A of the first embodiment.

図６は、比較例２に係る検波回路の回路図である。図６の回路は、図５に示した実施の形態２のものと比較して、ＧＭＲ素子フルブリッジ回路への入力電圧が信号生成部１８ｂの出力電圧ＶEXTから電源電圧Ｖccに替わった点と、差動増幅器１７の出力端子とローパスフィルタ１８ａの入力端子との間に乗算器１８ｃが追加された点と、信号生成部１８ｂの出力電圧ＶEXTが乗算器１８ｃに入力される点で相違し、その他の点で一致する。 FIG. 6 is a circuit diagram of a detection circuit according to Comparative Example 2. The circuit of FIG. 6 is different from that of the second embodiment shown in FIG. 5 in that the input voltage to the GMR element full bridge circuit is changed from the output voltage V EXT of the signal generator 18b to the power supply voltage Vcc. The difference is that a multiplier 18c is added between the output terminal of the differential amplifier 17 and the input terminal of the low-pass filter 18a, and that the output voltage V EXT of the signal generator 18b is input to the multiplier 18c. , Match in other respects.

以下、図５及び図６の各回路において、ＧＭＲ素子フルブリッジ回路への入力磁界をＨIN、ＧＭＲ素子１５ａ、１５ｄの抵抗値をＲMR+、ＧＭＲ素子１５ｂ、１５ｃの抵抗値をＲMR-、差動増幅器１７の反転入力端子の電圧をＶa、非反転入力端子の電圧をＶb、差動増幅器１７の出力端子の電圧をＶdiff、信号生成部１８ｂの出力電圧をＶEXT、コイル１２に流れる電流をＩEXT、乗算器１８ｃの出力電圧をＶMULTI（図６のみ）、ローパスフィルタ１８ａの出力電圧をＶoutとする。前述の式１〜式３は、図５及び図６の各回路においても共通である。 5 and 6, the input magnetic field to the GMR element full bridge circuit is H IN , the resistance values of the GMR elements 15a and 15d are R MR+ , the resistance values of the GMR elements 15b and 15c are R MR− , The voltage at the inverting input terminal of the differential amplifier 17 is Va, the voltage at the non-inverting input terminal is Vb, the voltage at the output terminal of the differential amplifier 17 is Vdiff, the output voltage of the signal generator 18b is V EXT , and the current flowing through the coil 12 is Is I EXT , the output voltage of the multiplier 18c is V MULTI (only in FIG. 6), and the output voltage of the low-pass filter 18a is V out. Equations 1 to 3 described above are common to the circuits of FIGS. 5 and 6.

図６に示す比較例２の検波回路では、差動増幅器１７の反転入力端子の電圧Ｖa、非反転入力端子の電圧Ｖbは、

Figure 2020085574
Figure 2020085574
 となる。差動増幅器１７の出力電圧Ｖdiffは、
Ｖdiff＝Ａdiff（Ｖａ−Ｖｂ） 式１７
と表される。Ａdiffは、差動増幅器１７のゲイン（定数）である。式１５〜式１７より、
Figure 2020085574
 となる。乗算器１８ｃの出力電圧ＶMULTIは、
ＶMULTI＝Ｖdiff×ＶEXT 式１９
で表される。式１８より、
Figure 2020085574
 となる。さらに、式３より、
Figure 2020085574
 となる。 In the detection circuit of Comparative Example 2 shown in FIG. 6, the voltage Va at the inverting input terminal and the voltage Vb at the non-inverting input terminal of the differential amplifier 17 are
Figure 2020085574
Figure 2020085574
 Becomes The output voltage Vdiff of the differential amplifier 17 is
Vdiff=Adiff(Va-Vb) Equation 17
Is expressed as Adiff is the gain (constant) of the differential amplifier 17. From Equation 15 to Equation 17,
Figure 2020085574
 Becomes The output voltage V MULTI of the multiplier 18c is
V MULTI =V diff ×V EXT formula 19
It is represented by. From Equation 18,
Figure 2020085574
 Becomes Furthermore, from Equation 3,
Figure 2020085574
 Becomes

図５に示す実施の形態２の検波回路１Ｂでは、差動増幅器１７の反転入力端子の電圧Ｖa、非反転入力端子の電圧Ｖbは、

Figure 2020085574
Figure 2020085574
 となる。差動増幅器１７の出力電圧Ｖdiffは、式１７、式２２、式２３より、
Figure 2020085574
 となる。さらに、式３より、
Figure 2020085574
 となる。 In the detection circuit 1B of the second embodiment shown in FIG. 5, the voltage Va at the inverting input terminal and the voltage Vb at the non-inverting input terminal of the differential amplifier 17 are
Figure 2020085574
Figure 2020085574
 Becomes The output voltage Vdiff of the differential amplifier 17 is calculated from the equations 17, 22, and 23.
Figure 2020085574
 Becomes Furthermore, from Equation 3,
Figure 2020085574
 Becomes

このように、図５に示す検波回路１Ｂにおける差動増幅器１７の出力電圧Ｖdiff（式２５）は、図６に示す比較例２の検波回路における乗算器１８ｃの出力電圧ＶMULTI（式２１）と比例する。よって、図５に示す検波回路１Ｂでは、差動増幅器１７の出力電圧Ｖdiffをローパスフィルタ１８ａに通した後の信号（センサ出力電圧Ｖout）は、ＧＭＲ素子フルブリッジ回路に印加される磁界信号を同期検波した結果の信号となる。すなわち、図５に示す検波回路１Ｂは、図６に示す比較例２と異なり乗算器１８ｃを有さないにもかかわらず、差動増幅器１７の出力電圧Ｖdiffとして乗算済みの信号が得られることから、乗算器１８ｃを有さずに同期検波が可能である。 Thus, the output voltage Vdiff (Equation 25) of the differential amplifier 17 in the detection circuit 1B shown in FIG. 5 is equal to the output voltage V MULTI (Equation 21) of the multiplier 18c in the detection circuit of Comparative Example 2 shown in FIG. Proportional. Therefore, in the detection circuit 1B shown in FIG. 5, the signal (sensor output voltage Vout) after passing the output voltage Vdiff of the differential amplifier 17 through the low-pass filter 18a synchronizes the magnetic field signal applied to the GMR element full bridge circuit. The signal is the result of detection. That is, unlike the comparative example 2 shown in FIG. 6, the detection circuit 1B shown in FIG. 5 does not have the multiplier 18c, but a signal that has been multiplied as the output voltage Vdiff of the differential amplifier 17 can be obtained. The synchronous detection is possible without the multiplier 18c.

実施の形態１と同様に、ＨIN及びＶEXTの時間変化を考慮して、ＨIN＝ＨIN・sin（ωt＋θ）（式１０）、ＶEXT＝ＶEXT・sin（ωt）（式１１）として計算すると、

Figure 2020085574
Figure 2020085574
 となる。このＶoutが検波後の信号強度出力となる。 In the same manner as in the first embodiment, H IN =H IN ·sin(ωt+θ) (Equation 10) and V EXT =V EXT ·sin(ωt) (Equation 11) in consideration of the temporal changes of H IN and V EXT. When calculated as
Figure 2020085574
Figure 2020085574
 Becomes This Vout becomes the signal strength output after detection.

本実施の形態によれば、信号生成部１８ｂの出力する交番電圧ＶEXTをＧＭＲ素子フルブリッジ回路に動作電圧として供給するため、ＧＭＲ素子フルブリッジ回路の出力電圧（Ｖa−Ｖb）は乗算済みの信号（ＨIN×ＶEXTに比例する電圧）となる。このため、乗算のための専用回路（例えば図６に示す比較例２の検波回路の乗算器１８ｃ）を設けずに、ＧＭＲ素子フルブリッジ回路に印加される磁界信号の同期検波が可能となる。よって、本実施の形態の検波回路１Ｂは、乗算のための専用回路が不要な分、小型かつ低コストなものとなる。 According to the present embodiment, since the alternating voltage V EXT output from the signal generator 18b is supplied to the GMR element full bridge circuit as the operating voltage, the output voltage (Va-Vb) of the GMR element full bridge circuit has already been multiplied. It becomes a signal (voltage proportional to H IN ×V EXT ). Therefore, the synchronous detection of the magnetic field signal applied to the GMR element full bridge circuit can be performed without providing a dedicated circuit for multiplication (for example, the multiplier 18c of the detection circuit of Comparative Example 2 shown in FIG. 6). Therefore, the detection circuit 1B of the present embodiment is small in size and low in cost because a dedicated circuit for multiplication is unnecessary.

（実施の形態３）
図７は、本発明の実施の形態３に係る検波回路１Ｃの回路図である。以下、図５に示した実施の形態２の検波回路１Ｂとの相違点を中心に説明する。検波回路１Ｃは、差動増幅器１７の出力端子とローパスフィルタ１８ａの入力端子との間に、磁界発生導体としてのコイル１２を有する。コイル１２の一端は、差動増幅器１７の出力端子に接続される。コイル１２の他端は、ローパスフィルタ１８ａの入力端子に接続される。 (Embodiment 3)
FIG. 7 is a circuit diagram of the detection circuit 1C according to the third embodiment of the present invention. Hereinafter, differences from the detection circuit 1B of the second embodiment shown in FIG. 5 will be mainly described. The detection circuit 1C has a coil 12 as a magnetic field generating conductor between the output terminal of the differential amplifier 17 and the input terminal of the low pass filter 18a. One end of the coil 12 is connected to the output terminal of the differential amplifier 17. The other end of the coil 12 is connected to the input terminal of the low pass filter 18a.

コイル１２は、差動増幅器１７の出力電流が流れることにより、受信部１３を磁気平衡状態にする負帰還磁界を発生する。磁気平衡状態は、ＧＭＲ素子１５ａ〜１５ｄの位置における磁界の感磁方向成分が所定値（例えばゼロ）の状態である。抵抗１９は、本実施の形態ではコイル１２に流れる電流を電圧に変換する電流電圧変換手段であって、コイル１２の他端とグランドとの間に設けられる。検波回路１Ｃでは、受信部１３を磁気平衡状態とするためにコイル１２に流れる電流（負帰還電流ＩFB）を利用して、入力磁界ＨINを検波する。すなわち、実施の形態１及び２の検波回路１Ａ、１Ｂでは入力磁界ＨINの検出方式が磁気比例式であるのに対し、本実施の形態の検波回路１Ｃでは入力磁界ＨINの検出方式が磁気平衡式である。 When the output current of the differential amplifier 17 flows, the coil 12 generates a negative feedback magnetic field that brings the receiving unit 13 into a magnetic equilibrium state. The magnetic equilibrium state is a state in which the magnetic sensitive component of the magnetic field at the positions of the GMR elements 15a to 15d has a predetermined value (for example, zero). In this embodiment, the resistor 19 is a current-voltage conversion unit that converts a current flowing through the coil 12 into a voltage, and is provided between the other end of the coil 12 and the ground. In the detection circuit 1C, the input magnetic field H IN is detected by using the current (negative feedback current I FB ) flowing in the coil 12 to bring the receiving unit 13 into the magnetic equilibrium state. That is, in the detection circuits 1A and 1B of the first and second embodiments, the detection method of the input magnetic field H IN is magnetic proportional, whereas in the detection circuit 1C of the present embodiment, the detection method of the input magnetic field H IN is magnetic. It is a balanced type.

検波回路１Ｃにおいて、負帰還電流ＩFBは、入力磁界ＨIN及びＧＭＲ素子フルブリッジ回路への入力電圧ＶEXTの積と一対一でリニアに対応し、
ＩFB＝α×ＨIN×ＶEXT 式２８
の関係が成り立つ。αは、差動増幅器１７のゲインやコイル１２とＧＭＲ素子フルブリッジ回路との磁気結合度によって決まる定数である。ローパスフィルタ１８ａへの入力電圧Ｖdiffは、抵抗１９の抵抗値Ｒsを用いて、
Ｖdiff＝Ｒs×ＩFB
＝Ｒs×α×ＨIN×ＶEXT 式２９
となり、実施の形態２の検波回路１Ｂにおける上記式２５と比例する値となる。 In the detection circuit 1C, the negative feedback current I FB linearly corresponds to the product of the input magnetic field H IN and the input voltage V EXT to the GMR element full bridge circuit in a one-to-one correspondence,
I FB =α×H IN ×V EXT Equation 28
The relationship is established. α is a constant determined by the gain of the differential amplifier 17 and the degree of magnetic coupling between the coil 12 and the GMR element full bridge circuit. The input voltage Vdiff to the low-pass filter 18a is calculated by using the resistance value Rs of the resistor 19.
Vdiff=Rs×I FB
=Rs×α×H IN ×V EXT Formula 29
And has a value proportional to the above equation 25 in the detection circuit 1B of the second embodiment.

図８は、比較例３に係る検波回路の回路図である。図８の回路は、図７に示した実施の形態３のものと比較して、ＧＭＲ素子フルブリッジ回路への入力電圧が信号生成部１８ｂの出力電圧ＶEXTから電源電圧Ｖccに替わった点と、コイル１２とローパスフィルタ１８ａの入力端子との間に乗算器１８ｃが追加された点と、信号生成部１８ｂの出力電圧ＶEXTが乗算器１８ｃに入力される点で相違し、その他の点で一致する。 FIG. 8 is a circuit diagram of a detection circuit according to Comparative Example 3. The circuit of FIG. 8 differs from the circuit of the third embodiment shown in FIG. 7 in that the input voltage to the GMR element full bridge circuit is changed from the output voltage V EXT of the signal generator 18b to the power supply voltage Vcc. , A point that a multiplier 18c is added between the coil 12 and the input terminal of the low-pass filter 18a and a point that the output voltage V EXT of the signal generator 18b is input to the multiplier 18c. Match.

比較例３の検波回路では、
ＩFB＝α×ＨIN×Ｖcc 式３０
Ｖdiff＝Ｒs×ＩFB
＝Ｒs×α×ＨIN×Ｖcc 式３１
ＶMULTI＝Ｖdiff×ＶEXT
＝Ｒs×α×ＨIN×Ｖcc×ＶEXT 式３２
である。 In the detection circuit of Comparative Example 3,
I FB =α×H IN ×Vcc Formula 30
Vdiff=Rs×I FB
=Rs×α×H IN ×Vcc Formula 31
V MULTI =Vdiff×V EXT
=Rs×α×H IN ×Vcc×V EXT Equation 32
Is.

本実施の形態によれば、信号生成部１８ｂの出力する交番電圧ＶEXTをＧＭＲ素子フルブリッジ回路に動作電圧として供給するため、差動増幅器１７の出力電圧Ｖdiff（式２９）が、図８に示す比較例３の検波回路における乗算器１８ｃの出力電圧ＶMULTI（式３２）と比例する。よって、図７に示す検波回路１Ｃでは、差動増幅器１７の出力電圧Ｖdiffをローパスフィルタ１８ａに通した後の信号（センサ出力電圧Ｖout）は、ＧＭＲ素子フルブリッジ回路に印加される磁界信号を同期検波した結果の信号となる。すなわち、図７に示す検波回路１Ｃは、図８に示す比較例３と異なり乗算器１８ｃを有さないにもかかわらず、差動増幅器１７の出力電圧Ｖdiffとして乗算済みの信号が得られることから、乗算器１８ｃを有さずに同期検波が可能である。したがって、本実施の形態の検波回路１Ｃは、乗算のための専用回路が不要な分、小型かつ低コストなものとなる。 According to the present embodiment, since the alternating voltage V EXT output from the signal generator 18b is supplied to the GMR element full bridge circuit as the operating voltage, the output voltage Vdiff (equation 29) of the differential amplifier 17 is shown in FIG. It is proportional to the output voltage V MULTI (equation 32) of the multiplier 18c in the detection circuit of Comparative Example 3 shown. Therefore, in the detection circuit 1C shown in FIG. 7, the signal (sensor output voltage Vout) after passing the output voltage Vdiff of the differential amplifier 17 through the low pass filter 18a synchronizes the magnetic field signal applied to the GMR element full bridge circuit. The signal is the result of detection. That is, unlike the comparative example 3 shown in FIG. 8, the detection circuit 1C shown in FIG. 7 does not have the multiplier 18c, but a signal that has been multiplied as the output voltage Vdiff of the differential amplifier 17 can be obtained. The synchronous detection is possible without the multiplier 18c. Therefore, the detection circuit 1C of the present embodiment is small in size and low in cost because a dedicated circuit for multiplication is unnecessary.

以上、実施の形態を例に本発明を説明したが、実施の形態の各構成要素や各処理プロセスには請求項に記載の範囲で種々の変形が可能であることは当業者に理解されるところである。以下、変形例について触れる。 Although the present invention has been described with the embodiment as an example, it will be understood by those skilled in the art that various modifications can be made to each component and each process of the embodiment within the scope of the claims. By the way. Hereinafter, modified examples will be described.

受信部１３は、１つのＧＭＲ素子と１つの固定抵抗とがハーフブリッジ接続されたものであってもよい。受信部１３が実施の形態１のようなＧＭＲ素子ハーフブリッジ回路である場合においても、入力磁界ＨINの検出方式を磁気平衡式としてもよい。磁気感応素子は、ＧＭＲ素子等の磁気抵抗効果素子に限定されず、ホール素子等の他の種類のものであってもよい。 The receiving unit 13 may be one in which one GMR element and one fixed resistor are half-bridge connected. Even when the receiving unit 13 is the GMR element half bridge circuit as in the first embodiment, the detection method of the input magnetic field H IN may be the magnetic balance method. The magnetically sensitive element is not limited to a magnetoresistive effect element such as a GMR element, but may be another type such as a Hall element.

１Ａ〜１Ｃ 検波回路、１３ 受信部、１５ａ〜１５ｄ ＧＭＲ素子（磁気抵抗効果素子）、１７ 差動増幅器、１８ａ ローパスフィルタ、１８ｂ 信号生成部（電圧印加部）、１８ｃ 乗算器、１９ 抵抗 1A-1C detection circuit, 13 receiving part, 15a-15d GMR element (magnetoresistive effect element), 17 differential amplifier, 18a low pass filter, 18b signal generation part (voltage application part), 18c multiplier, 19 resistance

Claims (4)

少なくとも１つの磁気感応素子を含む受信部と、
前記受信部を構成する磁気感応素子に所定周波数の交番電圧を印加する電圧印加部と、
前記受信部の出力信号を通すローパスフィルタと、を備える、検波回路。 A receiver including at least one magnetically sensitive element,
A voltage applying section for applying an alternating voltage of a predetermined frequency to the magnetically sensitive element forming the receiving section,
A low-pass filter that passes the output signal of the receiving unit. 前記受信部は、ブリッジ接続された複数の磁気感応素子からなるブリッジ回路を含む、請求項１に記載の検波回路。 The detection circuit according to claim 1, wherein the receiving unit includes a bridge circuit including a plurality of magnetically sensitive elements connected in a bridge. 前記受信部は、前記ブリッジ回路の出力電圧が入力される差動増幅器を有する、請求項２に記載の検波回路。 The detection circuit according to claim 2, wherein the reception unit includes a differential amplifier to which the output voltage of the bridge circuit is input. 前記差動増幅器から電流を供給され、前記ブリッジ回路を磁気平衡状態にする負帰還磁界を発生する磁界発生導体と、
前記差動増幅器から前記磁界発生導体に供給される電流を電圧に変換して前記ローパスフィルタに出力する電流電圧変換手段と、を備える、請求項３に記載の検波回路。 A magnetic field generating conductor that is supplied with a current from the differential amplifier and generates a negative feedback magnetic field that brings the bridge circuit into a magnetic equilibrium state,
4. The detection circuit according to claim 3, further comprising a current-voltage conversion unit that converts a current supplied from the differential amplifier to the magnetic field generation conductor into a voltage and outputs the voltage to the low-pass filter.
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JP2018146303A (en) * 2017-03-02 2018-09-20 Tdk株式会社 Magnetic sensor
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JP2000055998A (en) * 1998-08-05 2000-02-25 Tdk Corp Magnetic sensor device and current sensor device
WO2015056397A1 (en) * 2013-10-17 2015-04-23 公立大学法人大阪市立大学 Electric current measurement apparatus and electric current measurement method
JP2015219061A (en) * 2014-05-15 2015-12-07 Tdk株式会社 Magnetic field detection sensor and magnetic field detection device using the same
JP2016114408A (en) * 2014-12-12 2016-06-23 アルプス電気株式会社 Magnetic field detection device
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