JP2018129683A - Filter circuit, multiplexer, and module - Google Patents

Abstract

PROBLEM TO BE SOLVED: To increase the steepness of cutoff characteristics.SOLUTION: A filter circuit includes: a first element which is any one of a capacitor and an inductor connected in series between an input terminal T1 and an output terminal T2; a second element which is connected between the input terminal and the output terminal in parallel with the first element and also which is the other of the capacitor and the inductor different from the first element; a third element which is connected between the input terminal and the output terminal in parallel with the first element and in series with the second element and also which is the other of the capacitor and the inductor different from the first element; and an acoustic wave resonator R1 one end of which is connected to a first node N1 between the second element and the third element and the other end of which is connected to a ground terminal.SELECTED DRAWING: Figure 4

Description

  The present invention relates to a filter circuit, a multiplexer, and a module, for example, a filter circuit, a multiplexer, and a module having an acoustic wave resonator.

  As the low-pass filter (LPF) and the high-pass filter (HPF), a filter in which an inductor and a capacitor are combined (that is, LC filter) is used. The LC filter is configured by laminating ceramic layers, for example. It is known to connect a capacitor and an inductor to an acoustic wave filter (for example, Patent Document 1).

JP 2010-411141 A

  In the LC filter, the steepness of the cutoff characteristic between the pass band and the stop band and the insertion loss in the pass band are in a trade-off relationship. Therefore, if the insertion loss of a desired pass band is ensured, the steepness of the cutoff characteristic is deteriorated.

  The present invention has been made in view of the above problems, and an object thereof is to increase the steepness of the cutoff characteristic.

  The present invention provides a first element that is one of a capacitor and an inductor connected in series between an input terminal and an output terminal, and in parallel with the first element between the input terminal and the output terminal. A second element different from the first element among a capacitor and an inductor, and connected in parallel with the first element and in series with the second element between the input terminal and the output terminal; A third element which is the other of the capacitor and the inductor different from the first element, one end is connected to a first node between the second element and the third element, and the other end is connected to a ground terminal. A filter circuit comprising an acoustic wave resonator.

  In the above configuration, the attenuation pole formed by the first element, the second element, and the third element may be a low-pass filter having a higher frequency than the attenuation pole formed by the acoustic wave resonator.

  In the above configuration, the attenuation pole formed by the first element, the second element, and the third element may be a high-pass filter having a lower frequency than the attenuation pole formed by the acoustic wave resonator.

  In the above configuration, between the input terminal and a second node to which the first element and the second element are connected in common, and a third node to which the first element and the third element are connected in common A matching circuit may be provided at least one of the output terminal and the output terminal.

  In the above configuration, the first element includes a fourth element and a fifth element connected in series between the input terminal and the output terminal, and the filter circuit has one end connected to the fourth element and the first element. A sixth node connected to a fourth node between the five elements, the other end connected to a ground terminal, and a sixth element that is the same as the first element among the capacitor and the inductor may be provided.

  In the above configuration, the acoustic wave resonator may include a plurality of acoustic wave resonators connected in series or in parallel between the first node and the ground terminal.

  The said structure WHEREIN: The said acoustic wave resonator can be set as the structure containing IDT.

  In the above configuration, the acoustic wave resonator may include a piezoelectric thin film resonator.

  The present invention is a multiplexer having the filter circuit.

  The present invention is a module having the filter circuit.

  According to the present invention, the steepness of the cutoff characteristic can be enhanced.

1A is a circuit diagram of an LPF according to Comparative Example 1, FIG. 1B is a diagram showing pass characteristics, and FIG. 1C is a Smith chart showing reflection characteristics. 2A is a circuit diagram of an HPF according to Comparative Example 1, FIG. 2B is a diagram showing pass characteristics, and FIG. 2C is a Smith chart showing reflection characteristics. 3A is a plan view of the surface acoustic wave resonator, and FIG. 3B is a cross-sectional view of the piezoelectric thin film resonator. FIG. 4A is a circuit diagram of the LPF according to the first embodiment, FIG. 4B is a diagram showing pass characteristics, and FIG. 4C is a Smith chart showing reflection characteristics. FIG. 5 is a circuit diagram illustrating an equivalent circuit of the filter circuit according to the first embodiment. FIG. 6A is a circuit diagram of an LPF according to the first modification of the first embodiment, FIG. 6B is a diagram showing the pass characteristics, and FIG. 6C is a Smith chart showing the reflection characteristics. FIG. 7A is a circuit diagram of an LPF according to the second modification of the first embodiment, FIG. 7B is a diagram showing pass characteristics, and FIG. 7C is a Smith chart showing reflection characteristics. FIG. 8A is a circuit diagram of an LPF according to the third modification of the first embodiment, FIG. 8B is a diagram showing the pass characteristics, and FIG. 8C is a Smith chart showing the reflection characteristics. FIG. 9A and FIG. 9B are circuit diagrams of a filter circuit according to a fourth modification of the first embodiment. FIG. 10A is a circuit diagram of an HPF according to the second embodiment, FIG. 10B is a diagram showing the pass characteristics, and FIG. 10C is a Smith chart showing the reflection characteristics. FIG. 11A is a circuit diagram of an HPF according to the first modification of the second embodiment, FIG. 11B is a diagram showing pass characteristics, and FIG. 11C is a Smith chart showing reflection characteristics. FIG. 12A is a circuit diagram of an HPF according to the second modification of the second embodiment, FIG. 12B is a diagram showing the pass characteristics, and FIG. 12C is a Smith chart showing the reflection characteristics. FIG. 13A is a circuit diagram of an HPF according to a third modification of the second embodiment, FIG. 13B is a diagram showing the pass characteristics, and FIG. 13C is a Smith chart showing the reflection characteristics. FIG. 14A is a circuit diagram of an HPF according to a fourth modification of the second embodiment, FIG. 14B is a diagram showing the pass characteristics, and FIG. 14C is a Smith chart showing the reflection characteristics. FIG. 15 is a circuit diagram of a diplexer according to the third embodiment. FIG. 16 is a circuit diagram of a front-end circuit including a module according to the fourth embodiment. FIG. 17A to FIG. 17C are cross-sectional views of modules according to the fourth embodiment.

[Comparative Example 1]
Comparative Example 1 is an example of an LC filter. 1A is a circuit diagram of an LPF according to Comparative Example 1, FIG. 1B is a diagram showing pass characteristics, and FIG. 1C is a Smith chart showing reflection characteristics. As shown in FIG. 1A, capacitors C01 and C02 are connected in series between terminals T1 and T2. Inductors L01 and L01 are connected in series between terminals T1 and T2. Capacitors C01 and C02 and inductors L01 and L02 are connected in parallel. A capacitor C03 is connected between the node N1 between the inductors L01 and L02 and the ground terminal.

The transmission characteristic S21 from the terminal T1 to T2 and the reflection characteristic S11 from the terminal T1 were simulated. The simulation conditions in FIGS. 1B and 1C are as follows.
Terminals T1, T2: 50Ω termination C01, C02: 1.8pF
C03: 1.6 pF
L01, L02: 2nH

In FIG. 1B and FIG. 1C, the frequency with the largest S21 is denoted by m1, and the frequency at the bottom of the attenuation pole that is higher than the frequency m1 and closest to m1 is denoted by m2. The same applies to the following LPFs. The size of S21 and the size / angle of S11 at frequencies m1 and m2 are as follows.
Frequency m1 = 1.980 GHz, S21 = −0.00 dB, S11 = 0.02 / −6.4 °
Frequency m2 = 2.650 GHz, S21 = −97.8 dB, S11 = 1.0 / 0 °

  At the frequency m1, S21 is large (that is, the loss is small), and S11 is located almost at the center of the Smith chart. That is, the high frequency signal input from the terminal T1 is hardly reflected and attenuated by the LPF and is output from the terminal T2. At the frequency m2, S21 is small (ie, the attenuation is large), and the size of S11 is approximately 1. That is, the high frequency signal input from the terminal T1 is almost reflected or attenuated by the LPF and hardly output from the terminal T2. The difference between the frequencies m1 and m2 is 670 MHz.

  2A is a circuit diagram of an HPF according to Comparative Example 1, FIG. 2B is a diagram showing pass characteristics, and FIG. 2C is a Smith chart showing reflection characteristics. As shown in FIG. 2A, an inductor L03 is connected between the node N1 and the ground terminal. Other circuits are the same as those of the LPF, and the description thereof is omitted.

The simulation conditions in FIGS. 2A and 2B are as follows.
Terminals T1, T2: 50Ω termination C01, C02: 1.8pF
L01, L02: 2nH
L03: 2.2nF

2 (b) and 2 (c), the frequency with the largest S21 is denoted by m1, and the frequency at the bottom of the attenuation pole that is lower than the frequency m1 and closest to m1 is denoted by m2. The same applies to the following HPFs. The size of S21 and the size / angle of S11 at frequencies m1 and m2 are as follows.
Frequency m1 = 3.620 GHz, S21 = −0.00 dB, S11 = 0.00 / 3.0
Frequency m2 = 2.650 GHz, S21 = −98.1 dB, S11 = 1.0 / 0 °
The difference between the frequencies m1 and m2 is 970 MHz.

  As in Comparative Example 1, in the LC filter, the frequency between the frequencies m1 and m2 is several hundreds of MHz, and the cutoff characteristic between the pass band and the stop band is not steep. When trying to bring the frequencies m1 and m2 closer, S21 at the frequency m1 becomes smaller (that is, the loss becomes larger). Thus, if it is attempted to secure insertion loss in the pass band, the steepness of the cutoff characteristic between the pass band and the stop band cannot be improved.

  Example 1 is an example of a filter circuit having an LPF function. An example of an acoustic wave resonator used in the embodiment will be described. 3A is a plan view of the surface acoustic wave resonator, and FIG. 3B is a cross-sectional view of the piezoelectric thin film resonator. As shown in FIG. 3A, an IDT (Interdigital Transducer) 51 and a reflector 52 are formed on the piezoelectric substrate 50. The IDT 51 has a pair of comb-shaped electrodes 51a facing each other. The comb electrode 51a includes a plurality of electrode fingers 51b and a bus bar 51c that connects the plurality of electrode fingers 51b. The reflectors 52 are provided on both sides of the IDT 51. The IDT 51 excites a surface acoustic wave on the piezoelectric substrate 50. The piezoelectric substrate 50 is, for example, a lithium tantalate substrate or a lithium niobate substrate. The IDT 51 and the reflector 52 are made of, for example, an aluminum film or a copper film. The piezoelectric substrate 50 may be bonded to the lower surface of a support substrate such as a sapphire substrate, an alumina substrate, a spinel substrate, or a silicon substrate. A protective film or a temperature compensation film that covers the IDT 50 and the reflector 52 may be provided.

  As shown in FIG. 3B, a piezoelectric film 57 is provided on the substrate 55. A lower electrode 56 and an upper electrode 58 are provided so as to sandwich the piezoelectric film 57. A gap 59 is formed between the lower electrode 56 and the substrate 55. The lower electrode 56 and the upper electrode 58 excite elastic waves in the thickness longitudinal vibration mode in the piezoelectric film 57. The lower electrode 56 and the upper electrode 58 are metal films such as a ruthenium film. The piezoelectric film 57 is, for example, an aluminum nitride film. The substrate 55 is, for example, a silicon substrate, a sapphire substrate, an alumina substrate, a spinel substrate, or a glass substrate.

  FIG. 4A is a circuit diagram of the LPF according to the first embodiment, FIG. 4B is a diagram showing pass characteristics, and FIG. 4C is a Smith chart showing reflection characteristics. As shown in FIG. 4A, a capacitor C1 is connected in series between terminals T1 and T2. An inductor L2 is connected in parallel with the capacitor C1 between the terminals T1 and T2. An inductor L3 is connected between the terminals T1 and T2 in parallel with the capacitor C1 and in series with the inductor L2. Node N1 is a node between inductors L2 and L3. The node N2 is a node to which the capacitor C1 and the inductor L2 are connected in common. The node N3 is a node to which the capacitor C1 and the inductor L3 are connected in common. One end of the acoustic wave resonator R1 is connected to the node N1, and the other end is connected to the ground terminal. The acoustic wave resonator R1 is a one-port resonator. An LC parallel resonant circuit 10 in which inductors L2 and L3 and a capacitor C1 are connected in parallel is formed between nodes N2 and N3.

The simulation conditions in FIGS. 4B and 4C are as follows.
Terminals T1, T2: 50Ω termination C1: 1.5pF
L2, L3: 1.5 nH
R1: The simulation was performed using a piezoelectric thin film resonator having a resonance frequency of 2.26 GHz and an anti-resonance frequency of 2.33 GHz.

  As shown in FIG. 4B, the attenuation pole A1 formed mainly by the resonance frequency of the acoustic wave resonator R1 is formed at the frequency m2. The attenuation poles A2 and A3 formed mainly by the capacitor C1 and the inductors L2 and L3 (LC parallel resonance circuit 10) are formed on the higher frequency side than the attenuation pole A1.

  For example, when the capacitance of the capacitor C1 is made constant and the inductance of the inductor L1 is increased, the attenuation poles A2 and A3 are shifted to the lower frequency side, and the attenuation pole A1 is shifted to the higher frequency side. When the inductance of the inductor L1 is made constant and the capacitance of the capacitor C1 is increased, the attenuation pole A2 is shifted to the lower frequency side, and the attenuation poles A1 and A3 are shifted to the higher frequency side. Thus, the attenuation pole can be formed at a desired frequency by appropriately setting the capacitor C1 and the inductor L1.

  Since the attenuation pole A1 is mainly formed by the resonance frequency of the elastic wave resonator R1, the cutoff characteristic can be made steep. When the attenuation poles A1 and A2 approach each other and the attenuation poles A1 and A2 form one attenuation pole, the steepness of the attenuation pole A1 cannot be obtained. For this reason, it is preferable that the attenuation poles A1 and A2 each have a minimum.

In FIG. 4B and FIG. 4C, the magnitude of S21 and the magnitude / angle of S11 at frequencies m1 and m2 are as follows.
Frequency m1 = 2.223 GHz, S21 = −0.55 dB, S11 = 0.06 / 18 °
Frequency m2 = 2.314 GHz, S21 = −37.1 dB, S11 = 0.98 / 13 °

  In the first embodiment, the loss at the frequency m1 is as good as −0.55 dB, and the difference between the frequencies m1 and m2 is 91 MHz, which is smaller than that of the first comparative example by one digit. In this way, the insertion loss of the pass band can be reduced and the steepness of the cutoff characteristic between the pass band and the stop band can be improved.

  FIG. 5 is a circuit diagram illustrating an equivalent circuit of the filter circuit according to the first embodiment. As shown in FIG. 5, the acoustic wave resonator R1 is equivalently represented by capacitors C10 and C11 and an inductor L11. The attenuation poles A1 to A3 are determined by the combined impedance of the capacitors C1, C10 and C11 and the inductors L2, L3 and L11. Therefore, although the frequency at the bottom of the attenuation pole A1 is formed by the resonance frequency of the elastic wave resonator R1, it is different from the single resonance frequency of the elastic wave resonator R1. The bottom frequency of the attenuation poles A2 and A3 is formed by the capacitor C1 and the inductors L2 and L3, but is influenced by the acoustic wave resonator R1.

[Modification 1 of Example 1]
The first modification of the first embodiment is an example in which the inductor and the capacitor of the LC parallel resonance circuit 10 are replaced. FIG. 6A is a circuit diagram of an LPF according to the first modification of the first embodiment, FIG. 6B is a diagram showing the pass characteristics, and FIG. 6C is a Smith chart showing the reflection characteristics. As shown in FIG. 6A, compared to the first embodiment, the capacitor C1, the inductors L2 and L3 are replaced with the inductor L1, the capacitors C2 and C3, respectively. Other configurations are the same as those of the first embodiment, and the description thereof is omitted.

The simulation conditions in FIGS. 6B and 6C are as follows.
Terminals T1, T2: 50Ω termination L1: 1.5nH
C2, C3: 5.5 pF
R1: The simulation was performed using a piezoelectric thin film resonator having a resonance frequency of 2.26 GHz and an anti-resonance frequency of 2.33 GHz.

As shown in FIG. 6B, attenuation poles A1 and A2 are formed as in the first embodiment. The attenuation pole A3 is not formed. In FIG. 6B and FIG. 6C, the size of S21 and the size / angle of S11 at frequencies m1 and m2 are as follows.
Frequency m1 = 2.210 GHz, S21 = −0.31 dB, S11 = 0.8 / 23 °
Frequency m2 = 2.289 GHz, S21 = -30.7 dB, S11 = 0.97 / 69 °
The difference between the frequencies m1 and m2 is 79 MHz.

  In the first modification of the first embodiment, even if the inductor and the capacitor of the LC parallel resonant circuit 10 are replaced, the insertion loss of the pass band can be reduced and the steepness of the cutoff characteristic between the pass band and the stop band can be improved. .

[Modification 2 of Embodiment 1]
A second modification of the first embodiment is an example in which a matching circuit is provided. FIG. 7A is a circuit diagram of an LPF according to the second modification of the first embodiment, FIG. 7B is a diagram showing pass characteristics, and FIG. 7C is a Smith chart showing reflection characteristics. As shown in FIG. 7A, compared to the first embodiment, a matching circuit 12 is provided between the terminal T1 and the inductor L3. The matching circuit 12 includes an inductor L7 connected in series between the terminal T1 and the node N2, and a capacitor C7 shunt-connected to the node N2. A matching circuit 14 is provided between the inductor L3 and the terminal T2. The matching circuit 14 includes an inductor L8 connected between the node N3 and the terminal T2, and a capacitor C8 shunt-connected to the node N3. Other configurations are the same as those of the first embodiment, and the description thereof is omitted.

The simulation conditions in FIGS. 7B and 7C are as follows.
Terminals T1, T2: 50Ω termination C1: 1.5pF
C7, C8: 0.8 pF
L2, L3: 1.5 nH
L7, L8: 2.2nH
R1: The simulation was performed using a piezoelectric thin film resonator having a resonance frequency of 2.26 GHz and an anti-resonance frequency of 2.33 GHz.

As shown in FIG. 7B, attenuation poles A1 to A3 are formed as in the first embodiment. In FIG. 7B and FIG. 7C, the size of S21 and the size / angle of S11 at frequencies m1 and m2 are as follows.
Frequency m1 = 2.215 GHz, S21 = −0.40 dB, S11 = 0.04 / −57 °
Frequency m2 = 2.314 GHz, S21 = −34.1 dB, S11 = 0.97 / −68 °
The difference between the frequencies m1 and m2 is 99 MHz.

  In the second modification of the first embodiment, by providing the matching circuits 12 and 14, the insertion loss in the pass band can be made smaller than that in the first embodiment. In addition, the pass band can be widened. Further, the attenuation amount of the stop band can be made larger than that of the first embodiment.

[Modification 3 of Embodiment 1]
Modification 3 of the first embodiment is an example in which the capacitor C1 is divided. FIG. 8A is a circuit diagram of an LPF according to the third modification of the first embodiment, FIG. 8B is a diagram showing the pass characteristics, and FIG. 8C is a Smith chart showing the reflection characteristics. As shown in FIG. 8A, the capacitor C1 is divided into capacitors C4 and C5 in series as compared with the second modification of the first embodiment. Node N4 is a node between capacitors C4 and C5. Capacitor C6 has one end connected to node N4 and the other end connected to the ground terminal. Other configurations are the same as those of the second modification of the first embodiment, and the description thereof is omitted.

The simulation conditions in FIGS. 8B and 8C are as follows.
Terminals T1, T2: 50Ω termination C4, C5: 3.3 pF
C6: 0.5 pF
C7, C8: 0.6 pF
L2, L3: 1.6 nH
L7, L8: 2.7 nH
R1: The simulation was performed using a piezoelectric thin film resonator having a resonance frequency of 2.26 GHz and an anti-resonance frequency of 2.33 GHz.

As shown in FIG. 8B, attenuation poles A1 to A3 are formed as in the first embodiment. In FIG. 8B and FIG. 8C, the size of S21 and the size / angle of S11 at frequencies m1 and m2 are as follows.
Frequency m1 = 2.232 GHz, S21 = −0.39 dB, S11 = 0.04 / −90 °
Frequency m2 = 2.350 GHz, S21 = -31.8 dB, S11 = 0.97 / −114 °
The difference between the frequencies m1 and m2 is 118 MHz.

  In the third modification of the first embodiment, the amount of attenuation in the stop band can be made larger than that in the second modification of the first embodiment.

[Modification 4 of Example 1]
FIG. 9A and FIG. 9B are circuit diagrams of a filter circuit according to a fourth modification of the first embodiment. As shown in FIG. 9A, the acoustic wave resonator R1 may be divided in series into a plurality of acoustic wave resonators R1a and R1b. As shown in FIG. 9B, the acoustic wave resonator R1 may be divided into a plurality of acoustic wave resonators R1a and R1b in parallel. Other configurations are the same as those of the first embodiment, and the description thereof is omitted.

  Example 2 is an example of a filter circuit having an HPF function. FIG. 10A is a circuit diagram of an HPF according to the second embodiment, FIG. 10B is a diagram showing the pass characteristics, and FIG. 10C is a Smith chart showing the reflection characteristics. As shown in FIG. 10A, the circuit is the same as that of the first embodiment.

The simulation conditions in FIGS. 10B and 10C are as follows.
Terminals T1, T2: 50Ω termination C1: 3.9pF
L2, L3: 0.9nH
R1: The simulation was performed using a piezoelectric thin film resonator having a resonance frequency of 2.26 GHz and an anti-resonance frequency of 2.33 GHz.

  As shown in FIG. 10B, an attenuation pole A1 formed mainly from the resonance frequency of the acoustic wave resonator R1 is formed at the frequency m2. An attenuation pole A2 formed mainly from the capacitor C1 and the inductors L2 and L3 (LC parallel resonant circuit 10) is formed on the low frequency side of the attenuation pole A1.

In FIG. 10B and FIG. 10C, the size of S21 and the size / angle of S11 at frequencies m1 and m2 are as follows.
Frequency m1 = 2.287 GHz, S21 = −0.19 dB, S11 = 0.02 / −98 °
Frequency m2 = 2.202 GHz, S21 = −27.0 dB, S11 = 0.95 / −108 °

  In the second embodiment, the loss at the frequency m1 is as small as −0.19 dB, and the difference between the frequencies m1 and m2 is 85 MHz, which is smaller than that of the first comparative example by almost one digit. In this way, the insertion loss of the pass band can be reduced and the steepness of the cutoff characteristic between the pass band and the stop band can be improved.

[Modification 1 of Embodiment 2]
The first modification of the second embodiment is an example in which the inductor and the capacitor of the LC parallel resonance circuit 10 are replaced. FIG. 11A is a circuit diagram of an HPF according to the first modification of the second embodiment, FIG. 11B is a diagram showing pass characteristics, and FIG. 11C is a Smith chart showing reflection characteristics. As shown in FIG. 11A, compared to the first embodiment, the capacitor C1, the inductors L2 and L3 are replaced with the inductor L1, the capacitors C2 and C3, respectively. Other configurations are the same as those of the second embodiment, and the description thereof is omitted.

The simulation conditions in FIGS. 11B and 11C are as follows.
Terminals T1, T2: 50Ω termination C2, C3: 7.1 pF
L1: 2nH
R1: The simulation was performed using a piezoelectric thin film resonator having a resonance frequency of 2.26 GHz and an anti-resonance frequency of 2.33 GHz.

As shown in FIG. 11B, attenuation poles A1 and A2 are formed as in the second embodiment. In FIG. 11B and FIG. 11C, the size of S21 and the size / angle of S11 at frequencies m1 and m2 are as follows.
Frequency m1 = 2.291 GHz, S21 = −0.26 dB, S11 = 0.03 / −83 °
Frequency m2 = 2.237 GHz, S21 = −27.7 dB, S11 = 0.96 / −110 °
The difference between the frequencies m1 and m2 is 54 MHz.

  In the first modification of the second embodiment, even if the inductor and the capacitor of the LC parallel resonant circuit 10 are replaced, the insertion loss of the pass band can be reduced and the steepness of the cutoff characteristic between the pass band and the stop band can be improved. .

[Modification 2 of Embodiment 2]
A second modification of the second embodiment is an example in which a matching circuit is provided in the second embodiment. FIG. 12A is a circuit diagram of an HPF according to the second modification of the second embodiment, FIG. 12B is a diagram showing the pass characteristics, and FIG. 12C is a Smith chart showing the reflection characteristics. As shown in FIG. 12A, compared with the second embodiment, a matching circuit 12 is provided between the terminal T1 and the inductor L2. The matching circuit 12 includes an inductor L7 that is shunt-connected to the node N2. A matching circuit 14 is provided between the inductor L3 and the terminal T2. The matching circuit 14 includes an inductor L8 that is shunt-connected to the node N3. Other configurations are the same as those of the second embodiment, and the description thereof is omitted.

The simulation conditions in FIGS. 12B and 12C are as follows.
Terminals T1, T2: 50Ω termination C1: 3.9pF
L2, L3: 0.9nH
L7, L8: 2.5nH
R1: The simulation was performed using a piezoelectric thin film resonator having a resonance frequency of 2.26 GHz and an anti-resonance frequency of 2.33 GHz.

As shown in FIG. 12B, attenuation poles A1 and A2 are formed as in the second embodiment. In FIG. 12B and FIG. 12C, the size of S21 and the size / angle of S11 at frequencies m1 and m2 are as follows.
Frequency m1 = 2.260 GHz, S21 = −0.19 dB, S11 = 0.01 / −8 °
Frequency m2 = 2.202 GHz, S21 = −18.6 dB, S11 = 0.88 / 9 °
The difference between the frequencies m1 and m2 is 58 MHz.

  In the second modification of the second embodiment, by providing the matching circuits 12 and 14, the insertion loss in the pass band can be made smaller than that in the second embodiment. In addition, the pass band can be widened. Further, the attenuation amount of the stop band can be made larger than that of the first embodiment.

[Modification 3 of Embodiment 2]
The third modification of the second embodiment is an example in which a matching circuit is provided in the first modification of the second embodiment. FIG. 13A is a circuit diagram of an HPF according to a third modification of the second embodiment, FIG. 13B is a diagram showing the pass characteristics, and FIG. 13C is a Smith chart showing the reflection characteristics. As shown in FIG. 13A, a matching circuit 12 is provided between the terminal T1 and the node N2 as compared with the first modification of the second embodiment. A matching circuit 12 is provided between the terminal T1 and the node N2. The matching circuits 12 and 14 are the same as in the second modification of the second embodiment. Other configurations are the same as those of the second modification of the second embodiment, and the description thereof is omitted.

The simulation conditions in FIGS. 13B and 13C are as follows.
Terminals T1, T2: 50Ω termination L1: 2.0nH
C2, C3: 7.1 pF
L7, L8: 2.6 nH
R1: The simulation was performed using a piezoelectric thin film resonator having a resonance frequency of 2.26 GHz and an anti-resonance frequency of 2.33 GHz.

As shown in FIG. 13B, attenuation poles A1 and A2 are formed as in the second embodiment. In FIG. 13B and FIG. 13C, the magnitude of S21 and the magnitude / angle of S11 at frequencies m1 and m2 are as follows.
Frequency m1 = 2.276 GHz, S21 = −0.20 dB, S11 = 0.07 / 50 °
Frequency m2 = 2.237 GHz, S21 = −18.9 dB, S11 = 0.88 / −6 °
The difference between the frequencies m1 and m2 is 39 MHz.

  In the third modification of the second embodiment, by providing the matching circuits 12 and 14, the insertion loss in the passband can be made smaller than that of the first modification of the second embodiment. In addition, the pass band can be widened. Further, the attenuation amount of the stop band can be made larger than that of the first embodiment.

[Modification 4 of Example 2]
A fourth modification of the second embodiment is an example in which the inductor L1 is divided. FIG. 14A is a circuit diagram of an HPF according to a fourth modification of the second embodiment, FIG. 14B is a diagram showing the pass characteristics, and FIG. 14C is a Smith chart showing the reflection characteristics. As shown in FIG. 14A, the inductor L1 is divided into inductors L4 and L5 in series as compared with the third modification of the second embodiment. The inductor L6 has one end connected to the node N4 and the other end connected to the ground terminal. The matching circuit 12 has a capacitor C7 connected between the terminal T1 and the node N2. The matching circuit 14 has a capacitor C8 connected between the terminal T2 and the node N3. Other configurations are the same as those of the third modification of the second embodiment, and the description thereof is omitted.

The simulation conditions in FIGS. 14B and 14C are as follows.
Terminals T1, T2: 50Ω termination L4, L5: 1.3 nH
L6: 3.8 nH
L7, L8: 2.9 nH
C2, C3: 4.7 pF
C7, C8: 1.4 pF
R1: The simulation was performed using a piezoelectric thin film resonator having a resonance frequency of 2.26 GHz and an anti-resonance frequency of 2.33 GHz.

As shown in FIG. 14B, attenuation poles A1 and A2 are formed as in the second embodiment. In FIG. 14B and FIG. 14C, the size of S21 and the size / angle of S11 at frequencies m1 and m2 are as follows.
Frequency m1 = 2.400 GHz, S21 = −0.12 dB, S11 = 0.01 / 78 °
Frequency m2 = 2.220 GHz, S21 = -21.1 dB, S11 = 0.90 / -139 °
The difference between the frequencies m1 and m2 is 180 MHz.

  In the fourth modification of the second embodiment, the amount of attenuation in the stop band can be made larger than that in the third modification of the second embodiment.

[Modification 5 of Embodiment 2]
As in the fourth modification of the first embodiment, also in the HPF, as shown in FIG. 9A, the elastic wave resonator R1 may be divided in series into a plurality of elastic wave resonators R1a and R1b. As shown in FIG. 9B, the acoustic wave resonator R1 may be divided into a plurality of acoustic wave resonators R1a and R1b in parallel.

  According to the first and second embodiments and the modification thereof, the LC parallel resonant circuit 10 is connected in series between the terminal T1 (input terminal) and the terminal T2 (output terminal). One end of the acoustic wave resonator R1 is connected to the node N1 (first node) and the other end is connected to the ground terminal. Here, as an element forming the LC parallel resonance circuit 10, a capacitor C1 or an inductor L1 having one end connected to a node N2 (second node) and the other end connected to a node N3 (third node) is a first element. The capacitor C2 or the inductor L2 having one end connected to the node N2 and the other end connected to the node N1 is defined as a second element. The inductor L3 or the capacitor C3 having one end connected to the node N1 and the other end connected to the node N3 is defined as a third element. In order to form the LC parallel resonant circuit 10 from the first element to the third element, the first element is a capacitor C1 as in the first embodiment and its modifications 2 to 4 and the second embodiment and its modification 2, respectively. In this case, the second element and the third element become inductors L2 and L3, respectively. As in the first modification of the first embodiment and the first, third, and fourth modifications of the second embodiment, when the first element is an inductor L1, the second and third elements are capacitors C2 and C3, respectively.

  As a result, as in Examples 1 and 2 and its modifications, the insertion loss at the frequency m1 (pass band) is made comparable to that in Comparative Examples 1 and 2, and the cutoff characteristic between the pass band and the stop band is steep. Can be improved.

  As in the first embodiment and its modification, the attenuation poles A2 and A3 formed by the first element, the second element, and the third element have a higher frequency than the attenuation pole A1 formed by the acoustic wave resonator. Thereby, a filter circuit functioning as an LPF can be realized.

  As in the second embodiment, the attenuation pole A2 formed by the first element, the second element, and the third element has a lower frequency than the attenuation pole A1 formed by the acoustic wave resonator. As a result, a filter circuit that functions as an HPF can be realized.

  A band pass filter may be formed by combining LPF and HPF.

  The attenuation pole A1 is mainly formed by the resonance frequency of the acoustic wave resonator R1. However, it is affected by the LC parallel resonant circuit 10 and the like. For this reason, the frequency at the bottom of the attenuation pole A1 is different from the resonance frequency when the acoustic wave resonator R1 is used alone. The attenuation poles A2 and A3 are mainly formed from the LC parallel resonant circuit 10. However, it is affected by the elastic wave resonator R1. For this reason, the frequency at the bottom of the attenuation poles A2 and A3 is different from the resonance frequency when the LC parallel resonance circuit 10 is single.

  Since the attenuation pole A1 is mainly formed by the resonance frequency of the elastic wave resonator R1, the steepness of the cutoff characteristic between the pass band and the stop band can be improved. In addition, since the attenuation poles A1 and A2 have different minimums, the steepness of the cutoff characteristic can be further improved. Further, the attenuation amount of the stop band can be increased by the attenuation pole A2.

  As in the first and second embodiments and the modifications thereof, the capacitances of the capacitors C2 and C3 may be substantially the same. The inductances of the inductors L2 and L3 may be substantially the same. That is, the reactance of the first element and the reactance of the second element may be substantially equal.

  The capacitances of the capacitors C2 and C3 may be different from each other. The inductances of the inductors L2 and L3 may be different from each other. That is, the reactance of the first element and the reactance of the second element may be different from each other.

  In addition to the LC parallel resonance circuit 10, another LC parallel resonance circuit having a resonance frequency different from that of the LC parallel resonance circuit 10 may be provided between the terminals T1 and T2. Thereby, the stop band can be widened.

  As in the first embodiment and its modifications 1 to 3 and the second embodiment and its modifications 1 to 4, there may be one elastic wave resonator R1 connected in series between the node N1 and the ground terminal. As in the fourth modification of the first embodiment and the fifth modification of the second embodiment, a plurality of acoustic wave resonators R1a and R1b may be connected in series between the node N1 and the ground terminal. A plurality of acoustic wave resonators R1a and R1b may be connected in parallel between the node N1 and the ground terminal.

  When the plurality of acoustic wave resonators R1a and R1b are provided, the resonance frequencies of the plurality of acoustic wave resonators R1a and R1b may all be substantially equal. At least one resonance frequency of the plurality of elastic wave resonators R1a and R1b may be different from the resonance frequency of the other elastic wave resonators. By making the resonant frequencies of the acoustic wave resonators R1a and R1b different, the stop band can be widened.

  As in the second and third modifications of the first embodiment and the second to fourth modifications of the second embodiment, a matching circuit is provided between at least one of the terminal T1 and the node N1 and between the node N2 and the terminal T2. You may have. Thereby, the insertion loss of the pass band can be suppressed and the pass band can be widened.

  As in the third modification of the first embodiment, when the first element is a capacitor, the first element includes capacitors C4 (fourth element) and C5 (fifth element) connected in series between the terminal T1 and the terminal T2. Element). The capacitor C6 has one end connected to the node N4 (fourth node) and the other end connected to the ground terminal. As in the fourth modification of the second embodiment, when the first element is an inductor, the first element includes inductors L4 (fourth element) and L5 (fifth element) connected in series between the terminal T1 and the terminal T2. Element). The inductor L6 has one end connected to the node N4 (fourth node) and the other end connected to the ground terminal. Thereby, the attenuation characteristic of the stop band can be improved.

  The acoustic wave resonators R1, R1a, and R1b may include an IDT as shown in FIG. 3A, or may include a piezoelectric thin film resonator as shown in FIG.

  When the acoustic wave resonator is a piezoelectric thin film resonator, two acoustic wave resonators R1a and R1b are provided in series or in parallel between the node N1 and the ground terminal. The capacitance, resonance frequency, and anti-resonance frequency of the acoustic wave resonators R1a and R1b are made substantially equal. Further, when viewed from the node N1 or the ground terminal, the polarization directions of the piezoelectric films of the acoustic wave resonators R1a and R1b are opposite to each other. Thereby, the second harmonic generated by the acoustic wave resonators R1a and R1b can be suppressed.

  FIG. 15 is a circuit diagram of a diplexer according to the third embodiment. As shown in FIG. 15, the LPF 60 is connected between the common terminal Ant and the terminal TL. An HPF 62 is connected between the common terminal Ant and the terminal TH. The LPF 60 is the filter circuit of the first embodiment and its modification, and the HPF 62 is the filter circuit of the second embodiment and its modification. Thereby, the steepness of the cutoff characteristics of the LPF 60 and the HPF 62 can be improved. Therefore, it is possible to perform demultiplexing with a narrow band interval. One of the LPF 60 and the HPF 62 may be the filter circuit of the first or second embodiment. In addition to the diplexer, a multiplexer such as a duplexer, a triplexer, or a quadplex can be realized by combining the filter circuits of the first and second embodiments and the modification examples thereof. Thereby, it is possible to perform demultiplexing with a narrow band interval as in a system using carrier aggregation or MIMO (Multiple-Input and Multiple-Output).

  FIG. 16 is a circuit diagram of a front-end circuit including a module according to the fourth embodiment. As shown in FIG. 16, the common terminal Ant of the diplexer 30 is connected to the antenna 42. The terminals TL and TH of the diplexer 30 are connected to the duplexer 34 via the switch 32, respectively. The RX terminal of the duplexer 34 is connected to an RFIC (Radio Frequency Integrated Circuit) 40 (broken line). The TX terminal of the duplexer 34 is connected to an RFIC (Radio Frequency Integrated Circuit) 40 via a switch 36 and a power amplifier 38 (solid line).

  The power amplifier 38 amplifies the transmission signal output from the RFIC 40. The switch 36 outputs the amplified transmission signal to any one transmission terminal of one or a plurality of duplexers 34. The duplexer 34 outputs a signal in the transmission band among the high-frequency signals input to the transmission terminal to the switch 32 and suppresses signals of other frequencies. The duplexer 34 outputs a signal in the reception band among the high frequency signals input from the switch 32 to the RFIC 40 and suppresses signals of other frequencies. The RFIC 40 includes a low noise amplifier and amplifies a signal in the reception band.

  The switch 32 connects one of one or a plurality of duplexers 34 to the terminal TL or TH. The diplexer 30 outputs or inputs a low-band signal input or output to the terminal TL to the common terminal Ant, and does not input or output a high-band signal to the terminal TL.

  The diplexer 30 outputs or inputs a signal to the common terminal Ant in the low band input or output to the terminal TL, and does not input or output a high band signal to the terminal TL. The diplexer 30 outputs or inputs a signal to the common terminal Ant in the high band input or output to the terminal TH and does not input or output a low band signal to the terminal TH.

  In FIG. 16, the diplexer 30 can be the diplexer of the third embodiment. Further, the filter included in the duplexer 34 can be the filter circuits of the first and second embodiments and the modifications thereof.

  FIG. 17A to FIG. 17C are cross-sectional views of modules according to the fourth embodiment. As shown in FIG. 17A, the acoustic wave resonator 22, the multilayer filter 24, and the chip component 26 are mounted on the printed board 20. A resin sealing portion 28 is provided on the printed circuit board 20 so as to cover the acoustic wave resonator 22, the multilayer filter 24, and the chip component 26. The printed board 20 is a board in which wiring is formed on an insulating board such as glass epoxy resin. The resin sealing portion 28 is a mold resin such as an epoxy resin.

  The acoustic wave resonator 22 is provided with at least a part of the filters of the acoustic wave resonators R1, R1a and R1b and / or the duplexer 34 in the first and second embodiments and modifications thereof. The multilayer filter 24 is provided with at least a part of the LC parallel resonant circuit 10 according to the first and second embodiments and the modifications thereof, and / or at least a part of the diplexer 30 and the duplexer 34. The chip component 26 is at least a part of the capacitors and inductors of the LC parallel resonant circuit 10 and / or the diplexer 30 of the first and second embodiments and modifications thereof.

  As shown in FIG. 17B, a ceramic substrate 20 a such as LTCC (Low Temperature Co-fired Ceramic) or HTCC (High Temperature Co-fired Ceramic) may be used instead of the printed circuit board 20. Other configurations are the same as those in FIG.

  As shown in FIG. 17C, the multilayer filter 24 may not be provided. Other configurations are the same as those in FIG.

  The module may include at least one of the diplexer 30 and the duplexer 34 shown in FIG.

  Although the embodiments of the present invention have been described in detail above, the present invention is not limited to such specific embodiments, and various modifications and changes can be made within the scope of the gist of the present invention described in the claims. It can be changed.

10 LC parallel resonant circuit 12, 14 Matching circuit

[Modification 2 of Embodiment 1]
A second modification of the first embodiment is an example in which a matching circuit is provided. FIG. 7A is a circuit diagram of an LPF according to the second modification of the first embodiment, FIG. 7B is a diagram showing pass characteristics, and FIG. 7C is a Smith chart showing reflection characteristics. As shown in FIG. 7A, compared to the first embodiment, a matching circuit 12 is provided between the terminal T1 and the inductor L2 . The matching circuit 12 includes an inductor L7 connected in series between the terminal T1 and the node N2, and a capacitor C7 shunt-connected to the node N2. A matching circuit 14 is provided between the inductor L3 and the terminal T2. The matching circuit 14 includes an inductor L8 connected between the node N3 and the terminal T2, and a capacitor C8 shunt-connected to the node N3. Other configurations are the same as those of the first embodiment, and the description thereof is omitted.

[Modification 3 of Embodiment 2]
The third modification of the second embodiment is an example in which a matching circuit is provided in the first modification of the second embodiment. FIG. 13A is a circuit diagram of an HPF according to a third modification of the second embodiment, FIG. 13B is a diagram showing the pass characteristics, and FIG. 13C is a Smith chart showing the reflection characteristics. As shown in FIG. 13A, a matching circuit 12 is provided between the terminal T1 and the node N2 as compared with the first modification of the second embodiment. A matching circuit 14 is provided between the terminal T2 and the node N3 . The matching circuits 12 and 14 are the same as in the second modification of the second embodiment. Other configurations are the same as those of the second modification of the second embodiment, and a description thereof will be omitted.

Claims (10)

A first element that is one of a capacitor and an inductor connected in series between an input terminal and an output terminal;
A second element that is connected in parallel with the first element between the input terminal and the output terminal, and is the other of the capacitor and the inductor different from the first element;
A third element that is connected in parallel with the first element and in series with the second element between the input terminal and the output terminal and is the other of the capacitor and the inductor different from the first element;
An acoustic wave resonator having one end connected to a first node between the second element and the third element and the other end connected to a ground terminal;
A filter circuit comprising:   2. The filter circuit according to claim 1, wherein the attenuation pole formed by the first element, the second element, and the third element is a low-pass filter having a higher frequency than the attenuation pole formed by the acoustic wave resonator.   2. The filter circuit according to claim 1, wherein the attenuation pole formed by the first element, the second element, and the third element is a high-pass filter having a lower frequency than the attenuation pole formed by the acoustic wave resonator.   Between the input terminal and a second node to which the first element and the second element are commonly connected, and to a third node and the output terminal to which the first element and the third element are commonly connected. 4. The filter circuit according to claim 1, further comprising a matching circuit between at least one of the two. The first element includes a fourth element and a fifth element connected in series between the input terminal and the output terminal,
The filter circuit has one end connected to a fourth node between the fourth element and the fifth element, the other end connected to a ground terminal, and is the same as the first element among a capacitor and an inductor. The filter circuit according to claim 1, further comprising a sixth element.   The filter circuit according to claim 1, wherein the acoustic wave resonator includes a plurality of acoustic wave resonators connected in series or in parallel between the first node and the ground terminal.   The filter circuit according to claim 1, wherein the acoustic wave resonator includes an IDT.   The filter circuit according to claim 1, wherein the acoustic wave resonator includes a piezoelectric thin film resonator.   A multiplexer having the filter circuit according to claim 1. A module comprising the filter circuit according to claim 1.

Images