JP5613813B2 - Duplexer - Google Patents

Description

  The present invention relates to a duplexer used in an electric circuit such as a communication device.

  With the rapid spread of wireless devices such as mobile phones, the demand for duplexers is rapidly expanding. For example, there is a strong demand for a duplexer using an elastic wave device that is small and has high steepness.

  In recent years, a circuit configuration as shown in FIG. 1 has been proposed as a ladder filter (see, for example, Patent Documents 1 to 4). In the configuration of FIG. 1, an inductance LP is connected in parallel to the series resonator S1. Thereby, an attenuation pole can be generated in a region having a frequency lower than that of the filter pass band.

Japanese Patent Laid-Open No. 2003-332885 JP 2003-69382 A JP 2004-135322 JP JP 2004-242281 A

  For example, a duplexer can be configured by combining the above-described filter shown in FIG. 1 (hereinafter referred to as a self filter) and another filter having a different pass band (hereinafter referred to as a counterpart filter). When this duplexer is configured, it is possible to obtain excellent demultiplexing characteristics by adjusting the band in which the attenuation pole of the own filter appears to match the pass band of the counterpart filter.

  However, since the attenuation pole of the self-filter is generated by a resonator with an inductor connected in parallel, if this attenuation pole is placed in the passband of the counterpart filter, the spurious of the resonator with the inductance connected in parallel also passes through the counterpart filter. Placed in the band. Thereby, there is a problem that the loss of the passband of the counterpart filter increases.

  SUMMARY OF THE INVENTION An object of the present invention is to provide a duplexer that suppresses filter loss.

  The duplexer disclosed in the present application includes two or more filters connected to a common terminal and having different pass bands. At least one of the two or more filters includes a plurality of series resonators connected in series to a line connecting an input terminal and an output terminal of the filter, and a parallel resonator connected in parallel to the line. An inductance is connected in parallel to at least one of the resonators. The series resonator in which the inductance is connected in parallel is divided into a plurality of resonators connected in series.

  According to the present disclosure, it is possible to provide a duplexer in which the loss of a filter is suppressed.

The figure which shows the structure of the conventional ladder type filter Diagram for explaining ladder filter Diagram for explaining ladder filter Diagram for explaining ladder filter A diagram showing a structural example of a SAW resonator Diagram showing a configuration example of a boundary wave resonator The figure which shows the structural example of a Lamb wave resonator The figure which shows the structural example of FBAR Configuration example for preventing the dissipation of vibration energy from the excitation part in FBAR 1 is a circuit diagram showing a configuration example of a duplexer according to a first embodiment. Graph showing an example of pass characteristics of filters F1 and F2 The figure for explaining the filter characteristic when the inductance is connected in parallel to the series resonator of the ladder type filter Diagram showing an example of ripple generation The figure which shows the circuit structural example of the duplexer in 2nd Embodiment. The figure which shows the structural example at the time of using an elastic wave resonator as series resonator S1. FIG. 14 is a graph showing characteristics obtained in the duplexer having the circuit configuration shown in FIG. A graph showing the reflection coefficient Γ of the reception filter F2 viewed from the transmission filter F2. The figure which shows the structural example of resonator S11-S13 in case the series resonators S1-S7 shown in FIG. 14 are FBAR. The circuit diagram which shows the structural example of the splitter concerning 3rd Embodiment Diagram showing an example of FBAR manufacturing process The graph which shows the result of measuring the response when the axial ratio is changed variously for the FBAR having an elliptical shape The graph which shows the result of having measured the Q value in the resonance point when changing axial ratio variously about FBAR which has an elliptical shape It is a top view which shows an example of a structure of the filter F2 shown in FIG. An example of an RF module using the duplexer in the first to third embodiments Diagram showing a configuration example of communication equipment Circuit configuration diagram of the duplexer in the fifth embodiment The figure which shows an example of a structure in the case of chip-forming the transmission filter F1 and reception filter which are shown in FIG. The figure which shows the structural example of a duplexer package The figure which shows the structural example at the time of mounting the duplexer package 37 and an inductor on a board | substrate. The figure which shows the structural example of the components which formed inductor L1, L2, L3 on the quartz substrate

  Embodiments of the present invention will be specifically described below with reference to the drawings.

  First, examples of a resonator, a filter, and a duplexer that can be used in this embodiment will be described. The duplexer includes a transmission filter and a reception filter. As a technique for realizing these filters using an acoustic wave element, for example, ladder filters are widely accepted. The ladder filter is a high frequency filter configured by connecting two resonators having different resonance frequencies in a ladder shape. Here, the ladder type filter will be described with reference to FIGS.

2A is a series resonator S connected in series between the input terminal Tin and the output terminal Tout, and FIG. 2B is parallel to the line connecting the input terminal Tin and the output terminal Tout. The parallel resonator P connected to is shown. The series resonator S has a resonance frequency frs and an anti-resonance frequency fas. The parallel resonator P has a resonance frequency of frp and an anti-resonance frequency of fap. FIG. 2C is a graph showing frequency characteristics of the series resonator S and the parallel resonator P. In the example shown in FIG. 2C, fap and frs have substantially the same value.

  FIG. 3A is a diagram showing a circuit configuration of a single-stage filter (a pair of ladder circuits) in which the series resonator S is arranged in the series arm and the parallel resonator P is arranged in the parallel arm. When fap and frs are substantially the same value, the filter characteristics as shown in FIG. 3B are realized by arranging the series resonator S and the parallel resonator P as shown in FIG. .

  The ladder filter is configured by connecting a pair of ladder circuits shown in FIG. 3A in multiple stages. FIG. 4A is a circuit diagram illustrating a configuration example of a ladder filter. In the example shown in FIG. 4A, adjacent ladder circuits are connected in a mirror-inverted form to prevent reflection between the stages. In FIG. 4A, the capacitance of the series-arm resonator is Cs, and the capacitance of the parallel-arm resonator is Cp.

  In the multistage connection of the filters as shown in FIG. 4A, there are locations where the same type of resonators are connected in series in the series arm and locations where the same type of resonators are connected in parallel in the parallel arm. In the series arm and the parallel arm, adjacent resonators of the same type can be collectively constituted by one resonator. For example, as shown in FIG. 4B, adjacent resonators of the same type may be capacitively combined as one resonator. That is, two resonators (each capacitance = Cs) adjacent to each other in the series arm can be configured by one resonator having a capacitance Cs / 2. Two resonators (respective capacitances = Cp) adjacent to each other in the parallel arm can be constituted by one resonator having a capacitance 2Cp.

As an acoustic wave device constituting a ladder filter, a surface acoustic wave (SAW) resonator, a piezoelectric thin film resonator (for example, FBAR: Film Bulk Acoustic Resonator) and the like are widely used.

FIG. 5A is a plan view showing a structural example of the SAW resonator, and FIG. 5B is a cross-sectional view. The SAW resonator includes an IDT (InterDigitated Transducer) 2 serving as an excitation electrode on a piezoelectric substrate 10 and grating reflectors 3 disposed at both ends thereof. An input terminal in and an output terminal out are connected to the IDT by a wiring pattern. The IDT is also called a comb-shaped electrode, and has, for example, a plurality of electrode fingers arranged in parallel at equal intervals, and a bus bar connecting these electrode fingers. The direction in which the electrode fingers are arranged is the propagation direction of the elastic wave.

  As variations of the SAW resonator, for example, there are a boundary wave resonator and a Lamb wave resonator. FIG. 6A is a plan view showing a configuration example of the boundary wave resonator, and FIG. 6B is a cross-sectional view thereof. In the example shown in FIG. 6, the boundary wave resonator is provided with a dielectric film on the piezoelectric substrate 10 so as to cover the IDT 2 and the reflector 3. Fig.7 (a) is a top view which shows the structural example of a Lamb wave resonator, FIG.7 (b) is the sectional drawing. In the example shown in FIG. 7, the IDT 2 and the reflector 3 are provided on the piezoelectric film 16 provided on the substrate 11. A gap 9 is formed between the piezoelectric film 16 and the substrate 11 under the portion (excitation region) where the IDT 2 and the reflector 3 are provided. In the present application, the boundary wave resonator and the Lamb wave resonator are collectively referred to as a SAW resonator, a boundary wave resonator, and a resonator having an IDT such as a Lamb wave resonator as a surface acoustic wave resonator. To do.

  FIG. 8A is a plan view showing a configuration example of the FBAR, and FIG. 8B is a cross-sectional view thereof. In the example shown in FIG. 8, the FBAR includes a piezoelectric film 17 provided on the substrate 10, and an upper electrode 19 and a lower electrode 18 that sandwich the piezoelectric film 17. A region where the upper electrode 19 and the lower electrode 18 face each other with the piezoelectric film 17 interposed therebetween is an excitation portion (resonance region). Immediately below the excitation part is an air gap 9a to prevent the dissipation of vibration energy to the substrate.

  The air gap in the FBAR is of the type formed between the exciter and the substrate in addition to the air gap (via hole) that penetrates the substrate below the exciter as shown in FIG. 8B. A void is included.

  As a method for forming the cavity, for example, a method of forming the gap 9b by processing the substrate surface as shown in FIG. 9A, or a gap 9c on the dome on the substrate as shown in FIG. 9B. There is an air bridge system to be formed.

  Further, for example, as shown in FIG. 9C, in order to prevent the dissipation of vibration energy from the excitation unit, an acoustic multilayer film in which a high impedance layer and a low impedance layer are alternately laminated is used instead of the gap. May be provided. This configuration is called SMR (Solidly Mounted Resonator). In the present application, the piezoelectric thin film resonator includes both FBAR and SMR.

(First embodiment)
FIG. 10 is a circuit diagram illustrating a configuration example of a duplexer according to the first embodiment. The duplexer shown in FIG. 10 includes two filters F1 and F2 having different passbands connected to a common terminal Pcom. The filter F2 includes a plurality of series resonators S1 to Sn connected in series on a line connecting the common terminal Pcom of the filter F2 and the input / output terminal P2 of the filter F2, and a parallel resonator connected in parallel to the line. Including P1 to Pm. An inductance is connected in parallel to at least one of the series resonators S1 to Sn (here, S1 and S4 as an example). The series resonator S1 is divided into a plurality of resonators S11 to S13 connected in series.

  Thus, an attenuation pole can be generated outside the passband of the filter F2 by connecting the inductor in parallel to at least one of the series resonators S1 to Sn. Here, an excellent demultiplexing characteristic can be obtained by adjusting the attenuation band to be included in the pass band of the filter F1 (the counterpart filter). Therefore, a duplexer with reduced filter loss can be obtained. FIG. 11 is a graph showing an example of the pass characteristics of the filters F1 and F2 when the attenuation pole G is included in the passband of the counterpart filter.

  In the present embodiment, the series resonator S1 to which the inductors are connected in parallel is divided (series division) by the plurality of resonators S11 to S13 connected in series, and therefore, the spurious filter F1 in the resonator S1. The influence on the passing characteristics is suppressed. By thus dividing the resonators in series, the capacity, that is, the area of each resonator can be increased. Generally, spurious generated by a resonator is generated by the interaction between a wave propagating in the resonator and the end face of the resonator. Therefore, by enlarging the size of the resonator through series division of the resonators, it is possible to secure a sufficient distance that the wave can propagate in the plane direction without being affected by the end face of the resonator, and to suppress spurious.

  The divided resonators S11 to S13 preferably have different shapes. Thereby, since the band in which the spurious is generated is dispersed, the influence of the spurious on the pass characteristic of the filter can be further suppressed.

  Moreover, it is preferable that the capacitances of the divided resonators S11 to S13 are equal. Thereby, since the electrical energy supplied to each of the divided resonators S11 to S13 becomes substantially constant, the nonlinear response can be suppressed.

  When the resonators S11 to S13 are elastic wave resonators each having an IDT provided on the piezoelectric substrate and reflectors disposed on both sides thereof, the opening lengths of the resonators S11 to S13 are made different from each other. Can do. Accordingly, it is possible to disperse the frequency band in which spurious is generated in the resonators S11 to S13. Furthermore, from the viewpoint of equalizing the electrical energy of the resonators S11 to S13, it is desirable that the product of the logarithm and the aperture length of each resonator S11 to S13 is constant.

  When the resonators S11 to S13 are piezoelectric thin film resonators each having a piezoelectric film and an upper electrode and a lower electrode facing each other with the piezoelectric film interposed therebetween, the resonance regions in the resonators S11 to S13 have different shapes. be able to. Here, the resonance region is a region where the upper electrode and the lower electrode face each other with the piezoelectric film interposed therebetween. Accordingly, it is possible to disperse the frequency band in which spurious is generated in the resonators S11 to S13.

  Alternatively, the resonant frequencies of the divided resonators S11 to S13 may be different from each other. Thereby, the frequency band in which spurious is generated in the resonators S11 to S13 can be dispersed, and the influence of spurious on the filter F1 can be suppressed. For example, the resonance frequency can be changed by changing the membrane pressure of FABAR or changing the area of the resonance region.

  Note that the filter F2 shown in FIG. 10 is an example of a ladder type filter, but the effect similar to the above can be obtained even with a lattice type filter. Further, it is preferable from the viewpoint of filter characteristics that the series resonator to which the inductance is connected in parallel is a series resonator at the end of the ladder type filter or the lattice type filter as in the configuration example shown in FIG. . However, the series resonator to which the inductance is connected in parallel is not necessarily limited to the resonator at the end.

[Description of effects]
Next, the effect of the duplexer according to the present embodiment will be described. FIG. 12 is a diagram for explaining filter characteristics when an inductance is connected in parallel to a series resonator of a ladder type filter. The top graph in FIG. 12 shows the filter characteristics of the ladder type filter shown on the left. The second graph from the top shows the admittance characteristics of the parallel resonator of the ladder filter. The third graph from the top shows the admittance characteristics of the series resonator of the ladder filter. The fourth graph from the top shows the admittance characteristics of a series resonator in which inductances are connected in parallel in the ladder filter.

  In the example shown in FIG. 12, the resonance point of the parallel resonator is f2 and the antiresonance point is f3, and the resonance point of the series resonator is f4 and the antiresonance point is f5. When the inductance is connected in parallel to the series resonator, an antiresonance point is generated at f1 due to the antiresonance of the inductance connected in parallel with the capacitance of the resonator. This anti-resonance point generates an attenuation pole in the filter. Here, the antiresonance point f5 of the series resonator itself is shifted to the high frequency band f6 due to the interaction with the inductance connected in parallel. Further, the resonance point f4 of the series resonator itself does not fluctuate due to the addition of inductance.

  In the example shown in FIG. 12, the resonance point f4 of the series resonator to which the inductance is connected in parallel is not the same as the antiresonance point f3 of the parallel resonator, and f4 and f3 are arranged in the vicinity so that the passband is obtained. Can be widened.

  When the ladder type filter shown in FIG. 12 is adopted as one of the filters of the duplexer, the generated attenuation pole is arranged in the passband of the counterpart filter, so that the demultiplexing characteristic can be improved. it can.

  For example, it is desirable to arrange the attenuation pole arranged in the pass band of the counterpart filter at the center frequency of the counterpart filter pass band. However, for example, WCDMA Band 2 (transmission band: 1850 MHz to 1910 MHz, reception band: 1930 MHz to 1990 MHz) and Band 3 (transmission band: 1710 MHz to 1785 MHz, reception band: 1805 MHz to 1880 MHz) In the case of a specification in which the pass band of the transmission filter and the pass band of the reception filter are close to each other, in order to make it compatible with the steepness of the low frequency end of the own filter pass band, the attenuation pole is more It is desirable to arrange it closer to the high frequency end.

  Here, when the resonator to which the inductance is connected in parallel has spurious, spurious is also generated in the filter characteristics. However, the frequency of spurious generation differs greatly between the pass band and the non-pass band of the filter. This is because, in the passband, the electrical signal propagates mainly in the resonator, which is an elastic wave element, as mechanical vibration, and thus is strongly influenced by spurious due to unnecessary vibration in the elastic wave element. On the other hand, in the non-pass band, the electrical signal is mainly propagated by electromagnetic / electrostatic coupling between the signal wirings, so that the degree of influence of unnecessary vibration in the acoustic wave element is reduced.

  In the acoustic wave element, spurious is generated in a region below the resonance point. For example, the spurious component is arranged in the vicinity of the antiresonance point of the parallel resonator by disposing the resonance point of the series resonator in the non-pass band of the filter. It may be possible to suppress the influence on the filter characteristics.

  However, in the duplexer, when the attenuation pole is arranged in the counterpart band, the spurious of the series resonator to which the inductance is connected in parallel is arranged in the pass band of the counterpart filter. In this case, a ripple is generated in the pass characteristic of the counterpart filter. Therefore, it is difficult to suppress ripples by adjusting the resonance point and antiresonance point of the resonator. Specific examples will be described below.

  FIG. 13 is a diagram illustrating an example of ripple generation in the passband of the counterpart filter caused by the spurious of the series resonator having inductances connected in parallel. The top graph in FIG. 13 is a graph showing an example of admittance characteristics of a series resonator in which inductances are connected in parallel. In this example, spurious is generated between the resonance point and the antiresonance point. The second graph from the top shows the frequency characteristic (solid line) of the attenuation amount of the own filter and the frequency characteristic (dotted line) of the reflection coefficient of the own filter viewed from the other filter. Also in the reflection coefficient, a lip is generated at a position corresponding to the spurious. The third graph from the top shows the frequency characteristic (solid line) of the attenuation amount of the counterpart filter and the frequency characteristic (dotted line) of the reflection coefficient of the own filter viewed from the counterpart filter. In this example, ripples are generated in the passband of the counterpart filter.

  Thus, when the resonator to which the inductance is connected has a spurious, as shown in FIG. 13, the total reflection condition of the reflection coefficient of the own filter viewed from the other filter is locally disturbed and a ripple is generated. Is done. Due to the influence of this ripple, a ripple is also generated in the passband of the counterpart filter.

  In order to keep the loss of the other filter low in the pass band of the other band, it is desirable that the reflection coefficient of the own filter viewed from the other filter is in total reflection (0 dB) as well as the loss of the other filter itself. . However, for example, in the specifications where the pass band of the transmission filter and the reception filter are close, such as WCDMA Band 2 and Band 3, the spurious of the resonator in which the inductance is connected in parallel is arranged in the pass band of the counterpart filter. It is difficult to suppress the ripple by adjusting the frequency arrangement such as the resonance point and the antiresonance point.

  On the other hand, in the duplexer according to the present embodiment, as described above, the series resonator S1 in which the inductors are connected in parallel is divided into the plurality of resonators S11 to S13 connected in series. The influence of the spurious on the pass characteristic of the filter F1 is suppressed. Therefore, even when the attenuation pole of the own filter is arranged in the pass band of the counterpart filter, the ripple in the pass band of the counterpart filter can be suppressed.

(Second Embodiment)
FIG. 14 is a diagram illustrating a circuit configuration example of the duplexer according to the second embodiment. The circuit of FIG. 14 can be used, for example, for a WCDMA Band 2 duplexer. In the example shown in FIG. 14, a transmission filter F1 and a reception filter F2 are connected to a common terminal Ant (port1) connected to the antenna. The reception filter F2 is connected to the balun circuit B1. The balun circuit B1 is a balanced / unbalanced converter, and converts the single output of the reception filter F2 into a balanced output. In the present embodiment, the reception filter F2 having the resonator S1 having inductances connected in parallel is the self filter, and the transmission filter F1 is the counterpart filter.

  The reception filter F2 forms a seven-stage ladder type filter by the series arm serial resonators S1 to S4 and the parallel arm parallel resonators P1 to P4. An inductance L1 is connected in parallel to the first-stage series resonator S1. As in the first embodiment, the series resonator S1 is divided into a plurality of resonators connected in series.

  For example, a surface acoustic wave resonator can be used as the acoustic wave element constituting the series resonators S1 to S7. FIG. 15 is a diagram illustrating a configuration example when a surface acoustic wave resonator is used as the series resonator S1. FIG. 15A shows a configuration example in which all the divided resonators have the same shape. Each of the resonators S11 to S13 includes IDTs 2a to 2c and reflectors 3a to 3c on both sides thereof. The IDTs 2 a to 2 c of the resonators S 11 to S 13 are connected in series by the electrode 4. The opening lengths of the IDTs 2a to 2c and the number of pairs of electrode fingers of the resonators S11 to S13 are the same.

  FIG. 15B is a configuration example when the divided resonators have different shapes. In the resonators S11 to S13 in FIG. 15B, the opening lengths of the IDTs 2a to 2c and the number of electrode fingers of the resonators S11 to S13 are different. In FIG. 15B, the opening lengths of IDTs 2a to 2c are indicated by k1 to k3, respectively, and the number of electrode fingers of IDTs 2a to 2c is indicated by t1 to t3.

  The spurious included in the series resonator S1 to which the inductance L1 of the reception filter F2 is connected in parallel appears as a ripple in the pass characteristic of the transmission filter F1. Therefore, as shown in FIGS. 15A and 15B, the resonator S1 having the inductance L1 connected in parallel is divided into the resonators S11 to S13 connected in series to suppress this spurious. can do. Further, as shown in FIG. 15B, this spurious can be further suppressed by dividing the resonator S1 and making the number of electrode fingers and the opening length of each of the divided resonators different. it can.

  Here, when the resonator S1 to which the inductance L1 of the reception filter F2 is connected in parallel is in the first stage, in particular, in order to suppress the nonlinear response, the electric energy supplied to each of the divided resonators S11 to S13 is It is desirable to be constant. Therefore, it is desirable that the capacitances of the resonators S11 to S13 are the same. Specifically, the product of the logarithm and the aperture length of each resonator S11 to S13 can be the same. That is, the logarithm and the aperture length of each resonator S11 to S13 can be adjusted so that k1 * t1 = k2 * t2 = k3t3.

[Description of effects]
FIG. 16 is a graph showing characteristics obtained in the duplexer having the circuit configuration shown in FIG. The graph shown in FIG. 16A shows an example of the transmission characteristic S12 of the transmission filter F1 and the reflection coefficient Γ of the reception filter F2 viewed from the transmission filter F1. FIG. 16B is an enlarged view of the end of the transmission filter pass band in FIG. S12 and Γ shown in the graph of FIG. 16 can be calculated using, for example, an electromagnetic field simulation by a finite element method.

Than 16, in the pass characteristics S ₁₂ of the transmitting filter F1, the passband ends, it is confirmed that the ripple R1 is generated. The frequency at which the ripple R1 is generated substantially matches the ripple generation frequency in the reflection coefficient of the reception filter F2.

FIG. 17A shows the case where the shapes of the resonators S11 to S13 are different as shown in FIG. 15B (D), and the shape of the resonators S11 to S13 as shown in FIG. It is a graph which shows the reflection coefficient (GAMMA) of the reception filter F2 seen from the transmission filter F2 in the same case (W). 17 (b) is a graph showing transmission characteristics S ₁₂ of the transmission filter F1 in the same case as when the shape of each resonator S11~S13 are different. Here, the graphs of FIG. 17A and FIG. 17B are analysis results when the logarithm and aperture length of each resonator are the values shown in Table 1 below.

図１７（ａ）および図１７（ｂ）のグラフより、受信フィルタＦ２の分轄された共振器Ｓ１１〜Ｓ１３の開口長および対数を異ならせることによって、受信フィルタＦ２の反射係数が改善され、送信フィルタＦ１の通過帯域におけるリップルが軽減されることを確認することができる。

From the graphs of FIG. 17A and FIG. 17B, the reflection coefficient of the reception filter F2 is improved by changing the aperture length and the logarithm of the resonators S11 to S13 to which the reception filter F2 is divided, and the transmission filter It can be confirmed that the ripple in the passband of F1 is reduced.

[Configuration example for FBAR]
FIG. 18 is a diagram illustrating a configuration example of the resonators S11 to S13 when the series resonators S1 to S7 illustrated in FIG. 14 are FBARs. FIGS. 18A to 18C are views of the FBAR as viewed from above. A region (resonance region) where the upper electrode and the lower electrode face each other with the piezoelectric film interposed therebetween is indicated by hatching. In the example shown in FIGS. 18A to 18C, the divided resonators S11 to S13 have different resonance region shapes. In the example shown in FIG. 18A, the resonance regions of the resonators S11 to S13 are all square. The areas (S1, S2, S3) of the resonance regions of the resonators S11 to S13 are all the same (S1 = S2 = S3). Thereby, the electrostatic capacitance of each resonator S11-S13 becomes the same.

  In the example shown in FIG. 18B, the resonance region of the resonators S11 to S13 is an ellipse. In the resonance regions of the resonators S11 to S13, the ratio between the major axis a and the minor axis b is different from each other. As a result, the frequency at which spurious is generated is different for each of the resonators S11 to S13. Therefore, the spurious of the resonators S11 to S13 is dispersed, and the influence of the spurious on the pass characteristic of the transmission filter F1 is reduced.

  Here, in the resonators S11 to S13, the capacitances of the resonators S11 to S13 can be made the same by forming the products of the major axis and the minor axis of the ellipse in the resonance region to be the same. it can. That is, the major axes a1, a2, a3 and the minor axes b1, b2, b3 of the ellipses of the resonance regions in the resonators S11 to S13 are determined so that a1 × b1 = a2 × b2 = a3 × b3. It is preferable.

  In the example shown in FIG. 18C, the resonance region of the resonators S11 to S13 is a rectangle. The ratio of the long side a and the short side b of the rectangle is different for each of the resonators S11 to S13. Thereby, the spurious of resonator S11-S13 is disperse | distributed, and the influence which the spurious has on the passage characteristic of transmission filter F1 is suppressed. Further, the product of the long side a and the short side b is the same (a1 × b1 = a2 × b2 = a3 × b3). As described above, the rectangular major axes a1, a2, and a3 and the minor axes b1, b2, and b3 of the resonance regions in the resonators S11 to S13 are determined, so that the capacitances of the resonators S11 to S13 have the same capacitance. Can be.

  As illustrated in FIG. 18, even when the resonator used for the filter is an FBAR, it is possible to adjust the spurious depending on the shape of the FBAR. Also in the FBAR, it is desirable that the capacitances of the resonators to be divided are the same, and it is desirable that the area of the region where the upper electrode and the lower electrode face each other is equal.

(Third embodiment)
FIG. 19 is a circuit diagram illustrating a configuration example of the duplexer according to the third embodiment. The duplexer shown in FIG. 19 includes two filters F1 and F2 having different pass bands connected to a common terminal Pcom. The filter F2 includes a plurality of series resonators S1 to S4 connected in series on a line connecting the common terminal Pcom of the filter F2 and the input / output terminal P2 of the filter F2, and a parallel resonator connected in parallel to the line. Includes P1-P4. Inductances L1 and L2 are connected in parallel to at least one of the series resonators S1 to S4 (here, S1 and S4 as an example). The series resonators S1 to S4 are piezoelectric thin film resonators (for example, FBAR) each having a resonance region in which an upper electrode and a lower electrode face each other with a piezoelectric film interposed therebetween. The ratio (A / B) between the longest width A and the shortest width B in the electrode plane direction of the resonance region of the series resonator S1 to which the inductance L1 is connected in parallel is determined in the resonance regions of the other series resonators S2 to S4. It is smaller than the ratio (A / B) between the longest width A and the shortest width B.

  Here, an example of the manufacturing process of the FBAR will be described with reference to FIGS. Referring to FIG. 20A, for example, a Ru film as a lower electrode 18 is formed on a substrate 10 mainly made of Si by using a sputtering method in an Ar gas atmosphere under a pressure of 0.6 to 1.2 Pa. Form. Thereafter, the lower electrode 18 is formed into a predetermined shape using an exposure technique and an etching technique.

Referring to FIG. 20B, an AlN film as a piezoelectric film 17 is formed on the substrate 10 and the lower electrode 18 by using a sputtering method in an Ar / N ₂ mixed gas atmosphere under a pressure of about 0.3 Pa. . A Ru film is formed as an upper electrode 19 on the piezoelectric film 17 using a sputtering method in an Ar gas atmosphere under a pressure of 0.6 to 1.2 Pa.

  Referring to FIG. 20C, a region where the lower electrode 18 and the upper electrode 19 face each other with the piezoelectric film 17 interposed therebetween by etching from the back surface of the substrate 10 using a deep-RIE (reactive dry etching) method. A gap 9a is formed in the substrate 10 under the region. FBAR is formed by the above process.

Next, the relationship between the shape of the resonance region of the FBAR and the spurious and the relationship between the shape of the resonance region and the loss will be described. FIG. 21 shows the results of measuring the response of the FBAR having an elliptical resonance region when the axial ratio of the resonance region is variously changed. Here, the major axis of the ellipse forming the resonance region is a, the minor axis is b, and the axial ratio is a / b. In the measurement, S ₁₁ (reflection coefficient) was measured for a two-terminal resonator, and the dotted line in the figure represents the resonance point of the FBAR. The measurement result shown in FIG. 21 is a measurement result of FBAR having a film configuration of Ru upper electrode 250 nm, AlN piezoelectric thin film 1200 nm, Ru lower electrode 250 nm, and a capacitance of 0.75 pF. 21A shows the measurement results when the axial ratio is 10: 5, FIG. 21B shows the measurement results when the axial ratio is 8: 5, and FIG. 21C shows the measurement results when the axial ratio is 6: 5. ing. It can be seen that the smaller the axial ratio, the smaller the spurious below the resonance point.

  In FBAR, spurious are generated mainly by Lamb waves that propagate in the electrode plane. Spurious is less attenuated as the propagation distance is shorter, and its magnitude becomes more prominent. Therefore, in the case of an elliptical resonator, the length of the short axis with the shortest propagation distance greatly affects the size of the spurious. When comparing with the same electrode area, the smaller the axial ratio, the longer the short axis, and the spurious is suppressed and evaluated.

  FIG. 22 shows the result of measuring the Q value at the resonance point when the axial ratio is variously changed for the FBAR having an elliptical shape. From the result of FIG. 22, it is confirmed that the Q at the resonance point is larger as the axial ratio is larger. In a ladder type filter composed of a series resonator and a parallel resonator, it is important that the series resonator has a high resonance Q. When the FBAR is used, it is desirable that the axial ratio of the series resonator is large.

  Here, the series resonator S1 to which the inductance L1 is connected in parallel does not necessarily have a high Q value at the resonance point by disposing the resonance point in the vicinity of the antiresonance point of the parallel resonator. Rather, when a duplexer is assumed, it is desirable that the spurious that adversely affects the reflection coefficient is sufficiently suppressed. Thus, it can be said that there is a trade-off relationship between the spurious and the loss for the preferable shape of the resonance region of the series resonator S1. Therefore, it is preferable to configure the resonance region so that the axial ratio of the series resonator S1 to which the inductance L1 is connected in parallel is smaller than the axial ratio of the other series resonators.

  As described above, regarding the series resonator S1 to which the inductance L1 is connected in parallel, the axial ratio can be adjusted from the viewpoint that spurious should be more important than loss in terms of frequency. As a result, it is possible to design appropriately considering both the reduction of spurious and the Q value. Although the present embodiment has been described with respect to an elliptical FBAR, the same effect can be obtained with respect to a rectangular shape and other polygonal shapes.

  FIG. 23 is a plan view showing an example of the configuration of the filter F2 shown in FIG. The series resonators S1 to S4 and the parallel resonators P1 to P3 are formed in a resonance region where the upper electrode 4 and the lower electrode 2 face each other with the piezoelectric film 3 interposed therebetween. In FIG. 23, the position of the lower electrode 2 is indicated by a dotted line. These seven resonators have, for example, a dome-shaped cavity (not shown) between the substrate and the resonator. For the parallel resonators P1 to P3 of the parallel arm, 110 titanium (Ti) may be provided on the 300 nm ruthenium (Ru) film of the upper electrode.

  In the example shown in FIG. 23, the lower electrode 2 of the first stage series resonator S1 is electrically connected to the pad 5a above the piezoelectric film 3, and is between the upper electrode 4 of the series resonator S1 and the upper electrode 4 of S2. The wiring pattern is electrically connected to the pad 5b. An inductance L1 is connected between the pad 5a and the pad 5b.

The ratio (a ₁ / b ₁ ) between the major axis a ₁ and the minor axis b ₁ in the resonance region of the series resonator S1 to which the inductance L1 is connected in parallel is the major axis a in the resonance region of the other series resonators S2 to S4. ₂ and the minor axis b ₂ ratio (a ₂ / b ₂ ). In this way, by making the axial ratio of the series resonator with the inductance connected in parallel to the axial ratio of other series resonators, the trade-off between the resonator spurious and the loss resonator shape is taken into consideration. Appropriate design becomes possible.

(Fourth embodiment)
FIG. 24 is an example of an RF module using the duplexer in the first to third embodiments. In the example shown in FIG. 24, the duplexer is used as a part of a filter bank module (duplexer bank module). The RF module shown in FIG. 24 includes a switch module 6, a duplexer bank module 7, and an amplifier module 8. The duplexer bank module 7 includes a plurality of duplexers D1, D2, D3,... With different frequency bands. The switch module 6 switches a duplexer that handles signals transmitted and received by the antenna. That is, the switch module 6 selects which one of the duplexers D1, D2, D3,. The amplifier module 8 amplifies signals output from the duplexers D1, D2, D3,... Or signals input to the duplexers D1, D2, D3,.

  The duplexers D1, D2, D3,... Each include a transmission filter F1 and a reception filter 2. The duplexers shown in the first to third embodiments are used for the duplexers D1, D2, D3,. Thus, an RF module having excellent characteristics can be obtained by configuring an RF module using a duplexer having excellent demultiplexing characteristics while suppressing filter loss.

  The module using the duplexer is not limited to the example shown in FIG. For example, a case where the duplexer is combined with an amplifier module (Amp. Module) or a switch module (SW. Module) to be modularized can be assumed. Furthermore, a communication device including a duplexer is also included in the embodiment of the present invention.

  FIG. 25 is a diagram illustrating a configuration example of a communication device. In the communication device 50 shown in FIG. 25, a communication module 60, an RFIC 53, and a baseband IC 54 are provided on a module substrate 51. The duplexer shown in the above embodiment can be used for the communication module 60. For example, the communication module 60 may include the RF module shown in FIG.

  The transmission terminal Tx of the communication module 60 is connected to the RFIC 53, and the reception terminal Rx is also connected to the RFIC 53. The RFIC 53 is connected to the baseband IC 54. The RFIC 53 can be formed of a semiconductor chip and other components. The RFIC 53 is integrated with a circuit including a reception circuit for processing a reception signal input from a reception terminal and a transmission circuit for processing a transmission signal.

  The baseband IC 54 can also be realized by a semiconductor chip and other components. The baseband IC 54 includes a circuit for converting the reception signal received from the reception circuit included in the RFIC 53 into an audio signal and packet data, and a transmission circuit included in the RFIC 53 by converting the audio signal and packet data into a transmission signal. Is integrated with the circuit for output.

  Although not shown, output devices such as a speaker and a display are connected to the baseband IC 54, and output audio signals and packet data converted from received signals by the baseband IC 54, to the user of the communication device 50. Can be recognized. In addition, input devices included in the communication device 50 such as a microphone and a button are also connected to the baseband IC 54, and the baseband IC 54 can convert voice and data input from the user into transmission signals. . Note that the configuration of the communication device 50 is not limited to the example shown in FIG.

(Fifth embodiment)
In the present embodiment, a mounting example of the duplexer described in the first to fourth embodiments will be described. Here, as an example, a case where the duplexer having the circuit configuration shown in FIG. 26 is mounted will be described. In the duplexer shown in FIG. 26, the transmission filter F1 and the reception filter F2 are connected to the antenna terminal Ant. The transmission filter F1 is a ladder type filter composed of series resonators S1 to S4 and parallel resonators P1 to P2. The reception filter F2 includes a ladder filter LF1, a lumped constant balun B1, and a lattice filter LF2. The ladder type filter LF1 includes series resonators S5 to S7 and parallel resonators P3 to P5.

  The ladder filter LF1 inputs a signal from the antenna terminal Ant via a single terminal and outputs the signal to the balun B1. The balun B1 converts the single signal from the ladder filter LF1 into a balance signal. The lattice filter LF2 is a balance input / output filter that inputs a balance signal and outputs a balance signal. An inductance L1 is connected in parallel to the first stage series resonator S5 of the ladder filter LF1. The series resonator S5 and the inductance L1 form a matching circuit. As in the first and second embodiments, the series resonator S5 may be divided into a plurality of resonators, or may have a different shape from other series resonators as in the third embodiment. Good. In the present embodiment, in order to simplify the description, the division of the resonator and the change in shape are not shown.

  FIG. 27A is a diagram illustrating an example of a configuration when the transmission filter F1 illustrated in FIG. 26 is formed into a chip. In the example shown in FIG. 27A, series resonators S1 to S4 and parallel resonators P1 and P2 are formed by the IDT and the reflector on the piezoelectric substrate, and further, bumps and transmission terminals connected to the antenna terminal Ant Bumps connected to Tx and ground (GND) terminals are formed. These resonators S1 to S4, P1, P2, bumps and ground terminals are connected by a wiring pattern on the piezoelectric substrate.

  FIG. 27B is a diagram illustrating an example of a configuration when the reception filter F2 illustrated in FIG. Similarly, in the example shown in FIG. 27B, resonators S5 to S10 and P3 to P8 are formed by the IDT and the reflector on the piezoelectric substrate, and further, bumps connected to the antenna terminal Ant, and for the matching circuit The bumps connecting the inductors L1, the bumps connecting the inductors L2, L3 of the balun B1, the bumps connected to the receiving terminals Rx1, Rx2, and the ground terminal are formed. In the example shown in FIG. 27B, the matching circuit inductor L1 and the balun 2 inductors L2 and L3 are provided outside the chip.

  FIG. 28 is a diagram illustrating a configuration example of a duplexer package in which the transmission filter chip of FIG. 27A and the reception filter chip of FIG. 27B are packaged. In the example shown in FIG. 28, a transmission filter chip 33 and a reception filter chip 34 are flip-chip mounted on a ceramic package 36 having a cavity 29. The transmission filter chip 33 is a chip (FIG. 27A) that forms the transmission filter F1, and the reception filter chip 34 is a chip (FIG. 27B) that forms the reception filter F2. Each of these chips and the ceramic package 36 are electrically connected by, for example, Au bumps. A metal lid 27 is provided as a cap on the ceramic package 36 on which these chips are mounted. Thereby, the cavity 29 is hermetically sealed. The package configuration is not limited to the example shown in FIG. For example, an IPD chip constituting the balun B1 may be separately prepared and mounted in the cavity 29.

  FIG. 29 is a diagram illustrating a configuration example when the duplexer package 37 and the inductor illustrated in FIG. 28 are mounted on a substrate. In the example shown in FIG. 29, the duplexer package 37 is mounted on the printed circuit board 38. On the printed board 38, chip components such as inductors L1, L2, and L3 are also mounted. The inductors L 1, L 2, L 3 and the bumps of the duplexer package 37 are connected by the wiring pattern on the printed circuit board 38. Further, the transmission terminal Tx, the reception terminals Rx1, Rx2, and the ground terminal of the duplexer package 37 are also drawn out by the wiring pattern on the printed circuit board 38.

  In addition, the mounting form of a splitter is not restricted to the said example. For example, the inductors L1, L2, and L3 can be combined into parts. FIG. 30 is a diagram illustrating a configuration example of a component in which inductors L1, L2, and L3 are formed on a quartz substrate. In the example shown in FIG. 30, three spiral coils are formed on a quartz substrate 41, and the terminals of each spiral coil are pads.

  Regarding the above embodiment, the following additional notes are disclosed.

(Appendix 1)
Comprising two or more filters connected to a common terminal and having different passbands;
At least one of the two or more filters includes a plurality of series resonators connected in series to a line connecting the input terminal and the output terminal of the filter, and a parallel resonator connected in parallel to the line,
An inductance is connected in parallel to at least one of the series resonators,
The series resonator in which the inductance is connected in parallel is divided into a plurality of resonators connected in series.

(Appendix 2)
The duplexer according to appendix 1, wherein the divided resonators have different shapes.

(Appendix 3)
The duplexer according to appendix 1 or 2, wherein capacitances of the plurality of divided resonators are equal.

(Appendix 4)
The plurality of divided resonators are surface acoustic wave resonators each having an IDT provided on a piezoelectric substrate and reflectors disposed on both sides thereof.
The duplexer according to any one of appendices 1 to 3, wherein opening lengths of the plurality of distributed surface acoustic wave resonators are different from each other.

(Appendix 5)
The divided plurality of resonators are piezoelectric thin film resonators each having a piezoelectric film and an upper electrode and a lower electrode facing each other with the piezoelectric film interposed therebetween,
The duplexer according to any one of appendices 1 to 3, wherein shapes of regions in which the upper electrode and the lower electrode face each other across the piezoelectric film are different from each other in each of the divided piezoelectric thin film resonators. .

(Appendix 6)
The divided plurality of resonators are piezoelectric thin film resonators each having a piezoelectric film and an upper electrode and a lower electrode facing each other with the piezoelectric film interposed therebetween,
4. The duplexer according to any one of appendices 1 to 3, wherein the resonance frequencies of the piezoelectric thin film resonators are different from each other.

(Appendix 7)
With two or more filters with different passbands,
At least one of the two or more filters includes a plurality of series resonators connected in series to a line connecting the input terminal and the output terminal of the filter, and a parallel resonator connected in parallel to the line,
An inductance is connected in parallel to at least one of the series resonators,
The series resonator is a piezoelectric thin film resonator having a resonance region in which an upper electrode and a lower electrode face each other with a piezoelectric film interposed therebetween,
The ratio (A / B) of the longest width A and the shortest width B in the electrode plane direction of the resonance region of the series resonator in which the inductance is connected in parallel is the longest width A in the resonance region of the other series resonator. Is the smaller than the ratio (A / B) of the shortest width B.

(Appendix 8)
The filter having the series resonator is a ladder filter or a lattice filter,
The duplexer according to any one of appendices 1 to 7, wherein the series resonator in which the inductances are connected in parallel is the foremost resonator of the ladder filter or the lattice filter.

(Appendix 9)
A communication module including the duplexer according to any one of appendices 1 to 8.

(Appendix 10)
A communication device including the duplexer according to any one of appendices 1 to 8.

Claims (2)

With two or more filters with different passbands,
At least one of the two or more filters includes a plurality of series resonators connected in series to a line connecting the input terminal and the output terminal of the filter, and a parallel resonator connected in parallel to the line,
An inductance is connected in parallel to at least one of the series resonators,
The series resonator is a piezoelectric thin film resonator having a resonance region in which an upper electrode and a lower electrode face each other with a piezoelectric film interposed therebetween,
The ratio (A / B) of the longest width A and the shortest width B in the electrode plane direction of the resonance region of the series resonator in which the inductance is connected in parallel is the longest width A in the resonance region of the other series resonator. Is the smaller than the ratio (A / B) of the shortest width B. The filter having the series resonator is a ladder filter or a lattice filter,
2. The duplexer according to claim 1, wherein the series resonator to which the inductance is connected in parallel is a resonator at the foremost stage of the ladder filter or the lattice filter.

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