JP2017158245A - Motor controller and electrically-driven power steering device with the same mounted therein - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a power controller which prevents interference even if single system drive and double system drive are switched, by interposing such a filter that transfer function properties of the single system drive and the double system drive become equal when controlling drive of a motor including two system coils with two control sections each including a motor counter electromotive voltage compensation function, and an electrically-driven power steering device which improves a steering feeling.SOLUTION: A motor controller comprises control sections 1 and 2 each for controlling driving of a motor including two system coils for each of the two system coils. The control sections 1 and 2 control driving of the motor under current feedback control based on a current command value and compensate for a motor counter electromotive voltage in accordance with a counter electromotive voltage compensation signal based on a motor angle or a motor angular velocity. In each path for compensating for the motor counter electromotive voltage, an LPF for noise cancellation is provided and in a feedback path, an angle feedback filter with which transfer function properties of the single system drive and the double system drive become equal is provided.SELECTED DRAWING: Figure 13

Description

  The present invention is equipped with a motor control device that drives and controls a motor having two windings with two control units, and at least a current command value calculated based on a steering torque. The present invention relates to an electric power steering device that applies assist force. In particular, each of the two control units is equipped with a low-pass filter (LPF) for noise removal and has a back electromotive force compensation function that compensates for the motor back electromotive force (EMF). The present invention relates to a motor control device in which an angle feedback filter serving as a function characteristic is provided in a feedback path to suppress interference due to switching, and an electric power steering device equipped with the motor control device to improve steering feeling.

  An electric power steering device (EPS) equipped with a motor control device applies steering assist force (assist force) to the steering mechanism of the vehicle by the rotational force of the motor, and the driving force of the motor controlled by the inverter is A steering assist force is applied to the steering shaft or the rack shaft by a transmission mechanism such as a gear. Such a conventional electric power steering apparatus performs feedback control of the motor current in order to accurately generate the torque of the steering assist force. In feedback control, the motor applied voltage is adjusted so that the difference between the steering assist command value (current command value) and the motor current detection value is small. The adjustment of the motor applied voltage is generally performed by PWM (pulse width). This is done by adjusting the duty of modulation) control.

The general configuration of the electric power steering apparatus will be described with reference to FIG. 1. A column shaft (steering shaft, handle shaft) 2 of a handle 1 is a reduction gear 3, universal joints 4a and 4b, a pinion rack mechanism 5, a tie rod 6a, 6b is further connected to the steering wheels 8L and 8R via hub units 7a and 7b. Further, the column shaft 2 is provided with a torque sensor 10 for detecting the steering torque Th of the handle 1 and a steering angle sensor 14 for detecting the steering angle θ, and a motor 20 for assisting the steering force of the handle 1 is a reduction gear. 3 is connected to the column shaft 2 through 3. The control unit (ECU) 30 that controls the electric power steering apparatus is supplied with electric power from the battery 13 and also receives an ignition key signal via the ignition key 11. The control unit 30 calculates a current command value of an assist (steering assist) command based on the steering torque Th detected by the torque sensor 10 and the vehicle speed Vs detected by the vehicle speed sensor 12, and compensates the current command value. The current supplied to the EPS motor 20 is controlled by the voltage control command value Vref subjected to.
Note that the steering angle θ is detected from the steering angle sensor 14, and the steering angle can be acquired from a rotation sensor such as a resolver connected to the motor 20.
The control unit 30 is connected to a CAN (Controller Area Network) 40 that transmits and receives various types of vehicle information, and the vehicle speed Vs can also be received from the CAN 40. The control unit 30 can be connected to a non-CAN 41 that exchanges communications, analog / digital signals, radio waves, and the like other than the CAN 40.

  The control unit 30 is mainly composed of a CPU (including an MPU, MCU, etc.). FIG. 2 shows general functions executed by a program inside the CPU.

  Referring to FIG. 2, the control unit 30 will be described. The steering torque Th detected by the torque sensor 10 and the vehicle speed Vs detected by the vehicle speed sensor 12 (or from the CAN 40) are a current command for calculating a current command value Iref1. The value is input to the value calculation unit 31. The current command value calculation unit 31 calculates a current command value Iref1, which is a control target value of the current supplied to the motor 20, using an assist map or the like based on the input steering torque Th and vehicle speed Vs. The current command value Iref1 is input to the current limiting unit 33 via the adding unit 32A, and the current command value Irefm with the maximum current limited is input to the subtracting unit 32B, and the deviation ΔI (= Irefm−Im) is calculated, and the deviation ΔI is input to the PI control unit 34 for improving the characteristics of the steering operation. The voltage control command value Vref whose characteristics are improved by the PI control unit 34 is input to the PWM control unit 35, and the motor 20 is further PWM driven via the inverter 36. The current value Im of the motor 20 is detected by the motor current detector 37 and fed back to the subtraction unit 32B. The inverter 36 is composed of an FET bridge circuit as a semiconductor switching element.

  A rotation sensor 21 such as a resolver is connected to the motor 20, a motor rotation angle θ is output from the rotation sensor 21, and a motor speed ω is calculated by a motor speed calculation unit 22.

  Further, the compensation signal CM from the compensation signal generation unit 38 is added to the addition unit 32A, and the compensation of the steering system system is performed by adding the compensation signal CM to improve the convergence property, the inertia property, and the like. ing. The compensation signal generator 38 adds the self-aligning torque (SAT) 38-1 and the inertia 38-2 by the adder 38-4, and further adds the convergence 38-3 to the addition result by the adder 38-5. The addition result of the adder 38-5 is used as the compensation signal CM.

  In such a motor control device of an electric power steering device, since the motor 20 generates a counter electromotive voltage when the motor is driven, compensation for suppressing or attenuating the motor counter electromotive voltage is required. The reason will be described below.

A control system in which the motor 20 is driven from the current command value Iref is represented by a transfer function as shown in FIG. The current command value Iref is input to the subtraction unit 104 via the control filter (G _FF ) 101, and a deviation e _{1 from the} actual motor current Imn is calculated. The deviation e ₁ is input to the subtraction unit 105 through the control filter (G _FB ), the back electromotive voltage EMF of the motor 20 is subtracted by the subtraction unit 105, and the difference e ₃ is obtained as the electric system characteristic unit 110 (1 / (L · s + R)), and further, the torque constant Kt [Nm / A] is inputted to the mechanical system characteristic section 120 (1 / (J · s + D)). The counter electromotive voltage EMF is obtained by multiplying the motor angular velocity (motor rotational speed) ωm, which is an output of the mechanical characteristic unit 120, by a counter electromotive voltage constant Ke [V / (rad / s)]. The motor current Im from the electric system characteristic unit 110 is detected and fed back, but actually, the current detection noise Ni is mixed and fed back as the actual motor current Imn.

The electrical system characteristic unit 110 is expressed by the following equation 1, and the mechanical system characteristic unit 120 is expressed by the following equation 2. L in the electric system characteristic unit 110 is motor correlation inductance [H], R is motor correlation resistance [Ω], J in the mechanical system characteristic unit 120 is motor moment of inertia [Kg · m ² ], and D is motor viscosity coefficient [ Nm / (rad / s)].
In order to set the system from the current command value Iref to the motor current Im as a primary filter (1 / (T ₄ · s + 1)) whose frequency band is easy to control as shown in FIG. 4, T _{1 to} T ₄ are time constants. The transfer function of the control filter (G _FF ) 101 is set by the following formula 1, and the transfer function of the control filter (G _FB ) 102 is set by the following formula 2.

即ち、電流指令値Ｉｒｅｆからモータ電流Ｉｍまでの系を“１／（Ｔ_４・ｓ＋１）”（数３）とし、時定数Ｔ_４の設定で電流指令値Ｉｒｅｆからモータ電流Ｉｍまでの特性である電流制御帯域を決定（１次フィルタ）するために、制御フィルタ（Ｇ_ＦＦ）１０１及び制御フィルタ（Ｇ_ＦＢ）１０２の特性を決定する。

That is, the system from the current command value Iref to the motor current Im is “1 / (T ₄ · s + 1)” (Equation 3), and the characteristic from the current command value Iref to the motor current Im is set by setting the time constant T _4. In order to determine the current control band (primary filter), the characteristics of the control filter (G _FF ) 101 and the control filter (G _FB ) 102 are determined.

そして、制御フィルタ（Ｇ_ＦＢ）１０２を含む電流制御のフィードバック閉ループの系では、逆起電圧ＥＭＦとＥＭＦ補償を０として、数４及び図５に示すように、外乱の電流検出ノイズ（Ｎｉ）除去用の２次フィルタ（時定数Ｔ_２，Ｔ_３）と１次位相進みフィルタ（Ｔ_１）となる構成を採る。１次位相進みフィルタ（Ｔ_１）の項がないと、制御フィルタ（Ｇ_ＦＦ）１０１の分子の微分項が２次になってしまい、非常に不安定になるからである。要するに、制御フィルタ（Ｇ_ＦＦ）１０１は、電流指令値Ｉｒｅｆからモータ電流Ｉｍまでの特性を決めるフィルタであり、制御フィルタ（Ｇ_ＦＢ）１０２は、電流検出ノイズＮｉを除去するフィルタを含めた、電流指令値Ｉｒｅｆからモータ電流Ｉｍまでの特性を決めるフィルタと言える。

In the current control feedback closed loop system including the control filter (G _FB ) 102, the back electromotive force EMF and EMF compensation are set to 0, and the disturbance current detection noise (Ni) is removed as shown in Equation 4 and FIG. The second-order filter (time constants T ₂ and T ₃ ) and the first-order phase advance filter (T ₁ ) are used. This is because if there is no term of the primary phase advance filter (T ₁ ), the differential term of the numerator of the control filter (G _FF ) 101 becomes second order and becomes very unstable. In short, the control filter (G _FF ) 101 is a filter that determines the characteristics from the current command value Iref to the motor current Im, and the control filter (G _FB ) 102 is a current including a filter that removes the current detection noise Ni. It can be said that the filter determines the characteristics from the command value Iref to the motor current Im.

上記数１及び数４をＧ_ＦＢについて解くと、制御フィルタ（Ｇ_ＦＢ）１０２は上記数２となる。

When Equation 1 and Equation 4 are solved for G _FB , the control filter (G _FB ) 102 becomes Equation 2 above.

As described above, the setting of the control filter (G _FF ) 101 and the control filter (G _FB ) 102 does not consider the mixing of the motor back electromotive force EMF and EMF compensation. However, the generation of the motor back electromotive voltage EMF affects the actual current, and thus becomes a serious problem in accurately controlling the motor output. As such a countermeasure, compensation of the motor back electromotive force voltage is conceivable. For example, in Japanese Patent Laid-Open No. 2013-21870 (Patent Document 1), the back electromotive force voltage of the motor is estimated based on the voltage command value and the current detection value, The estimated back electromotive force value is added to the current command value. In Japanese Patent Laid-Open No. 2012-236472 (Patent Document 2), a counter electromotive voltage is estimated based on the rotational angular velocity, and the counter electromotive voltage compensation control value is calculated by multiplying the estimated counter electromotive voltage by a compensation coefficient. A voltage command value is obtained by adding the electromotive force compensation control value to the basic voltage.

  Further, in recent years, there has been a great demand for voice of clients for multi-system control including safety of an electric power steering device and functional sustainability in the event of an EPS failure. In that case, in order to increase the safety of the motor, a motor with multiple motor windings has appeared. FIG. 6 shows a Y-connected three-phase motor. One system is composed of a U-phase winding UW1, a V-phase winding VW1, and a W-phase winding WW1, and the other one is a U-phase winding UW2, V It consists of a phase winding VW2 and a W-phase winding WW2. The motor is driven by passing a three-phase current through the windings UW1 to WW1 or the windings UW2 to WW2. FIG. 7 shows a Δ-connected three-phase motor. One system is constituted by a U-phase winding UW1, a V-phase winding VW1, and a W-phase winding WW1, and the other one is a U-phase winding UW2. , V-phase winding VW2 and W-phase winding WW2. The motor is driven by passing a three-phase current through the windings UW1 to WW1 or the windings UW2 to WW2.

  Moreover, as a control structure, a control part is provided for every two system windings, and a motor is driven by each control part. In the case of such a two-system control configuration, there is a difference in motor torque between when both systems drive simultaneously driving the two systems and when one system fails when one system fails. May cause discomfort to the driver. The reason is that although the motor outputs are the same (one output), one signal of back electromotive voltage compensation is fed back to each control system, and thus interferes with each other.

  As the two control systems, for example, an apparatus disclosed in Japanese Patent Laid-Open No. 2008-271624 (Patent Document 3) has been proposed. The device of Patent Document 3 controls the phase difference change driving means so as to eliminate the deviation according to the deviation between the target value of the induced voltage constant parameter of the motor and the observed value, and based on the deviation, Means for determining the presence or absence of an abnormality in the phase difference changing operation, and the integral value of the deviation is used to determine the presence or absence of the abnormality.

JP 2013-2119880 A JP 2012-236472 A JP 2008-271624 A

  However, both of the back electromotive force compensations shown in Patent Documents 1 and 2 are compensated on the assumption that noise is not mixed. However, since noise is actually mixed, the current command value is compared with the current command value. There is a problem that the motor output cannot be accurately followed.

  Further, in the device of Patent Document 3, since the deviation is integrated, an error may be accumulated, and noise control and compensation of the motor back electromotive voltage are not taken into consideration, so that the balance of the control system may be lost. There is sex.

  The present invention has been made under the circumstances as described above, and an object of the present invention is to compensate for a motor back electromotive voltage and to control a motor having two system windings with two functions having a motor back electromotive force compensation function. In the case where the drive control is performed by a part, even if the single-system drive and the dual-system drive are switched to each other by inserting a filter that has the same transfer function characteristics in the single-system drive and the dual-system drive in the control system. An object of the present invention is to provide a motor control device that does not interfere with each other and to provide an electric power steering device with improved steering feeling.

  The present invention includes control units 1 and 2 that drive and control a motor having two system windings for each of the two system windings, and each of the control units 1 and 2 performs current feedback control based on a current command value. And a motor control device that compensates the motor back electromotive force with a back electromotive force compensation signal based on the motor angle or the motor angular speed, and the above object of the present invention is to provide a path for compensating the motor back electromotive force. This is achieved by providing an LPF for noise removal and an angle feedback filter having the same transfer function characteristics in the single-system drive and the two-system drive in the feedback path.

  The object of the present invention is to provide a coefficient unit that distributes the current command value in half, and drives the control units 1 and 2 by the distributed current command value from the coefficient unit. Or the control units 1 and 2 that input the current command value have a coefficient that halves the current command value, or the current command value is a dq-axis current command value. The control unit is a two-phase feedback vector control system, or the current command value is a dq-axis current command value, and the control unit is a three-phase feedback vector control system. Effectively achieved.

  According to the motor control device of the present invention, the back electromotive force voltage of the motor having the two-system winding is compensated, and the two control systems for driving the two-system winding are identical in the single-system driving and the both-system driving. Since the angle filter that becomes the transfer function characteristic is provided in the feedback path, there is no interference even in single-system drive at the time of failure (including abnormality) or in both-system drive at normal time. It becomes possible to balance the characteristics.

  In addition, since torque ripple is suppressed by the single-system control and the two-system control, the steering feeling can be further improved if it is mounted on the electric power steering device.

It is a lineblock diagram showing an outline of an electric power steering device. It is a block diagram which shows the structural example of the control unit (ECU) of an electric power steering apparatus. It is a block diagram of the motor control apparatus which shows the influence of a motor back electromotive voltage. It is a frequency response figure for demonstrating the characteristic of a control filter. It is a frequency response figure for demonstrating the characteristic of a control filter. It is a diagram of a Y connection motor having two system windings. It is a diagram of a Δ connection motor having two windings. It is a block diagram which shows the structural example of the motor control apparatus which has a back electromotive force compensation function. It is a block diagram which shows the structural example of the control system of the motor which has 2 type | system | group winding. It is a characteristic view which shows the example of a characteristic (simulation) of a conventional apparatus. It is a frequency response figure which shows the characteristic example (both system drive and single system drive) of the conventional apparatus. It is a frequency response figure which shows the example of the characteristic of the phase of a conventional apparatus (both system drive and single system drive). It is a block block diagram which shows an example of embodiment of this invention. It is a frequency response figure which shows the effect of this invention. It is a block diagram which shows the structural example of the vector control system (3 phase FB) which can apply this invention. It is a block diagram which shows the structural example of the vector control system (2 phase FB) which can apply this invention.

  In the present invention, compensation for the motor back electromotive force including noise is performed. First, back electromotive force compensation will be described.

  FIG. 8 shows a configuration example in the case where the back electromotive force EMF is compensated based on the motor angle θm detected by a rotation sensor (for example, a resolver), corresponding to FIG. Although the motor angle θm is differentiated by the differentiating unit 130, the rotation sensor (for example, the resolver) actually includes noise Nr, and the motor angle θmr to which the noise Nr is added (mixed) by the adding unit 133 is differentiated. Input to the unit 130. The motor angular velocity ωn differentiated by the differentiating unit 130 is input to a low-pass filter (LPF) 131 for noise removal whose transfer function is expressed by Equation 5, and the motor angular velocity ωn ′ from which the noise Nr has been removed by the LPF 131 is the inverse. The back electromotive force compensation constant Ke ′ is multiplied by the back electromotive force compensation constant unit 132 and input to the adder 106 as a back electromotive voltage compensation signal EMFc.

Incidentally, omega _L of LPF131 have a relation of ω _L = 2πf _c relative to the cutoff frequency _{f c} (e.g. 40 Hz).

この逆起電圧補償はモータ角度θｍ（モータ角速度ωｍ）に依存しており、ノイズＮｒの混入がなければＬＰＦ１３１は不要である。しかし、実際には回転センサから混入するノイズＮｒの影響があり、その影響を低減するＬＰＦ１３１を具備している。フィルタ処理を行わないと、電動パワーステアリング装置では操舵感が悪化してしまうからである。このＬＰＦ１３１のフィルタ処理により、モータ回転数の周波数が上がると、モータ逆起電圧ＥＭＦに対して位相が遅れ、完全にモータ逆起電圧ＥＭＦを相殺できない可能性がある。この相殺誤差が外乱として電流制御系内に混入して、フィードバック制御の役割である実モータ電流Ｉｍを電流指令値Ｉｒｅｆに追従させることができなくなる。逆起電圧補償が完全であれば、下記数６の関係となるべきである。

This back electromotive force compensation depends on the motor angle θm (motor angular velocity ωm), and the LPF 131 is not required unless noise Nr is mixed. However, there is actually an influence of noise Nr mixed from the rotation sensor, and the LPF 131 for reducing the influence is provided. This is because if the filter process is not performed, the steering feeling is deteriorated in the electric power steering apparatus. If the frequency of the motor rotation speed is increased by the filtering process of the LPF 131, the phase is delayed with respect to the motor counter electromotive voltage EMF, and the motor counter electromotive voltage EMF may not be completely canceled. This cancellation error is mixed into the current control system as a disturbance, and the actual motor current Im, which is the role of feedback control, cannot follow the current command value Iref. If the back electromotive force compensation is complete, the relationship of the following formula 6 should be satisfied.

(Equation 6)
EMFc-EMF = 0
However, the above equation 6 does not hold due to the mixing of noise Nr. In the first place, the main role of the current feedback control is to flow the actual current Im without delay according to the current command value Iref. In the control by the filter process, the followability can be improved as the control band is higher, but the sensitivity to noise is increased, and the steering feeling is deteriorated in the electric power steering apparatus.

  On the other hand, the configuration for controlling a motor having two windings is, for example, FIG. A motor 100 having an output shaft 111 includes two systems of Y-connected motor windings 101 and 102, and a rotation sensor 110 is connected to the motor 100. The ECU 200 that drives and controls the motor 100 inputs a power source, a steering torque Th, a steering angle θ, a vehicle speed Vs, an ignition state, and a rotation angle θe, and is connected to the CAN 40 and the non-CAN 41. The ECU 200 includes an MCU 201 that performs overall control and calculation, a GDM 211 that drives the motor winding 101 according to the current command value 1 from the MCU 201, and a GDM 212 that drives the motor winding 102 according to the current command value 2 from the MCU 201. It has. The GDMs 211 and 212 include a motor relay (contact type or semiconductor type), a monitor / diagnostic interface circuit, an inverter, and the like. Current / voltage detection values of the GDMs 211 and 212 are input to the MCU 201. That is, the MCU 201 has a correction calculation function between the current command value 1 and the current command value 2.

  In such a two-system winding motor and two-system control system configuration, when one system fails (including an abnormality), the assist control is continued to some extent by the other system control system that has not failed. It is possible. However, when two systems are used, there is a possibility that a difference in motor torque occurs between both systems and one system, which may give the driver discomfort. This is because the motor back electromotive force compensation signal is fed back to each system and interferes with each other.

  FIG. 10 shows such a state. When a transition is made from the two-system drive (two-system drive) to the one-system drive (one-system drive), a step (motor output torque) occurs as shown in part A of FIG. Further, when a transition is made from the one-system drive to the both-system drive, a step is generated as shown in part B of FIG. The occurrence of such a step is an operation to be avoided in the motor control device and the electric power steering device.

  The occurrence of a step at the time of switching in FIG. 10 is such that when the current command value is 40 [A], the gain characteristic when driving both systems is as shown by the black line in FIG. 11, and the phase characteristic is the black line in FIG. On the other hand, the gain characteristic at the time of one-system drive is as shown by the gray line in FIG. 11, and the phase characteristic is as shown by the gray line in FIG. 12, and has an inflection point. Because.

  Therefore, in the present invention, an angle filter that cancels out the error of the motor back electromotive force compensation and does not interfere with each other even when switching between the one-system drive and the two-system drive is provided in the feedback path of the motor control.

  A configuration example of the present invention will be described with reference to FIG.

  A motor having two windings (mechanical system characteristic unit 310) is provided with a two-system control system of control units 320 and 330, and is controlled by driving both systems of the control units 320 and 330 in a normal state. When the control system becomes faulty (including abnormality), one system drive is performed in the other control system. Further, when the failure is recovered, the single system drive is restored to the dual system drive.

  The current command value Iref is input to the subtractor 301, and the subtractor 301 calculates a deviation Iref from the current value If from the angle feedback (FB) filter 300. The deviation Irefe is input to the coefficient units 321 and 331 of the control units 320 and 330, and the switching signal SW is input to the coefficient units 321 and 331. The coefficient parts 321 and 331 are each set to a coefficient “1/2” in normal time (normal time), and the same current command value halved is distributed to the two control parts 320 and 330. When one control system fails, the coefficient of the coefficient part of the failed system is set to “0”, and the coefficient of the normal system is set to “1”. That is, the current command values Ic1 and Ic2 that are the outputs of the coefficient units 321 and 331 are expressed by the following equation (7).

(Equation 7)
Normal time: Ic1 = 1/2 × Irefe
Ic2 = 1/2 × Irefe
At failure (system 1 failure):
Ic1 = 0 × Iref = 0
Ic2 = 1 × Irefe = Irefe
At failure (system 2 failure):
Ic1 = 1 × Irefe = Irefe
Ic2 = 0 × Iref = 0

Current command value Ic1 and Ic2 are input to the control filter _(G FF) 322 and 332, respectively represented by the above Expression 1, the current command value Ig1 and Ig2 from control filter _(G FF) 322 and 332, respectively subtracting unit 323 and 333 are input. The motor currents Im1 and Im2 from the electric system characteristic units 327 and 337 are respectively fed back to the subtraction units 323 and 333, and the deviation currents Id1 and Id2 calculated by the subtraction units 323 and 333 are expressed by the above formula 2. Input to the represented control filters (G _FB ) 324 and 334.

The current command values Ie1 and Ie2 from the control filters (G _FB ) 324 and 334 are respectively input to the adders 325 and 335, added to the back electromotive force compensation signal EMFc, and the added current command values Ib1 and Ib2 are subtracted, respectively. Input to the units 326 and 336. A sub electromotive voltage EMF obtained by multiplying the motor angular velocity ωm by the counter electromotive voltage constant Ke is subtracted and input to the subtracting units 326 and 336, and a current command value Ir1 obtained by subtracting the counter electromotive voltage EMF is expressed by the following equation (8). A current command value Ir2 input to the electrical system characteristic unit 327 and subtracted from the back electromotive force EMF is input to the electrical system characteristic unit 337 represented by the following formula 9.

電気系特性部３２７からのモータ電流Ｉｍ１は、トルク定数部３２８でトルク定数Ｋｔ１を乗算されてトルクτ１として加算部３１３に入力され、電気系特性部３３７からのモータ電流Ｉｍ２は、トルク定数部３３８でトルク定数Ｋｔ２を乗算されてモータトルクτ２として加算部３１３に入力される。加算部３１３の加算で求められたモータトルクτは機械系特性部３１０に入力され、機械系特性部３１０の出力であるモータ角速度ωｍが、逆起電圧定数部３１２で逆起電圧定Ｋｅを乗算され、逆起電圧定Ｋｅを乗算された逆起電圧ＥＭＦが減算部３２６及び３３６に減算入力される。

The motor current Im1 from the electric system characteristic unit 327 is multiplied by the torque constant Kt1 by the torque constant unit 328 and input to the addition unit 313 as the torque τ1, and the motor current Im2 from the electric system characteristic unit 337 is input to the torque constant unit 338. Is multiplied by the torque constant Kt2 and input to the adder 313 as the motor torque τ2. The motor torque τ obtained by the addition of the addition unit 313 is input to the mechanical system characteristic unit 310, and the motor angular velocity ωm that is the output of the mechanical system characteristic unit 310 is multiplied by the back electromotive voltage constant Ke by the back electromotive voltage constant unit 312. Then, the counter electromotive voltage EMF multiplied by the counter electromotive voltage constant Ke is subtracted and input to the subtracting units 326 and 336.

  Further, the motor angular velocity ωm is integrated by the integrating unit 311 and output as the motor angle θm. Noise Nr is mixed into the motor angle θm and differentiated by the differentiating unit 314. The motor angular velocity ωn from the differentiating unit 314 is input to the LPF 315 expressed by the above equation 5, and the filtered angular velocity ωn ′ is multiplied by the counter electromotive voltage compensation constant Ke ′ by the counter electromotive voltage compensation constant unit 316 to obtain the counter electromotive voltage. The compensation signal EMFc is input to the adders 325 and 336.

The motor angle θm is fed back to the subtractor 301 via the angle feedback filter (G _ANG ) 300.

  With such a configuration, the transfer function from the current command value Iref to the motor torque τ1 of the system 1 is the following formula 10, and the transfer function from the current command value Iref to the motor torque τ2 of the system 2 is the following formula 11.

そして、通常時の両系統駆動時には系統１と系統２は同一状態で駆動され、下記数１２で表わされる。

When both systems are driven at normal time, the system 1 and the system 2 are driven in the same state, and are expressed by the following equation (12).

ここで、便宜的にＬ_１＝Ｌ_２＝Ｌ，Ｒ_１＝Ｒ_２＝Ｒ，Ｋ_ｔ１＝Ｋ_ｔ２＝Ｋｔとすれば、数１０〜数１２の分母の（）を０にすることにより（数１３）、片系統駆動と両系統駆動での特性を同じにすることができる。（）を０とするには、角度ＦＢフィルタ（Ｇ_ＡＮＧ）３００が下記数１４となれば良い。

Here, for convenience, if L ₁ = L ₂ = L, R ₁ = R ₂ = R, K _t1 = K _t2 = Kt, the denominator of tens to tens 12 is set to 0 ( (13) The characteristics of the single-system drive and the double-system drive can be made the same. In order to set () to 0, the angle FB filter (G _ANG ) 300 may be expressed by the following equation (14).

角度ＦＢフィルタ（Ｇ_ＡＮＧ）３００の伝達関数を数１４とすることにより、片系統駆動においても両系統駆動においても、モータトルクτは数１５となる。

By setting the transfer function of the angle FB filter (G _ANG ) 300 to _Expression 14, the motor torque τ is expressed by Expression 15 in both single-system driving and both-system driving.

この結果、両系統駆動から片系統駆動に切り換えても、逆に片系統駆動から両系統駆動に切り換えても特性が同じになり、干渉がなく、変曲点の少ない円滑な特性となる。電動パワーステアリング装置においては、片系統制御と両系統制御でトルクリップルが抑制されるので、操舵フィーリングを一層向上することができる。

As a result, the characteristics are the same even when switching from both-system driving to single-system driving, or conversely switching from one-system driving to both-system driving, and there is no interference and smooth characteristics with few inflection points. In the electric power steering device, torque ripple is suppressed by single-system control and double-system control, and thus the steering feeling can be further improved.

FIG. 14 compares the single-system drive and both-system drive with the characteristics based on the insertion of the angle FB filter (G _ANG ) 300 of the present invention. FIG. 14 (A) shows the gain characteristics, and FIG. ) Indicates phase characteristics. As shown in FIG. 14, it can be seen that smooth characteristics with few inflection points are obtained by the insertion of the angle FB filter (G _ANG ) 300.

  Since the present invention can also be applied to a vector control system that drives and controls a brushless motor, the vector control system will be described.

The vector control system of FIG. 15, and the current command value calculating section 220 is provided to correct by calculating the d-axis current command value i _d and the q-axis current command value i _q, steering the current command value calculating section 220 The torque Th, the vehicle speed Vs, the motor angle (rotation angle) θ _e , and the motor angular velocity ω calculated by the angular velocity calculation unit 226 are input from the rotation sensor 110 connected to the motor 100. D-axis current command value i _d and the q-axis current command value i _q calculated by the current command value calculating unit 220 is input to the 2-phase / 3-phase conversion unit 221, three-phase current in synchronization with the motor angle theta _e The command values are converted into Iuref, Ivref, and Iwref. Three-phase current command values Iuref, Ivref, Iwref are input to a subtractor 222 (222u, 222v, 222w), and deviations ΔIu, ΔIv, ΔIw from motor currents Imu, Imv, Imw detected by the current detection circuit 225A are obtained. Calculated. The calculated deviations ΔIu, ΔIv, ΔIw are input to the PI control unit 223, and the current-controlled three-phase voltage control command values Vuref, Vvref, Vwref are input to the PWM control unit 224, and are calculated by the PWM control unit 224. The motor 100 is driven via the inverter 225 based on each phase duty.

Also, in the vector control system of FIG. 16, the current detection circuit 225A with the detected three-phase motor current Imu, Imv, 3-phase converted to 2-phase in synchronism with the motor angle theta _e the Imw / 2 phase converter 227 Is provided. D-axis current command value computed is corrected by the current command value calculating section 220 _{i d} and the q-axis current command value _{i q} is input to the subtraction unit 222 (222d, 222q), 3-phase / 2-phase conversion by the subtraction unit 222 Deviations Δi _d and Δi _q from the two-phase currents Imd and Imq from the unit 227 are calculated. The deviations Δi _d and Δi _q are input to the two-phase / three-phase conversion unit 221, and the converted three-phase current command values Iuref, Ivref, Iwref are input to the PI control unit 223, and thereafter the same as in FIG. Actions are performed.

  The control system in FIG. 15 is a three-phase feedback vector control system that feeds back three-phase motor currents Imu, Imv, and Imw. The control system in FIG. 16 has three-phase motor currents Imu, Imv, and Imw of 2. This is a two-phase feedback vector control system which is converted into phase currents Imd and Imq and fed back. The present invention can be applied to both the three-phase feedback vector control system and the two-phase feedback vector control system, and other control systems may be used.

In the above-described embodiment, the motor having the two-system winding is driven and controlled by separate GDMs, but may be integrated in hardware. Moreover, although the switchable coefficient part is provided in the preceding stage of each system, a variable gain may be provided in the control filter (G _FF ).

1 Handle 2 Column shaft (steering shaft, handle shaft)
10 Torque Sensor 12 Vehicle Speed Sensor 20, 100 Motor 30, 200 Control Unit (ECU)
31, 220 Current command value calculation unit 34, 223 PI control unit 35, 224 PWM control unit 36, 225 Inverter 110, 327, 337 Electric system characteristic unit 120, 310 Mechanical system characteristic unit 131, 315 LPF
201 MCU
211, 212 GDM
221 2-phase / 3-phase converter 227 3-phase / 2-phase converter 300 Angle feedback filters 320, 330 Control units 321, 331 Coefficient unit

Claims (6)

Control units 1 and 2 that drive and control a motor having two windings for each of the two windings are provided, and each of the control units 1 and 2 controls driving of the motor by current feedback control based on a current command value. In addition, in the motor control device that compensates for the motor back electromotive force by the back electromotive force compensation signal based on the motor angle or the motor angular speed,
An LPF for noise removal is provided in each of the paths for compensating the motor back electromotive voltage, and an angle feedback filter having the same transfer function characteristics in the single system drive and both system drives is provided in the feedback path. A motor control device. The coefficient part which distributes the said current command value to 1/2 each is provided, The said control parts 1 and 2 are driven by the distribution current command value from the said coefficient part. Motor control device. The motor control device according to claim 1, wherein the control units 1 and 2 that input the current command value have a coefficient that reduces the current command value to ½. The motor control device according to claim 1, wherein the current command value is a dq-axis current command value, and the control unit is a two-phase feedback vector control system. The motor control device according to claim 1, wherein the current command value is a dq-axis current command value, and the control unit is a three-phase feedback vector control system. An electric power steering device equipped with the motor control device according to claim 1.

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