JP6648592B2 - Motor control device and electric power steering device equipped with the same - Google Patents

Description

  The present invention provides a motor control device that drives a motor with current feedback based on a current command value, controls the current command value by following a motor actual current, and includes the motor control device, and at least a current command calculated based on a steering torque. The present invention relates to an electric power steering device that applies an assist force by a motor to a steering system of a vehicle according to a value. In particular, a motor control device that cancels a back electromotive voltage at the time of driving a motor without changing the sensitivity to noise and inserts an error compensation filter for canceling a compensation error that performs noise compensation and LPF compensation in a feedforward path. And an electric power steering device equipped with the same.

  2. Description of the Related Art An electric power steering device (EPS) equipped with a motor control device applies a steering assisting force (assisting force) to a steering mechanism of a vehicle by a rotational force of a motor. A steering assist force is applied to the steering shaft or the rack shaft by a transmission mechanism such as a gear. Such a conventional electric power steering device performs feedback control of a motor current in order to accurately generate a torque of a steering assist force. The feedback control is to adjust the motor applied voltage so that the difference between the steering assist command value (current command value) and the detected motor current value is small. The adjustment of the motor applied voltage is generally performed by PWM (pulse width). Modulation) control by adjusting the duty.

  A general configuration of the electric power steering apparatus will be described with reference to FIG. 1. 6b, and further connected to steered wheels 8L, 8R via hub units 7a, 7b. The column shaft 2 is provided with a torque sensor 10 for detecting the steering torque Th of the steering wheel 1 and a steering angle sensor 14 for detecting the steering angle θ, and a motor 20 for assisting the steering force of the steering wheel 1 is provided with a reduction gear. 3 is connected to the column shaft 2. The control unit (ECU) 30 that controls the electric power steering device is supplied with electric power from the battery 13 and receives an ignition key signal via the ignition key 11. The control unit 30 calculates a current command value of an assist (steering assist) command based on the steering torque Th detected by the torque sensor 10 and the vehicle speed Vs detected by the vehicle speed sensor 12, and compensates for the current command value. The current supplied to the EPS motor 20 is controlled by the voltage control command value Vref subjected to the above.

  The steering angle θ is detected from the steering angle sensor 14, and the steering angle can be obtained from a rotation sensor such as a resolver connected to the motor 20.

  The control unit 30 is connected to a CAN (Controller Area Network) 40 for transmitting and receiving various information of the vehicle, and the vehicle speed Vs can be received from the CAN 40. The control unit 30 can also be connected to a non-CAN 41 other than the CAN 40 for transmitting and receiving communications, analog / digital signals, radio waves, and the like.

  The control unit 30 is mainly composed of a CPU (including an MPU and an MCU), and FIG. 2 shows general functions executed by a program in the CPU.

  The control unit 30 will be described with reference to FIG. 2. The steering torque Th detected by the torque sensor 10 and the vehicle speed Vs detected by the vehicle speed sensor 12 (or from the CAN 40) are calculated based on the current command for calculating the current command value Iref1. The value is input to the value calculator 31. The current command value calculation unit 31 calculates a current command value Iref1, which is a control target value of the current supplied to the motor 20, using an assist map or the like based on the input steering torque Th and vehicle speed Vs. The current command value Iref1 is input to the current limiting unit 33 via the adding unit 32A, and the current command value Irefm of which the maximum current is limited is input to the subtracting unit 32B, and the deviation ΔI (=) from the motor current value Im being fed back. Irefm−Im) is calculated, and the deviation ΔI is input to the PI control unit 34 for improving the characteristics of the steering operation. The voltage control command value Vref whose characteristics have been improved by the PI control unit 34 is input to the PWM control unit 35, and the motor 20 is PWM-driven via the inverter 36. The current value Im of the motor 20 is detected by the motor current detector 37 and is fed back to the subtractor 32B. The inverter 36 is configured by an FET bridge circuit as a semiconductor switching element.

  A rotation sensor 21 such as a resolver is connected to the motor 20. The rotation sensor 21 outputs a motor rotation angle θ, and a motor speed ω is calculated by a motor speed calculation unit 22.

  Further, the compensation signal CM from the compensation signal generator 38 is added to the adder 32A, and the characteristics of the steering system are compensated by adding the compensation signal CM, so that the convergence and the inertia characteristics are improved. ing. The compensation signal generation unit 38 adds the self-aligning torque (SAT) 38-1 and the inertia 38-2 in the addition unit 38-4, and further adds the convergence 38-3 to the addition result in the addition unit 38-5. Then, the addition result of the addition section 38-5 is used as the compensation signal CM.

  In such an electric power steering apparatus, since the motor 20 generates a back electromotive voltage when the motor is driven, compensation for suppressing or attenuating the motor back electromotive voltage is necessary. The reason will be described below.

FIG. 3 shows a system in which the motor 20 is driven from the current command value Iref by a transfer function. The current command value Iref is inputted to the subtraction unit 104 via the control filter _(G FF) 101, the deviation _{e 1} between the actual motor current Im is calculated. Deviation _{e 1} is input to the subtraction unit 105 via the control filter _{(G FB),} the counter electromotive voltage EMF of the motor 20 is subtracted by the subtraction unit 105, an electrical system characteristics unit 110 of the difference _{e 3} the motor 20 (1 / (L · s + R)), and further input to the mechanical characteristic unit 120 (1 / (J · s + D)) via a torque constant Kt [Nm / A]. The back electromotive voltage EMF is obtained by multiplying the motor angular velocity (motor rotation speed) ωm output from the mechanical characteristic unit 120 by the back electromotive voltage constant Ke [V / (rad / s)]. The motor current Im from the electric system characteristic unit 110 is detected and fed back. However, actually, the current detection noise Ni is mixed and fed back as the actual motor current Imn.

L of an electrical system characteristics 110 mode tie inductance [H], R is the motor resistance [Omega], the J of the mechanical system characteristics unit 120 motor inertia moment [Kg · m ^2], D is the motor coefficient of viscosity [Nm / (rad / s)].

The system from the current command value Iref until the motor current Im, FIG. 4 to control the frequency band as shown in easily primary filter _{(1 / (T 4 · s} + 1)) and to _order, time constant T 1 through T ₄ The transfer function of the control filter (G _FF ) 101 is set by the following equation 1, and the transfer function of the control filter (G _FB ) 102 is set by the following equation 2.

That is, the system from the current command value Iref to the motor current Im is represented by “1 / (T ₄ · s + 1)” (Equation 3), and the characteristics from the current command value Iref to the motor current Im by setting the time constant T _4. In order to determine the current control band (primary filter), the characteristics of the control filter (G _FF ) 101 and the control filter (G _FB ) 102 are determined.

Then, in the current control feedback closed loop system including the control filter (G _FB ) 102, the back electromotive force EMF is set to 0, and as shown in Expression 4 and FIG. A configuration is adopted that serves as a next-order filter (time constants T ₂ and T ₃ ) and a first-order phase advance filter (T ₁ ). This is because if there is no term of the first-order phase advance filter (T ₁ ), the differential term of the numerator of the control filter (G _FF ) 101 becomes second-order and becomes very unstable. In short, the control filter (G _FF ) 101 is a filter for determining characteristics from the current command value Iref to the motor current Im, and the control filter (G _FB ) 102 is a filter including a filter for removing the current detection noise Ni. It can be said that this is a filter that determines characteristics from the command value Iref to the motor current Im.

When Equations 1 and 4 are solved for G _FB , the control filter (G _FB ) 102 becomes Equation 2 above.

As described above, the setting of the control filter (G _FF ) 101 and the control filter (G _FB ) 102 does not take into account the mixing of the motor back electromotive voltage EMF. However, since the generation of the motor back electromotive voltage EMF affects the current command value, it is a serious problem in accurately controlling the motor output. As such a countermeasure, compensation of a motor back electromotive voltage can be considered. For example, in Japanese Patent Application Laid-Open No. 2013-219870 (Patent Document 1), a back electromotive voltage of a motor is estimated based on a voltage command value and a current detection value. The back electromotive force estimated value is added to the current command value. In Japanese Patent Application Laid-Open No. 2012-236472 (Patent Document 2), a back electromotive force is estimated based on a rotational angular velocity, and a back electromotive voltage compensation control value is calculated by multiplying the estimated back electromotive voltage by a compensation coefficient. The electromotive voltage compensation control value is added to the basic voltage to obtain a voltage command value.

JP 2013-72970 A JP 2012-236472 A

  Although the back electromotive force compensation as described above is compensated for on the assumption that the detection noise of the position and the rotation speed does not mix, the noise is actually mixed. There is a problem that the motor output cannot be followed.

  SUMMARY OF THE INVENTION The present invention has been made in view of the circumstances described above, and an object of the present invention is to provide a feedforward path and a characteristic of a back electromotive voltage and a back electromotive voltage compensation path without delay over a wide range without changing sensitivity to noise. It is an object of the present invention to provide a motor control device that makes the actual current follow the current command value reliably, and also provides an electric power steering device with improved steering feeling.

  The present invention relates to a motor control device that controls driving of a motor by current feedback control based on a current command value and compensates for a motor back electromotive voltage by a back electromotive force compensation signal based on a motor angle or a motor angular velocity. An object is to provide an LPF for removing noise in a path for compensating the motor back electromotive voltage, and to provide a reverse between the motor back electromotive voltage and the back electromotive voltage compensation signal in a feed forward path for the motor control. This is achieved by providing an error compensation filter for canceling the electromotive voltage compensation error.

  The object of the present invention is that the motor control is dq-axis vector control, or the current command value is a dq-axis current command value, and the control unit is a two-phase feedback vector control system. Alternatively, the current command value is a dq-axis current command value, and the control unit is a three-phase feedback vector control system, which is more effectively achieved. By mounting, an electric power steering device with a good steering feeling can be provided.

  According to the motor control device according to the present invention, in the motor control device having the function of compensating the motor back electromotive voltage at the time of driving the motor, the motor control device includes a reverse feed, which is a difference between the motor back electromotive voltage and the back electromotive voltage compensation signal, in the feedforward path. Since the error compensation filter that cancels out the electromotive voltage compensation error is interposed, the tracking characteristics of the motor can be improved without delay without changing the sensitivity to noise.

1 is a configuration diagram illustrating an outline of an electric power steering device. FIG. 2 is a block diagram illustrating a configuration example of a control unit (ECU) of the electric power steering device. It is a block diagram which shows the path | route of generation | occurrence | production of a back electromotive voltage in a motor control apparatus (transfer function). FIG. 4 is a frequency response diagram for explaining characteristics of a control filter. FIG. 4 is a frequency response diagram for explaining characteristics of a control filter. FIG. 2 is a block diagram illustrating a configuration example of a vector control system (three-phase FB) to which the present invention can be applied. FIG. 3 is a block diagram illustrating a configuration example of a vector control system (two-phase FB) to which the present invention can be applied. It is a block diagram which shows the example of a structure of the back electromotive force compensation in a motor control apparatus (transfer function). FIG. 4 is a diagram illustrating frequency characteristics of gain / phase and a frequency characteristic showing an effect of the present invention when an LPF is used for back electromotive force compensation. FIG. 8 is a frequency characteristic diagram showing details of FIG. 7. FIG. 1 is a block diagram illustrating an example of an embodiment of the present invention. FIG. 4 is a frequency characteristic diagram showing characteristics before compensation according to the present invention. FIG. 3 is a frequency characteristic diagram showing characteristics of compensation according to the present invention.

The motor control device according to the present invention compensates for the generation of the back electromotive voltage at the time of driving the motor without changing the sensitivity to the noise, and feeds the error compensation filter for compensating error compensation for noise compensation and LPF feedforward. Inserted in the route. This allows the actual motor current to accurately follow the current command value without delay without changing the sensitivity to noise. The present invention can also be applied to a vector control system for driving and controlling a brushless motor. The control system will be described.

The vector control system of FIG. 6, and the current command value calculating section 220 is provided to correct by calculating the d-axis current command value i _d and the q-axis current command value i _q, steering the current command value calculating section 220 The torque Th, the vehicle speed Vs, the motor angle (rotation angle) θ _e from the rotation sensor 100A connected to the motor 100, and the motor angular speed ω calculated by the angular speed calculation unit 226 are input. D-axis current command value i _d and the q-axis current command value i _q calculated by the current command value calculating unit 220 is input to the 2-phase / 3-phase conversion unit 221, three-phase current in synchronization with the motor angle theta _e It is converted into command values Iuref, Ivref, Iwref. The three-phase current command values Iuref, Ivref, Iwref are input to the subtraction unit 222 (222u, 222v, 222w), and deviations ΔIu, ΔIv, ΔIw from the motor currents Imu, Imv, Imw detected by the current detection circuit 225A are calculated. Is calculated. The calculated deviations ΔIu, ΔIv, ΔIw are input to the PI control unit 223, and the current-controlled three-phase voltage control command values Vuref, Vvref, Vwref are input to the PWM control unit 224, and are calculated by the PWM control unit 224. The motor 100 is driven via the inverter 225 based on the duty of each phase.

  Although the current detection circuit 225A is provided in the inverter 225 in FIG. 6, the current detection circuit 225A can also detect a supply line to the motor 100 or the like.

Also, in the vector control system of FIG. 7, the current detection circuit 225A with the detected three-phase motor current Imu, Imv, 3-phase converted to 2-phase in synchronism with the motor angle theta _e the Imw / 2 phase converter 227 Is provided. D-axis current command value computed is corrected by the current command value calculating section 220 _{i d} and the q-axis current command value _{i q} is input to the subtraction unit 222 (222d, 222q), 3-phase / 2-phase conversion by the subtraction unit 222 2-phase current Imd from part 227, the deviation .DELTA.i _d with imq, .DELTA.i _q is calculated. Deviation .DELTA.i _d, .DELTA.i _q are input to the 2-phase / 3-phase conversion unit 221, converted three-phase current command values Iuref, Ivref, Iwref is input to the PI control unit 223, since, like the case of FIG. 6 Operation is performed.

  The control system of FIG. 6 is a three-phase feedback vector control system in which three-phase motor currents Imu, Imv, Imw are fed back. The control system of FIG. 7 has two-phase motor currents Imu, Imv, Imw of two. This is a two-phase feedback vector control system that is converted into phase currents Imd and Imq and fed back. The present invention can be applied to both the three-phase feedback vector control system and the two-phase feedback vector control system.

  FIG. 8 shows a configuration example in which the back electromotive force EMF is compensated based on the motor angle θm detected by a rotation sensor (for example, a resolver), corresponding to FIG. Although the motor angle θm is differentiated by the differentiator 130, the rotation sensor (for example, a resolver) actually contains noise Nr, and the motor angle θmr to which the noise Nr is added (mixed) by the adder 133 is differentiated. Input to the unit 130. The motor angular velocity ωn differentiated by the differentiator 130 is input to a noise removal low-pass filter (LPF) 131 whose transfer function is expressed by Equation 5, and the motor angular velocity ωn ′ from which the noise Nr has been removed by the LPF 131 is inversed. The voltage is multiplied by the back electromotive force compensation constant Ke ′ in the electromotive force compensation constant unit 132 and input to the adding unit 106 as the back electromotive force compensation signal EMFc.

Incidentally, omega _L of LPF131 have a relation of ω _L = 2πf _c relative to the cutoff frequency _{f c} (e.g. 40 Hz).

This back electromotive force compensation depends on the motor angle θm (motor angular velocity ωm), and the LPF 131 is not required unless the noise Nr is mixed. However, there is actually an effect of noise Nr mixed from the rotation sensor, and the LPF 131 is provided to eliminate or reduce the effect. This is because the steering feeling is deteriorated in the electric power steering apparatus unless the filter processing is performed.

When the frequency of the motor rotation speed increases due to the filter processing of the LPF 131, the phase is delayed with respect to the motor back electromotive voltage EMF, and there is a possibility that the motor back electromotive voltage EMF cannot be completely canceled. This cancellation error enters the current control system as a disturbance, and it becomes impossible to cause the actual motor current Im, which plays a role of feedback control, to follow the current command value Iref. If the back electromotive force compensation is perfect, the relationship of the following equation 6 should be satisfied.
(Equation 6)
EMFc−EMF = 0
However, Equation 6 does not hold due to the mixing of the noise Nr. In the first place, the main role of the current feedback control is to flow the actual current without delay according to the current command value. In the control by the filter processing, the followability can be improved as the control band is higher, but the sensitivity to noise is increased, and the steering feeling is deteriorated in the electric power steering device.

  9 and 10 show frequency characteristics when back electromotive force compensation is performed using an LPF. FIGS. 9 (A) and 10 (A) show the frequency characteristics of the gain. , The gain is concavely reduced in the frequency range of 10 to 100 Hz. 9 (B) and 10 (B) show the frequency characteristics of the phase. Before the countermeasure according to the present invention, the back electromotive voltage EMF cannot be completely compensated, and the compensation cancellation error is generated as a disturbance in the control loop. As a result, the detection current cannot follow the current command value.

As a countermeasure, in the present invention, an error compensating filter (G _EC ) 200 such that the tracking characteristic becomes the target LPF characteristic (= 1 / (T ₄ · s + 1)) is interposed in the feedforward path. FIG. 11 shows the configuration corresponding to FIG.

The transfer function of the control filter (G _FF ) 101 is represented by Equation 1, and the transfer function of the control filter (G _FB ) 102 is represented by Equation 2.

  Here, when the range from the current command value Iref to the motor current Im output by the electric system characteristic unit 110 is represented by a transfer function, the following equation 7 is obtained.

By substituting the _{GFF of the} above _equation 1 and the _GFB of the above equation 2 into the above _equation 7, and transforming it, the equation 8 is obtained.

Here, by designing the error compensating filter _GEC as shown in Expression 9, Expression 7 becomes Expression 10 below.

Therefore, it is possible to obtain desired characteristics by adjusting the constant T ₄ when the number 10.

9 and 10 compare the mechanism of the problem that the detected current does not follow the current command value at the time of steering with a Bode diagram. Before the countermeasure, the back electromotive force compensation logic has an LPF, I cannot fully compensate. This canceling error enters the control loop as a disturbance, and as a result, the detected current cannot follow the current command value. However, by providing an error compensation filter 200 (G _EC ) in the feedforward path for canceling the back electromotive voltage compensation error, which is the difference between the motor back electromotive voltage and the back electromotive voltage compensation signal, the characteristics before the measures are improved. You can see that it is done. That is, both the gain characteristic and the phase characteristic are target characteristics.

  FIG. 12 shows the frequency characteristic in a state where the error compensation filter 200 is not inserted. By inserting the error compensation filter 200 shown in FIG. 13 having a characteristic opposite to this characteristic into the feedforward path, FIG. In addition, the characteristics after the measures shown in FIG. 10 can be obtained.

1 Handle 2 Column shaft (steering shaft, handle shaft)
10 Torque sensor 12 Vehicle speed sensor 20, 100 Motor 30 Control unit (ECU)
31, 220 Current command value calculation unit 34, 223 PI control unit 35, 224 PWM control unit 36, 225 Inverter 101, 102 Control filter 110 Electric system characteristic unit 120 Mechanical system characteristic unit 131 Low pass filter (LPF)
200 Error compensation filter 221 2-phase / 3-phase converter 227 3-phase / 2-phase converter

Claims (5)

In a motor control device that drives and controls the motor by current feedback control based on a current command value and compensates the motor back electromotive voltage by a back electromotive force compensation signal based on the motor angle or motor angular velocity,
An LPF for removing noise is provided in a path for compensating for the back electromotive voltage of the motor, and a back electromotive voltage which is a difference between the motor back electromotive voltage and the back electromotive voltage compensation signal is provided in a feed forward path for motor control. A motor control device comprising an error compensation filter for canceling a compensation error. The motor control device according to claim 1, wherein the control of the motor is dq-axis vector control. The motor control device according to claim 2, wherein the current command value is a dq-axis current command value, and the control unit is a two-phase feedback vector control system. The motor control device according to claim 2, wherein the current command value is a dq-axis current command value, and the control unit is a three-phase feedback vector control system. An electric power steering device equipped with the motor control device according to claim 1.