JP2017055475A - Dc power supply unit and air conditioner - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a DC power supply unit capable of preventing degradation or breakage in element characteristics caused by an inrush current and capable of achieving high efficiency and suppression of a harmonic current.SOLUTION: A DC power supply unit 1 includes: a bridge rectifier circuit 10 with diodes D1, D2 and MOSFETs (Q1, Q2); a reactor L1 provided between an AC power supply VS and the bridge rectifier circuit 10; a smoothing capacitor C1 connected on the output side of the bridge rectifier circuit 10 to smooth a voltage; a converter control unit 18 controlling the MOSFETs (Q1, Q2); and an inrush current prevention circuit U1 protecting a circuit element from an inrush current running in applying a voltage of the AC power supply VS.SELECTED DRAWING: Figure 1

Description

  The present invention relates to a DC power supply device that converts an AC voltage into a DC voltage, and an air conditioner using the DC power supply device.

Trains, automobiles, air conditioners, and the like are equipped with a DC power supply device that converts an AC voltage into a DC voltage. And the direct-current voltage output from a direct-current power supply device is converted into the alternating voltage of a predetermined frequency with an inverter, and this alternating voltage is applied to loads, such as a motor. Such a DC power supply is required to improve power conversion efficiency and save energy.
Therefore, a synchronous rectifier circuit having a MOSFET (Metal-Oxide-Semiconductor-Field-Efect-Transistor) in a circuit has been proposed in a DC power supply device that converts an AC power supply into a DC power supply as in Patent Document 1.

JP 2008-61412 A

  By the way, such a DC power supply device includes a smoothing capacitor and a reactor for smoothing the voltage after rectifying the AC voltage. Usually, before the circuit is powered on, the smoothing capacitor is not charged. Therefore, when an AC voltage is applied to the DC power supply device in this state, an inrush current flows through the circuit. For this reason, the inrush current may cause deterioration of element characteristics or in the worst case, destruction of the element. This inrush current increases as the capacity of the smoothing capacitor increases. Also, the inrush current increases as the reactor inductance decreases.

  In addition to energy saving, DC power supply devices are required to reduce harmonic currents from the viewpoint of protection of electronic equipment and power distribution / reception facilities. For this purpose, power source power factor must be improved.

  SUMMARY OF THE INVENTION It is an object of the present invention to provide a direct-current power supply device that can suppress harmonic currents with high efficiency while preventing deterioration and destruction of element characteristics due to inrush current, and an air conditioner using the direct-current power supply device. And

In order to solve the above problems, a DC power supply device according to the present invention includes a rectifier circuit connected to an AC power supply and having first to fourth diodes, and the third diode as a parasitic diode, or the first The third diode is connected in parallel, has a withstand voltage characteristic in the direction in which the third diode is turned off, and has a saturation voltage lower than the forward voltage drop of the first to fourth diodes. 1 switching element and the fourth diode as a parasitic diode, or connected in parallel to the fourth diode, having a withstand voltage characteristic in a direction in which the fourth diode is turned off, And a second switching element having a saturation voltage lower than a forward voltage drop of the first to fourth diodes, and provided between the AC power supply and the rectifier circuit. A reactor connected to the output side of the rectifier circuit, a smoothing capacitor for smoothing a voltage applied from the rectifier circuit, control means for controlling the first and second switching elements, and the AC power supply And a protection circuit that protects the circuit element from an inrush current that flows when a voltage is applied.
Other means will be described in the embodiment for carrying out the invention.

  ADVANTAGE OF THE INVENTION According to this invention, it becomes possible to provide the direct-current power supply device which can suppress a harmonic current highly efficiently and can suppress the deterioration and destruction of the element characteristic by rush current, and the air conditioner using this direct-current power supply device.

It is a schematic block diagram which shows the DC power supply device in this embodiment. It is the figure which showed the path | route through which an inrush current flows, when there is no inrush current prevention circuit and an alternating current power supply voltage is a positive polarity. It is the figure which showed the operation | movement waveform of each block when there is no inrush current prevention circuit. It is the figure which showed the drain-source voltage-drain reverse current characteristic of MOSFET. It is the figure which showed the operation | movement waveform of each block in case there exists an inrush current prevention circuit. It is the figure which showed the electric current path | route which flows into a circuit, when an alternating current power supply voltage is a positive polarity and an inrush prevention relay will be in an ON state. It is the figure which showed the electric current path | route which flows into a circuit, when an alternating current power supply voltage is a positive polarity, and a power relay will be in an ON state after an inrush prevention relay is switched from ON to OFF. It is the figure which showed the electric current path | route which flows into a circuit, when an alternating current power supply voltage is a negative polarity and an inrush prevention relay will be in an ON state. It is the figure which showed the electric current path | route which flows into a circuit, when an alternating current power supply voltage is a negative polarity and a power relay turns into an ON state after an inrush prevention relay changes from ON to OFF. It is the figure which showed the electric current path | route which flows into a circuit, when diode rectification is performed when an alternating current power supply voltage is a positive polarity. It is the figure which showed the electric current path | route which flows into a circuit, when diode rectification is performed in the case where an alternating current power supply voltage has a negative polarity. It is the figure which showed the electric current path | route which flows into a circuit, when synchronous rectification is performed in the case where an alternating current power supply voltage is a positive polarity. It is the figure which showed the electric current path | route which flows into a circuit, when synchronous rectification is performed in the case where alternating current power supply voltage is a negative polarity. FIG. 6 is a waveform diagram of a power supply voltage, a circuit current, and a MOSFET driving pulse during synchronous rectification. It is the figure which showed the electric current path | route which flows into a circuit when an AC power supply voltage is a positive polarity, when a circuit is short-circuited. It is the figure which showed the electric current path | route which flows into a circuit when an AC power supply voltage is a negative polarity, when a circuit is short-circuited. FIG. 6 is a waveform diagram of a power supply voltage, a circuit current, and a MOSFET drive pulse when a power factor correction current is passed. FIG. 5 is a waveform diagram of a power supply voltage, a circuit current, and a MOSFET drive pulse when high-speed switching is performed. It is the figure which showed the relationship of the duty of MOSFET at the time of performing high-speed switching. It is the figure which showed the duty relationship of MOSFET when performing high-speed switching and considering dead time. It is the figure which showed the relationship between the alternating current power supply voltage at the time of performing high-speed switching, and a circuit current. It is the figure which showed the duty of MOSFET when the amount of delay of the current phase by a reactor is considered when the alternating current power supply voltage is positive polarity. It is the figure explaining the outline | summary of the partial switching. It is the figure explaining switching of the operation mode of the direct-current power supply device according to the magnitude | size of load. It is a figure explaining the current waveform in the case of switching from partial switching to high-speed switching. It is a front view of the indoor unit of the air conditioner in this embodiment, an outdoor unit, and a remote control. It is the schematic explaining the mode that the operation mode of a direct-current power supply device and the operation area | region of an air conditioner were switched according to the magnitude | size of load. This is a MOSFET driving circuit by a bootstrap system. This is a MOSFET drive circuit using a transformer. It is the figure which showed the mode of charge of a bootstrap circuit.

Hereinafter, embodiments for carrying out the present invention will be described in detail with reference to the drawings.
FIG. 1 is a configuration diagram of a DC power supply device 1 according to the present embodiment.
As shown in FIG. 1, the DC power supply 1 is a converter that converts an AC power supply voltage Vs supplied from an AC power supply VS into a DC voltage Vd and outputs the DC voltage Vd to a load H (inverter, motor, etc.). is there. The DC power supply device 1 has an input side connected to an AC power source VS and an output side connected to a load H.

The DC power supply device 1 includes a reactor L1, a smoothing capacitor C1, diodes D1, D2, D3, and D4, MOSFETs (Q1, Q2) that are switching elements, and a shunt resistor R1. The diodes D1, D2, D3, D4 and the MOSFETs (Q1, Q2) constitute the bridge rectifier circuit 10.
The MOSFETs (Q1, Q2) are switching elements, the diode D3 is a parasitic diode of the MOSFET (Q1), and the diode D4 is a parasitic diode of the MOSFET (Q2). Further, the saturation voltage of the MOSFET (Q1) and the MOSFET (Q2) is lower than the forward voltage drop of the diodes D1, D2 and the parasitic diodes D3, D4.

  The DC power supply device 1 further includes a current detection unit 11, a gain control unit 12, an AC voltage detection unit 13, a zero cross determination unit 14, a load detection unit 15, a boost ratio control unit 16, and a DC voltage detection unit. 17, a converter control unit 18, and a power supply circuit 19. The power supply circuit 19 generates a power supply voltage 15V for driving the MOSFETs (Q1, Q2) and a control voltage 5V for driving a control IC (not shown) such as a microcomputer from the DC voltage.

  The diodes D1, D2 and the MOSFETs (Q1, Q2) are bridge-connected. The anode of the diode D1 is connected to the cathode of the diode D2, and the connection point N1 is connected to one end of the AC power supply VS via the wiring hb.

The source of the MOSFET (Q1) is connected to the drain of the MOSFET (Q2). The source of the MOSFET (Q1) is connected to one end of the AC power supply VS via the connection point N2, the wiring ha, and the reactor L1.
The anode of the diode D2 is connected to the source of the MOSFET (Q2).
The drain of the MOSFET (Q1) is connected to the cathode of the diode D1.

  The cathode of the diode D1 and the drain of the MOSFET (Q1) are connected to the positive electrode of the smoothing capacitor C1 and one end of the load H through the wiring hc. Furthermore, the sources of the diode D2 and the MOSFET (Q2) are connected to the negative electrode of the smoothing capacitor C1 and the other end of the load H through the shunt resistor R1 and the wiring hd, respectively.

  Reactor L1 is provided on wiring ha, that is, between AC power supply VS and bridge rectifier circuit 10. The reactor L1 stores electric power supplied from the AC power source VS as energy, and further boosts the energy by releasing this energy.

  The smoothing capacitor C1 smoothes the voltage rectified through the diode D1 and the MOSFET (Q1) to obtain a DC voltage Vd. The smoothing capacitor C1 is connected to the output side of the bridge rectifier circuit 10, and the positive electrode side is connected to the wiring hc and the negative electrode side is connected to the wiring hd.

  The MOSFETs (Q1, Q2) that are switching elements are on / off controlled by a command from the converter control unit 18 to be described later. By using MOSFETs (Q1, Q2) as switching elements, switching can be performed at a high speed, and so-called synchronous rectification control can be performed by passing a current through a MOSFET with a small voltage drop. Conduction loss can be reduced.

  By using a super junction MOSFET having a small on-resistance as the MOSFETs (Q1, Q2), it is possible to further reduce conduction loss. Here, a reverse recovery current is generated in the parasitic diode of the MOSFET during a circuit short-circuit operation. In particular, the parasitic diode of the super junction MOSFET has a problem that the reverse recovery current is larger than that of a normal MOSFET parasitic diode and the switching loss is large. Therefore, switching loss can be reduced by using a MOSFET having a short reverse recovery time (trr) as the MOSFET (Q1, Q2).

Since diodes D1 and D2 do not generate a reverse recovery current even during active operation, it is preferable to select diodes D1 and D2 having a low forward voltage. For example, it is possible to reduce the conduction loss of the circuit by using a general rectifier diode or a high breakdown voltage Schottky barrier diode.
The shunt resistor R1 has a function of detecting an instantaneous current flowing through the circuit.
The current detection unit 11 has a function of detecting an average current flowing through the circuit.

  The gain control unit 12 has a function of controlling a current control gain Kp determined from the circuit current effective value Is and the DC voltage boost ratio a. At this time, by controlling Kp × Is to a predetermined value, the DC voltage Vd can be boosted a times from the AC power supply voltage Vs.

  The AC voltage detector 13 detects an AC power supply voltage Vs applied from the AC power supply VS, and is connected to the wirings ha and hb. The AC voltage detection unit 13 outputs the detected value to the zero cross determination unit 14.

  The zero cross determination unit 14 has a function of determining whether the polarity of the AC power supply voltage Vs detected by the AC voltage detection unit 13 has been switched, that is, whether the zero cross point has been reached. The zero cross determination unit 14 is a polarity detection unit that detects the polarity of the AC power supply voltage Vs. For example, the zero-cross determination unit 14 outputs a signal “1” to the converter control unit 18 while the AC power supply voltage Vs is positive, and outputs “1” to the converter control unit 18 when the AC power supply voltage Vs is negative. A 0 'signal is output.

  The load detection unit 15 is configured by a shunt resistor (not shown), for example, and has a function of detecting a current flowing through the load H. When the load H is an inverter or a motor, the rotation speed of the motor or the applied voltage of the motor may be calculated based on the load current detected by the load detection unit 15. Also, the modulation factor of the inverter may be calculated from the DC voltage detected by the DC voltage detector 17 described later and the applied voltage of the motor. The load detection unit 15 outputs the detection value (current, motor rotation speed, modulation rate, etc.) to the boost ratio control unit 16.

  The step-up ratio control unit 16 selects the step-up ratio a of the DC voltage Vd from the detection value of the load detection unit 15 and outputs the selection result to the converter control unit 18. Then, the converter controller 18 performs switching control by outputting a drive pulse to the MOSFETs (Q1, Q2) so as to boost the DC voltage Vd to the target voltage.

  The DC voltage detector 17 detects the DC voltage Vd applied to the smoothing capacitor C1, and its positive side is connected to the wiring hc and its negative side is connected to the wiring hd. DC voltage detection unit 17 outputs the detected value to converter control unit 18. Note that the detection value of the DC voltage detection unit 17 is used to determine whether or not the voltage value applied to the load H has reached a predetermined target value.

  The block M including the converter control unit 18 is, for example, a microcomputer (not shown), reads out a program stored in a ROM (Read Only Memory), develops it in a RAM (Random Access Memory), and a CPU (Central Processing Unit) executes various processes. Based on information input from the current detection unit 11 or the shunt resistor R1, the gain control unit 12, the zero cross determination unit 14, the step-up ratio control unit 16, and the DC voltage detection unit 17, the converter control unit 18 performs MOSFET (Q1, Control on / off of Q2). The processing executed by converter control unit 18 will be described later.

The inrush current prevention circuit U1 has a configuration in which an inrush current prevention relay SW1 and an inrush current prevention resistor R3 are connected in series with the power relay SW2, and has a function of a protection circuit for reducing the inrush current.
Hereinafter, protection control for inrush current by the inrush current prevention circuit U1 will be described.

FIG. 2 is a diagram showing the path of the input current is flowing through the circuit when the power relay SW2 is turned on with the power supply voltage having a positive polarity and no inrush current prevention circuit U1.
That is, the current flows in the order of AC power supply VS → reactor L1 → diode D3 → smoothing capacitor C1 → shunt resistor R1 → diode D2 → current detection unit 11 → AC power supply VS.

3A to 3E are waveform diagrams showing the operation state of each block in the case shown in FIG.
FIG. 3A shows on / off of the power relay SW2.
FIG. 3B shows the DC voltage Vd.
FIG. 3C shows the input current is (instantaneous value).
FIG. 3D shows a 15V power supply applied by the power supply circuit 19.
FIG. 3E shows the voltage at the microcomputer reset terminal.
First, at the moment when the power relay SW2 is turned on (see FIG. 3A), a large input current (inrush current) is (see FIG. 3C) flows, and the smoothing capacitor C1 is charged. At this time, the DC voltage Vd is a surge generated as shown in FIG. Thereafter, 15V (see FIG. 3D) and 5V are generated by the power supply circuit 19, the microcomputer is reset (see FIG. 3E), and the DC power supply device 1 becomes operable.

This inrush current varies depending on the capacitance of the smoothing capacitor C1 and the inductance value of the reactor L1. Specifically, the inrush current increases as the capacitance of the smoothing capacitor C1 increases and the inductance of the reactor L1 decreases.
If this inrush current is excessive, the characteristics of the diodes D1 and D2 and the MOSFETs (Q1 and Q2) are deteriorated, which may lead to destruction in the worst case.
Therefore, the present embodiment is characterized by the following two points in order to avoid such a state.

The first characteristic point is that the MOSFETs (Q1, Q2) are turned off during inrush current flow, and current is passed through the parasitic diodes D3, D4.
The DC power supply device 1 of the present embodiment has a configuration in which MOSFETs (Q1, Q2) are assembled vertically. As another form that changes to this configuration, there can be considered two lower configurations in which the MOSFET (Q1) is disposed at the position of the diode D2 of the present embodiment and the diode D2 is disposed at the position of the MOSFET (Q1) of the present embodiment. In the case of this configuration, when a short circuit operation described later is performed, a short circuit current directly flows through the ground (wiring hd) without the reactor L1 between the AC power supply VS and the MOSFETs (Q1, Q2). Noise becomes excessive.

  In order to avoid this, in the DC power supply device 1 of the present embodiment, the MOSFETs (Q1, Q2) are arranged vertically. However, in the case of this configuration, reverse recovery current (Irr) is generated in the parasitic diode D3 of the MOSFET (Q1) when the circuit short circuit operation is performed as described above. The switching loss increases as the reverse recovery current increases and the switching frequency increases. Therefore, in the direct-collection power supply device of this embodiment, a high-speed trr type MOSFET having a small Irr, that is, a fast reverse recovery time (trr) is used for the MOSFET (Q1).

However, the high-speed trr MOSFET is characterized by a large loss at a large current.
FIG. 4 is a characteristic diagram of the drain voltage-drain reverse current of the MOSFET.
Here, the drain voltage is a voltage when the drain is viewed from the MOSFET source. The drain reverse current means a current flowing from the source to the drain of the MOSFET. The solid line is the data for the high-speed trr type MOSFET, and the alternate long and short dash line is the data for the low-speed trr type MOSFET. It can be seen that the characteristics of the low-speed trr type and the high-speed trr type are reversed at the intersection M1 (the point where the drain reverse current is about 3 A). For example, when the drain reverse current is 100 A, the drain-source voltage is higher in the high-speed trr type MOSFET than in the low-speed trr type MOSFET. That is, it can be said that the high-speed trr type has a larger loss at a large current. In other words, it can be said that the high-speed trr type is more easily broken by the inrush current.
That is, in the case of the configuration in which the MOSFETs (Q1, Q2) are arranged above and below as in the present embodiment, the device characteristics are easily deteriorated or destroyed by the inrush current as compared with the lower two configurations described above. It is necessary to ensure protection against
Therefore, in the present embodiment, the MOSFETs (Q1, Q2) are turned off when inrush current flows, and the inrush current is passed through the parasitic diodes D3, D4. That is, the diode rectification operation using the diodes D1 to D4 is performed without turning on the MOSFETs (Q1, Q2).

The broken line in FIG. 4 indicates data in a state where 15 V is applied to the gate, that is, the MOSFET is in the on state, in the high-speed trr type element.
Compared with the data of the solid line in the gate voltage 0V and OFF state described above, the characteristics are reversed at the intersection M2 (drain reverse current is about 60 A). For example, when the drain reverse current is 100 A, the drain-source voltage is smaller at Vg = 0V than at Vg = 15V, and therefore the loss is smaller at Vg = 0V. Therefore, in the case of a large current, the MOSFET has an effect of preventing special deterioration and destruction of the element due to the inrush current because the gate voltage is 0 V, that is, the inrush current is passed through the parasitic diode in the off state. .

The second feature point is reduction of inrush current by the inrush current prevention circuit U1.
The DC power supply device 1 of this embodiment includes an inrush current prevention circuit U1 for reducing the inrush current shown in FIG.

FIGS. 5A to 5F are time charts when the inrush current prevention circuit U1 is operated.
FIG. 5A shows on / off of the inrush current prevention relay SW1.
FIG. 5B shows on / off of the power relay SW2.
FIG. 5C shows the DC voltage Vd.
FIG. 5D shows the input current is (instantaneous value).
FIG. 5E shows a 15 V power supply applied by the power supply circuit 19.
FIG. 5F shows the voltage at the microcomputer reset terminal.
First, as shown in FIG. 5A, the inrush current prevention relay SW1 is turned on for a predetermined time T1. This is called an inrush current protection operation. At this time, the path of the current flowing through the circuit is as shown in FIG. That is, the current flows in the order of AC power supply VS → inrush current prevention relay SW1 → current prevention resistance R3 → reactor L1 → diode D3 → smoothing capacitor C1 → shunt resistance R1 → diode D2 → current detection unit 11 → AC power supply VS.
An inrush current prevention resistor R3 is installed at the tip of the inrush current prevention relay SW1, and the electric charge is charged to the smoothing capacitor C1 with a slower time constant than the circuit operation of FIG. 2 through the inrush current prevention resistor R3. Then, as shown in FIG. 5 (c), the DC voltage Vd rises gently, 15V (see FIG. 5 (e)) and 5V are generated by the power supply circuit 19, and the microcomputer is reset (see FIG. 3 (f)). The At this time, the smoothing capacitor C1 is not fully charged.
Then, after the predetermined time T1 has elapsed, the remaining charge is charged by turning on the power relay SW2 (see FIG. 5B) and turning off the inrush current prevention relay SW1 (see FIG. 5A). The smoothing capacitor C1 is fully charged. The subsequent steps are called normal operation.
At this time, the path of the current flowing through the circuit is as shown in FIG. That is, the current flows in the order of AC power supply VS → power relay SW2 → reactor L1 → diode D3 → smoothing capacitor C1 → shunt resistor R1 → diode D2 → current detection unit 11 → AC power supply VS.
Thereafter, the inrush current prevention relay SW1 is always turned off. As described above, since the power relay SW2 is turned on in a state where the smoothing capacitor C1 is charged in advance, the inrush current shown in FIG. 5D is smaller than the inrush current shown in FIG. It is possible to prevent a large current from flowing.
The reason why the inrush current prevention relay SW1 is turned off before the smoothing capacitor C1 is fully charged is to prevent the inrush current prevention resistor R3 from generating heat.

FIG. 8 is a diagram illustrating a current path that flows through the circuit when the inrush current prevention relay is turned on when the AC power supply voltage has a negative polarity.
That is, the current flows in the order of AC power supply VS → current detection unit 11 → diode D1 → smoothing capacitor C1 → shunt resistor R1 → diode D4 → reactor L1 → current prevention resistor R3 → inrush current prevention relay SW1 → AC power supply VS. Through the inrush current prevention resistor R3, the electric charge is charged to the smoothing capacitor C1 with a time constant slower than the circuit operation of FIG. 2, and the DC voltage Vd rises gently as shown in FIG. 19 generates 15V (see FIG. 5 (e)) and 5V, and the microcomputer is reset (see FIG. 3 (f)).

FIG. 9 is a diagram illustrating a current path that flows through the circuit when the power relay is turned on after the inrush current prevention relay SW1 is turned from on to off when the AC power supply voltage has a negative polarity.
That is, the current flows in the order of AC power supply VS → current detection unit 11 → diode D1 → smoothing capacitor C1 → shunt resistor R1 → diode D4 → reactor L1 → power relay SW2 → AC power supply VS. Since the power relay SW2 is turned on with the smoothing capacitor C1 charged in advance, the inrush current shown in FIG. 5 (d) is smaller than the inrush current shown in FIG. 3 (c), and thus an excessive current flows. It is possible to prevent this.

  As described above, the DC power supply device 1 according to the present embodiment turns on the AC power by turning off the MOSFETs (Q1, Q2) when the inrush current flows and installing the inrush current prevention circuit U1. It is possible to prevent the characteristic deterioration and destruction of the MOSFETs (Q1, Q2) due to the inrush current derived from time to time.

  For example, by setting the resistance value of the inrush current prevention resistor R3 to 500Ω or more and the on time of the inrush current prevention relay SW1 to 1.2 seconds or more, it is possible to protect circuit elements such as MOSFETs and diodes from the inrush current. is there.

  Next, the operation mode of the DC power supply device 1 of the present embodiment will be described. Hereinafter, the DC power supply device 1 is driven in the operation mode to be described after the inrush current protection control is performed after the AC power supply voltage Vs is applied. That is, at this time, the inrush current prevention relay SW1 is always off and the power relay SW2 is always on.

  The operation modes of the DC power supply device 1 are roughly classified into four modes: a diode rectification mode, a synchronous rectification mode, a partial switching mode, and a high-speed switching mode. The partial switching mode and the high-speed switching mode are modes in which the converter performs an active operation (power factor improving operation), and the DC voltage Vd is boosted and the power factor is improved by passing a power factor improving current through the bridge rectifier circuit 10. It is a mode to perform. For example, when a load such as an inverter or a motor is large, it is necessary to boost the DC voltage Vd. Further, as the load increases and the current flowing through the DC power supply device 1 increases, the harmonic current also increases. Therefore, in the case of a high load, it is necessary to boost the voltage in the partial switching mode or the high-speed switching mode to reduce the harmonic current, that is, improve the power factor of the power supply input.

≪Diode rectification mode≫
The diode rectification mode is a mode in which full-wave rectification is performed using four diodes D1 to D4. In this mode, the MOSFET (Q1) and the MOSFET (Q2) are off.
FIG. 10 shows a current path that flows through the circuit when diode rectification is performed when the AC power supply voltage Vs has a positive polarity.
In FIG. 10, current flows in the direction indicated by the broken-line arrow during a period in which the AC power supply voltage Vs is a positive half cycle. That is, the current flows in the order of AC power supply VS → reactor L1 → parasitic diode D3 → smoothing capacitor C1 → shunt resistor R1 → diode D2 → AC power supply VS.
FIG. 11 shows a current path that flows through the circuit when diode rectification is performed when the AC power supply voltage Vs has a negative polarity.
In FIG. 11, current flows in a direction indicated by a broken-line arrow during a half cycle in which the AC power supply voltage Vs is negative. That is, the current flows in the order of AC power supply VS → diode D1 → smoothing capacitor C1 → shunt resistor R1 → parasitic diode D4 → reactor L1 → AC power supply VS.

≪Synchronous rectification mode≫
In order to perform high-efficiency operation with respect to the diode rectification described above, synchronous rectification control is performed by switching the MOSFETs (Q1, Q2) according to the polarity of the AC power supply voltage Vs.

FIG. 12 is a diagram illustrating a current path that flows through the circuit when synchronous rectification is performed when the AC power supply voltage Vs has a positive polarity.
In FIG. 12, a current flows in a direction indicated by a broken-line arrow during a period in which the AC power supply voltage Vs is a positive half cycle. That is, the current flows in the order of AC power supply VS → reactor L1 → MOSFET (Q1) → smoothing capacitor C1 → shunt resistor R1 → diode D2 → AC power supply VS. At this time, the MOSFET (Q2) is always off, and the MOSFET (Q1) is always on. If the MOSFET (Q1) is not in the ON state, current flows through the parasitic diode D3 of the MOSFET (Q1) as in the above-described diode rectification operation. However, since the forward voltage drop of the parasitic diode of the MOSFET is usually large, a large conduction loss occurs. Therefore, the conduction loss can be reduced by turning on the MOSFET (Q1) and allowing a current to flow through the on-resistance portion of the MOSFET (Q1). This is the principle of so-called synchronous rectification control. Note that the on-operation start timing of the MOSFET (Q1) is determined from the zero cross timing at which the polarity of the AC power supply voltage Vs switches from negative to positive. The timing at which the MOSFET (Q1) is turned off is the timing at which the polarity of the AC power supply voltage Vs is switched from positive to negative.

FIG. 13 is a diagram illustrating a current path that flows through the circuit when synchronous rectification is performed when the AC power supply voltage Vs has a negative polarity.
In FIG. 13, current flows in the direction indicated by the broken-line arrow during the half cycle of the negative AC power supply voltage Vs. That is, current flows in the order of AC power supply VS → diode D1 → smoothing capacitor C1 → shunt resistor R1 → MOSFET (Q2) → reactor L1 → AC power supply VS. At this time, the MOSFET (Q1) is always off, and the MOSFET (Q2) is always on. Note that the on-operation start timing of the MOSFET (Q2) is determined from the zero cross timing at which the polarity of the AC power supply voltage Vs is switched from positive to negative. The timing at which the MOSFET (Q2) is turned off is the timing at which the polarity of the AC power supply voltage Vs is switched from negative to positive.
By operating the DC power supply device 1 as described above, high-efficiency operation is possible.

FIGS. 14A to 14D are waveform diagrams of the AC power supply voltage Vs, the circuit current is, and the MOSFET drive pulse at the time of synchronous rectification.
FIG. 14A shows the waveform of the instantaneous value vs of the AC power supply voltage, and FIG. 14B shows the waveform of the circuit current is. FIG. 14C shows a drive pulse waveform of the MOSFET (Q1), and FIG. 14D shows a drive pulse waveform of the MOSFET (Q2).
As shown in FIG. 14A, the instantaneous value vs of the AC power supply voltage has a substantially sinusoidal waveform.
As shown in FIG. 14C, the drive pulse of the MOSFET (Q1) is at the H level when the polarity of the AC power supply voltage Vs is positive, and at the L level when negative.
As shown in FIG. 14D, the drive pulse of the MOSFET (Q2) is inverted from the drive pulse of the MOSFET (Q1). When the polarity of the AC power supply voltage Vs is positive, the drive pulse is L level. Becomes H level.
As shown in FIG. 14B, the circuit current is flows when the AC power supply voltage Vs reaches a predetermined amplitude, that is, when the AC power supply voltage Vs is larger than the DC voltage Vd.

≪Overview of switching operation≫
Next, a high-speed switching operation for boosting the DC voltage Vd and improving the power factor will be described.
In this operation mode, the MOSFETs (Q1, Q2) are controlled to be switched at a certain switching frequency, the circuit is short-circuited via the reactor L1 (hereinafter referred to as power factor correction operation), and a short-circuit current (hereinafter referred to as power factor) is supplied to the circuit. The DC voltage Vd is boosted and the power factor is improved. First, the operation when the power factor correction current is passed will be described.

When the AC power supply voltage Vs is synchronously rectified in a positive cycle, the current flow is as shown in FIG. 12, and the operation of the MOSFETs (Q1, Q2) is as described above. At this time, as shown in FIG. 14B, the circuit current is is distorted with respect to the power supply voltage. This is because the current flows only when the DC voltage Vd becomes smaller than the AC power supply voltage Vs, and the characteristic of the reactor L1.
Therefore, the power factor correction current is passed through the circuit a plurality of times, and the power factor is improved by bringing the circuit current close to a sine wave, thereby reducing the harmonic current.

FIG. 15 is a diagram showing a path of the power factor correction current isp that flows when the MOSFET (Q2) is turned on in a positive cycle of the power supply voltage.
The path of the power factor correction current isp is in the order of AC power supply VS → reactor L1 → MOSFET (Q2) → diode D2 → AC power supply VS. At this time, the energy represented by the following formula (1) is stored in the reactor L1. This energy is discharged to the smoothing capacitor C1, so that the DC voltage Vd is boosted.

  The flow of current when synchronous rectification is performed in a cycle where the AC power supply voltage Vs is negative is as shown in FIG. 13, and the operation of the MOSFETs (Q1, Q2) is as described above.

FIG. 16 is a diagram illustrating a path when the power factor correction current isp is caused to flow by turning on the MOSFET (Q1) in a cycle in which the power supply voltage is negative.
The current path is in the order of AC power supply VS → diode D1 → MOSFET (Q1) → reactor L1 → AC power supply VS. Also at this time, as described above, energy is stored in the reactor L1, and the DC voltage Vd is boosted by the energy.

FIGS. 17A to 17D are waveform diagrams of the AC power supply voltage Vs, the circuit current is, and the MOSFET driving pulse when the power factor correction current is passed twice (referred to as two shots).
FIG. 17A shows the waveform of the instantaneous value vs of the AC power supply voltage, and FIG. 9B shows the waveform of the circuit current is. FIG. 9C shows a drive pulse waveform of the MOSFET (Q1), and FIG. 17D shows a drive pulse waveform of the MOSFET (Q2).
As shown in FIG. 17A, the instantaneous value vs of the AC power supply voltage has a substantially sinusoidal waveform.

As shown in FIG. 17C, the drive pulse of the MOSFET (Q1) becomes H level when the polarity of the AC power supply voltage Vs is positive, and further becomes two L level pulses at a predetermined timing. It becomes L level when the polarity of the AC power supply voltage Vs is negative, and further becomes two H level pulses at a predetermined timing.
As shown in FIG. 17C, the drive pulse for the MOSFET (Q2) is inverted from the drive pulse for the MOSFET (Q1). This is because the power factor correction operation and synchronous rectification are combined. For example, when the AC power supply voltage Vs has a positive polarity, the MOSFET (Q2) is turned on to perform the power factor correction operation. Thereafter, after the MOSFET (Q1) is turned off, the section in which the MOSFET (Q2) is turned on is a synchronous rectification operation. In this way, by combining the power factor improvement operation and the synchronous flow operation, high efficiency operation is possible while improving the power factor.

  As shown in FIG. 17B, the circuit current is rises when the AC power supply voltage Vs is positive and the drive pulse of the MOSFET (Q2) becomes H level, and the AC power supply voltage Vs is negative and It rises when the drive pulse of the MOSFET (Q1) becomes H level. Thereby, a power factor is improved.

  For example, when the AC power supply voltage Vs is positive, the current path during the power factor correction operation is as shown in FIG. The current path when the MOSFET (Q2) is turned off and the MOSFET (Q1) is turned on to switch to the synchronous rectification operation is as shown in FIG.

  Note that this power factor correction operation and the above-described diode rectification operation may be combined. That is, when the AC power supply voltage Vs has a positive polarity, the current path during the power factor correction operation is as shown in FIG. FIG. 7 shows a current path when the parasitic diode D3 is turned on after the MOSFET (Q2) is turned off to switch to the diode rectification operation.

18A to 18D are waveform diagrams of the AC power supply voltage Vs, the circuit current is, and the MOSFET drive pulse when high-speed switching is performed.
18A shows the waveform of the instantaneous value vs of the AC power supply voltage, and FIG. 18B shows the waveform of the circuit current is. FIG. 18C shows the drive pulse waveform of the MOSFET (Q1), and FIG. 18D shows the drive pulse waveform of the MOSFET (Q2).
As shown in FIG. 18A, the instantaneous value vs of the AC power supply voltage has a substantially sinusoidal waveform.

  In the high-speed switching operation, for example, when the power supply voltage has a positive polarity, the power factor correction current isp is caused to flow by turning on the MOSFET (Q2) and turning off the MOSFET (Q1) during the power factor correction operation. . Next, the MOSFET (Q2) is turned off and the MOSFET (Q1) is turned on. Thus, the reason why the MOSFETs (Q1, Q2) are switched on and off in accordance with the presence / absence of the power factor correction operation is that synchronous rectification is performed. In order to simply perform a high-speed switching operation, the MOSFET (Q1) is always in an off state, and the MOSFET (Q2) may be switched at a constant frequency.

However, at this time, if the MOSFET (Q1) is also in the off state when the MOSFET (Q2) is off, the current flows through the parasitic diode D3 of the MOSFET (Q1). As described above, since the forward voltage drop of the parasitic diode is large, a large conduction loss occurs. Therefore, in this embodiment, when the MOSFET (Q2) is off, the conduction loss is reduced by performing synchronous rectification by turning on the MOSFET (Q1).
The circuit current is (instantaneous value) flowing through the DC power supply device 1 can be expressed by the following equation (2).

Furthermore, when this equation (2) is rewritten, the following equation (3) is obtained.

Equation (4) shows the relationship between the circuit current is (instantaneous value) and the circuit current effective value Is. Note that is (instantaneous value) is a value detected by the shunt resistor R1, and the circuit current effective value Is is a value detected by the current detection unit 11.

When formula (3) is modified and formula (4) is substituted, formula (5) below is obtained.

When the reciprocal of the step-up ratio is the right side, the following equation (6) is obtained.

Furthermore, the duty d of the MOSFET can be expressed as in Expression (7).

  As described above, by controlling Kp × Is shown in the equation (6), the voltage can be boosted to a times the effective value of the AC power supply voltage Vs, and the duty d (conductivity) of the MOSFET at that time is expressed by the equation It can be given by (7).

  FIG. 19 is a diagram illustrating the on-duty relationship of the drive pulses of the MOSFET (Q2) and the MOSFET (Q1) in the half cycle of the power supply voltage (positive polarity). The vertical axis in FIG. 19 indicates the on-duty, and the horizontal axis indicates the time corresponding to a half cycle of the positive polarity power supply voltage.

  The on-duty of the drive pulse of the MOSFET (Q1) indicated by the broken line is proportional to the AC power supply voltage Vs. The on-duty of the driving pulse of the MOSFET (Q2) indicated by a two-dot chain line is obtained by subtracting the on-duty of the driving pulse of the MOSFET (Q1) from 1.0.

In FIG. 19, as shown by the equation (7), as the circuit current is increases, the duty d of the drive pulse of the MOSFET (Q2) that performs the switching operation in order to flow the power factor correction current decreases. As is is smaller, the duty d of the driving pulse of the MOSFET (Q2) is larger. The duty d of the drive pulse of the MOSFET (Q1) on the side where the synchronous rectification is performed has a reverse characteristic to the duty d of the drive pulse of the MOSFET (Q2).
Actually, it is necessary to consider the dead time in order to avoid the vertical short circuit.

FIG. 20 is a diagram in which the on-duty of the drive pulse of the MOSFET (Q2) in consideration of the dead time in the half cycle of the power supply voltage (positive polarity) is additionally written with a solid line. The vertical axis in FIG. 12 indicates the on-duty, and the horizontal axis indicates the time corresponding to the positive half cycle of the AC power supply voltage Vs.
As described above, when a predetermined dead time is given, the duty of the driving pulse of the MOSFET (Q2) is reduced by this dead time.

  FIG. 21 is a diagram showing the relationship between the instantaneous value vs of the AC power supply voltage Vs and the circuit current is (instantaneous value). The solid line indicates the instantaneous value vs of the AC power supply voltage Vs, and the broken line indicates the instantaneous value of the circuit current is. The horizontal axis of FIG. 21 shows the time for a half cycle of the positive polarity power supply voltage.

As shown in FIG. 21, the instantaneous value vs of the AC power supply voltage Vs and the circuit current is (instantaneous value) are both substantially sinusoidal due to the high-speed switching control, so that the power factor can be improved.
The duty d _Q2 of the MOSFET (Q2) is shown in the following equation (8).

The duty d _Q1 of the MOSFET (Q1) is shown in the following equation (9).

  Further, looking at the relationship between the power supply voltage and the current, the circuit current is is controlled in a sine wave shape, so that the power factor is good. This assumes a state where the inductance of the reactor L1 is small and there is no phase lag of the current with respect to the power supply voltage. If the inductance of the reactor L1 is large and the current phase is delayed with respect to the voltage phase, the duty d may be set in consideration of the current phase.

  FIG. 22 is a diagram showing the duty of MOSFET (Q2) when the delay of the current phase due to reactor L1 is taken into account when AC power supply voltage Vs is positive. The vertical axis in FIG. 22 indicates the duty of the MOSFET (Q2), and the horizontal axis indicates the time corresponding to a half cycle of the positive polarity power supply voltage.

  The solid line indicates the duty of MOSFET (Q2) when the current phase delay due to reactor L1 is not taken into consideration. The broken line indicates the duty of MOSFET (Q2) when the delay of the current phase due to reactor L1 is taken into consideration. By controlling in this way, even if the inductance of the reactor L1 is large, the current can be controlled in a sine wave shape.

  In the above, the case where it implements combining high speed switching and synchronous rectification has been demonstrated. As described above, high-speed switching and diode rectification may be combined. That is, when the AC power supply voltage Vs has a positive polarity, the MOSFET (Q1) is always turned off and only the MOSFET (Q2) is switched at high speed. Even if control is performed in this manner, an effect of improving the power factor can be obtained.

≪Partial switching operation≫
As described above, the circuit current is can be shaped into a sine wave by performing a high-speed switching operation, and a high power factor can be ensured. However, the switching loss increases as the switching frequency increases.

Since the harmonic current increases as the circuit input increases, it becomes difficult to satisfy the regulation value of the higher-order harmonic current in particular. Therefore, it is necessary to ensure a high power factor as the input current increases. Conversely, when the input is small, the harmonic current is also small, so there is a case where it is not necessary to secure a power factor more than necessary. In other words, it can be said that the harmonic current may be reduced by securing an optimum power factor while considering the efficiency in accordance with the load condition.
Therefore, in order to improve the power factor while suppressing an increase in switching loss, a partial switching operation may be performed.

  The partial switching operation is not a power factor correction operation at a predetermined frequency as in a high-speed switching operation, but a direct current by performing a plurality of power factor correction operations at a predetermined phase in a half cycle of the AC power supply voltage Vs. This is an operation mode for boosting the voltage Vd and improving the power factor. Compared with the case of high-speed switching operation, the switching loss of the MOSFETs (Q1, Q2) is reduced, so that the switching loss can be reduced. Hereinafter, the partial switching operation will be described with reference to FIG.

FIGS. 23A to 23D are diagrams showing the relationship between the drive pulse of the MOSFET (Q1), the AC power supply voltage Vs, and the circuit current is in the cycle in which the AC power supply voltage Vs is positive.
FIG. 23A shows the instantaneous value vs of the AC power supply voltage, and FIG. 23B shows the circuit current is. FIG. 23C shows a driving pulse for the MOSFET (Q2), and FIG. 23D shows a driving pulse for the MOSFET (Q1).
As shown in FIG. 23A, the instantaneous value vs of the AC power supply voltage is substantially sinusoidal.
The alternate long and short dash line in FIG. 23B shows the ideal circuit current is in a substantially sine wave shape. At this time, the power factor is most improved.

  Here, for example, when the point P1 on the ideal current is considered, the slope at this point is set to di (P1) / dt. Next, the slope of the current when the MOSFET (Q2) is turned on for a time ton1_Q2 from the state where the current is zero is set to di (ton1_Q2) / dt. Furthermore, after the power is turned on for the time ton1_Q2, the current gradient when the power is turned off for the time toff_Q2 is set to di (toff1_Q2) / dt. At this time, control is performed so that the average value of di (ton1_Q2) / dt and di (toff1_Q2) / dt is equal to the slope di (P1) / dt at the point P1.

  Next, similarly to the point P1, the slope of the current at the point P2 is set to di (P2) / dt. The slope of the current when the MOSFET (Q2) is turned on over time ton2_Q2 is set as di (ton2_Q2) / dt, and the slope of the current when turned off over time toff2_Q2 is set as di (toff2_Q2) / dt. . As in the case of the point P1, the average value of di (ton2_Q2) / dt and di (toff2_Q2) / dt is set equal to the slope di (P2) / dt at the point P2. This is repeated thereafter. At this time, it can be approximated to an ideal sine wave as the switching frequency of the MOSFET (Q2) is increased.

The reason why the switching of the MOSFET (Q1) and the MOSFET (Q2) is switched complementarily is that the partial switching operation and the synchronous rectification operation are performed in combination.
In some cases, partial switching operation and diode rectification operation may be combined.

≪Switching control mode≫
The DC power supply device 1 of the present embodiment can perform diode rectification control, synchronous rectification control, partial switching control, and high-speed switching control. For example, depending on the equipment to be used, required performance may change depending on the load condition, such as a high efficiency priority area, a boosting and power factor improvement priority area, and the like. Therefore, by selectively switching the modes for performing the four controls described above based on predetermined threshold information, it is possible to achieve both higher efficiency and lower harmonic current more optimally.
Hereinafter, the mode switched by each control method is shown.

FIG. 24 is a diagram illustrating switching of the operation mode of the DC power supply device according to the size of the load. In this figure, the first threshold is abbreviated as “threshold # 1”, and the second threshold is abbreviated as “threshold # 2”. Also, the first to eighth control methods are simply abbreviated as “# 1” to “# 8”.
The first control method switches between a mode in which synchronous rectification control is performed and a mode in which synchronous rectification control and partial switching control are performed simultaneously, based on predetermined first threshold information. In the drawing, the partial switching control is abbreviated as “partial SW”.

  The second control method switches between a mode in which synchronous rectification control is performed and a mode in which synchronous rectification control and high-speed switching control are performed at the same time based on predetermined first threshold information. In the drawings, the high-speed switching control is abbreviated as “high-speed SW”.

  The third control method includes a mode in which synchronous rectification control is performed based on predetermined first and second threshold information, a mode in which synchronous rectification control and partial switching control are simultaneously performed, synchronous rectification control and high-speed switching. This is to switch between the modes in which the control is performed simultaneously.

  The fourth control method is to switch between a mode in which synchronous rectification control is performed and a mode in which diode rectification control and partial switching control are simultaneously performed based on predetermined first threshold information.

  The fifth control method switches between a mode in which synchronous rectification control is performed and a mode in which diode rectification control and high-speed switching control are performed simultaneously based on predetermined first threshold information.

  The sixth control method includes a mode for performing synchronous rectification control, a mode for simultaneously performing diode rectification control and partial switching control, diode rectification control, and high speed based on predetermined first and second threshold information The mode for switching the switching control simultaneously is switched.

  The seventh control method includes a mode in which synchronous rectification control is performed, a mode in which diode rectification control and partial switching control are simultaneously performed, synchronous rectification control and high speed based on first and second threshold information determined in advance. The mode for switching the switching control simultaneously is switched.

  The eighth control method includes a mode in which synchronous rectification control is performed, a mode in which synchronous rectification control and partial switching control are simultaneously performed, diode rectification control, and high-speed control based on predetermined first and second threshold information. The mode for switching the switching control simultaneously is switched.

  For example, if the main purpose is to improve efficiency, reduce harmonic current, or boost the voltage, the first to third control methods may be used. Further, efficiency is not so much a priority, and if the main purpose is to reduce harmonic current or boost the voltage, the mode may be switched in the fourth to sixth control methods. For example, when a partial switching operation or a high-speed switching operation is combined with a synchronous rectification operation, it is necessary to control two MOSFETs in an AC power supply voltage half cycle, which makes the control complicated. However, in combination with diode rectification, the number of MOSFETs to be controlled is one in a half cycle, which leads to simplification of control. In short, optimal control may be selected as necessary, such as efficiency, harmonic reduction, and controllability.

  Note that threshold information serving as a trigger for control switching includes, for example, a circuit current detected by the current detection unit 11. Alternatively, the load information detected by the load detection unit 15 may be used. As the load information, for example, when the load H is a motor or an inverter, a motor current, a motor rotation speed, a modulation rate, a DC voltage, or the like may be used.

  Further, when the control is switched between the two modes as in the first, second, fourth, and fifth control methods, the threshold information may be one (first threshold information). When switching between the three modes as in the third, sixth, seventh, and eighth control methods, two pieces of threshold information (first threshold information and second threshold information) are prepared. Furthermore, the first threshold information and the second threshold information are related to the magnitude of the load. That is, there is a relationship that the first threshold information is larger than the second threshold information.

  For example, in the third control method, a synchronous rectification operation is performed in a region below the first threshold, a synchronous rectification operation + partial switching operation is performed in a region greater than or equal to the first threshold and less than the second threshold, In the region above the threshold value, the synchronous rectification operation + high-speed switching operation is performed. The same applies to other modes.

Further, when switching from the partial switching operation to the high-speed switching operation as in the third and sixth to eighth control methods, the DC voltage Vd may fluctuate. This is because the power factor is better at the time of high speed switching than at the time of partial switching, so that the DC voltage Vd is boosted too much if switching is made to be the same as the current amplitude of the partial switching.
In order to avoid this, it is preferable to switch by adjusting the on-time so that the peak of the current during the high-speed switching operation becomes smaller than the current during the partial switching operation at the moment of switching.

FIG. 25 is a diagram illustrating current waveforms when switching from partial switching to high-speed switching.
FIG. 25A schematically shows the AC power supply voltage Vs and the circuit current is during the partial switching control.
FIG. 25B schematically shows the AC power supply voltage Vs and the circuit current is when switching to high-speed switching control. The peak of the circuit current is at this time is smaller than the peak of the circuit current is shown in FIG. In this way, it is possible to suppress fluctuations in the DC voltage by adjusting and switching on time. This is because the power factor is good at the time of high-speed switching with respect to partial switching, so that the current becomes small. That is, if the switching is performed so that the current amplitude is the same as that of the partial switching, the DC voltage is excessively boosted. Thereby, it is possible to suppress the fluctuation of the DC voltage Vd.

Similarly, when switching from high-speed switching to partial switching, it is possible to prevent the DC voltage Vd from decreasing by adjusting the ON time so that the amplitude of the current increases in contrast to the previous case. .
Furthermore, the control can be stably switched by switching each control at the timing of the power supply voltage zero cross.

<Operation of air conditioner and DC power supply>
FIG. 26 is a front view of the indoor unit, the outdoor unit, and the remote controller of the air conditioner according to the present embodiment.
As shown in FIG. 26, the air conditioner A is a so-called room air conditioner, and includes an indoor unit 100, an outdoor unit 200, a remote controller Re, and a DC power supply (not shown) (see FIG. 1). The indoor unit 100 and the outdoor unit 200 are connected by a refrigerant pipe 300, and air-conditions the room in which the indoor unit 100 is installed by a known refrigerant cycle. The indoor unit 100 and the outdoor unit 200 transmit and receive information to and from each other via a communication cable (not shown). Further, the outdoor unit 200 is connected by wiring (not shown), and an AC voltage is supplied through the indoor unit 100. The DC power supply device is provided in the outdoor unit 200 and converts AC power supplied from the indoor unit 100 side into DC power.

  The remote controller Re is operated by the user and transmits an infrared signal to the remote controller transmission / reception unit Q of the indoor unit 100. The contents of the infrared signal are commands such as an operation request, a change in set temperature, a timer, an operation mode change, and a stop request. The air conditioner A performs air conditioning operations such as a cooling mode, a heating mode, and a dehumidifying mode based on these infrared signal commands. Moreover, the indoor unit 100 transmits data such as room temperature information, humidity information, and electricity bill information from the remote control transmission / reception unit Q to the remote control Re.

  The installation positions of the inrush current prevention circuit U1 and the inrush current prevention relay SW1 of the present embodiment are not limited to the indoor unit 100 and the outdoor unit 200, respectively, and may be installed on either the indoor unit or the outdoor unit. .

  An operation flow of the DC power supply device mounted on the air conditioner A will be described. The direct-current power supply device performs high-efficiency operation, reduces harmonic current by improving the power factor, and boosts the direct-current voltage Vd. As described above, the operation mode includes four operation modes of diode rectification operation, synchronous rectification operation, high-speed switching operation, and partial switching operation.

  For example, when an inverter or a motor of the air conditioner A is considered as the load H, the DC power supply device may be operated in the synchronous rectification mode if the load is small and operation with an emphasis on efficiency is necessary.

  If the load becomes large and it is necessary to increase the voltage and secure the power factor, it is preferable to cause the DC power supply device to perform a high-speed switching operation. In addition, as in the rated operation of the air conditioner A, if the load is not so large but it is necessary to ensure the pressure increase or the power factor, the partial switching operation may be performed. Note that either diode rectification or synchronous rectification may be combined during partial switching and high-speed switching.

FIG. 27 is a schematic diagram illustrating a state in which the operation mode of the DC power supply device 1 and the operation region of the air conditioner A are switched according to the size of the load.
In the case where the load is provided with threshold values # 1 and # 2 and the air conditioner A is considered as a device, the DC power supply device performs synchronous rectification in an intermediate region where the load is small, and partial switching (diode rectification or Synchronous rectification is combined), and high-speed switching (combining either diode rectification or synchronous rectification) is performed as necessary.

  In a low temperature heating operation region where the load is higher than that of the rated operation, the DC power supply device 1 performs high-speed switching, and performs partial switching (combining either diode rectification or synchronous rectification) as necessary.

  As described above, the DC power supply device can reduce the harmonic current while performing high-efficiency operation by switching to the optimum operation mode according to the operation region of the air conditioner A.

  When the load H is an inverter, a motor, or the like, the parameters that determine the magnitude of the load may include a current flowing through the inverter or the motor, an inverter modulation rate, and a motor rotation speed. Further, the magnitude of the load H may be determined by the circuit current is flowing through the DC power supply device. Moreover, you may judge the magnitude | size of load with a DC voltage.

  For example, if the magnitude of the load is equal to or less than threshold value # 1, the DC power supply device performs synchronous rectification, and if it exceeds threshold value # 1, performs partial switching (combining either diode rectification or synchronous rectification). Or, if the magnitude of the load exceeds the threshold value # 2, the DC power supply performs high-speed switching (combining either diode rectification or synchronous rectification), and partial switching (diode rectification or synchronization) if the threshold value # 2 or less. Combine any one of rectification).

  As described above, the DC power supply device can reduce the harmonic current while performing high-efficiency operation by switching to the optimum operation mode corresponding to the size of the load.

In the present embodiment, an example in which a super junction MOSFET is used as the MOSFET (Q1, Q2) has been described. By using a switching element using SiC (Silicon Carbide) -MOSFET or GaN (Gallium nitride) as the MOSFET (Q1, Q2), it is possible to realize further high-efficiency operation.

Thus, by providing the air conditioner A with the DC power supply device of the present embodiment, the energy efficiency (that is, APF) is high, and the reliability can be improved. Even if the DC power supply device of the present embodiment is mounted on a device other than an air conditioner, it is possible to improve the efficiency and reliability.
Further, the DC power supply device 1 of the present embodiment uses a bootstrap system shown in FIG. 28 as a drive circuit for the MOSFETs (Q1, Q2).

FIG. 28 shows a drive circuit for MOSFETs (Q1, Q2) by a bootstrap system.
A first terminal of the drive circuit K2 is connected to the gate of the MOSFET (Q2) via a resistor Rg21, and a resistor Rg22 is further connected between the gate and the source. The second terminal of the drive circuit K2 is connected to the source of the MOSFET (Q2) and the ground. A drive power supply E of 15 V is applied between the power supply terminal and the second terminal of the drive circuit K2. The drive circuit K2 applies a predetermined gate voltage between the first terminal and the second terminal in accordance with a signal from an input terminal (not shown) to drive the MOSFET (Q2).

  A first terminal of the drive circuit K1 is connected to the gate of the MOSFET (Q1) via a resistor Rg11, and a resistor Rg12 is further connected between the gate and the source. The second terminal of the drive circuit K1 is connected to the source of the MOSFET (Q1). A smoothing capacitor C2 is connected between the power supply terminal and the second terminal of the drive circuit K1, and 15V of the drive power supply E is further applied through the resistor R4 and the diode D5. The drive circuit K1 drives a MOSFET (Q1) by applying a predetermined gate voltage between the first terminal and the second terminal in accordance with a signal from an input terminal (not shown).

When the MOSFET (Q2) is on, a current flows through a path indicated by a broken line. The voltage at the node N2 becomes equal to the grant voltage, and 15 V is applied to the smoothing capacitor C2. As a result, power can be supplied to the drive circuit K1.
By adopting the bootstrap method, it is possible to control the MOSFETs (Q1, Q2) while suppressing an increase in the number of components and the circuit scale.

  As a driving circuit for the MOSFETs (Q1, Q2), a circuit using a transformer as shown in FIG. 29, which will be described later, is conceivable. However, the number of components is increased, the circuit scale is increased, and the cost is increased. It gets worse.

FIG. 29 shows a MOSFET drive circuit using a transformer Tr.
A first terminal of the drive circuit K2 is connected to the gate of the MOSFET (Q2) via a resistor Rg21, and a resistor Rg22 is further connected between the gate and the source. The second terminal of the drive circuit K2 is connected to the source of the MOSFET (Q2). The secondary winding of the transformer Tr is connected to the power supply terminal and the second terminal of the drive circuit K2 via the rectifier circuit 32 and the three-terminal regulator 22. As a result, a predetermined power supply voltage is applied to the drive circuit K2.

  A first terminal of the drive circuit K1 is connected to the gate of the MOSFET (Q1) via a resistor Rg11, and a resistor Rg12 is further connected between the gate and the source. The second terminal of the drive circuit K1 is connected to the source of the MOSFET (Q1). The secondary winding of the transformer Tr is connected to the power supply terminal and the second terminal of the drive circuit K1 via the rectifier circuit 32 and the three-terminal regulator 22. As a result, a predetermined power supply voltage is applied to the drive circuit K1.

FIG. 30 shows how the bootstrap circuit is charged.
When the AC power supply voltage Vs is applied in the direction of the thick arrow, the connection point N2 has a low potential. At this time, a charge current flows through the smoothing capacitor C2 in the direction of the broken line to charge the charge. The drive circuit K1 can drive the MOSFET (Q1) by the electric charge charged in the smoothing capacitor C2.

  The present invention is not limited to the embodiments described above, and includes various modifications. For example, the above-described embodiment has been described in detail for easy understanding of the present invention, and is not necessarily limited to the one having all the configurations described. A part of the configuration of one embodiment can be replaced with the configuration of another embodiment, and the configuration of another embodiment can be added to the configuration of one embodiment. Moreover, it is also possible to add, delete, and replace other configurations for a part of the configuration of each embodiment.

  For example, the number and configuration of the gate resistors in the gate drive circuit shown in FIG. 28 are not limited to the configuration shown in FIG. For example, a diode for extracting charges may be provided in the lines of the resistors Rg11 and Rg21 when the MOSFETs (Q1, Q2) are turned off to increase the off speed.

  Furthermore, although a high-speed trr type element is used as the MOSFET (Q1, Q2), a high-efficiency operation is possible by using an element having a trr of 300 ns or less.

The effect of synchronous rectification increases as the on-resistance of the MOSFETs (Q1, Q2) decreases. Specifically, high-efficiency operation is possible when the on-resistance is 0.1Ω or less.
The bridge rectifier circuit 10 is not limited to the configuration of the parasitic diodes of the MOSFETs (Q1, Q2) and the diodes D1, D2, but is configured by connecting the diodes in parallel to the MOSFETs (Q1, Q2) and combining the diodes D1, D2. May be.

  A part or all of the above-described configurations, functions, processing units, processing means, and the like may be realized by hardware such as an integrated circuit. Each of the above-described configurations, functions, and the like may be realized by software by a processor interpreting and executing a program that realizes each function. Information such as programs, tables, and files for realizing each function can be stored in a recording device such as a memory or a hard disk, or a recording medium such as a flash memory card or a DVD (Digital Versatile Disk).

  In each embodiment, the control lines and information lines indicate what is considered necessary for the explanation, and not all the control lines and information lines on the product are necessarily shown. Actually, it may be considered that almost all the components are connected to each other.

DESCRIPTION OF SYMBOLS 10 Bridge rectifier circuit 11 Current detection part R1 Shunt resistance 12 Gain control part 13 AC voltage detection part 14 Zero cross determination part 15 Load detection part 16 Boost ratio control part 17 DC voltage detection part 18 Converter control part 19 Power supply circuit Vs AC power supply C1 Smoothing Capacitor D1, D2 Diode D3, D4 Diode (parasitic diode)
ha, hb, hc, hd wiring L1 reactor Q1, Q2 MOSFET
U1 Inrush current prevention circuit (Protection circuit)

Claims (10)

A rectifier circuit connected to an AC power source and having first to fourth diodes;
The third diode is included as a parasitic diode, or is connected in parallel to the third diode, has a withstand voltage characteristic in a direction in which the third diode is turned off, and the first to second A first switching element having a saturation voltage lower than the forward voltage drop of the four diodes;
The fourth diode is included as a parasitic diode, or is connected in parallel to the fourth diode, has a withstand voltage characteristic in a direction in which the fourth diode is turned off, and the first to first A second switching element having a saturation voltage lower than the forward voltage drop of the four diodes;
A reactor provided between the AC power supply and the rectifier circuit;
A smoothing capacitor connected to the output side of the rectifier circuit and smoothing a voltage applied from the rectifier circuit;
Control means for controlling the first and second switching elements;
A protection circuit that protects the circuit element from an inrush current that flows when the voltage of the AC power supply is applied;
A DC power supply device comprising: The protection circuit includes a first power relay, an inrush current prevention circuit having an inrush current prevention resistor connected in series with the first power relay, and a second connected in parallel with the inrush current prevention circuit. Consists of power relay,
The DC power supply device according to claim 1. The control means turns off the first and second switching elements at the time of inrush current flow at the time of voltage application of the AC power supply.
The DC power supply device according to claim 2, wherein A power supply circuit for generating 5V voltage and 15V voltage;
A bootstrap circuit for driving the first and second switching elements;
Is further provided,
After the power supply circuit is activated, the protection circuit transitions from an inrush current protection operation to a normal operation.
The DC power supply device according to claim 3. The control means includes
Diode rectification control using the first to fourth diodes, synchronous rectification control for switching the first switching element and the second switching element in synchronization with the polarity of the voltage of the AC power supply, The partial switching control in which the control for partially shorting the reactor to the AC power supply is repeated a plurality of times during a half cycle, or the high-speed switching control for shorting the reactor at a predetermined frequency over the entire AC cycle is predetermined. Based on the threshold information
The DC power supply device according to any one of claims 1 to 4, wherein The control means is based on threshold information determined in advance.
Switching between the mode for performing the synchronous rectification control and the mode for simultaneously performing the synchronous rectification control and the partial switching control;
Switching between a mode for performing the synchronous rectification control and a mode for simultaneously performing the synchronous rectification control and the high-speed switching control;
Switching between a mode for performing the synchronous rectification control, a mode for simultaneously performing the synchronous rectification control and the partial switching control, and a mode for simultaneously performing the synchronous rectification control and the high-speed switching control;
Switching between the mode for performing the synchronous rectification control and the mode for simultaneously performing the diode rectification control and the partial switching control,
Switching between a mode for performing the synchronous rectification control and a mode for simultaneously performing the diode rectification control and the high-speed switching control;
Switching between a mode for performing the synchronous rectification control, a mode for simultaneously performing the diode rectification control and the partial switching control, and a mode for simultaneously performing the diode rectification control and the high-speed switching control;
Switching between a mode for performing the synchronous rectification control, a mode for simultaneously performing the diode rectification control and the partial switching control, and a mode for simultaneously performing the synchronous rectification control and the high-speed switching control;
Switching between a mode for performing the synchronous rectification control, a mode for simultaneously performing the synchronous rectification control and the partial switching control, and a mode for simultaneously performing the diode rectification control and the high-speed switching control is performed. To
The DC power supply device according to claim 5, wherein: The control means switches control at a zero cross of an AC voltage applied by the AC power source.
The DC power supply device according to claim 5 or 6, wherein In the case of switching from the partial switching control to the high-speed switching control, or in the case of switching from the high-speed switching control to the partial switching control, the control means switches so that the output DC voltage maintains a predetermined value.
The DC power supply device according to claim 5 or 6, wherein The first and second switching elements are switching elements using any one of a super junction MOSFET (Metal-Oxide-Semiconductor Field-Effect Transistor), SiC-MOSFET, and GaN (Gallium nitride).
The direct current power supply device according to any one of claims 1 to 8, wherein the direct current power supply device is provided. A DC power supply device according to any one of claims 1 to 9, comprising:
An air conditioner characterized by that.