WO2021038867A1

WO2021038867A1 - Direct-current power supply device, motor drive device, air blower, compressor, and air conditioner

Info

Publication number: WO2021038867A1
Application number: PCT/JP2019/034261
Authority: WO
Inventors: 和徳畠山; 啓介植村
Original assignee: 三菱電機株式会社
Priority date: 2019-08-30
Filing date: 2019-08-30
Publication date: 2021-03-04
Also published as: JPWO2021038867A1; JP7162746B2

Abstract

This direct-current power supply device (50) is provided with: a reactor (2); a converter (3) which is connected to an alternating-current power supply (1) via the reactor (2); a short circuit (330) that has a short-circuiting switching element (331), that is connected between input terminals of the converter (3), and that short-circuits power supply voltage via the reactor (2); a smoothing capacitor (4) which is connected between output terminals of the converter (3); and a control unit (10) which controls the operation of the converter (3) on the basis of bus voltage and the power supply voltage. Further, the control unit (10) has a plurality of operational modes in which the converter (3) is caused to operate in different modes through combinations of conduction of switching elements (Q1-Q4) and conduction of the short-circuiting switching element (331).

Description

DC power supply, motor drive, blower, compressor and air conditioner

The present invention includes a DC power supply device that converts an AC voltage output from an AC power supply into a DC voltage and applies it to a load, a motor drive device that drives a motor that is a load, a blower and a compressor equipped with a motor drive device, and , With respect to an air conditioner equipped with a blower or compressor.

The following Patent Document 1 describes a connection point between a first diode and a second diode, and a connection between a first metal oxide semiconductor field effect transistor (Metal Oxide Semiconductor Field Effect Transistor: MOSFET) and a second MOSFET. A DC power supply device is disclosed in which an AC power supply is connected via a reactor and the AC voltage of the AC power supply is converted into a DC voltage by switching between the first MOSFET and the first MOSFET. The first diode and the first MOSFET are elements connected to the positive electrode side of the smoothing capacitor, and the second diode and the second MOSFET are elements connected to the negative electrode side of the smoothing capacitor. The first and second diodes and the first and second MOSFETs are bridge-connected to form a rectifier.

In the DC power supply device described in Patent Document 1, the first MOSFET is turned on at the timing when the current flows through the parasitic diode of the first MOSFET, and the second MOSFET is operated at the timing when the current flows through the parasitic diode of the second MOSFET. To operate. This technique is called synchronous rectification. The DC power supply is controlled with high efficiency by synchronous rectification.

Further, Patent Document 1 discloses a configuration in which a short-circuit circuit is provided on the input side of the rectifier, which is connected in parallel to the rectifier and for short-circuiting the output of the AC power supply via the reactor. A short-circuit switching element is connected to the short-circuit circuit, and when the short-circuit switching element is turned on, the output of the AC power supply is short-circuited by the short-circuit circuit.

Japanese Unexamined Patent Publication No. 2011-151984

It is described that in the DC power supply device described in Patent Document 1, synchronous rectification operation that realizes improvement of power factor and reduction of harmonic current is possible by operating a short circuit. However, in Patent Document 1, the short-circuit circuit is operated only once every half cycle, and it cannot be said that the improvement of the power factor and the suppression of the power supply harmonics are sufficient.

The present invention has been made in view of the above, and an object of the present invention is to obtain a DC power supply device capable of achieving both efficiency improvement by synchronous rectification, power factor improvement, and power supply harmonic suppression.

In order to solve the above-mentioned problems and achieve the object, the DC power supply device according to the present invention includes a reactor and four unidirectional elements connected by a bridge, and is connected to an AC power supply via the reactor to be connected to an AC power supply. It includes a converter that converts a power supply voltage, which is an AC voltage output from, into a DC voltage and applies it to a load. Further, the DC power supply device includes a short-circuit circuit having a short-circuit switching element and connected between the input terminals of the converter to perform a power short-circuit operation of short-circuiting the power supply voltage via a reactor by turning on the short-circuit switching element. Further, the DC power supply unit includes a smoothing capacitor connected between the output terminals of the converter, a first physical quantity detector for detecting a first physical quantity indicating an operating state on the output side of the converter, and an operation on the input side of the converter. It includes a second physical quantity detecting unit that detects a second physical quantity that represents a state, and a control unit that inputs the first and second physical quantities and controls the operation of the converter. Two of the four unidirectional elements in the converter are connected in series to form the first leg, and the remaining two unidirectional elements are connected in series to form the second leg. At least two unidirectional elements in the first and second legs connected to the positive side of the smoothing capacitor, or two in the first and second legs connected to the negative side of the smoothing capacitor. A switching element is connected in parallel to each of the unidirectional element, the two unidirectional elements in the first leg, or the two unidirectional elements in the second leg. The control unit further has a plurality of operation modes in which the conduction of the switching element and the continuity of the short-circuit switching element are combined to operate the converter in different operation modes.

According to the DC power supply device according to the present invention, it is possible to achieve both efficiency improvement by synchronous rectification, power factor improvement and power supply harmonic suppression.

The figure which shows the structural example of the motor drive device which includes the DC power supply device which concerns on Embodiment 1. Schematic cross-sectional view showing a schematic structure of a MOSFET used in the converter of the first embodiment. The first figure which shows the path of the electric current flowing through the converter in Embodiment 1. The second figure which shows the path of the current flowing through the converter in Embodiment 1. FIG. 3 shows a path of a current flowing through the converter according to the first embodiment. FIG. 4 shows a path of a current flowing through a converter according to the first embodiment. The figure explaining the feature of the operation mode in Embodiment 1. The figure which shows the 1st example of the operation waveform at the time of operating in the operation mode shown in FIG. The figure which shows the structural example of the gate drive circuit in Embodiment 1. The figure which shows the 2nd example of the operation waveform at the time of operating in the operation mode shown in FIG. The figure which shows the 3rd example of the operation waveform at the time of operating in the operation mode shown in FIG. The figure which shows the loss characteristic of the MOSFET used in the DC power supply device which concerns on Embodiment 1. The figure which shows the timing which the control part turns on a switching element in the DC power supply device which concerns on Embodiment 1. Flow chart used to explain the operation of the main part in the first embodiment A block diagram showing an example of a hardware configuration that embodies the function of the control unit according to the first embodiment. A block diagram showing another example of a hardware configuration that embodies the function of the control unit according to the first embodiment. The figure which shows the structure of the air conditioner which concerns on Embodiment 2.

The DC power supply device, motor drive device, blower, compressor, and air conditioner according to the embodiment of the present invention will be described below with reference to the accompanying drawings. The present invention is not limited to the embodiments shown below. Further, in the following, the electrical connection will be described simply as "connection".

Embodiment 1.
FIG. 1 is a diagram showing a configuration example of a motor drive device 100 including a DC power supply device 50 according to the first embodiment. The DC power supply device 50 according to the first embodiment is a power supply device that converts a power supply voltage, which is an AC voltage output from a single-phase AC power supply 1, into a DC voltage and applies it to a load 12. Further, the motor drive device 100 according to the first embodiment is a drive device that converts the DC power output from the DC power supply device 50 into AC power and supplies the converted AC power to the motor 500 to drive the motor 500. is there.

As shown in FIG. 1, the motor drive device 100 according to the first embodiment includes a DC power supply device 50, a control unit 10, and a load 12 as main components.

The DC power supply device 50 includes a reactor 2, a converter 3, a gate drive circuit 15 which is a first drive circuit, a smoothing capacitor 4, a voltage detection unit 5, a current detection unit 6, a voltage detection unit 7, and the like. A power supply circuit 14 which is a control power supply and a short-circuit circuit 330 are provided. One end of the reactor 2 is connected to the AC power supply 1, and the other end of the reactor 2 is connected to the converter 3. The reactor 2 temporarily stores the electric power supplied from the AC power source 1. The converter 3 converts the AC voltage output from the AC power supply 1 into a DC voltage and outputs the AC voltage to the

DC bus

16a and 16b. The

DC bus lines

16a and 16b are electrical wirings that connect the converter 3 and the load 12. The voltage between the DC bus 16a and the DC bus 16b is called the "bus voltage".

The short circuit 330 is arranged between the reactor 2 and the converter 3. Further, the short circuit circuit 330 is connected between the input terminals of the converter 3. The short-circuit circuit 330 includes a short-circuit switching element 331 and a diode bridge 332 connected in parallel to the short-circuit switching element 331. The short-circuit switching element 331 includes a transistor 331a and a diode 331b connected in parallel to the transistor 331a. There is no operational problem even if the diode 331b is not mounted. An example of the transistor 331a is an insulated gate bipolar transistor (IGBT) (indicated Gate Bipolar Transistor: IGBT) (not shown). MOSFETs may be used instead of the IGBTs. When the transistor 331a is a MOSFET, the parasitic diode of the MOSFET may be used as the diode 331b.

The short-circuit circuit 330 performs a power supply short-circuit operation that short-circuits the AC voltage applied via the reactor 2 by turning on the short-circuit switching element 331.

The load 12 includes a gate drive circuit 17, which is a second drive circuit, an inverter 18, a current detection unit 9, and a motor 500. Among the components of the load 12, the gate drive circuit 17, the inverter 18, and the current detection unit 9, excluding the motor 500, are the components of the motor drive device 100. The inverter 18 converts the DC voltage output from the DC power supply device 50 into an AC voltage applied to the motor 500 and outputs the AC voltage. Examples of equipment on which the motor 500 is mounted are blowers, compressors or air conditioners.

Note that FIG. 1 shows an example in which the device connected to the inverter 18 is the motor 500, but the present invention is not limited to this. The device connected to the inverter 18 may be any device to which AC power is input, and may be a device other than the motor 500.

The converter 3 includes a first leg 31 and a second leg 32. The first leg 31 and the second leg 32 are connected in parallel. In the first leg 31, the first upper arm element 311 and the first lower arm element 312 are connected in series. In the second leg 32, the second upper arm element 321 and the second lower arm element 322 are connected in series. The other end of the reactor 2 is connected to a connection point 3a between the first upper arm element 311 and the first lower arm element 312 in the first leg 31. The connection point 3b between the second upper arm element 321 and the second lower arm element 322 is connected to the other end of the AC power supply 1. In the converter 3, the connection points 3a and 3b form an AC terminal.

In FIG. 1, the reactor 2 is connected between one end of the AC power supply 1 and the connection point 3a, but is connected between another end of the AC power supply 1 and the connection point 3b. May be good.

In the converter 3, the side where the connection points 3a and 3b are located is called the "AC side", the AC voltage output from the AC power supply 1 is called the "power supply voltage", and the cycle of the power supply voltage is called the "power supply cycle". is there.

The first upper arm element 311 includes a switching element Q1 and a diode D1 connected in parallel to the switching element Q1. The first lower arm element 312 includes a switching element Q2 and a diode D2 connected in parallel to the switching element Q2. The second upper arm element 321 includes a switching element Q3 and a diode D3 connected in parallel to the switching element Q3. The second lower arm element 322 includes a switching element Q4 and a diode D4 connected in parallel to the switching element Q4.

The diodes D1 and D4 are unidirectional so that forward current flows when the polarity of the power supply voltage is positive, that is, the side connected to the reactor 2 has a higher potential than the side not connected to the reactor 2. It is an element. The diodes D2 and D3 are unidirectional so that a forward current flows when the polarity of the power supply voltage is negative, that is, the side not connected to the reactor 2 has a higher potential than the side connected to the reactor 2. It is an element.

Note that FIG. 1 discloses a configuration in which switching elements Q1, Q2, Q3, and Q4 are connected in parallel to each of the diodes D1, D2, D3, and D4, but the present invention is not limited to this. Switching elements may be connected to each of the two diodes connected to the positive side of the smoothing capacitor 4, that is, the diode D1 in the first leg 31 and the diode D3 in the second leg 32. Alternatively, switching elements may be connected to each of the two diodes connected to the negative side of the smoothing capacitor 4, that is, the diode D2 in the first leg 31 and the diode D4 in the second leg 32. Alternatively, a switching element may be connected to each of the two diodes in the first leg 31, that is, the diodes D1 and D2. Alternatively, a switching element may be connected to each of the two diodes in the second leg 32, that is, the diodes D3 and D4.

Further, in FIG. 1, MOSFETs are illustrated for each of the switching elements Q1, Q2, Q3, and Q4, but the limitation is not limited to MOSFETs. The MOSFET is a switching element capable of passing a current in both directions between the drain and the source. Any switching element may be used as long as it is a switching element capable of bidirectionally flowing a current between the first terminal corresponding to the drain and the second terminal corresponding to the source, that is, a bidirectional element.

Further, "parallel" here means that the first terminal corresponding to the drain of the MOSFET and the cathode of the diode are connected, and the second terminal corresponding to the source of the MOSFET and the anode of the diode are connected. To do. As the diode, a parasitic diode that the MOSFET itself has inside may be used. Parasitic diodes are also called body diodes.

Further, the switching elements Q1, Q2, Q3 and Q4 are not limited to MOSFETs formed of silicon-based materials, and are wide-band such as _{silicon carbide (SiC), gallium nitride (GaN), gallium oxide (Ga 2} O _{3) or diamond.} It may be a MOSFET formed of a bandgap (Wide Band Gap) semiconductor.

In general, WBG semiconductors have higher withstand voltage and heat resistance than silicon semiconductors. Therefore, by using a WBG semiconductor for at least one of the switching elements Q1, Q2, Q3, and Q4, the withstand voltage resistance and the allowable current density of the switching element are increased, and the semiconductor module incorporating the switching element is miniaturized. it can.

Further, as the switching elements Q1, Q2, Q3, and Q4, a MOSFET having a Super Junction (SJ) structure may be used instead of the WBG semiconductor. By using the SJ-MOSFET, it is possible to suppress the demerit of the WBG semiconductor that the capacitance is high and recovery is likely to occur while taking advantage of the low on-resistance which is the merit of the SJ-MOSFET.

Return to the explanation in Fig. 1. The positive side of the smoothing capacitor 4 is connected to the DC bus 16a on the high potential side. The DC bus 16a is drawn from the connection point 3c between the first upper arm element 311 in the first leg 31 and the second upper arm element 321 in the second leg 32. The negative side of the smoothing capacitor 4 is connected to the DC bus 16b on the low potential side. The DC bus 16b is drawn from the connection point 3d between the first lower arm element 312 in the first leg 31 and the second lower arm element 322 in the second leg 32. In the converter 3, the connection points 3c and 3d form a DC terminal. Further, in the converter 3, the side where the connection points 3c and 3d are located may be referred to as the "DC side".

The output voltage of the converter 3 is applied to both ends of the smoothing capacitor 4. The smoothing capacitor 4 is connected to the

DC bus lines

16a and 16b. The smoothing capacitor 4 smoothes the output voltage of the converter 3. The voltage smoothed by the smoothing capacitor 4 is applied to the inverter 18.

The voltage detection unit 5 detects the power supply voltage and outputs the detected value Vs of the power supply voltage to the control unit 10. The power supply voltage is an absolute value of the instantaneous voltage of the AC power supply 1. The effective value of the instantaneous voltage may be used as the power supply voltage.

The current detection unit 6 detects the power supply current, which is the AC current flowing between the AC power supply 1 and the converter 3, and outputs the detected value Is of the power supply current to the control unit 10. An example of a current detector used in the current detection unit 6 is an AC current transformer (Alternating Current Current Transformer: ACCT). The voltage detection unit 7 detects the bus voltage and outputs the detected value Vdc of the bus voltage to the control unit 10.

The bus voltage is a physical quantity that represents the operating state of the DC side, that is, the output side of the converter 3. The power supply voltage is a physical quantity representing the operating state of the AC side, that is, the input side of the converter 3. In order to distinguish between these two physical quantities, the bus voltage may be referred to as a "first physical quantity" and the power supply voltage may be referred to as a "second physical quantity". Further, the voltage detection unit 7 that detects the bus voltage may be called a "first physical quantity detection unit", and the voltage detection unit 5 that detects the power supply voltage may be called a "second physical quantity detection unit".

The power supply circuit 14 is connected to both ends of the smoothing capacitor 4. The power supply circuit 14 uses the voltage of the smoothing capacitor 4 to generate low-voltage DC voltages such as 5V, 12V, 15V, and 24V. The low-voltage DC voltage is generated by utilizing the electric charge accumulated in the smoothing capacitor 4. A low-voltage DC voltage is applied to each part of the supply destination as an operating voltage. The power supply circuit 14 outputs, for example, a DC voltage of 5 V to the control unit 10, the current detection unit 6, and the like. In the control unit 10, a DC voltage of 5 V is applied to a processor (not shown) in FIG.

The inverter 18 includes a leg 18A in which the upper arm element 18UP and the lower arm element 18UN are connected in series, a leg 18B in which the upper arm element 18VP and the lower arm element 18VN are connected in series, and the upper arm element 18WP and the lower. It includes a leg 18C in which an arm element 18WN is connected in series. The

legs

18A, 18B and 18C are connected in parallel to each other.

FIG. 1 illustrates a case where the upper arm elements 18UP, 18VP, 18WP and the lower arm elements 18UN, 18VN, 18WN are IGBTs, but the present invention is not limited to this. Instead of the IGBT, a MOSFET or an integrated gate commutated thyristor (IGCT) may be used.

The upper arm element 18UP includes a transistor 18a and a diode 18b connected in parallel to the transistor 18a. The other upper arm elements 18VP and 18WP and the lower arm elements 18UN, 18VN and 18WN have the same configuration. The term "parallel" as used herein means that the anode side of the diode is connected to the first terminal corresponding to the emitter of the IGBT, and the cathode side of the diode is connected to the second terminal corresponding to the collector of the IGBT.

Note that FIG. 1 has a configuration including three legs in which the upper arm element and the lower arm element are connected in series, but the configuration is not limited to this. The number of legs may be four or more. Further, the circuit configuration shown in FIG. 1 is adapted to the motor 500, which is a three-phase motor. When the motor 500 is a single-phase motor, the inverter 18 is also configured to correspond to the single-phase motor. Specifically, the configuration is provided with two legs in which the upper arm element and the lower arm element are connected in series. When the motor 500 is either a single-phase motor or a three-phase motor, one leg may be composed of a plurality of pairs of upper and lower arm elements.

When the transistor 18a of the upper arm elements 18UP, 18VP, 18WP and the lower arm elements 18UN, 18VN, 18WN is a MOSFET, the upper arm elements 18UP, 18VP, 18WP and the lower arm elements 18UN, 18VN, 18WN are silicon carbide, gallium nitride. It may be formed of a system material or a WBG semiconductor such as diamond. If a MOSFET formed of a WBG semiconductor is used, the effects of withstand voltage and heat resistance can be enjoyed.

The connection point 26a between the upper arm element 18UP and the lower arm element 18UN is connected to the first phase (for example, U phase) of the motor 500, and the connection point 26b between the upper arm element 18VP and the lower arm element 18VN is the first phase of the motor 500. It is connected to the second phase (for example, V phase), and the connection point 26c between the upper arm element 18WP and the lower arm element 18WN is connected to the third phase (for example, W phase) of the motor 500. In the inverter 18, the

connection points

26a, 26b, and 26c form an AC terminal.

The current detection unit 9 detects the motor current flowing between the inverter 18 and the motor 500, and outputs the detected value Iuvw of the motor current to the control unit 10.

The control unit 10 controls each switching element in the converter 3 based on the detection value Vs of the voltage detection unit 5, the detection value Is of the current detection unit 6, and the detection value Vdc of the voltage detection unit 7. S311 to S322 and a short-circuit control signal S331 for controlling the short-circuit switching element 331 of the short-circuit circuit 330 are generated.

The control signal S311 is a control signal for controlling the switching element Q1, and the control signal S322 is a control signal for controlling the switching element Q4. The switching elements Q2 and Q3 are also controlled by the control signal from the control unit 10. The control signals S311 to S322 generated by the control unit 10 are input to the gate drive circuit 15.

In the following, the operation of each arm element according to the control signals S311 to S322 is appropriately referred to as "switching operation". Further, the operation of the short-circuit switching element 331 according to the short-circuit control signal S331 is appropriately referred to as "short-circuit switching operation".

Further, the control unit 10 is provided with each switching element in the inverter 18 so that the motor 500 rotates at a desired rotation speed based on the detection value Vdc of the voltage detection unit 7 and the detection value Iuvw of the current detection unit 9. Control signals S1 to S6 for controlling the above are generated. The inverter 18 has a three-phase circuit configuration, and has six switching elements corresponding to the three-phase circuit configuration. Further, six control signals S1 to S6 are generated corresponding to the six switching elements. The control signals S1 to S6 generated by the control unit 10 are input to the gate drive circuit 17.

The gate drive circuit 15 generates drive pulses G311 to G322 for driving each switching element in the converter 3 based on the control signals S311 to S322. The drive pulse G311 is a drive pulse for driving the switching element Q1, and the drive pulse G322 is a drive pulse for driving the switching element Q4. The switching elements Q2 and Q3 are also driven by the drive pulse from the gate drive circuit 15.

The gate drive circuit 17 generates drive pulses G1 to G6 for driving each switching element in the inverter 18 based on the control signals S1 to S6.

Note that, in FIG. 1, the control unit 10 is provided inside the motor drive device 100 as a common control unit for controlling the short-circuit circuit 330, the DC power supply device 50, and the load 12, but the configuration is not limited to this. Individual control units that control each of the DC power supply device 50 and the load 12 may be configured, and each control unit may be provided inside each of the DC power supply device 50 and the load 12. Further, the control unit that controls the short-circuit circuit 330 is generally provided in the control unit that controls the DC power supply device 50.

Next, the basic operation of the motor drive device 100 according to the first embodiment will be described. First, in the first leg 31, the first upper arm element 311 and the first lower arm element 312 operate so as to be complementary or not turned on at the same time. That is, when one of the first upper arm element 311 and the first lower arm element 312 is on, the other is off. As described above, the first upper arm element 311 and the first lower arm element 312 are controlled by the control signals S311 and S312 generated by the control unit 10. An example of the control signals S311 and S312 is a pulse width modulation (PWM) signal.

In order to prevent a short circuit of the smoothing capacitor 4 via the AC power supply 1 and the reactor 2, when the absolute value of the detected value Is of the power supply current output from the AC power supply 1 is equal to or less than the current threshold value, the first upper arm element 311 And the first lower arm element 312 are both turned off. Hereinafter, the short circuit of the smoothing capacitor 4 is referred to as a “capacitor short circuit”. Capacitor short circuit is a state in which the energy stored in the smoothing capacitor 4 is released and the current is regenerated in the AC power supply 1.

As described above, the second upper arm element 321 and the second lower arm element 322 constituting the second leg 32 are controlled by the control signals S321 and S322 generated by the control unit 10. The second upper arm element 321 and the second lower arm element 322 are basically turned on or off depending on the polarity of the power supply voltage, which is the polarity of the power supply voltage. Specifically, when the power supply voltage polarity is positive, the second lower arm element 322 is on and the second upper arm element 321 is off. When the power supply voltage polarity is negative, the second upper arm element 321 is on and the second lower arm element 322 is off.

Next, the relationship between the state of each arm element of the converter 3 in the first embodiment and the path of the current flowing through the motor drive device 100 according to the first embodiment will be described. In the following description, it is assumed that each arm element of the converter 3 is a MOSFET, and the diode of each arm element is a parasitic diode that the MOSFET itself has inside.

First, the structure of the MOSFET will be described with reference to FIG. FIG. 2 is a schematic cross-sectional view showing a schematic structure of a MOSFET used in the converter 3 of the first embodiment. FIG. 2 illustrates an n-type MOSFET.

In the case of an n-type MOSFET, a p-type semiconductor substrate 600 is used, as shown in FIG. A source electrode S, a drain electrode D, and a gate electrode G are formed on the semiconductor substrate 600. High-concentration impurities are ion-implanted into the portions in contact with the source electrode S and the drain electrode D to form an n-type region 601. Further, in the semiconductor substrate 600, an oxide insulating film 602 is formed between the portion where the n-type region 601 is not formed and the gate electrode G. That is, an oxide insulating film 602 is interposed between the gate electrode G and the p-type region 603 of the semiconductor substrate 600.

When a positive voltage is applied to the gate electrode G, electrons are attracted to the boundary surface between the p-shaped region 603 and the oxide insulating film 602 in the semiconductor substrate 600, and the boundary surface is negatively charged. Where the electrons are gathered, the density of the electrons becomes higher than the hole density and the electron is formed into an n-type. This n-shaped portion serves as a current path and is called a channel 604. Channel 604 is an n-type channel in the example of FIG. By controlling the MOSFET on, the flowing current flows through channel 604 more than the parasitic diode formed in the p-type region 603.

FIG. 3 is a first diagram showing a path of a current flowing through the converter 3 in the first embodiment. FIG. 3 shows a state in which the power supply voltage polarity is positive and the absolute value of the detected value Is of the power supply current is larger than the current threshold value. In this state, the first upper arm element 311 and the second lower arm element 322 are on, and the first lower arm element 312, the second upper arm element 321 and the short-circuit switching element 331 are off. At this time, the current flows in the order of the AC power supply 1, the reactor 2, the switching element Q1, the smoothing capacitor 4, the switching element Q4, and the AC power supply 1. As described above, in the first embodiment, the switching elements Q1 and Q4 corresponding to the diodes D1 and D4 are turned on at the timing when the current flows through the diodes D1 and D4 instead of passing the current through the diodes D1 and D4. , Has an operation mode in which a current flows through each channel. This operation is called "synchronous rectification operation" or "synchronous rectification". In FIG. 3, the MOSFETs that are turned on are indicated by circles. The same applies to the following figures. The details of the operation mode will be described later.

FIG. 4 is a second diagram showing the path of the current flowing through the converter 3 in the first embodiment. FIG. 4 shows a state in which the power supply voltage polarity is negative and the absolute value of the detected value Is of the power supply current is larger than the current threshold value. In this state, the first lower arm element 312 and the second upper arm element 321 are on, and the first upper arm element 311 and the second lower arm element 322 and the short-circuit switching element 331 are off. At this time, the current flows in the order of the AC power supply 1, the switching element Q3, the smoothing capacitor 4, the switching element Q2, the reactor 2, and the AC power supply 1. As described above, in the first embodiment, the synchronous rectification operation in which the current is passed through each channel of the switching elements Q3 and Q2 may be performed instead of passing the current through the diodes D3 and D2.

FIG. 5 is a third diagram showing the path of the current flowing through the converter 3 in the first embodiment. FIG. 5 shows a state in which the power supply voltage polarity is positive and the absolute value of the detected value Is of the power supply current is larger than the current threshold value. In this state, the short-circuit switching element 331 is on, and the first upper arm element 311, the first lower arm element 312, the second upper arm element 321 and the second lower arm element 322 are off. At this time, the current flows in the order of the AC power supply 1, the reactor 2, the diode bridge 332, the short-circuit switching element 331, the diode bridge 332, and the AC power supply 1. As a result, a power supply short-circuit path that does not pass through the smoothing capacitor 4 is formed. As in this example, the first embodiment provides a mode in which a power short-circuit path is formed by passing a current through the short-circuit switching element 331 and the diode bridge 332 without passing a current through each arm element.

FIG. 6 is a fourth diagram showing the path of the current flowing through the converter 3 in the first embodiment. FIG. 6 shows a state in which the power supply voltage polarity is negative and the absolute value of the detected value Is of the power supply current is larger than the current threshold value. In this state, the short-circuit switching element 331 is on, and the first upper arm element 311, the first lower arm element 312, the second upper arm element 321 and the second lower arm element 322 are off. At this time, the current flows in the order of the AC power supply 1, the diode bridge 332, the short-circuit switching element 331, the diode bridge 332, the reactor 2, and the AC power supply 1. As a result, a power supply short-circuit path that does not pass through the smoothing capacitor 4 is formed. As in this example, the first embodiment provides a mode in which a power short-circuit path is formed by passing a current through the short-circuit switching element 331 and the diode bridge 332 without passing a current through each arm element.

The control unit 10 can control the values of the power supply current and the bus voltage by controlling the switching of the current path described above. When the power supply voltage polarity is positive, the motor drive device 100 continuously switches between the operation shown in FIG. 3 and the operation shown in FIG. Further, when the power supply voltage polarity is negative, the motor drive device 100 continuously switches between the operation shown in FIG. 4 and the operation shown in FIG. As a result, it is possible to realize bus voltage control for increasing the bus voltage or suppressing the increase of the bus voltage, current control for improving the power factor and power supply harmonics, and synchronous rectification for improving the operating efficiency. it can.

Next, the operation mode used in the DC power supply device 50 of the first embodiment will be described with reference to FIGS. 7 and 8. FIG. 7 is a diagram illustrating the characteristics of the operation mode according to the first embodiment. FIG. 8 is a diagram showing a first example of an operation waveform when operated in the operation mode shown in FIG. 7.

FIG. 7 shows four operation modes: (a) rectification mode, (b) synchronous rectification mode, (c) low-speed switching mode, and (d) high-speed switching mode. Each operation mode is classified according to the combination of whether or not two controls, synchronous rectification and short-circuit switching operation, are performed. Synchronous rectification is as described above and is performed to improve operating efficiency. The short-circuit switching operation is performed to control the bus voltage, improve the force factor of the current flowing in and out of the converter 3, and suppress harmonics. The low-speed switching mode may be referred to as a "first switching mode", and the high-speed switching mode may be referred to as a "second switching mode". Further, the operation in the rectification mode may be called "diode rectification operation", and the operation in the synchronous rectification mode may be called "synchronous rectification operation". Further, the operation in the first switching mode may be referred to as "first switching operation", and the operation in the second switching mode may be referred to as "second switching operation".

The DC power supply device 50 according to the first embodiment has a rectification mode, and further has at least one operation mode of a synchronous rectification mode, a low-speed switching mode, and a high-speed switching mode. In addition, in applications or products that do not require boosting operation, it may not be necessary to have a low-speed switching mode and a high-speed switching mode.

FIG. 8A shows an operating waveform when operated in the rectified mode. Specifically, from the upper side, the waveforms of the power supply voltage, the power supply current, and the control signals S311 to S322 that control each of the switching elements Q1 to Q4 are shown. The same applies to other operation modes. In the rectification mode, since it is not necessary to control the switching elements Q1 to Q4 and the short-circuit switching element 331, there is an advantage that the consumption of the drive power supply for operating the gate drive circuit 15 and the drive power supply for operating the short-circuit switching element 331 can be suppressed. is there. Further, since it is not necessary to control the switching elements Q1 to Q4 and the short-circuit switching element 331, there is an advantage that the control is easy.

FIG. 8B shows an operation waveform when operated in the synchronous rectification mode. The synchronous rectification mode is an operation mode in which the corresponding switching element is turned on at the timing of flowing through the parasitic diode and is passed through to the channel side of the switching element. In the example of FIG. 8B, the switching elements Q1 and Q4 or the switching elements Q2 and Q3 are turned on at the timing of flowing through the parasitic diode. When the synchronous rectification mode is used, it is possible to improve the efficiency especially when the flowing current is small. In the synchronous rectification mode, the passing element is simply replaced with a switching element from the parasitic diode. Therefore, current control and bus voltage control are not performed.

FIG. 8C shows the operation waveform when operated in the low-speed switching mode. The low-speed switching mode is an operation mode in which the power supply voltage is short-circuited via the reactor 2 at least once in a half cycle of the power supply cycle, in other words, an operation mode in which the power supply short circuit is locally performed within the half cycle of the power supply cycle. In the example of FIG. 8C, the short-circuit switching element 331 performs two short-circuit operations every half cycle of the power supply voltage. By performing the short-circuit operation, energy is stored in the reactor 2. When the short-circuit operation is released after the energy is stored, the energy stored in the reactor 2 is transferred to the smoothing capacitor 4 and stored. As a result, the voltage of the smoothing capacitor 4, that is, the bus voltage can be boosted.

The boost amount of the bus voltage is adjusted by the bus voltage control. A proportional integration controller or the like is used for bus voltage control. In the bus voltage control, the operation of the converter 3 is controlled so that the detected value Vdc of the bus voltage approaches the target voltage. Further, in the bus voltage control, the short-circuit time when the power supply voltage is short-circuited via the reactor 2 is controlled. Further, in the bus voltage control, by changing the response time of the proportional integration controller, it is possible to suppress an excessive increase in the bus voltage that may occur due to the occurrence of load fluctuation.

In the low-speed switching mode, a short-circuit current can be passed by short-circuit operation. As a result, the power factor can be improved and the harmonic current can be suppressed by expanding the flow width of the power supply current. Regarding the improvement of the current waveform, the timing for performing the short-circuit operation may be determined in advance with reference to the zero crossing point of the power supply voltage, and may be referred to according to the load. Alternatively, the power supply current may be detected and the short circuit time may be controlled so that the detected current waveform approaches a sine wave. In the low-speed switching mode, since the short-circuit operation time is short, it is possible to suppress the generation of harmonic noise.

FIG. 8D shows an operation waveform when operated in the high-speed switching mode. The high-speed switching mode is an operation mode in which the power supply short-circuit operation described above is performed over the entire range of one cycle of the power supply voltage. The significance of the power short-circuit operation is the same as that of the low-speed switching mode. That is, energy is stored in the reactor 2 by performing the power supply short-circuit operation, and the energy stored in the reactor 2 is transferred to the smoothing capacitor 4 by releasing the short-circuit operation after the energy is stored. This makes it possible to boost the bus voltage. The control of the boost amount of the bus voltage can also be realized by the same control as in the low speed switching mode.

As described above, in the high-speed switching mode, the short-circuit operation is performed over the entire range of one cycle of the power supply voltage, so that the current flow width is wider than in the low-speed switching mode. As a result, it is possible to further improve the power factor and suppress the harmonic current as compared with the low-speed switching mode. Further, in the high-speed switching mode, the power factor can be controlled to a value close to 1. As a result, the load can be driven to the limit of the breaker capacity, especially on the high load side, and the power of the device can be increased.

Next, the effect when the gate drive circuit 15 shown in FIG. 1 is configured to include a bootstrap circuit will be described. Here, first, the bootstrap circuit will be described with reference to FIG. FIG. 9 is a diagram showing a configuration example of the gate drive circuit 15 according to the first embodiment.

In FIG. 9, the gate drive circuit 15 includes

drive circuits

51 and 52 and a bootstrap circuit 54. The drive circuit 51 is a drive circuit used when driving the first upper arm element 311 of the first leg 31. The drive circuit 52 is a drive circuit used when driving the first lower arm element 312 of the first leg 31. The second upper arm element 321 and the second lower arm element 322 of the second leg 32 are also driven by two similar drive circuits.

The bootstrap circuit 54 includes a resistor 54a, a diode 54b, and a capacitor 54c which is a bootstrap capacitor. A drive voltage is applied to the capacitor 54c from the drive power supply 55 via a series circuit of the resistor 54a and the diode 54b. In the bootstrap circuit 54 configured in this way, when the lower arm switching elements Q2 and Q4 are turned on, a current flows through the resistor 54a, the diode 54b, the capacitor 54c, and the lower arm switching elements Q2 and Q4, and the capacitor 54c becomes It will be charged. The charging voltage of the capacitor 54c is a gate drive voltage for driving the switching elements Q1 and Q3 of the upper arm.

As in the example of FIG. 9, in the gate drive circuit 15 provided with the bootstrap circuit 54, the gate drive voltage for driving the switching elements Q1 and Q3 of the upper arm turns on the switching elements Q2 and Q4 of the lower arm. Obtained by letting.

According to the switching patterns of FIGS. 8 (b) to 8 (d), among the switching elements Q1 to Q4, the switching elements Q2 and Q4, which are the lower arm elements, are alternately turned on for half a cycle of the power supply voltage. Be controlled. When the switching elements Q2 and Q4 of the lower arm operate, the capacitor 54c of the bootstrap circuit 54 is charged as described above. Therefore, if the control signals S311 to S322 as shown in FIG. 8 are used, it is possible to reliably generate a gate drive voltage for driving the switching elements Q1 and Q3 of the upper arm.

Further, FIG. 10 is a diagram showing a second example of the operation waveform when operated in the operation mode shown in FIG. 7. 10 (b) shows the control signals S311 and S322 and the short-circuit control signal S331 shown in FIG. 8 (b) as they are, and the control signal S312 and the control signal S321 are replaced. Even if they are replaced in this way, sufficient time is given for the switching elements Q2 and Q4 of the lower arm to operate on. As a result, it is possible to reliably generate a gate drive voltage for driving the switching elements Q1 and Q3 of the upper arm.

Further, FIG. 11 is a diagram showing a third example of the operation waveform when operated in the operation mode shown in FIG. 7. In FIGS. 11 (b) to 11 (d), at least the operations of the switching elements Q1 and Q2 are stopped over the entire range of one cycle of the power supply cycle. Even if the operations of the switching elements Q1 and Q2 are stopped over the entire range of one cycle of the power supply cycle, synchronous rectification is not performed, and there is no problem in the rectification operation. Further, as will be described in detail later, when the current flowing through each arm element is large, the loss is improved by passing a parasitic diode or a diode connected in parallel rather than passing through the channel of the MOSFET. There is. Therefore, the operations of FIGS. 8, 10 or 11 may be appropriately replaced depending on the degree of temperature rise of the switching element.

Further, in the switching patterns of FIGS. 11B to 11D, two or more switching elements are not turned on at the same time. Therefore, since the current is blocked by at least one of the diodes D1 to D4, it is possible to reliably prevent a capacitor short circuit.

Further, in the switching patterns of FIGS. 8, 10 and 11 (c) and 11 (d), a power short circuit is realized by using only the short circuit switching element 331 without using the power short circuit operation by the switching elements Q1 to Q4. it can. The effect of this control will be described below.

For example, when the switching elements Q1 and Q3 are turned on to perform a power short-circuit operation, if the switching element Q4 is turned on, a capacitor short-circuit occurs at the route of the smoothing capacitor 4, the switching element Q3, and the switching element Q4. Therefore, when the switching element Q3 is in the on state, the switching element Q4 needs to be in the off state. After that, when performing synchronous rectification, it is necessary to control the switching element Q3 to the off state and then turn on the switching element Q4, which complicates the control. On the other hand, if the power supply short circuit is realized by operating the short circuit switching element 331 of the short circuit circuit 330, the power supply short circuit operation and the synchronous rectification operation can be compatible with each other without operating the switching elements Q3 and Q4 in a complementary manner. .. Specifically, the short-circuit switching element 331 may be turned off, and then the switching elements Q1 to Q4 may be controlled to be turned on. When the short-circuit switching element 331 is controlled to be turned on a plurality of times, the switching elements Q1 to Q4 may be turned on at the timing when the short-circuit switching element 331 is turned off. In either case, the effect of synchronous rectification can be obtained.

Further, when the complementary operations of the switching elements Q1 and Q2 and the switching elements Q3 and Q4 are not performed, the dead time is a short-circuit prevention time for preventing the switching elements Q1 and Q2 and the switching elements Q3 and Q4 from being turned on at the same time. There is no need to provide. If the control does not provide a dead time, the consistency between the command value by the control and the actual command value is improved. This makes it possible to improve efficiency while improving controllability and control stability.

In particular, in the switching pattern of FIG. 11D, all the switching elements Q1 to Q4 in the converter 3 are off-operated over the entire range of one cycle of the power supply voltage. Thereby, controllability and control stability can be improved. Moreover, since the switching element does not operate on, it is possible to reliably prevent a capacitor short circuit.

Further, in the switching pattern of each (d) of FIGS. 8 and 11, the switching elements Q1 and Q3 of the upper arm are in the off operation over the entire range of one cycle of the power supply voltage. As a result, the power consumption of the drive circuit 51 that drives the switching elements Q1 and Q3 of the upper arm can be suppressed. Since the power consumption of the drive circuit 51 increases in proportion to the number of switchings, it is effective to reduce the power consumption and improve the efficiency by performing the second switching operation in which the number of switchings is large. Further, since the switching elements Q1 and Q2 and the switching elements Q3 and Q4 are not complementarily operated, the controllability and control stability can be improved.

Further, in FIG. 11C, when the short-circuit switching element 331 is in the first switching operation, each switching element of one of the first leg 31 and the second leg 32 is turned on. It is a switching pattern that does not work. Further, in FIG. 10D, when the short-circuit switching element 331 is in the second switching operation, each switching element of one of the first leg 31 and the second leg 32 is turned on. It is a switching pattern that does not work. Therefore, if any of these switching patterns is used, simple switching control can be performed, and controllability and control stability can be improved.

Note that the switching patterns shown in FIGS. 8, 10 and 11 are examples, and are not limited to these examples. The switching elements Q1 to Q4 and the short-circuit switching element 331 are operated in an arbitrary operation mode by arbitrarily combining the diode rectification operation, the synchronous rectification operation, the first switching operation, and the second switching operation shown in these figures. be able to.

Next, the loss characteristics of the MOSFET used in the motor drive device 100 will be described. FIG. 12 is a diagram showing the loss characteristics of the MOSFET used in the DC power supply device 50 of the first embodiment. In FIG. 12, the horizontal axis shows the current flowing through the MOSFET in the on state and the current flowing through the parasitic diode. The vertical axis shows the voltage required to pass a current through the switching element in the on state and the voltage required to pass a current through the parasitic diode.

In FIG. 12, the solid line represents the forward voltage of the parasitic diode. The parasitic diode forward voltage is an example of a current-voltage characteristic that represents the loss that occurs in a parasitic diode. Generally, a diode requires a large voltage because the loss is large when the current value is small, but when the current value is larger than a certain value, the rate of change of the loss is improved and the slope of the current-voltage characteristic is relaxed. .. This characteristic appears in the waveform shown by the solid line in FIG.

The broken line represents the MOSFET drain-source voltage, which is the voltage between the MOSFET drain and the source. The MOSFET drain-source voltage is an example of a current-voltage characteristic that represents a current flowing through a carrier of a switching element and a loss caused by the on-resistance of the switching element due to the current flowing. In a switching element such as a MOSFET, the voltage required to pass a current increases in a quadratic curve with respect to the current value. This characteristic appears in the waveform shown by the broken line in FIG.

In FIG. 12, at the cross point where the solid line and the broken line intersect, the current flowing through the parasitic diode and the voltage required to flow the current are equal to the current flowing through the MOSFET and the voltage required to flow the current. It is a point. In the first embodiment, the current value at the cross point where the two current-voltage characteristics of the parasitic diode and the switching element intersect is defined as the “second current threshold”. The above-mentioned current threshold value, that is, the current threshold value used when comparing the absolute value of the detected value Is of the power supply current is referred to as a "first current threshold value". In FIG. 12, the second current threshold value is represented by “Ith2”. The second current threshold is a value larger than the first current threshold.

Next, the timing at which the control unit 10 turns on / off the switching element using the first current threshold value and the second current threshold value in the synchronous rectification mode will be described. FIG. 13 is a diagram showing the timing at which the control unit 10 turns on the switching element in the DC power supply device 50 according to the first embodiment. In FIG. 13, the horizontal axis is time. The waveforms of the power supply voltage and the power supply current are shown in the upper part of FIG. In the lower part of FIG. 13, switching elements Q1 and Q2 are current-synchronized switching elements whose on / off is controlled according to the polarity of the power supply current, and switching elements Q3 and Q4 are turned on / off according to the polarity of the power supply voltage. Is shown to be a controlled voltage-synchronized switching element. Further, FIG. 13 shows the values of the first current threshold value Is1 and the second current threshold value Is2 together with the waveform of the power supply current. Although FIG. 13 shows one cycle of the AC power output from the AC power supply 1, the control unit 10 shall perform the same control as the control shown in FIG. 13 in the other cycles.

When the power supply voltage polarity is positive, the control unit 10 turns on the switching element Q4 and turns off the switching element Q3. Further, when the power supply voltage polarity is negative, the control unit 10 turns on the switching element Q3 and turns off the switching element Q4. In FIG. 13, the timing at which the switching element Q4 is turned from on to off and the timing at which the switching element Q3 is turned from off to on are the same timing, but the timing is not limited to this. The control unit 10 may provide a dead time during which the switching elements Q3 and Q4 are both turned off between the timing at which the switching element Q4 is turned from on to off and the timing at which the switching element Q3 is turned from off to on. Similarly, the control unit 10 provides a dead time during which the switching elements Q3 and Q4 are both turned off between the timing at which the switching element Q3 is turned from on to off and the timing at which the switching element Q4 is turned from off to on. May be good.

When the power supply voltage polarity is positive, the control unit 10 turns on the switching element Q1 when the absolute value of the power supply current becomes equal to or higher than the first current threshold value Is1. Further, when the absolute value of the power supply current exceeds the second current threshold value Is2, the switching element Q1 is turned off. After that, the control unit 10 turns on the switching element Q1 when the absolute value of the power supply current becomes small and the absolute value of the power supply current becomes equal to or less than the second current threshold value Is2. Further, when the absolute value of the power supply current becomes smaller than the first current threshold value Is1, the switching element Q1 is turned off. Further, when the power supply voltage polarity is negative, the control unit 10 turns on the switching element Q2 when the absolute value of the power supply current becomes equal to or higher than the first current threshold value Is1. Further, when the absolute value of the power supply current exceeds the second current threshold value Is2, the switching element Q2 is turned off. After that, the control unit 10 turns on the switching element Q2 when the absolute value of the power supply current becomes small and the absolute value of the power supply current becomes equal to or less than the second current threshold value Is2. Further, when the absolute value of the power supply current becomes smaller than the first current threshold value Is1, the switching element Q2 is turned off.

When the absolute value of the power supply current is equal to or less than the first current threshold value Is1, the control unit 10 controls so that the switching elements Q1 and Q3 do not turn on at the same time, and controls the switching elements Q2 and Q4 not to turn on at the same time. .. As a result, the control unit 10 can prevent a capacitor short circuit in the motor drive device 100.

By controlling the control unit 10 as described above, the motor drive device 100 can realize synchronous rectification by the switching elements Q1 and Q2 of the first leg 31. Specifically, when the absolute value of the power supply current is equal to or greater than the first current threshold value Is1 and equal to or less than the second current threshold value Is2, the control unit 10 supplies a current to the switching element Q1 or the switching element Q2 having a small loss in this range. Shed. Further, when the absolute value of the power supply current is larger than the second current threshold value Is2, the control unit 10 causes a current to flow through the diode D1 or the diode D2 having a small loss in this range. As a result, the motor drive device 100 can pass a current through an element having a small loss according to the current value, so that a decrease in efficiency can be suppressed and a highly efficient device with reduced loss can be obtained.

Note that the control unit 10 may perform a boosting operation by performing switching control in which the switching elements Q1 and Q2 are complementarily turned on and off during the period in which the switching element Q1 is turned on. Similarly, the control unit 10 may perform a boosting operation by performing switching control in which the switching elements Q1 and Q2 are complementarily turned on and off during the period in which the switching element Q2 is turned on.

That is, when the absolute value of the power supply current is equal to or higher than the first current threshold value Is1 and equal to or lower than the second current threshold value Is2, the control unit 10 has the first leg 31 and the second leg according to the polarity of the power supply current. The switching element of one of the switching elements Q1 and Q2 constituting the first leg 31 of one of the 32 is allowed to be turned on. Further, when the absolute value of the power supply current is smaller than the first current threshold value Is1 or larger than the second current threshold value Is2, the control unit 10 is the same one switching element as the above-mentioned switching elements Q1 and Q2. Prohibit turning on.

Specifically, when the polarity of the power supply current is positive and the absolute value of the power supply current is equal to or higher than the first current threshold value Is1 and equal to or lower than the second current threshold value Is2, the control unit 10 turns on the switching element Q1. Allow. When the absolute value of the power supply current is smaller than the first current threshold value Is1 or larger than the second current threshold value Is2, the switching element Q1 is prohibited from being turned on. When the polarity of the power supply current is positive and the absolute value of the power supply current is equal to or greater than the first current threshold value Is1 and equal to or less than the second current threshold value Is2, the control unit 10 switches the switching element Q1 during the off period. Turn on Q2. When the absolute value of the power supply current is smaller than the first current threshold value Is1 or larger than the second current threshold value Is2, turning on the switching element Q2 is also prohibited.

Further, the control unit 10 permits the switching element Q2 to be turned on when the polarity of the power supply current is negative and the absolute value of the power supply current is equal to or higher than the first current threshold value Is1 and equal to or lower than the second current threshold value Is2. .. When the absolute value of the power supply current is smaller than the first current threshold value Is1 or larger than the second current threshold value Is2, the switching element Q2 is prohibited from being turned on. Further, when the polarity of the power supply current is negative and the absolute value of the power supply current is equal to or higher than the first current threshold value Is1 and equal to or lower than the second current threshold value Is2, the control unit 10 is in the period when the switching element Q2 is off. The switching element Q1 is turned on. When the absolute value of the power supply current is smaller than the first current threshold value Is1 or larger than the second current threshold value Is2, the switching element Q1 is also prohibited from being turned on.

In this way, the control unit 10 allows the switching element to be turned on in a region where the absolute value of the power supply current is equal to or higher than the first current threshold value Is1 and the loss of the switching element is smaller than the loss of the parasitic diode. Further, the control unit 10 prohibits the switching element from being turned on in a region where the loss of the switching element is larger than the loss of the parasitic diode.

In the example of FIG. 13, the control unit 10 controls the on / off of the switching elements Q3 and Q4 according to the polarity of the power supply voltage, and controls the on / off of the switching elements Q1 and Q2 according to the polarity of the power supply current. However, it is not limited to this. The control unit 10 may control the on / off of the switching elements Q1 and Q2 according to the polarity of the power supply voltage, and may control the on / off of the switching elements Q3 and Q4 according to the polarity of the power supply current.

Further, the second current threshold value Is2 is, as described above, a current value when the voltage required for passing the current through the parasitic diode and the switching element becomes the same value, but is not limited to this. The second current threshold value Is2 may be a value determined according to the characteristics of the voltage required to pass a current through the parasitic diode and the characteristics of the voltage required to pass a current through the switching element.

For example, the second current threshold value Is2 is set to be larger than the current value when the voltage required to pass the current through the parasitic diode and the switching element becomes the same value according to the switching loss generated in the switching element. It may be a value. Thereby, it is possible to determine the second current threshold value Is2 in consideration of the switching element generated when the switching element is switched from on to off. In this case, the control unit 10 keeps the switching element on when the loss cannot be reduced by turning off the switching element even if the absolute value of the power supply current becomes larger while the switching element is on. To. As a result, the motor drive device 100 can further suppress a decrease in efficiency.

Further, the second current threshold value Is2 may be a value obtained by adding or subtracting a specified value with respect to the current value when the voltage required for passing the current through the parasitic diode and the switching element becomes the same value. .. As a result, the second current threshold value Is2 can be determined in consideration of the difference in characteristics due to the variation in the components of each element. In this case, the control unit 10 improves the reduction of loss as compared with the case where the second current threshold value Is2 is the current value when the voltage required for passing the current through the parasitic diode and the switching element becomes the same value. It may not be possible. However, the control unit 10 can reduce the loss as compared with the case where the switching element is continuously turned on even if the absolute value of the power supply current is further increased while the switching element is turned on.

FIG. 14 is a flowchart used for explaining the operation of the main part in the first embodiment. FIG. 14 shows a processing flow in which the control unit 10 of the motor drive device 100 controls the switching elements Q1 and Q2 on and off. Here, as an example, a case where the polarity of the power supply current is positive will be described.

The control unit 10 compares the absolute value | Is | of the detected value Is of the power supply current with the first current threshold value (step S21). When the absolute value | Is | is smaller than the first current threshold value (steps S21, No), the control unit 10 prohibits the switching element Q1 from being turned on (step S22). When the absolute value | Is | is equal to or greater than the first current threshold value (step S21, Yes), the control unit 10 compares the absolute value | Is | with the second current threshold value (step S23). When the absolute value | Is | is equal to or less than the second current threshold value (steps S23, No), the control unit 10 permits the switching element Q1 to be turned on (step S24). When the absolute value | Is | is larger than the second current threshold value (step S23, Yes), the control unit 10 prohibits the switching element Q1 from being turned on (step S22). After step S22 or step S24, the control unit 10 returns to step S21 and repeats the above process. When the polarity of the power supply current is negative, the control unit 10 performs the same processing as described above for the switching element Q2.

In step S21 described above, the case where the absolute value | Is | and the first current threshold value are equal is determined by "Yes", but it may be determined by "No". That is, the case where the absolute value | Is | and the first current threshold value are equal may be determined by either "Yes" or "No". Further, in step S23 described above, the case where the absolute value | Is | and the second current threshold value are equal is determined by "No", but it may be determined by "Yes". That is, the case where the absolute value | Is | and the second current threshold value are equal may be determined by either "Yes" or "No".

Next, the configuration of the switching element will be described. In the motor drive device 100, one of the methods for increasing the switching speed of the switching element is a method for reducing the gate resistance of the switching element. As the gate resistance becomes smaller, the charge / discharge time to the gate input capacitance becomes shorter, and the turn-on period and the turn-off period become shorter, so that the switching speed becomes faster.

However, there is a limit to reducing switching loss by reducing the gate resistance. Therefore, it is illustrated that the switching element is composed of a WBG semiconductor such as GaN or SiC. By using a WBG semiconductor for the switching element, the loss per switching can be further suppressed, the efficiency is further improved, and high frequency switching becomes possible. Further, by enabling high-frequency switching, the reactor 2 can be miniaturized, and the motor drive device 100 can be miniaturized and lightened. Further, by using the WBG semiconductor for the switching element, the switching speed is improved and the switching loss is suppressed. This makes it possible to simplify heat dissipation measures so that the switching element can continue to operate normally. Further, by using a WBG semiconductor for the switching element, the switching frequency can be set to a sufficiently high value, for example, 16 kHz or more. As a result, noise caused by switching can be suppressed.

Further, in a GaN semiconductor, two-dimensional electron gas is generated at the interface between the GaN layer and the aluminum gallium nitride layer, and the mobility of carriers is high due to this two-dimensional electron gas. Therefore, a switching element using a GaN semiconductor can realize high-speed switching. Here, when the AC power supply 1 is a commercial power supply of 50 Hz or 60 Hz, the audible range frequency is in the range of 16 kHz to 20 kHz, that is, in the range of 266 to 400 times the frequency of the commercial power supply. GaN semiconductors are suitable for switching at frequencies higher than this audible frequency. When switching elements Q1 to Q4 made of silicon (Si), which is the mainstream as a semiconductor material, are driven at a switching frequency of several tens of kHz or more, the ratio of switching loss becomes large, and heat dissipation measures are indispensable. On the other hand, the switching elements Q1 to Q4 made of the GaN semiconductor have a very small switching loss even when driven at a switching frequency of several tens of kHz or more, specifically, a switching frequency higher than 20 kHz. Therefore, heat dissipation measures are not required, or the size of the heat dissipation member used for heat dissipation measures can be reduced, and the motor drive device 100 can be made smaller and lighter. Further, since high frequency switching is possible, the reactor 2 can be miniaturized. The switching frequency is preferably 150 kHz or less in order to prevent the primary component of the switching frequency from entering the measurement range of the noise terminal voltage standard.

Further, since the WBG semiconductor has a smaller capacitance than the Si semiconductor, the generation of recovery current due to switching is small, and the generation of loss and noise due to recovery current can be suppressed. Therefore, the WBG semiconductor is suitable for high frequency switching.

Note that the on-resistance of SiC semiconductors is smaller than that of GaN semiconductors. Therefore, the first upper arm element 311 and the first lower arm element 312 of the first leg 31 having a larger number of switching times than the second leg 32 are made of a GaN semiconductor, and the second leg element 312 has a smaller number of switching times. The second upper arm element 321 and the second lower arm element 322 of the leg 32 may be made of a SiC semiconductor. As a result, the characteristics of the SiC semiconductor and the GaN semiconductor can be fully utilized. Further, by using the SiC semiconductor for the second upper arm element 321 and the second lower arm element 322 of the second leg 32, which has fewer switching times than the first leg 31, the second upper arm Of the losses of the element 321 and the second lower arm element 322, the conduction loss accounts for a large proportion, and the turn-on loss and the turn-off loss become small. Therefore, the increase in heat generation due to the switching of the second upper arm element 321 and the second lower arm element 322 is suppressed, and the second upper arm element 321 and the second lower arm element constituting the second leg 32 are suppressed. The chip area of 322 can be made relatively small. This makes it possible to effectively utilize a SiC semiconductor having a low yield at the time of chip manufacturing.

Further, an SJ-MOSFET having a super junction structure may be used for the second upper arm element 321 and the second lower arm element 322 of the second leg 32 having a small number of switchings. By using SJ-MOSFET, it is possible to suppress the demerit that the capacitance is high and recovery is likely to occur while taking advantage of the low on-resistance which is the merit of SJ-MOSFET. Further, by using the SJ-MOSFET, the manufacturing cost of the second leg 32 can be reduced as compared with the case of using the WBG semiconductor.

In addition, WBG semiconductors have higher heat resistance than Si semiconductors and can operate even at high junction temperatures. Therefore, by using the WBG semiconductor, the first leg 31 and the second leg 32 can be configured by a small chip having a large thermal resistance. In particular, SiC semiconductors, which have a low yield during chip manufacturing, can be used for small chips to reduce costs.

Further, since the WBG semiconductor suppresses the increase in the loss generated in the switching element even when driven at a high frequency of about 100 kHz, the loss reduction effect due to the miniaturization of the reactor 2 becomes large, and a wide output band, that is, a wide load Under the conditions, a highly efficient converter can be realized.

Further, the WBG semiconductor has higher heat resistance than the Si semiconductor and has a high heat generation allowable level for switching due to the bias of the loss between the arms, so that it is suitable for the first leg 31 in which the switching loss due to high frequency driving occurs.

As described above, according to the first embodiment, the voltage detection unit 7 which is the first physical quantity detection unit detects the bus voltage which is the first physical quantity which represents the operating state of the output side of the converter 3. The voltage detection unit 5, which is the second physical quantity detection unit, detects the power supply voltage, which is the second physical quantity indicating the operating state of the input side of the converter 3. The first and second physical quantities are input to the control unit 10. The control unit 10 controls the continuity of each switching element of the converter 3 based on the first and second physical quantities, and combines the continuity of the switching elements Q1 to Q4 with the continuity of the short-circuit switching element 331 to form the converter 3. It has a plurality of operation modes for operating in different operation modes. As a result, it is possible to achieve both efficiency improvement, power factor improvement, and power supply harmonic suppression by synchronous rectification.

Further, according to the first embodiment, when the short-circuit switching element 331 is in the second switching operation, the control unit 10 turns off all the switching elements Q1 to Q4 in the converter 3. As a result, controllability and control stability can be improved, and capacitor short circuits can be reliably prevented.

Further, according to the first embodiment, when the short-circuit switching element 331 is in the first switching operation, the control unit 10 receives one of the first leg 31 and the second leg 32 in the converter 3. The switching element of the leg is turned off. As a result, simple switching control can be performed, and controllability and control stability can be improved.

Further, according to the first embodiment, the gate drive circuit 15 which is a drive circuit for driving the converter 3 is a drive power source for driving the switching elements Q1 and Q3 of the upper arm connected to the positive side of the smoothing capacitor 4. The bootstrap circuit 54 is provided. When the short-circuit switching element 331 is in the second switching operation, the control unit 10 turns off the switching elements Q1 and Q3 of the upper arm in the converter 3. As a result, the power consumption of the bootstrap circuit 54 can be suppressed. Further, since it is not necessary to perform complementary operations of the switching elements Q1 and Q2 and the switching elements Q3 and Q4, controllability and control stability can be improved.

Next, the hardware configuration for realizing the function of the control unit 10 in the first embodiment will be described with reference to the drawings of FIGS. 15 and 16. FIG. 15 is a block diagram showing an example of a hardware configuration that embodies the function of the control unit 10 according to the first embodiment. FIG. 16 is a block diagram showing another example of a hardware configuration that embodies the function of the control unit 10 according to the first embodiment.

When the function of the control unit 10 according to the first embodiment is realized, as shown in FIG. 15, the processor 300 that performs the calculation, the memory 302 that stores the program read by the processor 300, and the input / output of the signal are input / output. It can be configured to include the interface 304 to be performed.

The processor 300 may be an arithmetic unit, a microprocessor, a microcomputer, a CPU (Central Processing Unit), or a DSP (Digital Signal Processor). Further, the memory 302 includes a non-volatile or volatile semiconductor memory such as a RAM (Random Access Memory), a ROM (Read Only Memory), a flash memory, an EPROM (Erasable Program ROM), and an EPROM (registered trademark) (Electrically EPROM). Examples thereof include magnetic disks, flexible disks, optical disks, compact disks, mini disks, and DVDs (Digital Versailles Disc).

The memory 302 stores a program that executes the function of the control unit 10 according to the first embodiment. The processor 300 sends and receives necessary information via the interface 304, the processor 300 executes a program stored in the memory 302, and the processor 300 refers to a table stored in the memory 302 to perform the above-described processing. It can be carried out. The calculation result by the processor 300 can be stored in the memory 302.

Further, when the function of the control unit 10 in the first embodiment is realized, the processing circuit 305 shown in FIG. 16 can also be used. The processing circuit 305 corresponds to a single circuit, a composite circuit, an ASIC (Application Specific Integrated Circuit), an FPGA (Field-Programmable Gate Array), or a combination thereof. The information input to the processing circuit 305 and the information output from the processing circuit 305 can be obtained via the interface 306. Even in the configuration using the processing circuit 305, some processing in the control unit 10 may be performed by the processor 300 having the configuration shown in FIG.

Embodiment 2.
In the second embodiment, an application example of the motor drive device 100 described in the first embodiment will be described. FIG. 17 is a diagram showing the configuration of the air conditioner 400 according to the second embodiment. The motor drive device 100 described in the first embodiment can be applied to products such as a blower, a compressor, and an air conditioner. In the second embodiment, as an application example of the motor drive device 100 according to the first embodiment, an example in which the motor drive device 100 is applied to the air conditioner 400 will be described.

In FIG. 17, a motor 500 is connected to the output side of the motor drive device 100, and the motor 500 is connected to the compression element 504. The compressor 505 includes a motor 500 and a compression element 504. The refrigeration cycle unit 506 is configured to include a four-way valve 506a, an indoor heat exchanger 506b, an expansion valve 506c, and an outdoor heat exchanger 506d.

The flow path of the refrigerant circulating inside the air conditioner 400 is from the compression element 504 via the four-way valve 506a, the indoor heat exchanger 506b, the expansion valve 506c, the outdoor heat exchanger 506d, and again via the four-way valve 506a. Therefore, it is configured to return to the compression element 504. The motor drive device 100 receives AC power from the AC power source 1 and rotates the motor 500. The compression element 504 executes a compression operation of the refrigerant by rotating the motor 500, and the refrigerant can be circulated inside the refrigeration cycle unit 506.

Further, in the air conditioner 400, the operation under the intermediate condition where the output is less than half of the rated output, that is, the low output condition is dominant throughout the year, so that the contribution to the annual power consumption under the intermediate condition is high. Become. Further, in the air conditioner 400, the rotation speed of the motor 500 tends to be low, and the bus voltage required to drive the motor 500 tends to be low. Therefore, it is effective from the viewpoint of system efficiency that the switching element used in the air conditioner 400 is operated in a passive state. Therefore, the motor drive device 100 capable of reducing the loss in a wide range of operation modes from the passive state to the high frequency switching state is useful for the air conditioner 400. The motor drive device also has a method called an interleave method, which is different from the method of the first embodiment. Although the reactor 2 can be miniaturized by the interleave method, it is not necessary to miniaturize the reactor 2 because the air conditioner 400 is often operated under intermediate conditions. On the other hand, from the viewpoint of harmonic suppression and power factor, the method of the first embodiment is more effective. Therefore, the motor drive device 100 according to the first embodiment is particularly useful in an air conditioner.

Further, since the motor drive device 100 according to the first embodiment can suppress the switching loss, the temperature rise of the motor drive device 100 is suppressed, and even if the size of the outdoor unit blower (not shown) is reduced, the motor drive device 100 It is possible to secure the cooling capacity of the substrate mounted on the. Therefore, the motor drive device 100 according to the first embodiment is suitable for an air conditioner 400 having high efficiency and a high output of 4.0 kW or more.

Further, by using the motor drive device 100 according to the first embodiment, the bias of heat generation between the legs is reduced. As a result, the reactor 2 can be downsized by driving the switching elements Q1 to Q4 at high frequencies, and an increase in the weight of the air conditioner 400 can be suppressed. Further, according to the motor drive device 100 according to the first embodiment, the switching loss is reduced, the energy consumption rate is low, and the highly efficient air conditioner 400 can be realized by high-frequency driving of the switching elements Q1 to Q4.

In the air conditioner 400, when a momentary power failure occurs, the operation of the converter 3 is stopped first, then the rotation of the compressor 500 is stopped, and finally the rotation of the fan is stopped. To operate. Generally, the driving energy of the fan is small, and the amount of heat generated by the fan is small. Therefore, the circuit components of the converter 3 and the inverter 18 can be cooled by the wind of the fan by finally stopping the rotation of the fan. In particular, when the temperature of the smoothing capacitor 4, which is a component of the converter 3, becomes high, the capacity decreases. Therefore, it is possible to extend the life of the smoothing capacitor 4 by appropriately cooling it even in the event of a momentary power failure.

The configuration shown in the above-described embodiment shows an example of the content of the present invention, can be combined with another known technique, and is configured without departing from the gist of the present invention. It is also possible to omit or change a part of.

1 AC power supply, 2 reactor, 3 converter, 3a, 3b, 3c, 3d, 26a, 26b, 26c connection point, 4 smoothing capacitor, 5,7 voltage detector, 6,9 current detector, 10 control unit, 12 load , 14 power supply circuit, 15, 17 gate drive circuit, 16a, 16b DC bus, 18 inverter, 18A, 18B, 18C leg, 18a, 331a transistor, 18b, 54b, 331b, D1, D2, D3, D4 diode, 18UN, 18VN, 18WN lower arm element, 18UP, 18VP, 18WP upper arm element, 31 first leg, 32 second leg, 50 DC power supply, 51, 52 drive circuit, 54 bootstrap circuit, 54a resistor, 54c capacitor, 55 drive power supply, 100 motor drive device, 300 processor, 302 memory, 304, 306 interface, 305 processing circuit, 311 first upper arm element, 312 first lower arm element, 321 second upper arm element, 322nd 2 lower arm element, 330 short circuit, 331 short switching element, 332 diode bridge, 400 air conditioner, 500 motor, 504 compression element, 505 compressor, 506 refrigeration cycle part, 506a four-way valve, 506b indoor heat exchanger, 506c expansion valve, 506d outdoor heat exchanger, 600 semiconductor substrate, 601,603 region, 602 oxide insulating film, 604 channel, D drain electrode, Q1, Q2, Q3, Q4 switching element, S source electrode.

Claims

With the reactor
A converter that has four unidirectional elements that are bridge-connected, is connected to an AC power supply via the reactor, converts the power supply voltage, which is the AC voltage output from the AC power supply, into a DC voltage, and applies it to the load. ,
A short-circuit circuit having a short-circuit switching element, which is connected between the input terminals of the converter and performs a power short-circuit operation of short-circuiting the power supply voltage via the reactor by turning on the short-circuit switching element.
A smoothing capacitor connected between the output terminals of the converter,
A first physical quantity detection unit that detects a first physical quantity that represents the operating state of the output side of the converter, and
A second physical quantity detector that detects a second physical quantity that represents the operating state of the input side of the converter, and
A control unit to which the first and second physical quantities are input and controls the operation of the converter,
With
Two of the four unidirectional elements in the converter are connected in series to form a first leg, and the remaining two unidirectional elements are connected in series. Make up the second leg,
At least two of the one-way elements in the first and second legs connected to the positive side of the smoothing capacitor, or two in the first and second legs connected to the negative side of the smoothing capacitor. Switching elements are connected in parallel to each of the one-way element, the two one-way elements in the first leg, or the two one-way elements in the second leg.
The control unit is a DC power supply device having a plurality of operation modes in which the continuity of the switching element and the continuity of the short-circuit switching element are combined to operate the converter in different operation modes.
A diode rectification operation in which a current is passed through the four unidirectional elements, a synchronous rectification operation in which a switching element corresponding to the unidirectional element is turned on at a timing when a current flows through the unidirectional element, and a power supply voltage. A first switching operation in which the power supply short-circuit operation is locally performed within half a cycle of the power supply cycle, which is one cycle, and a second switching operation in which the power supply short-circuit operation is performed over the entire range of one cycle of the power supply cycle. At least one of the actions and
The DC power supply device according to claim 1, wherein the converter is operated in a different operation mode by combining the above.
The DC power supply device according to claim 2, wherein when the short-circuit switching element is in the second switching operation, all the switching elements are not turned on.
2. When the short-circuit switching element is in the first switching operation or the second switching operation, the switching element of one of the first and second legs does not operate on. Or the DC power supply according to 3.
A drive circuit for driving the converter based on a control signal output from the control unit is provided.
The drive circuit includes a bootstrap circuit that is a drive power source for driving the switching element connected to the positive side of the smoothing capacitor.
The DC power supply device according to claim 2, wherein the switching element connected to the bootstrap circuit does not operate when the short-circuit switching element is in the second switching operation.
The DC power supply device according to any one of claims 1 to 5, wherein the switching element is a metal oxide semiconductor field effect transistor formed of a wide bandgap semiconductor.
The DC power supply device according to claim 6, wherein the wide bandgap semiconductor is silicon carbide, gallium nitride, gallium oxide, or diamond.
The DC power supply device according to any one of claims 1 to 5, wherein the switching element is a metal oxide semiconductor field effect transistor having a superjunction structure.
The DC power supply device according to any one of claims 6 to 8, wherein the unidirectional element is a parasitic diode of the metal oxide semiconductor field effect transistor.
The DC power supply device according to any one of claims 1 to 9, wherein the unidirectional element is a diode.
The DC power supply device according to any one of claims 1 to 10.
An inverter that converts the output voltage of the DC power supply device into an AC voltage and applies it to the motor provided in the load.
Motor drive device equipped with.
A blower including the motor driving device according to claim 11.
A compressor including the motor driving device according to claim 11.
An air conditioner including at least one of the blower according to claim 12 and the compressor according to claim 13.