JP2016220378A - Dc power supply device and air conditioner using the same - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a DC power supply device capable of securing the driving voltage for a high-side element and stably performing synchronous rectification control.SOLUTION: A DC power supply device has a full-wave rectification operation mode of converting an AC voltage input through a reactor L1 to a DC voltage by a synchronous rectification operation and a step-up chopper operation of first and second diodes D1, D2 and first and second MOSFETs Q1, Q2, the first MOSFET and the second MOSFET being switched so as to perform the synchronous rectification operation according to the polarity of the AC voltage, a partial switching operation mode in which on/off switching is complementarily performed by a predetermined number of times in a half cycle of the AC voltage, a high-speed switching operation mode in which on/off switching is complementarily performed at a predetermined frequency. The capacitor C2 of a bootstrap circuit serving as a control power source of the first MOSFET is charged when the second MOSFET is turned on.SELECTED DRAWING: Figure 1

Description

  The present invention relates to a DC power supply device that converts an AC voltage into a DC voltage, and an air conditioner using the same.

  In trains, automobiles, air conditioners and the like, a DC power supply device (DC power supply device) that converts an AC voltage into a DC voltage is mounted. And the direct-current voltage output from a direct-current power supply device is converted into the alternating voltage of a predetermined frequency with an inverter, and this alternating voltage is applied to loads, such as a motor.

  In such a DC power supply device, there is a demand for energy saving by increasing power conversion efficiency. As means for increasing the power conversion efficiency, for example, a conventional technique described in Patent Document 1 is known.

  In this prior art, a synchronous rectifier circuit using a MOSFET as a converter circuit has been proposed. The circuit configuration is such that the drain of the first MOSFET is connected to the positive side of the DC voltage and the source of the second MOSFET is connected to the negative side. The source of the first MOSFET and the drain of the second MOSFET are connected.

Japanese Patent No. 5349636

  In the configuration in which the first MOSFET is connected to the positive electrode side of the smoothing capacitor and the second MOSFET is connected to the negative electrode side of the smoothing capacitor to perform synchronous rectification as in the above prior art, it is connected to the positive electrode side of the smoothing capacitor. In order to drive the first MOSFET, a power supply circuit based on the source potential of the first MOSFET is required. However, the above prior art does not take into consideration the stable rectification control by securing the driving voltage of the first MOSFET, that is, the high-side element.

  Therefore, the present invention provides a direct-current power supply device that can secure a driving voltage for a high-side element and stably perform synchronous rectification control, and an air conditioner that is improved in reliability by using the direct-current power supply device.

  In order to solve the above-mentioned problem, a direct current power supply device according to the present invention has an anode of a first diode and a cathode of a second diode connected, a source of the first MOSFET and a drain of the second MOSFET connected, The cathode of the first diode and the drain of the first MOSFET are connected to the positive side of the smoothing capacitor, the anode of the second diode and the source of the second MOSFET are connected to the negative side of the smoothing capacitor, and the first diode An AC power supply is connected to a connection point between the first MOSFET and the second MOSFET, and a connection point between the first MOSFET and the second MOSFET via a reactor, and by switching of the first MOSFET and the second MOSFET, Converting an AC voltage of an AC power source into a DC voltage, the first MOSFET and the second MOSFE However, the full-wave rectification operation mode that is switched so as to perform synchronous rectification operation according to the polarity of the AC voltage, and the first MOSFET and the second MOSFET are complementary in a predetermined number of times in a half cycle of the AC voltage. A partial switching operation mode that is switched on and off; and a high-speed switching operation mode in which the first MOSFET and the second MOSFET are complementarily switched on and off at a predetermined frequency. A bootstrap circuit serving as a control power source for the MOSFET, and a capacitor of the bootstrap circuit is charged when the second MOSFET is switched on.

  In order to solve the above problems, an air conditioner according to the present invention includes an electric compressor in which a compressor is driven by an AC electric motor, an inverter circuit that supplies AC electric power to the electric compressor, and an inverter circuit as a load. A refrigerant that is compressed by an electric compressor circulates in the refrigeration cycle, and the DC power supply is the DC power supply according to the present invention.

  According to the present invention, the bootstrap circuit serving as the control power source of the first MOSFET is provided, and the capacitor of the bootstrap circuit is turned on in the full-wave rectification operation mode, the partial switching operation mode, and the high-speed switching operation mode. By being charged at the time, the driving voltage of the first MOSFET, which is the high-side element, can be secured, and the synchronous rectification control can be stably performed. This improves the efficiency of the DC power supply device and the air conditioner using the DC power supply device, and improves the reliability.

  Problems, configurations, and effects other than those described above will become apparent from the following description of embodiments.

BRIEF DESCRIPTION OF THE DRAWINGS It is a block diagram of the direct-current power supply device which is Example 1 of this invention. The path of the current flowing through the main circuit when the polarity of the AC power supply voltage is positive is shown. The path of the current flowing through the main circuit when the polarity of the AC power supply voltage is negative is shown. The path of the power factor correction current flowing through the main circuit when the polarity of the AC power supply voltage is positive is shown. The path of the power factor correction current flowing in the main circuit when the polarity of the AC power supply voltage is negative is shown. The relationship between the drive pulse of MOSFET, power supply voltage, and main circuit current in the positive cycle of AC power supply voltage is shown. The time change of the duty factor of MOSFET in the positive half cycle of alternating current power supply voltage is shown. The time change of the instantaneous value of the power supply voltage and the instantaneous value of the circuit current is shown. The time change of the duty ratio of MOSFET when current phase delay is considered and when it is not considered is shown. The path of the capacitor charging current in the bootstrap circuit is shown. When the polarity of the AC power supply voltage is positive, the path of the main circuit current and the path of the charging current in the bootstrap circuit when the capacitor of the bootstrap circuit is charged are shown. When the polarity of the AC power supply voltage is negative, the main circuit current path and the charging current path in the bootstrap circuit when the capacitor of the bootstrap circuit is charged are shown. FIG. 11 shows voltage waveforms of the main circuit unit and the AC power supply of the DC power supply device according to the first embodiment shown in FIG. FIG. 11 shows voltage waveforms of the main circuit unit and the AC power supply of the DC power supply device according to the first embodiment shown in FIG. 2 shows a gate drive circuit portion of a MOSFET in the first embodiment. The circuit part which operate | moves at the time of charge is shown. The circuit part which operate | moves at the time of discharge is shown. The circuit part which operate | moves at the time of discharge is shown. The charging time constant switching means for the capacitor of the bootstrap circuit is shown. The charging time constant switching means for the capacitor of the bootstrap circuit is shown. It is an air-conditioning cycle block diagram of the air conditioner of Example 2. It is an external view of the outdoor unit of the air conditioner of Example 2.

  Hereinafter, embodiments of the present invention will be described with reference to the drawings. In each figure, the same reference numerals indicate the same constituent elements or constituent elements having similar functions.

  FIG. 1 is a configuration diagram of a DC power supply device that is Embodiment 1 of the present invention.

  First, the circuit configuration and the configuration of each unit according to the first embodiment will be described with reference to FIG.

  The DC power supply device according to the first embodiment is a converter that converts an AC voltage supplied from an AC power supply Vs into a DC voltage and outputs the DC voltage to a load H (an inverter, a motor, or the like). The DC power supply device has an input side connected to an AC power source Vs and an output side connected to a load H.

  As shown in FIG. 1, the DC power supply includes a reactor L1, a smoothing capacitor C1, diodes D1 and D2, MOSFETs Q1 and Q2, a current detection unit 11, a current control gain control unit 12, and an AC voltage detection unit. 13, a zero cross determination unit 14, a load detection unit 15, a boost ratio control unit 16, a DC voltage detection unit 17, and a control unit 18.

  The anode of the diode D1 is connected to the cathode of the diode D2, and the connection point P1 is connected to the AC power source Vs via the wiring ha. Note that one end of the wiring ha is connected to the AC power supply Vs via the reactor L1, and the other end is connected to the connection point P1.

  The source of the MOSFET Q1 is connected to the drain of the MOSFET Q2, and the connection point P2 is connected to the AC power source Vs via the wiring hb. Note that one end of the wiring hb is connected to the AC power supply Vs via the current detection unit 11, and the other end is connected to the connection point P2.

  The anode of the diode D2 is connected to the source of the MOSFET Q2. The drain of MOSFET Q1 is connected to the cathode of diode D1. The cathode of the diode D1 and the drain of the MOSFET Q1 are connected to the positive electrode of the smoothing capacitor C1 via the wiring hc. Further, the anode of the diode D2 and the source of the MOSFET Q2 are connected to the negative electrode of the smoothing capacitor C1.

  The reactor L1 stores electric power supplied from the AC power supply Vs as energy, and further boosts the energy by releasing this energy. Reactor L1 is provided on wiring ha. That is, reactor L1 is connected between AC power supply Vs and connection point P1.

  The smoothing capacitor C1 smoothes the voltage rectified by the diodes D1 and D2 and the MOSFETs Q1 and Q2 into a DC voltage, and is connected to the output side of the DC power supply device via the wirings hc and hd.

  MOSFETs Q1 and Q2 are on / off controlled by a command from converter control unit 18 to be described later. In addition, by using a MOSFET (Metal Oxide Semiconductor Field Effect Transistor) as a switching element, switching can be performed at high speed, and synchronous rectification control is performed in which a rectification current is supplied to a MOSFET having a smaller voltage drop than a diode. Circuit loss can be reduced. Further, if a super junction MOSFET having a small on-resistance is used, conduction loss can be reduced.

  The MOSFETs Q1 and Q2 have parasitic diodes D11 and D21, respectively, inside the element. These parasitic diodes can be used as appropriate as diodes connected in reverse parallel to the MOSFET (for example, free-wheeling diodes). However, reverse recovery current is generated in the parasitic diodes D11 and D21 of the MOSFETs Q1 and Q2 in the first embodiment in a partial switching operation and a high-speed switching operation described later. Therefore, if a MOSFET having a small reverse recovery time (trr) is used, switching loss can be reduced. In addition, as for the diodes D1 and D2, since no reverse recovery current is generated in the circuit operation of the first embodiment, a diode having a small forward voltage can be used without considering the reverse recovery characteristics. For example, by using a general rectifier diode or a high breakdown voltage Schottky barrier diode, the conduction loss of the circuit can be reduced.

  The current detector 11 detects a current flowing through the wirings ha and hb, and is provided in the wiring hb. In the first embodiment, a current transformer (CT) is used as the current detector 11, but a Hall element, a shunt resistor, or the like may be used.

  The current control gain control unit 12 controls the current control gain Kp set from the circuit current effective value Is and the step-up ratio a. At this time, by controlling Kp × Is to a constant value, a DC voltage boosted to a times the AC power supply voltage can be obtained.

  The AC voltage detector 13 is connected to the wirings ha and hb and detects an AC power supply voltage applied from the AC power supply Vs. The AC voltage detection unit 13 outputs the voltage detection value to the zero cross determination unit 14. The zero-cross determination unit 14 determines whether the value of the AC power supply voltage value has been switched (that is, whether the zero-cross point has been reached) based on the voltage detection value by the AC voltage detection unit 13. For example, the zero cross determination unit 14 outputs a signal “1” to the control unit 18 while the AC power supply voltage is positive, and outputs a signal “0” to the converter control unit 18 while the power supply voltage is negative. Output. That is, the zero cross determination unit 14 switches the output signal from 1 to 0 or from 0 to 1 at the zero cross point of the AC power supply voltage.

  The load detector 15 is a shunt resistor, for example, and detects a load current supplied to the load H. The load detection unit 15 outputs the load current detection value to the boost ratio control unit 16. When the load H is a motor, a speed sensor such as a rotary encoder may be used as the load detector 15 to detect the rotational speed of the motor, and the load current value may be estimated from this rotational speed. In this case, the load current value is estimated by the load detection unit 15 itself or the subsequent step-up ratio control unit 16.

  The step-up ratio control unit 16 sets the DC voltage step-up ratio a based on the load current detection value by the load detection unit 15 and outputs the set step-up ratio a to the control unit 18. Then, the control unit 18 creates a drive pulse so as to boost the DC output voltage to the target voltage corresponding to the boost ratio a input from the boost ratio control unit 16, and outputs the drive pulse to the MOSFETs Q1 and Q2, whereby the MOSFET Q1 and Q2 switching control is performed. In the first embodiment, the control unit 18 creates a PWM pulse as a drive pulse by a PWM (Pulse Width Modulation) method.

  The DC voltage detector 17 detects the voltage (DC voltage) of the smoothing capacitor C1. The positive side and the negative side of the DC voltage detection unit 17 are connected to the wiring hc and the wiring hd, respectively. The DC voltage detection unit 17 outputs the DC voltage detection value to the control unit 18. Note that the DC voltage detection value of the DC voltage detection unit 17 is used by the control unit 18 to determine whether or not the DC voltage applied to the load H has reached a predetermined target value.

  The control part 18 is comprised by the microcomputer (Microcomputer), for example. In this case, a CPU (Central Processing Unit) reads a control program stored in a ROM (Read Only Memory), develops it in a RAM (Random Access Memory), and executes various processes for controlling the DC power supply device. . The control unit 18 turns the MOSFETs Q1 and Q2 on and off based on information input from the current detection unit 11, the current control gain control unit 12, the zero cross determination unit 14, the boost ratio control unit 16, and the DC voltage detection unit 17. Control.

  The drive circuit of the MOSFET Q1 includes a resistor R1, a diode D3, a capacitor C2, a drive circuit K1, a gate resistor Rg11, and a resistor Rg12. The drive circuit K1 uses the charged capacitor C2 as a DC voltage source to create a gate control signal for the MOSFET Q1 according to the drive pulse from the control unit 18. The gate control signal created by the drive circuit K1 is applied to the control terminal (gate terminal) of the MOSFET Q1 through the gate resistance Rg11 and the resistance Rg12.

  The drive circuit for the MOSFET Q2 includes a DC voltage source E, a drive circuit K2, a gate resistor Rg21, and a resistor Rg22. The drive circuit K2 uses the DC voltage source E as a control power supply to create a gate control signal for the MOSFET Q2 according to the drive pulse from the control unit 18. The gate control signal created by the drive circuit K1 is applied to the control terminal (gate terminal) of the MOSFET Q2 via the gate resistance Rg21 and the resistance Rg22.

  As the drive circuits K1 and K2, for example, a high voltage IC for driving a motor is used.

  In order to drive the MOSFET Q1 on the high side among the MOSFETs Q1 and Q2, a DC voltage source using the source of the MOSFET Q1 as a reference potential is used. Therefore, in the first embodiment, a bootstrap circuit including a resistor R1, a diode D3, and a capacitor C2 is used as a control power source for the MOSFET Q1. In this bootstrap circuit, when the low-side MOSFET Q2 is turned on, the capacitor C2 is charged by the DC voltage source E which is a control power source for the low-side MOSFET Q2. As described above, the charged capacitor C2 serves as a DC voltage source for driving the MOSFET Q1 on the high side. The detailed operation of the bootstrap circuit will be described later.

  Next, the operation of the DC power supply device according to the first embodiment will be described with reference to FIGS.

  As will be described later, the DC power supply device according to the first embodiment has three operation modes, that is, operation modes of full-wave rectification, partial switching, and high-speed switching. In these three operation modes, synchronous rectification control using MOSFETs Q1 and Q2 is used. The synchronous rectification control is a control means for reducing the conduction loss of the circuit by performing switching control of the MOSFETs Q1 and Q2 in accordance with the phase of the AC power supply voltage Vs.

  First, the synchronous rectification control in the first embodiment will be described.

  FIG. 2 shows a path of a current flowing through the main circuit of the DC power supply device when the polarity of the AC power supply voltage is positive (Vs> 0).

  As shown in FIG. 2, the main circuit current flows through a path (see an arrow in FIG. 2) that passes through the AC power source Vs, the reactor L1, the diode D1, the smoothing capacitor C1, the MOSFET Q2, and the AC power source Vs in order. At this time, the MOSFET Q1 is in an off state, and the MOSFET Q2 is in an on state. Here, the parasitic diode D21 of the MOSFET Q2 can also be a current path. However, since the MOSFET Q2 is in the on state, the main circuit current flows in the low-on-resistance region including the channel in the MOSFET Q2, and hardly flows through the parasitic diode D21. For this reason, the conduction loss of the main circuit part including the semiconductor element can be reduced.

  FIG. 3 shows a path of current flowing through the main circuit of the DC power supply device when the polarity of the AC power supply voltage is negative (Vs <0).

  As shown in FIG. 3, the main circuit current flows through a path (see an arrow in FIG. 3) that passes through the AC power source Vs, MOSFET Q1, smoothing capacitor C1, diode D2, reactor L1, and AC power source Vs in order. At this time, the MOSFET Q1 is in an on state, and the MOSFET Q2 is in an off state. Here, the parasitic diode D11 of the MOSFET Q1 can also be a current path. However, since the MOSFET Q1 is in the on state, the main circuit current flows in the low-on-resistance region including the channel in the MOSFET Q1, and hardly flows through the parasitic diode D11. For this reason, the conduction loss of the main circuit part including the semiconductor element can be reduced.

  Next, three operation modes of the DC power supply main circuit, that is, full-wave rectification, partial switching, and high-speed switching will be described.

  First, full-wave rectification will be described.

  In full-wave rectification, when the polarity of the AC power supply voltage is positive (Vs> 0), the MOSFET Q1 is always off, the MOSFET Q2 is always on, and the polarity of the AC power supply voltage is negative (Vs <0), MOSFET Q1 is always on, and MOSFET Q2 is always off. The main circuit current flows as shown in FIGS. 2 and 3 in the former and latter cases, respectively. That is, since the synchronous rectification control as described above is performed in the full-wave rectification, the loss of the main circuit of the DC power supply device can be reduced.

  The MOSFETs Q1 and Q2 are complementarily controlled to be turned on / off, but are switched between the on state and the off state at the zero cross timing determined by the zero cross determination unit 14. That is, the MOSFET Q1 is switched from the off state to the on state and the MOSFET Q2 is switched from the on state to the off state at the zero cross timing at which the polarity of the AC power supply voltage switches from positive to negative. In addition, at the zero cross timing when the polarity of the AC power supply voltage switches from negative to positive, the MOSFET Q1 is switched from the on state to the off state, and the MOSFET Q2 is switched from the off state to the on state.

  When the MOSFETs Q1 and Q2 are switched between the on state and the off state, depending on the switching characteristics of the MOSFETs Q1 and Q2, a period during which both the MOSFETs Q1 and Q2 are in the off state, that is, a dead time may be set. As a result, it is possible to prevent MOSFETs Q1 and Q2 from being simultaneously turned on and an excessive current flowing through MOSFETs Q1 and Q2.

  Next, partial switching will be described.

  In the first embodiment, when the power consumption of the load H increases, the DC voltage of the smoothing capacitor C1 is boosted. Also, since the current flowing through the main circuit has a distorted waveform with respect to the sine wave due to the characteristics of the reactor L1, the power output to the load H increases, and when the input current from the AC power source increases, the power factor decreases, For this reason, the harmonic current also increases. Therefore, in the first embodiment, the DC voltage is boosted and the power factor is improved by a partial switching operation in which complementary switching of the MOSFETs Q1 and Q2 is repeated a predetermined number of times in a half cycle of the AC power supply voltage.

  FIG. 4 shows a path of the power factor correction current flowing through the main circuit of the DC power supply device when the polarity of the AC power supply voltage is positive (Vs> 0). FIG. 5 shows the path of the power factor correction current flowing in the main circuit of the DC power supply device when the polarity of the AC power supply voltage is negative (Vs <0).

  As shown in FIG. 4, when the polarity of the AC power supply voltage is positive, the power factor improving current passes through the AC power supply Vs, the reactor L1, the diode D1, the MOSFET Q1, and the AC power supply Vs in this order (see arrows in FIG. 4). ) In the main circuit. At this time, MOSFET Q1 is in the on state and MOSFET Q2 is in the off state.

  Further, as shown in FIG. 5, when the polarity of the AC power supply voltage is negative (Vs <0), the power factor improving current passes through the AC power supply Vs, MOSFET Q2, diode D2, reactor L1, and AC power supply Vs in this order. (Refer to the arrow in FIG. 5), it flows in the main circuit. At this time, the MOSFET Q1 is in an off state, and the MOSFET Q2 is in an on state.

  When these power factor improving currents flow, the waveform of the input current from the AC power source Vs becomes close to a sine wave, so that the power factor on the AC power source side is improved.

  Further, when one of the MOSFET Q1 and the MOSFET Q2 is turned on and the other is turned off and the power factor improving current Isp flows through the main circuit, the reactor L1 has electric energy represented by the formula (1). Is accumulated.

W = (1/2) · L ₁ · I _sp ² (1)
Thereafter, when one is turned off and the other is turned on, the energy stored in the reactor L1 is released to the smoothing capacitor C1, thereby boosting the DC voltage.

  As described above, when Vs> 0, MOSFET Q1 operates as a boost chopper, and MOSFET Q2 operates as a synchronous rectifier. When Vs <0, the MOSFET Q2 operates as a step-up chopper, and the MOSFET Q1 operates as a synchronous rectifier.

  The partial switching operation will be described in more detail with reference to FIG.

  FIG. 6 shows the relationship between the drive pulses (gate control signals) of MOSFETs Q1 and Q2, the power supply voltage, and the main circuit current in the positive cycle of the AC power supply voltage Vs.

  The main circuit current can be brought close to an ideal current that is sinusoidal like the power supply voltage shown in FIG. 6 by the partial switching operation. The circuit operation in this case is as follows.

  At the point P1 on the ideal current, the slope of the ideal current is di (P1) / dt. Next, from the state where the main circuit current is zero, the MOSFET Q1 is given a delay from time 0, and then the slope of the main circuit current when it is turned on with the on width ton1_Q1 is di (ton1) / dt. Furthermore, after turning on with the time width ton1_Q1, the slope of the main circuit current when turning off for the time width toff1_Q1 at the time corresponding to the point P1 is di (toff1) / dt. At this time, the average value of di (ton1) / dt and di (toff1) / dt is made equal to the gradient di (P1) / dt of the ideal current at the point P1. This average value is, for example, an arithmetic average value.

  Next, similarly to the point P1, the slope of the ideal current at the point P2 is set to di (P2) / dt. The slope of the main circuit current when the MOSFET Q1 is turned off with the on width ton2_Q1 after the MOSFET Q1 is turned off for the time width toff1_Q1 is defined as di (ton2) / dt. Furthermore, after turning on with the time width ton2_Q1, the slope of the main circuit current when turning off for the time width toff2_Q1 at the time corresponding to the point P2 is di (toff2) / dt. As in the case of the point P1, the average value of di (ton2) / dt and di (toff2) / dt is made equal to the slope of the ideal current di (P2) / dt at the point P2. Thereafter, such an operation is repeated. At this time, the greater the number of switchings of the MOSFET Q1, the closer the waveform of the main circuit current is to a sine wave like an ideal current. That is, when the number of times of switching is increased, the power factor is improved and the harmonic current can be reduced.

  Further, as shown in FIG. 6, the MOSFET Q2 is turned on for a time width ton1_Q2, and thereafter is turned off for a time width toff1_Q2. At the timing when the MOSFET Q2 is turned off from on, the MOSFET Q1 is turned on during the time width ton1_Q1 as described above. Then, at the timing when the MOSFET Q1 is turned from on to off, the MOSFET Q2 is turned on during the time width ton2_Q2. Thereafter, the MOSFETs Q1 and Q2 are similarly turned on and off in a complementary manner. As a result, the DC power supply device according to the first embodiment performs the power factor correction operation by the switching control of the MOSFET Q1 and the boosting operation with the synchronous rectification operation by the MOSFET Q2 by the switching control of the MOSFETs Q1 and Q2. . Thereby, the efficiency of the DC power supply device can be improved.

  In the negative half cycle of the AC power supply voltage, as in the case of the positive half cycle described above, the main circuit current waveform is brought closer to a sine wave like an ideal current, the power factor is improved, and the harmonics are increased. Wave current can be reduced. The DC power supply device according to the first embodiment performs a power factor correction operation by switching control of the MOSFET Q2 (see FIG. 5), and a boosting operation accompanied by synchronous rectification operation by the MOSFET Q1 by switching control of the MOSFETs Q1 and Q2. To do. Thereby, the efficiency of the DC power supply device can be improved.

  Next, high-speed switching that is the third operation mode will be described.

  The direct-current power supply device according to the first embodiment performs a power factor correction operation and a boosting operation in the high-speed switching operation as in the partial switching operation. At this time, the MOSFETs Q1 and Q2 are complementarily controlled at a higher speed than the partial switching operation at a constant period, that is, at a predetermined frequency.

  When the polarity of the AC power supply voltage is positive (Vs> 0), during the power factor correction operation, the MOSFET Q1 is turned on and the MOSFET Q2 is turned off, so that a power factor correction current flows and electric energy is supplied to the reactor L1. Accumulate. Next, the boosting operation is performed by turning off the MOSFET Q1 and turning on the MOSFET Q2 to release the stored energy of the reactor L1 to the smoothing capacitor C1. Here, it is conceivable that the stored energy of the reactor L1 is discharged through the parasitic diode D21 of the MOSFET Q2 without turning on the MOSFET Q2. However, since the power loss of the main circuit increases due to the conduction loss of the parasitic diode D21, the power loss of the main circuit is reduced by performing the synchronous rectification operation by turning on the MOSFET Q2 in the first embodiment.

  When the polarity of the AC power supply voltage is a negative half cycle (Vs <0), by turning on MOSFET Q2 and turning off MOSFET Q1, a power factor correction current flows and electric energy is stored in reactor L1. To do. Next, the MOSFET Q2 is turned off, the MOSFET Q1 is turned on, and the stored energy of the reactor L1 is released to the smoothing capacitor C1, whereby the boosting operation is performed. Also in this case, if the stored energy of the reactor L1 is released through the parasitic diode D11, the power loss of the main circuit increases. Therefore, the power loss of the main circuit is reduced by performing the synchronous rectification operation by turning on the MOSFET Q1. The

  Hereinafter, specific means for switching control of MOSFETs Q1 and Q2 in the high-speed switching operation mode will be described.

  First, the instantaneous current is flowing through the DC power supply device can be expressed by Equation (2).

i _s = ^21/2 · V _s · sin (ωt) / (K _p · V _d ) (2)
Here, Vs is a power supply voltage effective value, Kp is a current control gain, and Vd is a DC voltage. Furthermore, Formula (3) is obtained by rewriting Formula (2).

K _p · i _s = 2 ^1/2 · V _s · sin (ωt) / V _d (3)
Further, the instantaneous current is is expressed by Expression (4), where Is is an effective current value.

i _s = ^21/2 · I _s · sin (ωt) (4)
Expression (5) is obtained from Expression (3) and Expression (4).

K _p = ^21/2 · V _s · sin (ωt) / (V _d × 2 ^1/2 · I _s · sin (ωt)) = (V _s / V _d ) / I _s (5)
In equation (5), (Vs / Vd) is the reciprocal of the step-up ratio (= (Vd / Vs)). Therefore, when the step-up ratio is a, equation (6) is obtained from equation (5).

K _p · I _s = 1 / a (6)
Furthermore, the MOSFET flow rate d (duty) can be expressed by equation (7).

d = 1−K _p · | i _s | = 1− | i _s | / (a · I _s ) (7)
As shown in the above equation (6), by controlling Kp · Is to a constant value, a DC voltage Vd boosted a times the power supply voltage effective value Vs is output. At that time, the conduction ratio d of the MOSFET performing the boost chopper operation is given by the above equation (7).

  FIG. 7a shows the time change of the conduction ratio (duty) of MOSFET Q1 and MOSFET Q2 in the positive half cycle of the AC power supply voltage, and FIG. 7b shows the time change of the power supply voltage instantaneous value vs and the circuit current instantaneous value is. Show. Moreover, FIG. 7c shows the time change of the duty ratio (duty) of MOSFET Q1 when the current phase delay is taken into consideration and when it is not taken into consideration. The frequency of the power supply voltage is 50 Hz, and the chopper frequency is 20 kHz.

  As shown in FIGS. 7a and 7b, the larger the instantaneous current is, the smaller the conduction ratio (duty) of the MOSFET Q1 that is switching-controlled for the power factor correction operation and the step-up chopper operation, and the smaller is is, The conduction ratio (duty) of MOSFET Q1 increases. The change in the duty ratio (duty) of the MOSFET Q2 that performs the synchronous rectification operation is opposite to the duty ratio (duty) of the MOSFET Q1. Therefore, the conduction ratio dQ1 of MOSFET Q1 and the conduction ratio dQ2 of MOSFET Q2 are expressed by Expression (8) and Expression (9), respectively.

dQ1 = 1−K _p · | i _s | (8)
dQ2 = 1-dQ1 (9)
FIG. 7a shows a case where the inductance of the reactor L1 is small and there is almost no phase delay of the current with respect to the power supply voltage. When the inductance of the reactor L1 is large and the current phase is delayed with respect to the voltage phase, the duty ratio (duty) may be set in consideration of the current phase as shown in FIG. 7c.

  Further, as shown in FIG. 7b, the power factor is improved because the waveform of the current is is controlled in a sine wave shape by high-speed switching.

  As described above, a boosted DC voltage can be obtained by high-speed switching while improving power factor and reducing harmonic current and improving efficiency by synchronous rectification.

  The three operation modes described above, full-wave rectification, partial switching, and high-speed switching, are selected according to the operating state of the device. At this time, you may control so that the control part 18 may switch an operation mode according to the driving | running state of an apparatus. For example, full-wave rectification with a light load that does not boost DC voltage, partial switching when the load increases and boosts and improves the power factor, and fast switching when the load increases and harmonic components increase The operation mode is switched according to the size of the load so as to perform switching.

  In addition, as an index for judging the magnitude of the load, when the load is a motor or an inverter, etc. And temperature information such as the temperature of the device.

  Next, the operation of the bootstrap circuit serving as the control power supply for the high-side MOSFET Q1 in the DC power supply device of the first embodiment will be described.

  FIG. 8 shows the path of the charging current of the capacitor C2 in the bootstrap circuit. When the capacitor C2 is charged, the diode D3 becomes conductive when the connection point P2 becomes a low level, and a charging current flows through a path indicated by a broken line in FIG. That is, the capacitor C2 is charged when the MOSFET Q2 is turned on during the period when the polarity of the AC power supply voltage is positive (Vs> 0) and during the period when the polarity of the AC power supply voltage is negative (Vs <0). That is, the capacitor C2 is charged when the MOSFET Q2 performs a synchronous rectification operation and a boost chopper operation (with a power factor correction operation).

  First, the circuit operation when the polarity of the AC power supply voltage is positive (Vs> 0) will be described with reference to FIG.

  FIG. 9 shows the main circuit current path (solid arrow) and the charging current in the bootstrap circuit when the capacitor C2 of the bootstrap circuit is charged when the polarity of the AC power supply voltage is positive (Vs> 0). The route (broken arrow) is shown.

  Before the AC power supply voltage Vs is input to the DC power supply device, the smoothing capacitor C1 is not charged. When the AC power supply voltage Vs (> 0) is input in this state, the main circuit passes through the AC circuit Vs, the reactor L1, the diode D1, the smoothing capacitor C1, the parasitic diode D21, and the AC power supply Vs in order. Current flows. As a result, the smoothing capacitor C1 is charged, and the smoothing capacitor C1 is fully charged. The smoothing capacitor C1 generates a DC voltage obtained by rectifying the AC power supply voltage Vs (when Vs = 200V, the DC voltage is 240V). At this time, since the connection point P2 becomes a low level, when the MOSFET Q2 is turned on for the synchronous rectification operation, the diode D3 becomes conductive, and the DC voltage source E, the resistor R1, the diode D3, the MOSFET Q2, and the DC voltage source. A current flows through the bootstrap circuit along a route passing through E in order. As a result, the capacitor C2 is charged. That is, the capacitor C2 is charged by the DC voltage source E serving as a control power source for the drive circuit K2 of the low-side MOSFET Q2. If the MOSFET Q2 is turned off after the capacitor C2 is charged, the charged capacitor C2 becomes a substantially floating DC voltage source, so that it is used as a control power source for the drive circuit K1 of the high-side MOSFET Q1. Can do.

  Next, circuit operation when the polarity of the AC power supply voltage is negative (Vs <0) will be described with reference to FIGS.

  FIG. 10 shows the main circuit current path (solid arrow) and the charging current in the bootstrap circuit when the capacitor C2 of the bootstrap circuit is charged when the polarity of the AC power supply voltage is negative (Vs <0). The route (broken arrow) is shown.

  As shown in FIG. 10, when the MOSFET Q2 is turned on, the connection point P2 becomes a low level, so that the diode D3 conducts and passes through the DC voltage source E, the resistor R1, the diode D3, the MOSFET Q2 and the DC voltage source E in this order. A current flows through the bootstrap circuit along the path. As a result, the capacitor C2 is charged. That is, the capacitor C2 is charged by the DC voltage source E serving as a control power source for the drive circuit K2 of the low-side MOSFET Q2. If the MOSFET Q2 is turned off after the capacitor C2 is charged, the charged capacitor C2 becomes a substantially floating DC voltage source, so that it is used as a control power source for the drive circuit K1 of the high-side MOSFET Q1. Can do.

  Here, when the polarity of the AC power supply voltage is negative (Vs <0) and the MOSFET Q2 is turned on to perform the boost chopper operation, as shown in FIG. 10, the main circuit includes the AC power supply Vs, the MOSFET Q2, The main circuit current flows through a path (solid arrow in FIG. 10) that passes through the diode D2, the reactor L1, and the AC power supply Vs in order. This main circuit current is a short-circuit current that flows when the AC power supply is short-circuited by turning on the MOSFET Q1. Therefore, in the following description, such a main circuit current is described as a short-circuit current.

  11 and 12 show voltage waveforms of the main circuit unit and the AC power supply of the DC power supply device according to the first embodiment shown in FIG. 11 and 12, the main circuit current (short-circuit current) path (solid arrow) and the charging current path (bootstrap circuit) (when the polarity of the AC power supply voltage is negative (Vs <0)) (Broken arrows).

  The magnitude of the short-circuit current varies depending on the timing at which the MOSFET Q2 is switched. As shown in FIG. 11, the short-circuit current increases at the timing near the AC power supply voltage peak value. In addition, as shown in FIG. 12, the short-circuit current decreases at the timing near the zero cross point of the AC power supply voltage.

  In order to prevent an excessive short-circuit current, it is preferable to switch the MOSFET Q2 when the AC power supply voltage is small, or to reduce the duty ratio of the MOSET Q2.

  Next, the charging / discharging operation of the capacitor C2 of the bootstrap circuit in the first embodiment will be described with reference to FIGS.

  FIG. 13a shows the gate drive circuit portion of the MOSFETs Q1 and Q2 in the first embodiment. Also in FIG. 13a, the charging current path (broken arrow in the figure) in the bootstrap circuit is shown as in FIGS. FIG. 13b shows the circuit portion that operates during charging. Further, FIGS. 13c and 13d show circuit portions that operate during discharge. In FIGS. 13c and 13d, the drive circuits K1 and K2 are omitted for simplicity.

  Hereinafter, a case where the operation mode is full-wave rectification will be described. The basic charge / discharge operation is the same for the other operation modes.

When the polarity of the AC power supply voltage is positive (Vs> 0), the MOSFET Q2 is turned on for synchronous rectification. This charges the capacitor C2 of the bootstrap circuit as described above and as shown in FIG. 13a. At this time, in FIG. 13b, and the resistance value of the resistor R1 and _{R 1,} the capacitance of the capacitor C2 and _{C 2,} When E the voltage of the DC voltage source E, the voltage _V c2 of the capacitor C2 (t) is Equation (10 ).

V _c2 (t) = E × (1−exp (−t / (C ₂ · R ₁ ))) (10)
As shown in Expression (10), the time constant when the capacitor C2 is charged is C ₂ · R ₁ . For example, when the DC voltage E = 15 V, C ₂ = 33 μF, and R ₁ = 22Ω, the time constant is 726 μs, and if the time is charged for about 5.32 ms, the battery is almost fully charged (Vc2 = about 14.99 V).

  Next, the discharge of the capacitor C2 will be described.

The charged capacitor C2 is discharged when the MOSFET Q1 is turned on for synchronous rectification when the polarity of the AC power supply voltage is negative (Vs <0). At this time, since the charge is charged in the gate capacitance of the MOSFET Q1, a discharge current flows mainly through the path (solid arrow in the figure) shown in FIG. 13c. In this case, in FIG. 13c, assuming that the gate capacitance of the MOSFET Q1 is C _{g (Q1)} , the resistance value of the gate resistance Rg11 is R _g11, and the voltage of the capacitor C2 is V _c2 (t), the gate voltage V of the MOSFET Q1 _{g (Q1)} is represented by Formula (11).

V _{g (Q1)} = V _c2 (t) × (1−exp (−t / (C _{g (Q1)} · R _g11 ))) (11)
As shown in the equation (11), the time constant at the time of gate charging of the MOSFET Q1 is C _{g (Q1)} · R _g11 . When the gate capacitance of the MOSFET Q1 is charged from the fully charged state of the capacitor C2, for example, when V _c2 (t) = 15 V, C _{g (Q1)} = 1000 pF, R _g11 = 400Ω, the time constant is 40 μs, and the capacitor About 3 μs after the start of C2 discharge, the charging of the gate capacitance is completed (V _{g (Q1)} = 15 V).

In addition, the voltage V _c2 (t) of the capacitor C2 at this time is expressed by Expression (12).

V _c2 (t) = E × exp (−t / (C ₂ · R _g11 )) (12)
As shown in Expression (12), the discharge time constant of the capacitor C2 is C ₂ · R _g11 . For example, when C ₂ = ₃₃ μF, R _g11 = 400Ω, and E = 15 V, the time constant is about 13.2 ms, and after about 3 μs after the charging of the gate capacitance of the MOSFET Q1, the voltage V _c2 of the capacitor C2 is It is approximately 15V.

Furthermore, after the charge is charged in the gate capacitance of MOSFET Q1, a discharge current flows through the path shown in FIG. 13d (solid arrow in the figure), and capacitor C2 is discharged. At this time, the resistance value of the gate resistor Rg12 and _{R g12,} when the voltage of the capacitor C2 when the charge on the gate capacitance of the MOSFET Q1 is fully charged and _{V c2,} the voltage _V c2 of the capacitor C2 '(t) is It is represented by Formula (13).

V _c2 ′ (t) = V _c2 × exp (−t / (C ₂ · (R _g11 + R _g12 )) (13)
As shown in Equation (13), the discharge time constant of the capacitor C2 is C ₂ · (R _g11 + R _g12 ). For _{example, C 2 = 33μF, R g11} = 400Ω, R g12 = 20kΩ, when V c2 = 15V, when the constant = 673.2ms. If the frequency of the AC power supply voltage is 50 Hz, the voltage of the capacitor C2 after 10 ms, which is a half cycle of the power supply voltage, is about 14.76V, which is a voltage drop of about 0.24V.

Further, for example, when the time constant is reduced with the resistors R _g11 = 100Ω and R _g12 = 10 kΩ, the full charge of the MOSFET Q1 becomes faster after about 0.78 μs, but the voltage of the capacitor C2 after 10 ms is about 11.16V. Thus, the voltage drop is about 3.84V. That is, the smaller the time constant during discharge, the greater the voltage drop across the capacitor C2 due to switching of the MOSFET Q1.

  When the operation mode is full-wave rectification, the MOSFET Q1 is switched only once within a half cycle period of the AC power supply voltage, so the voltage drop is about 0.24V in the above example. However, when the operation mode is partial switching and high-speed switching, switching of the MOSFET Q1 is performed a plurality of times, so that the gate capacitance of the MOSFET Q1 is charged and discharged a plurality of times. That is, the capacitor C2 is discharged a plurality of times, and the voltage drop of the capacitor C2 due to the discharge increases. For example, when the operation mode is high-speed switching with a switching frequency of 20 kHz (period 50 μs), the MOSFET Q1 performs switching 200 times within a half cycle period of the AC power supply voltage, that is, 10 ms. At this time, assuming that the duty ratio (duty) is 50% (ON time 25 μs), the voltage drop of the capacitor C2 becomes about 0.01 V by one charge / discharge of the MOSFET Q1, and the voltage drop is reduced by 200 times of switching. 2V.

  If the time constant is further reduced, the voltage drop is further increased. Therefore, the stability of the switching control cannot be ensured, and the controllability is lowered. If the voltage of the capacitor C2 is equal to or lower than the gate threshold voltage of the MOSFET Q1, the MOSFET Q1 cannot be switched.

For this reason, the voltage of the capacitor C2 is detected so that the voltage of the capacitor C2 does not fall below the predetermined value from the voltage value E, and when the voltage falls below a certain threshold value, the MOSFET Q1 is turned off for a predetermined period and the MOSFET Q2 is turned off. It is preferable to charge the capacitor C2 by turning it on. At this time, in order to suppress a short-circuit current flowing in the main circuit, it is preferable to reduce the current conduction rate (duty) of MOSFET Q2. However, as described above, since the charging time constant of the capacitor C2 is dependent on the resistance value R ₁ of the resistor R1 of the bootstrap circuit, the resistance value R ₁ is greater, it may not be performed promptly charged.

  Therefore, in the DC power supply device of the first embodiment, any one of charging time constant switching means for the capacitor C2 of the bootstrap circuit as shown in FIGS. 14a and 14b is applied. According to these means, when the capacitor C2 is charged by the bootstrap circuit, the time constant at the time of charging the capacitor C2 is switched to a value smaller than usual, so that the capacitor C2 can be quickly returned even if the duty ratio is reduced. Can be fully charged.

  In the means shown in FIG. 14a, the MOSFET Q3 is connected in parallel to the resistor R1, and the charging time constant is switched by switching the MOSFET Q3 in synchronization with the switching of the MOSFET Q2. The voltage of the capacitor C <b> 2 is divided by the resistors Ra and Rb, and the divided voltage is input to the microcomputer constituting the control unit 18. The microcomputer monitors the voltage of the capacitor C2 based on the divided voltage. When the control unit 18 determines that the voltage of the capacitor C2 is equal to or lower than a predetermined threshold value, the control unit 18 transmits a command to turn on / off the MOSFET Q3 to a gate drive circuit of the MOSFET Q3 (not shown). As a result, when MOSFET Q3 is turned on, the charging current flows through a path (broken arrow in the figure) that passes through DC voltage source E, MOSFET Q3, diode D3, capacitor C2, MOSFET Q2, and DC voltage source E in this order. The time constant during charging is switched to a value smaller than the value set by the resistor R1.

  For example, when the DC voltage source E is 15V, the threshold voltage of the capacitor C2 is 14V.

  In the means shown in FIG. 14b, an NTC (Negative Temperature Coefficient) thermistor TH1 having a negative temperature characteristic is connected in parallel to the resistor R1. When the temperature of the NTC thermistor TH1 rises due to the heat generated by the current and the resistance value of the NTC thermistor TH1 falls, the charging current becomes DC voltage source E, NTC thermistor TH, diode D3, capacitor C2, MOSFET Q2, and DC voltage source. Since the route also passes through the route E (broken arrow in the figure), the time constant during charging is smaller than the value set by the resistor R1. This means is suitable when the MOSFET Q2 is switched a plurality of times, such as partial switching or high-speed switching, and the amount of heat generated by the NTC thermistor TH is increased.

  As described above, according to the charging time constant switching means shown in FIGS. 14a and 14b, even when the voltage of the capacitor C2 becomes equal to or lower than a predetermined value, the voltage drop of the capacitor C2 is reduced by charging with the MOSFET Q2. This can prevent the DC power supply apparatus from performing stable synchronous rectification operation. Therefore, controllability of the DC power supply device is improved, and efficiency and reliability are improved.

  As described above, according to the first embodiment, the bootstrap circuit serving as the control power source of the first MOSFET is provided, and the capacitor of the bootstrap circuit includes the full-wave rectification operation mode, the partial switching operation mode, and the high-speed switching operation mode. In this case, when the second MOSFET is turned on, the driving voltage of the first MOSFET, which is the high-side element, can be secured, and the synchronous rectification control can be stably performed. Thereby, the efficiency of the DC power supply device is improved and the reliability is improved.

  Next, an air conditioner that is Embodiment 2 of the present invention will be described.

  The air conditioner of the second embodiment includes an electric compressor in which the compressor is driven by a three-phase AC motor, an inverter circuit that supplies a three-phase AC voltage (electric power) to the electric compressor, and the inverter circuit as a load. The DC power supply device according to the first embodiment is provided as the DC power supply device.

  Hereinafter, the air conditioner of the fifth embodiment will be further described with reference to FIGS. 15 and 16.

  FIG. 15 is a configuration diagram of an air-conditioning cycle of the air conditioner according to the second embodiment. The air conditioner includes an indoor heat exchanger 51, an outdoor heat exchanger 52, a compressor 53, an expansion valve 54, a four-way valve 55, an indoor blower fan 56, and an outdoor blower fan 57. The compressor 53, the outdoor heat exchanger 52, the outdoor blower fan (propeller fan) 57, and the expansion valve 54 are arranged in the outdoor unit (see FIG. 16), and the indoor heat exchanger 51 and the indoor blower fan 56 are indoor units (not shown). )).

  During the cooling operation, the high-temperature and high-pressure refrigerant discharged from the compressor 53 flows into the outdoor heat exchanger 52 through the four-way valve 55. The refrigerant flowing into the outdoor heat exchanger 52 is condensed and becomes liquid refrigerant by exchanging heat with outdoor air sent by the outdoor blower fan 57. The liquid refrigerant passes through the expansion valve 54 to become a low-temperature and low-pressure two-phase refrigerant and flows into the indoor heat exchanger 51. The low-temperature and low-pressure two-phase refrigerant that has flowed into the indoor heat exchanger 51 exchanges heat with the indoor air sent by the indoor blower fan 56. At this time, the indoor air sent to the indoor heat exchanger 51 is cooled by the low-temperature and low-pressure two-phase refrigerant that has flowed into the indoor heat exchanger 51, and is discharged into the room from a blower outlet (not shown). Since the air discharged into the room from the air outlet (not shown) is lower than the temperature of the air at the air inlet (not shown), the room temperature can be lowered. The refrigerant heat-exchanged by the indoor heat exchanger 51 returns to the compressor 53 again via the four-way valve 55.

  During the heating operation, the high-temperature and high-pressure refrigerant discharged from the compressor 53 flows into the indoor heat exchanger 51 through the four-way valve 55. Then, it passes through the four-way valve 55 and the outdoor heat exchanger 52 and returns to the compressor 53 via the four-way valve 55.

  FIG. 16 is an external view of an outdoor unit of an air conditioner according to the second embodiment. The space in the outdoor unit is divided into a compressor chamber A in which the compressor 53 is installed and a blower chamber B in which the outdoor blower fan 57 is installed, with the partition plate 70 interposed therebetween. An electrical component 79 such as a power supply device that drives the compressor 53 is accommodated in the electrical box and is located between the compressor 53 and the upper lid 80 and across the compressor chamber A and the blower chamber B. is set up. Further, the electrical component 79 is located above the partition plate 70 and is supported by the partition plate 70.

  The outdoor air is sucked from the rear side of the outdoor unit by the outdoor blower fan 57, passes through the outdoor heat exchanger 52, and then blown out from the front side (front panel 81) of the outdoor unit.

  The compressor 53, that is, the three-phase AC motor included in the electric compressor, is supplied with variable voltage and variable-frequency three-phase AC voltage (power) by the inverter circuit serving as a load of the DC power supply device according to the first embodiment. , The compressor 53 is driven by the three-phase AC motor. Thereby, the air conditioner performs cooling and heating operations as described above.

  According to the air conditioner of the second embodiment, the synchronous rectification control can be stably performed in the DC power supply device, and thus the air conditioner can be stably operated. Therefore, the energy efficiency (APF) of the air conditioner is improved and the reliability of the air conditioner is improved.

  In addition, this invention is not limited to the Example mentioned above, Various modifications are included. For example, the above-described embodiments have been described in detail for easy understanding of the present invention, and are not necessarily limited to those having all the configurations described. Further, it is possible to add, delete, and replace other configurations for a part of the configuration of each embodiment.

  For example, a super junction MOSFET or a SiC-MOSFET may be used as the MOSFET Q1 and the MOSFET Q2. Thereby, the efficiency of the DC power supply device is improved. Further, instead of MOSFET Q3 in FIG. 14a, other switching elements such as bipolar transistors may be used.

  Moreover, even if the direct-current power supply device of the present invention is mounted on equipment other than the air conditioner, efficiency and reliability are improved.

DESCRIPTION OF SYMBOLS 11 Current detection part 12 Current control gain control part 13 AC voltage detection part 14 Zero cross determination part 15 Load detection part 16 Boost ratio control part 17 DC voltage detection part 18 Control part 51 ... Indoor heat exchanger 52 ... Outdoor heat exchanger 53 ... Compressor 54 ... Expansion valve 55 ... Four-way valve 56 ... Indoor fan 57 ... Outdoor fan 70 ... Partition plate 79 ... Electrical component 80 ... Upper lid 81 ... Front panel Vs AC power supply C1 Smoothing capacitor D1, D2 Diodes ha, hb, hc , Hd wiring L1 reactor Q1, Q2 MOSFET
C2 Capacitor D3 Diode R1 Resistance Rg11, Rg12, R21, Rg22 Gate resistance E DC voltage source K1, K2 Drive circuit Q3 MOSFET
TH1 NTC thermistor

Claims (11)

The anode of the first diode and the cathode of the second diode are connected,
The source of the first MOSFET and the drain of the second MOSFET are connected,
The cathode of the first diode and the drain of the first MOSFET are connected to the positive side of the smoothing capacitor;
The anode of the second diode and the source of the second MOSFET are connected to the negative side of the smoothing capacitor;
An AC power supply is connected to a connection point between the first diode and the second diode and a connection point between the first MOSFET and the second MOSFET via a reactor.
In the DC power supply device that converts the AC voltage of the AC power supply into a DC voltage by switching the first MOSFET and the second MOSFET,
A full-wave rectification operation mode in which the first MOSFET and the second MOSFET are switched to perform a synchronous rectification operation according to the polarity of the AC voltage;
A partial switching operation mode in which the first MOSFET and the second MOSFET are complementarily switched on and off a predetermined number of times in a half cycle of the AC voltage;
A high-speed switching operation mode in which the first MOSFET and the second MOSFET are switched on and off complementarily at a predetermined frequency;
Have
Furthermore, a bootstrap circuit serving as a control power source for the first MOSFET is provided,
The DC power supply device, wherein a capacitor of the bootstrap circuit is charged when the second MOSFET is switched on. The DC power supply device according to claim 1,
In the partial switching operation mode and the high-speed switching mode, one and the other of the first MOSFET and the second MOSFET are switched so as to perform a step-up chopper operation and a synchronous rectification operation, respectively. Power supply. In the DC power supply device according to claim 2,
The DC power supply device, wherein the capacitor of the bootstrap circuit is charged when the second MOSFET performs the step-up chopper operation or the synchronous rectification operation. In the DC power supply device according to claim 3,
The DC power supply device according to claim 1, wherein the first MOSFET and the second MOSFET are switched for the step-up chopper operation near a zero cross point of the AC voltage. In the DC power supply device according to claim 2,
A DC power supply device, wherein a power factor correction current is caused to flow along with the step-up chopper operation. In the DC power supply device according to claim 2,
In the case of the high-speed switching mode, the conduction ratio of the one of the first MOSFET and the second MOSFET is increased as the main circuit current is increased, and the other is decreased as the main circuit current is increased. A direct current power supply device. In the DC power supply device according to claim 3,
A DC power supply device comprising time constant switching means for reducing a charging time constant of the bootstrap circuit. In the DC power supply device according to claim 7,
The DC power supply characterized in that the time constant switching means is a switching element connected in parallel with a resistor of the bootstrap circuit, and the switching element is turned on in accordance with a voltage of a capacitor of the bootstrap circuit. apparatus. In the DC power supply device according to claim 7,
The DC power supply device characterized in that the time constant switching means is an NTC thermistor connected in parallel to the resistance of the bootstrap circuit. The DC power supply device according to claim 1,
One of the full-wave rectification operation mode, the partial switching operation mode, and the high-speed switching operation mode is set according to the size of the load. An electric compressor whose compressor is driven by an AC motor;
An inverter circuit for supplying AC power to the electric compressor;
A DC power supply device having the inverter circuit as a load;
With
In the air conditioner in which the refrigerant compressed by the electric compressor circulates in an air conditioning cycle,
An air conditioner, wherein the DC power supply device is the DC power supply device according to claim 1.

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