JP7162747B2 - DC power supplies, motor drives, blowers, compressors and air conditioners - Google Patents

Description

The present invention provides a DC power supply device that converts an AC voltage output from an AC power supply into a DC voltage and applies it to a load, a motor drive device that drives a motor that is the load, a blower and a compressor that include the motor drive device, and , air conditioners with blowers or compressors.

Patent Document 1 below describes a connection point between a first diode and a second diode, and a connection between a first metal oxide semiconductor field effect transistor (MOSFET) and a second MOSFET. A DC power supply is disclosed in which an AC power supply is connected to points and via a reactor, and the AC voltage of the AC power supply is converted into a DC voltage by switching a first MOSFET and a second MOSFET. The first diode and the first MOSFET are elements connected to the positive side of the smoothing capacitor, and the second diode and the second MOSFET are elements connected to the negative side of the smoothing capacitor.

In the DC power supply device described in Patent Document 1, the first MOSFET is turned on at the timing when the current flows through the parasitic diode of the first MOSFET, and the second MOSFET is turned on at the timing when the current flows through the parasitic diode of the second MOSFET. turn on. This technique is called synchronous rectification. Synchronous rectification controls the DC power supply with high efficiency.

JP 2016-220378 A

In the synchronous rectification described in Patent Literature 1, each MOSFET is always turned on at the timing when the current flows through the parasitic diodes of the first and second MOSFETs. Here, the high efficiency of the synchronous rectification operation is based on the premise that the voltage drop of the parasitic diode connected in parallel is higher than the ON voltage of the MOSFET. However, depending on the operating state of the MOSFET, the on-voltage of the MOSFET may be higher than the voltage drop of the parasitic diode connected in parallel. Therefore, it is difficult to say that the synchronous rectification described in Patent Document 1 is optimized for synchronous rectification. Therefore, the technique of Patent Document 1 has room for improvement in terms of efficiency when applying synchronous rectification.

SUMMARY OF THE INVENTION It is an object of the present invention to provide a direct-current power supply device capable of further improving efficiency when applying synchronous rectification.

In order to solve the above-described problems and achieve the object, a DC power supply device according to the present invention includes a reactor and four unidirectional elements that are bridge-connected, and is connected to an AC power supply via the reactor. a converter that converts a power supply voltage, which is an AC voltage output from the converter, into a DC voltage and applies it to a load; and a smoothing capacitor that is connected between output terminals of the converter. Further, the DC power supply includes a first physical quantity detection unit that detects a first physical quantity representing an operating state on the output side of the converter, and a second physical quantity detecting unit that detects a second physical quantity representing the operating state on the input side of the converter. A physical quantity detection unit, and a control unit to which the first and second physical quantities are input and which controls the operation of the converter. Two of the four unidirectional elements in the converter are connected in series to form a first leg, and the remaining two unidirectional elements are connected in series to form a second leg. and at least two unidirectional elements in the first and second legs connected to the positive side of the smoothing capacitor, or two unidirectional elements in the first and second legs connected to the negative side of the smoothing capacitor A switching element is connected in parallel to each of the unidirectional element or the two unidirectional elements in the first leg or the two unidirectional elements in the second leg. The control unit further has a plurality of operation modes for operating the converter in different operation modes by controlling the conduction of the switching elements according to the ambient temperature of the switching elements, the outside air temperature of the converter, or an operation command to the converter.

Advantageous Effects of Invention According to the DC power supply device of the present invention, it is possible to achieve further efficiency improvement when applying synchronous rectification.

1 is a diagram showing a configuration example of a motor drive device including a DC power supply device according to Embodiment 1; FIG. Schematic cross-sectional view showing a schematic structure of a MOSFET used in the converter of Embodiment 1. FIG. A first diagram showing paths of currents flowing through the converter in the first embodiment A second diagram showing paths of currents flowing through the converter in the first embodiment A third diagram showing paths of currents flowing through the converter in the first embodiment A fourth diagram showing paths of currents flowing through the converter in the first embodiment A diagram for explaining features of operation modes according to the first embodiment. A diagram showing operation waveforms when operated in the operation mode shown in FIG. FIG. 4 shows loss characteristics of MOSFETs used in the motor drive device according to the first embodiment; FIG. 4 is a diagram showing timings at which a control unit turns on switching elements in the DC power supply device according to Embodiment 1; Flowchart used to explain the operation of the main part in Embodiment 1 A diagram showing the temperature characteristics of the forward current of a parasitic diode in a general MOSFET A diagram showing the temperature characteristics of the on-resistance of a general MOSFET A diagram showing how the cross points shown in FIG. 9 vary with temperature. FIG. 2 is a block diagram showing an example of a hardware configuration that implements the functions of the control unit according to Embodiment 1; FIG. 4 is a block diagram showing another example of the hardware configuration that implements the functions of the control unit according to Embodiment 1; A diagram showing a configuration example of a gate driver circuit in Embodiment 2 A diagram showing operation waveforms in the second embodiment corresponding to FIG. A diagram showing the configuration of an air conditioner according to Embodiment 3 A diagram showing features of operation modes in Embodiment 3

A DC power supply device, a motor drive device, a blower, a compressor, and an air conditioner according to embodiments of the present invention will be described below with reference to the accompanying drawings. It should be noted that the present invention is not limited by the embodiments shown below. Also, hereinafter, the electrical connection will be simply referred to as "connection".

Embodiment 1.
FIG. 1 is a diagram showing a configuration example of a motor drive device 100 including a DC power supply device 50 according to Embodiment 1. As shown in FIG. A DC power supply device 50 according to Embodiment 1 is a power supply device that converts a power supply voltage, which is an AC voltage output from a single-phase AC power supply 1 , into a DC voltage and applies the DC voltage to a load 12 . Further, the motor drive device 100 according to the first embodiment is a drive device that converts the DC power output from the DC power supply device 50 into AC power and supplies the converted AC power to the motor 500 to drive the motor 500. be.

A motor drive device 100 according to Embodiment 1 includes, as main components, a DC power supply device 50, a control unit 10, and a load 12, as shown in FIG.

The DC power supply device 50 includes a reactor 2, a converter 3, a gate drive circuit 15 which is a first drive circuit, a smoothing capacitor 4, a voltage detection unit 5, a current detection unit 6, a voltage detection unit 7, and a power supply circuit 14 that is a control power supply. One end of reactor 2 is connected to AC power supply 1 , and the other end of reactor 2 is connected to converter 3 . Reactor 2 temporarily stores power supplied from AC power supply 1 . The converter 3 converts the AC voltage output from the AC power supply 1 into a DC voltage and outputs the DC voltage to the DC buses 16a and 16b. DC buses 16 a and 16 b are electrical wiring that connects converter 3 and load 12 . The voltage between the DC bus 16a and the DC bus 16b is called "bus voltage".

The load 12 includes a gate drive circuit 17 as a second drive circuit, an inverter 18 , a current detector 9 and a motor 500 . Of the components of the load 12 , the gate drive circuit 17 , the inverter 18 and the current detector 9 are components of the motor drive device 100 except for the motor 500 . Inverter 18 converts the DC voltage output from DC power supply 50 into AC voltage to be applied to motor 500 and outputs the AC voltage. Examples of equipment in which the motor 500 is mounted are blowers, compressors, or air conditioners.

Although FIG. 1 shows an example in which the device connected to the inverter 18 is the motor 500, the device is not limited to this. A device connected to the inverter 18 may be a device to which AC power is input, and may be a device other than the motor 500 .

Converter 3 comprises a first leg 31 and a second leg 32 . The first leg 31 and the second leg 32 are connected in parallel. In the first leg 31, a first upper arm element 311 and a first lower arm element 312 are connected in series. In the second leg 32, a second upper arm element 321 and a second lower arm element 322 are connected in series. The other end of the reactor 2 is connected to a connection point 3 a between the first upper arm element 311 and the first lower arm element 312 on the first leg 31 . A connection point 3 b between the second upper arm element 321 and the second lower arm element 322 is connected to the other end of the AC power supply 1 . In converter 3, connection points 3a and 3b form AC terminals.

In FIG. 1, the reactor 2 is connected between one end of the AC power supply 1 and the connection point 3a, but is connected between another end of the AC power supply 1 and the connection point 3b. good too.

In the converter 3, the side on which the connection points 3a and 3b are located may be called the "AC side", the AC voltage output from the AC power supply 1 may be called the "power supply voltage", and the cycle of the power supply voltage may be called the "power cycle". be.

First upper arm element 311 includes switching element Q1 and diode D1 connected in parallel with switching element Q1. First lower arm element 312 includes switching element Q2 and diode D2 connected in parallel with switching element Q2. Second upper arm element 321 includes switching element Q3 and diode D3 connected in parallel to switching element Q3. The second lower arm element 322 includes a switching element Q4 and a diode D4 connected in parallel with the switching element Q4.

The diodes D1 and D4 are unidirectional diodes arranged so that forward current flows when the polarity of the power supply voltage is positive, that is, when the side connected to the reactor 2 has a higher potential than the side not connected to the reactor 2. element. The diodes D2 and D3 are unidirectional diodes arranged so that forward current flows when the polarity of the power supply voltage is negative, that is, when the side not connected to the reactor 2 has a higher potential than the side connected to the reactor 2. element.

Although FIG. 1 discloses a configuration in which switching elements Q1, Q2, Q3, and Q4 are connected in parallel to diodes D1, D2, D3, and D4, respectively, the configuration is not limited to this. A switching element may be connected to each of the two diodes connected to the positive side of the smoothing capacitor 4, that is, the diode D1 in the first leg 31 and the diode D3 in the second leg 32. Alternatively, a switching element may be connected to each of the two diodes connected to the negative side of the smoothing capacitor 4, that is, the diode D2 in the first leg 31 and the diode D4 in the second leg 32. Alternatively, a switching element may be connected to each of the two diodes in the first leg 31, namely the diodes D1 and D2. Alternatively, a switching element may be connected to each of the two diodes in the second leg 32, namely diodes D3 and D4.

Also, in FIG. 1, MOSFETs are illustrated as the switching elements Q1, Q2, Q3, and Q4, respectively, but they are not limited to MOSFETs. A MOSFET is a switching element that allows current to flow bidirectionally between a drain and a source. Any switching element may be used as long as it is a switching element that allows current to flow bidirectionally between a first terminal corresponding to the drain and a second terminal corresponding to the source, that is, a bidirectional element.

The term "parallel" used herein means that the first terminal corresponding to the drain of the MOSFET is connected to the cathode of the diode, and the second terminal corresponding to the source of the MOSFET is connected to the anode of the diode. do. A parasitic diode that the MOSFET itself has inside may be used as the diode. A parasitic diode is also called a body diode.

Further, the switching elements Q1, Q2, Q3, Q4 are not limited to MOSFETs made of a silicon-based material, and may be made of silicon carbide (SiC), gallium nitride (GaN), gallium oxide (Ga ₂ O ₃ ), or diamond. A MOSFET made of a wide band gap (WBG) semiconductor may also be used.

In general, WBG semiconductors have higher withstand voltage and heat resistance than silicon semiconductors. Therefore, by using a WBG semiconductor for at least one of the switching elements Q1, Q2, Q3, and Q4, the voltage resistance and allowable current density of the switching elements are increased, and the semiconductor module incorporating the switching elements is miniaturized. can.

Also, the switching elements Q1, Q2, Q3, and Q4 may use MOSFETs with a super junction (SJ) structure instead of the WBG semiconductors. By using the SJ-MOSFET, it is possible to take advantage of the low on-resistance which is an advantage of the SJ-MOSFET, while suppressing the disadvantage of the WBG semiconductor that the electrostatic capacity is high and recovery is likely to occur.

Returning to the description of FIG. The positive side of the smoothing capacitor 4 is connected to the DC bus 16a on the high potential side. The DC bus 16 a is drawn out from a connection point 3 c between the first upper arm element 311 on the first leg 31 and the second upper arm element 321 on the second leg 32 . The negative side of the smoothing capacitor 4 is connected to the DC bus 16b on the low potential side. The DC bus 16b is drawn out from a connection point 3d between the first lower arm element 312 on the first leg 31 and the second lower arm element 322 on the second leg 32 . In converter 3, connection points 3c and 3d form DC terminals. In addition, in the converter 3, the side on which the connection points 3c and 3d are located may be referred to as the "direct current side".

The output voltage of converter 3 is applied across smoothing capacitor 4 . The smoothing capacitor 4 is connected to DC buses 16a and 16b. Smoothing capacitor 4 smoothes the output voltage of converter 3 . The voltage smoothed by smoothing capacitor 4 is applied to inverter 18 .

The voltage detection unit 5 detects the power supply voltage and outputs the detected value Vs of the power supply voltage to the control unit 10 . The power supply voltage is the absolute value of the instantaneous voltage of the AC power supply 1 . Note that the effective value of the instantaneous voltage may be used as the power supply voltage.

Current detection unit 6 detects a power supply current, which is an alternating current flowing between AC power supply 1 and converter 3 , and outputs a detected value Is of the power supply current to control unit 10 . An example of a current detector used in the current detector 6 is an alternating current current transformer (ACCT). The voltage detection unit 7 detects a bus voltage and outputs a detected value Vdc of the bus voltage to the control unit 10 .

The bus voltage is a physical quantity representing the operating state of the DC side of the converter 3, that is, the output side. Also, the power supply voltage is a physical quantity representing the operating state of the AC side of the converter 3, that is, the input side. In order to distinguish between these two physical quantities, the bus voltage may be called the "first physical quantity" and the power supply voltage may be called the "second physical quantity". In addition, the voltage detection unit 7 that detects the bus voltage may be called the "first physical quantity detection unit", and the voltage detection unit 5 that detects the power supply voltage may be called the "second physical quantity detection unit".

The power supply circuit 14 is connected across the smoothing capacitor 4 . The power supply circuit 14 uses the voltage of the smoothing capacitor 4 to generate low-voltage DC voltages of 5V, 12V, 15V, and 24V. A low-voltage DC voltage is generated using the charge accumulated in the smoothing capacitor 4 . A low-voltage DC voltage is applied to each part of the supply destination as an operating voltage. The power supply circuit 14 outputs, for example, a DC voltage of 5V to the control section 10, the current detection section 6, and the like. In the control unit 10, a DC voltage of 5V is applied to a processor (not shown in FIG. 1).

The inverter 18 includes a leg 18A in which an upper arm element 18UP and a lower arm element 18UN are connected in series, a leg 18B in which an upper arm element 18VP and a lower arm element 18VN are connected in series, an upper arm element 18WP and a lower arm element 18WP. and a leg 18C to which the arm element 18WN is connected in series. Leg 18A, leg 18B and leg 18C are connected in parallel with each other.

FIG. 1 exemplifies the case where the upper arm elements 18UP, 18VP, 18WP and the lower arm elements 18UN, 18VN, 18WN are insulated gate bipolar transistors (IGBT), but the present invention is not limited to this. A MOSFET or an integrated gate commutated thyristor (IGCT) may be used instead of the IGBT.

Upper arm element 18UP includes transistor 18a and diode 18b connected in parallel with transistor 18a. Other upper arm elements 18VP, 18WP and lower arm elements 18UN, 18VN, 18WN have the same configuration. "Parallel" here means that the anode side of the diode is connected to the first terminal corresponding to the emitter of the IGBT, and the cathode side of the diode is connected to the second terminal corresponding to the collector of the IGBT.

Although FIG. 1 shows a configuration including three legs in which the upper arm element and the lower arm element are connected in series, the configuration is not limited to this. The number of legs may be four or more. Also, the circuit configuration shown in FIG. 1 is adapted to the motor 500 which is a three-phase motor. When the motor 500 is a single-phase motor, the inverter 18 is also configured to correspond to the single-phase motor. Specifically, it is configured to have two legs in which the upper arm element and the lower arm element are connected in series. It should be noted that one leg may be composed of a plurality of pairs of upper and lower arm elements regardless of whether the motor 500 is a single-phase motor or a three-phase motor.

When the transistors 18a of the upper arm elements 18UP, 18VP, 18WP and the lower arm elements 18UN, 18VN, 18WN are MOSFETs, the upper arm elements 18UP, 18VP, 18WP and the lower arm elements 18UN, 18VN, 18WN are silicon carbide and gallium nitride. It may be formed of a WBG semiconductor such as a base material or diamond. By using a MOSFET formed of a WBG semiconductor, it is possible to enjoy the effects of withstand voltage and heat resistance.

A connection point 26a between the upper arm element 18UP and the lower arm element 18UN is connected to the first phase (for example, U phase) of the motor 500, and a connection point 26b between the upper arm element 18VP and the lower arm element 18VN is connected to the first phase of the motor 500. 2 (eg, V phase), and the connection point 26c between the upper arm element 18WP and the lower arm element 18WN is connected to the third phase (eg, W phase) of the motor 500. FIG. In the inverter 18, the connection points 26a, 26b, 26c form AC terminals.

The current detection unit 9 detects motor current flowing between the inverter 18 and the motor 500 and outputs a detected value Iuvw of the motor current to the control unit 10 .

The control unit 10 generates a control signal for controlling each switching element in the converter 3 based on the detection value Vs of the voltage detection unit 5, the detection value Is of the current detection unit 6, and the detection value Vdc of the voltage detection unit 7. S311 to S322 are generated. The control signal S311 is a control signal for controlling the switching element Q1, and the control signal S322 is a control signal for controlling the switching element Q4. The switching elements Q2 and Q3 are also controlled by the control signal from the control section 10. FIG. Hereinafter, the operation of each arm element in accordance with the control signals S311 to S322 will be appropriately referred to as "switching operation". Control signals S 311 to S 322 generated by the control section 10 are input to the gate drive circuit 15 .

Based on the detected value Vdc of the voltage detector 7 and the detected value Iuvw of the current detector 9, the controller 10 controls each switching element provided in the inverter 18 so that the motor 500 rotates at a desired rotation speed. to generate control signals S1 to S6 for controlling the The inverter 18 has a three-phase circuit configuration and has six switching elements corresponding to the three-phase circuit configuration. Also, six control signals S1 to S6 are generated corresponding to the six switching elements. Control signals S 1 to S 6 generated by the control section 10 are input to the gate drive circuit 17 .

Gate drive circuit 15 generates drive pulses G311-G322 for driving respective switching elements in converter 3 based on control signals S311-S322. The drive pulse G311 is a drive pulse for driving the switching element Q1, and the drive pulse G322 is a drive pulse for driving the switching element Q4. The switching elements Q2 and Q3 are also driven by drive pulses from the gate drive circuit 15. FIG.

Gate drive circuit 17 generates drive pulses G1 to G6 for driving respective switching elements in inverter 18 based on control signals S1 to S6.

In FIG. 1, the controller 10 is provided inside the motor drive device 100 as a common controller that controls the DC power supply 50 and the load 12, but the configuration is not limited to this. Separate control units may be configured to control the DC power supply 50 and the load 12 , and the respective control units may be provided inside the DC power supply 50 and the load 12 .

Next, basic operation of the motor drive device 100 according to Embodiment 1 will be described. First, in the first leg 31, the first upper arm element 311 and the first lower arm element 312 operate in a complementary manner or in such a manner that they are not turned on at the same time. That is, when one of the first upper arm element 311 and the first lower arm element 312 is on, the other is off. As described above, the first upper arm element 311 and the first lower arm element 312 are controlled by control signals S311 and S312 generated by the control section 10. FIG. An example of the control signals S311 and S312 is a pulse width modulation (PWM) signal.

In order to prevent a short circuit of the smoothing capacitor 4 via the AC power supply 1 and the reactor 2, when the absolute value of the detected value Is of the power supply current output from the AC power supply 1 is equal to or less than the current threshold, the first upper arm element 311 and the first lower arm element 312 are both turned off. Below, the short circuit of the smoothing capacitor 4 is called "capacitor short circuit." A capacitor short-circuit is a state in which the energy stored in the smoothing capacitor 4 is released and current is regenerated in the AC power supply 1 .

As described above, the second upper arm element 321 and the second lower arm element 322 forming the second leg 32 are controlled by the control signals S321 and S322 generated by the control section 10. FIG. The second upper arm element 321 and the second lower arm element 322 are basically turned on or off according to the polarity of the power supply voltage. Specifically, when the power supply voltage polarity is positive, the second lower arm element 322 is on and the second upper arm element 321 is off. Also, when the power supply voltage polarity is negative, the second upper arm element 321 is on and the second lower arm element 322 is off.

Next, the relationship between the state of each arm element of converter 3 in the first embodiment and the path of the current flowing through motor drive device 100 in the first embodiment will be described. In the following description, each arm element of converter 3 is a MOSFET, and the diode of each arm element is a parasitic diode that the MOSFET itself has inside.

First, the structure of the MOSFET will be described with reference to FIG. FIG. 2 is a schematic cross-sectional view showing a schematic structure of a MOSFET used in converter 3 of the first embodiment. FIG. 2 illustrates an n-type MOSFET.

For an n-type MOSFET, a p-type semiconductor substrate 600 is used, as shown in FIG. A source electrode S, a drain electrode D and a gate electrode G are formed on the semiconductor substrate 600 . An n-type region 601 is formed in a portion in contact with the source electrode S and the drain electrode D by ion-implanting a high-concentration impurity. In addition, an oxide insulating film 602 is formed between the portion of the semiconductor substrate 600 where the n-type region 601 is not formed and the gate electrode G. As shown in FIG. That is, an oxide insulating film 602 is interposed between the gate electrode G and the p-type region 603 in the semiconductor substrate 600 .

When a positive voltage is applied to the gate electrode G, electrons are attracted to the interface between the p-type region 603 and the oxide insulating film 602 in the semiconductor substrate 600, and the interface is negatively charged. Where electrons gather, the electron density is higher than the hole density and the area becomes n-type. This n-type portion is called a channel 604 and serves as a passage for current. Channel 604 is an n-type channel in the example of FIG. By turning on the MOSFET, more current flows through the channel 604 than through the parasitic diode formed in the p-type region 603 .

FIG. 3 is a first diagram showing paths of current flowing through converter 3 in the first embodiment. FIG. 3 shows a state in which the polarity of the power supply voltage is positive and the absolute value of the detected value Is of the power supply current is greater than the current threshold. In this state, the first upper arm element 311 and the second lower arm element 322 are on and the first lower arm element 312 and the second upper arm element 321 are off. At this time, current flows in the order of the AC power supply 1, the reactor 2, the switching element Q1, the smoothing capacitor 4, the switching element Q4, and the AC power supply 1. As described above, the first embodiment has an operation mode in which the current is passed through the respective channels of the switching elements Q1 and Q4, rather than through the diodes D1 and D4. This operation is called "synchronous rectification". Note that in FIG. 3, the MOSFETs that are turned on are indicated by circles. The same applies to subsequent figures. Details of the operation modes will be described later.

FIG. 4 is a second diagram showing paths of current flowing through converter 3 in the first embodiment. FIG. 4 shows a state in which the polarity of the power supply voltage is negative and the absolute value of the detected value Is of the power supply current is greater than the current threshold. In this state, the first lower arm element 312 and the second upper arm element 321 are on and the first upper arm element 311 and the second lower arm element 322 are off. At this time, current flows in the order of AC power supply 1, switching element Q3, smoothing capacitor 4, switching element Q2, reactor 2, and AC power supply 1. As described above, in Embodiment 1, synchronous rectification may be performed in which current is passed through the respective channels of the switching elements Q3 and Q2 instead of passing the current through the diodes D3 and D2.

FIG. 5 is a third diagram showing paths of current flowing through converter 3 in the first embodiment. FIG. 5 shows a state in which the polarity of the power supply voltage is positive and the absolute value of the detected value Is of the power supply current is greater than the current threshold. In this state, the first lower arm element 312 and the second lower arm element 322 are on and the first upper arm element 311 and the second upper arm element 321 are off. At this time, current flows in the order of the AC power supply 1, the reactor 2, the switching element Q2, the switching element Q4, and the AC power supply 1. As a result, a power supply short-circuit path that does not pass through the smoothing capacitor 4 is formed. When forming the power supply short-circuit path shown in FIG. 5, it is essential to turn on the first lower arm element 312, but the second lower arm element 322 may be either on or off. In the first embodiment, as shown in FIG. 5, a mode is prepared in which a current is passed through the channel of the switching element Q4 instead of through the diode D4 to form a power supply short-circuit path.

FIG. 6 is a fourth diagram showing paths of current flowing through converter 3 in the first embodiment. FIG. 6 shows a state in which the polarity of the power supply voltage is negative and the absolute value of the detected value Is of the power supply current is greater than the current threshold. In this state, the first upper arm element 311 and the second upper arm element 321 are on, and the first lower arm element 312 and the second lower arm element 322 are off. At this time, current flows in the order of AC power supply 1, switching element Q3, switching element Q1, reactor 2, and AC power supply 1. As a result, a power supply short-circuit path that does not pass through the smoothing capacitor 4 is formed. When forming the power supply short-circuit path shown in FIG. 6, it is essential to turn on the first upper arm element 311, but the second upper arm element 321 may be turned on or off. In Embodiment 1, as shown in FIG. 6, a mode is provided in which a current is passed through the channel of the switching element Q3 instead of through the diode D3 to form a power supply short-circuit path.

The control unit 10 can control the values of the power supply current and the bus voltage by controlling the switching of the current paths described above. Motor drive device 100 continuously switches between the operation shown in FIG. 3 and the operation shown in FIG. 5 when the polarity of the power supply voltage is positive. Further, when the polarity of the power supply voltage is negative, motor drive device 100 continuously switches between the operation shown in FIG. 4 and the operation shown in FIG. As a result, it is possible to realize control for suppressing the rise of the bus voltage, current control for improving the power factor, and synchronous rectification for improving the operating efficiency.

Next, operation modes used in the DC power supply device 50 of the first embodiment will be described with reference to FIGS. 7 and 8. FIG. FIG. 7 is a diagram for explaining features of operation modes according to the first embodiment. FIG. 8 is a diagram showing operation waveforms when operated in the operation mode shown in FIG.

FIG. 7 describes four modes of operation: (a) rectification mode, (b) synchronous rectification mode, (c) slow switching mode, and (d) fast switching mode. Each operation mode is classified by a combination of whether three controls, synchronous rectification, current control, and bus voltage control, are performed or not. Synchronous rectification is as described above and is performed to improve operational efficiency. Bus voltage control is control for suppressing an increase in bus voltage. Current control is control for improving the power factor of the current flowing in and out of the converter 3 and suppressing harmonics. The low-speed switching mode may be called "first switching mode", and the high-speed switching mode may be called "second switching mode".

The DC power supply device 50 in Embodiment 1 has a rectification mode, and further has at least one operation mode of a synchronous rectification mode, a slow switching mode, and a high speed switching mode. Applications or products that do not require the boost operation may not necessarily have the slow switching mode and the fast switching mode.

FIG. 8(a) shows operating waveforms when operated in the rectification mode. Specifically, the waveforms of the power supply voltage, the power supply current, and the control signals S321 to S322 for controlling the switching elements Q1 to Q4 are shown from the top side. Other operating modes are similar. Since there is no need to control the switching elements in the rectification mode, there is an advantage that the power consumption of the power supply circuit 14 that operates the gate drive circuit 15 can be suppressed. In addition, there is an advantage that control is easy because there is no need to control switching elements.

FIG. 8B shows operating waveforms when operating in the synchronous rectification mode. The synchronous rectification mode is an operation mode in which the corresponding switching element is turned on at the timing when the parasitic diode is energized, and current is energized on the channel side of the switching element. In the example of FIG. 8B, the switching elements Q1 and Q4 or the switching elements Q2 and Q3 are controlled to be ON at the timing when the parasitic diode is energized. A synchronous rectification mode can be used to achieve high efficiency, especially when the currents flowing are small. In the synchronous rectification mode, the parasitic diode is replaced with a switching element as the energized element. Therefore, as shown in FIG. 7, current control and bus voltage control are not implemented.

FIG. 8(c) shows operation waveforms when operating in the low-speed switching mode. The low-speed switching mode is an operation mode in which the power supply voltage is short-circuited via the reactor 2 once or more per half cycle of the power supply voltage. In the example of FIG. 8C, two short-circuit operations are performed for each half cycle of the power supply voltage. Energy is stored in the reactor 2 by performing the short-circuit operation. When the short-circuit operation is released after the energy is accumulated, the energy accumulated in the reactor 2 is transferred to the smoothing capacitor 4 and accumulated. This makes it possible to boost the voltage of the smoothing capacitor 4, that is, the bus voltage.

The boost amount of the bus voltage is adjusted by bus voltage control. A proportional integral controller or the like is used for bus voltage control. In the bus voltage control, the operation of converter 3 is controlled so that the detected value Vdc of the bus voltage approaches the target voltage. Also, in the bus voltage control, the short-circuit time when the power supply voltage is short-circuited via the reactor 2 is controlled. Further, in the bus voltage control, by changing the response time of the proportional-integral controller, it is possible to suppress an excessive rise in the bus voltage that may occur due to the occurrence of load fluctuations.

In the slow switching mode, the short circuit action allows the short circuit current to flow. As a result, it is possible to improve the power factor and suppress the harmonic current by expanding the flow width of the power supply current. As for the improvement of the current waveform, it is also possible to predetermine the timing for performing the short-circuit operation with reference to the zero crossing point of the power supply voltage, and refer to it according to the load. Alternatively, the power supply current may be detected and the short-circuit time may be controlled so that the detected current waveform approaches a sine wave. In the low-speed switching mode, since the operation time for short-circuiting is short, it is possible to suppress the occurrence of harmonic noise.

FIG. 8(d) shows operation waveforms when operated in the high-speed switching mode. The high-speed switching mode is an operation mode in which the power supply voltage is short-circuited multiple times via the reactor 2 over the entire period of the power supply voltage. The significance of the short-circuit operation is the same as in the slow switching mode. That is, energy is accumulated in the reactor 2 by performing the short-circuit operation, and the energy accumulated in the reactor 2 is transferred to the smoothing capacitor 4 by canceling the short-circuit operation after the energy is accumulated. This makes it possible to boost the bus voltage. Control of the boost amount of the bus voltage can also be realized by the same control as in the low-speed switching mode.

As described above, in the high-speed switching mode, the short-circuiting operation is performed over the entire period of one cycle of the power supply voltage, so the width of the current flow is wider than in the low-speed switching mode. As a result, compared to the low-speed switching mode, it is possible to further improve the power factor and suppress the harmonic current. Also, in the fast switching mode, the power factor can be controlled to a value close to unity. As a result, especially on the high load side, the load can be driven up to the limit of the breaker capacity, and the power of the device can be increased.

Next, loss characteristics of the MOSFET used in the DC power supply device 50 will be described. FIG. 9 is a diagram showing loss characteristics of MOSFETs used in the DC power supply device 50 according to the first embodiment. In FIG. 9, the horizontal axis indicates the current flowing through the MOSFET in the ON state and the current flowing through the parasitic diode. Also, the vertical axis indicates the voltage required to cause current to flow through the switching element in the ON state and the voltage required to cause current to flow to the parasitic diode.

In FIG. 9, the solid line K1 represents the parasitic diode forward voltage. A parasitic diode forward voltage is an example of a current-voltage characteristic that represents the losses that occur in a parasitic diode. Diodes generally require a large voltage when the current value is small because the loss is large, but when the current value exceeds a certain value, the change rate of the loss is improved and the slope of the current-voltage characteristics is relaxed. . This characteristic appears in the waveform indicated by the solid line K1 in FIG.

Also, dashed line K2 represents the MOSFET drain-source voltage, which is the voltage between the drain and source of the MOSFET. A MOSFET drain-source voltage is an example of a current-voltage characteristic that represents current flowing in carriers of a switching element and loss caused by the on-resistance of the switching element due to the flow of the current. In a switching element such as a MOSFET, the voltage required to allow current to flow increases quadratically with respect to the current value. This characteristic appears in the waveform indicated by the dashed line K2 in FIG.

In FIG. 9, the cross points where the solid line K1 and the dashed line K2 intersect are the current flowing through the parasitic diode and the voltage required to cause the current to flow, the current flowing to the MOSFET and the voltage required to cause the current to flow, is the point at which are equal. In Embodiment 1, the current value at the cross point where the two current-voltage characteristics of the parasitic diode and the switching element intersect is the "second current threshold". Note that the above-described current threshold, that is, the current threshold used when comparing the absolute values of the detected value Is of the power supply current is called the "first current threshold". In FIG. 9, the second current threshold is represented by "Ith2". The second current threshold is a value greater than the first current threshold.

Next, the timing at which the control unit 10 turns on and off the switching element using the first current threshold and the second current threshold in the synchronous rectification mode will be described. FIG. 10 is a diagram showing timings at which the control unit 10 turns on the switching elements in the DC power supply device 50 according to the first embodiment. In FIG. 10, the horizontal axis is time. The upper part of FIG. 10 shows the waveforms of the power supply voltage and the power supply current. In the lower part of FIG. 10, the switching elements Q1 and Q2 are current synchronous switching elements whose ON/OFF is controlled according to the polarity of the power supply current, and the switching elements Q3 and Q4 are ON/OFF according to the polarity of the power supply voltage. is a controlled voltage synchronous switching element. FIG. 10 also shows the values of the first current threshold Ith1 and the second current threshold Ith2 along with the waveform of the power supply current. Although FIG. 10 shows one cycle of the AC power output from the AC power supply 1, the control unit 10 performs control similar to that shown in FIG. 10 also in other cycles.

When the polarity of the power supply voltage is positive, the control unit 10 turns on the switching element Q4 and turns off the switching element Q3. Further, when the polarity of the power supply voltage is negative, the control unit 10 turns on the switching element Q3 and turns off the switching element Q4. In FIG. 10, the timing when the switching element Q4 turns off from on and the timing when the switching element Q3 turns on from off are the same timing, but the timing is not limited to this. The control unit 10 may provide a dead time during which both the switching elements Q3 and Q4 are turned off between the timing when the switching element Q4 is turned off from on and the timing when the switching element Q3 is turned on from off. Similarly, the control unit 10 provides a dead time during which both the switching elements Q3 and Q4 are turned off between the timing when the switching element Q3 is turned off from on and the timing when the switching element Q4 is turned on from off. good too.

When the polarity of the power supply voltage is positive, the control unit 10 turns on the switching element Q1 when the absolute value of the power supply current becomes equal to or greater than the first current threshold value Ith1. Furthermore, when the absolute value of the power supply current exceeds the second current threshold Ith2, the switching element Q1 is turned off. After that, when the absolute value of the power supply current becomes smaller and the absolute value of the power supply current becomes equal to or less than the second current threshold Ith2, the control unit 10 turns on the switching element Q1. Furthermore, when the absolute value of the power supply current becomes smaller than the first current threshold Ith1, the switching element Q1 is turned off. Further, when the polarity of the power supply voltage is negative, the control unit 10 turns on the switching element Q2 when the absolute value of the power supply current becomes equal to or greater than the first current threshold value Ith1. Furthermore, when the absolute value of the power supply current exceeds the second current threshold Ith2, the switching element Q2 is turned off. After that, when the absolute value of the power supply current becomes smaller and the absolute value of the power supply current becomes equal to or less than the second current threshold Ith2, the control unit 10 turns on the switching element Q2. Furthermore, when the absolute value of the power supply current becomes smaller than the first current threshold Ith1, the switching element Q2 is turned off.

When the absolute value of the power supply current is equal to or less than the first current threshold Ith1, the control unit 10 controls the switching elements Q1 and Q3 so as not to turn on simultaneously, and controls the switching elements Q2 and Q4 not to turn on simultaneously. . Thereby, the control unit 10 can prevent a capacitor short circuit in the motor drive device 100 .

With the control of the control unit 10 as described above, the motor drive device 100 can realize synchronous rectification by the switching elements Q1 and Q2 of the first leg 31 . Specifically, when the absolute value of the power supply current is greater than or equal to the first current threshold value Ith1 and less than or equal to the second current threshold value Ith2, the control unit 10 allows the current to flow through the switching element Q1 or the switching element Q2, which has a small loss within this range. flow. Further, when the absolute value of the power supply current is greater than the second current threshold Ith2, the control unit 10 causes the current to flow through the diode D1 or the diode D2, which has a small loss within this range. As a result, the motor drive device 100 can pass a current through an element with a small loss according to the current value, so that a decrease in efficiency can be suppressed, and a highly efficient device with a reduced loss can be obtained.

Note that the control unit 10 may perform the step-up operation by performing switching control to complementarily turn on and off the switching elements Q1 and Q2 during the period in which the switching element Q1 is turned on. Similarly, the control unit 10 may perform switching control to complementarily turn on and off the switching elements Q1 and Q2 during the period in which the switching element Q2 is turned on to perform the boosting operation.

That is, when the absolute value of the power supply current is greater than or equal to the first current threshold Ith1 and less than or equal to the second current threshold Ith2, the control unit 10 controls the first leg 31 and the second leg according to the polarity of the power supply current. One of the switching elements Q1 and Q2 forming the first leg 31 of 32 is permitted to be turned on. Further, when the absolute value of the power supply current is smaller than the first current threshold Ith1 or larger than the second current threshold Ith2, the control unit 10 selects one switching element out of the switching elements Q1 and Q2 as described above. turn on is prohibited.

Specifically, when the polarity of the power supply current is positive and the absolute value of the power supply current is equal to or greater than the first current threshold Ith1 and equal to or less than the second current threshold Ith2, the control unit 10 turns on the switching element Q1. allow When the absolute value of the power supply current is smaller than the first current threshold Ith1 or larger than the second current threshold Ith2, the switching element Q1 is prohibited from being turned on. When the polarity of the power supply current is positive and the absolute value of the power supply current is equal to or greater than the first current threshold Ith1 and equal to or less than the second current threshold Ith2, the control unit 10 turns off the switching element Q1 during the OFF period. Turn on Q2. If the absolute value of the power supply current is smaller than the first current threshold Ith1 or larger than the second current threshold Ith2, the switching element Q2 is also prohibited from being turned on.

Further, when the polarity of the power supply current is negative and the absolute value of the power supply current is equal to or greater than the first current threshold Ith1 and equal to or less than the second current threshold Ith2, the control unit 10 permits turning on of the switching element Q2. . If the absolute value of the power supply current is smaller than the first current threshold Ith1 or larger than the second current threshold Ith2, the switching element Q2 is prohibited from being turned on. Further, when the polarity of the power supply current is negative and the absolute value of the power supply current is equal to or greater than the first current threshold Ith1 and equal to or less than the second current threshold Ith2, the control unit 10 controls Switching element Q1 is turned on. When the absolute value of the power supply current is smaller than the first current threshold Ith1 or larger than the second current threshold Ith2, the switching element Q1 is also prohibited from being turned on.

Thus, the control unit 10 allows the switching element to be turned on in a region where the absolute value of the power supply current is equal to or greater than the first current threshold Ith1 and the loss of the switching element is smaller than the loss of the parasitic diode. Further, the control unit 10 prohibits the switching element from turning on in a region where the loss of the switching element is greater than the loss of the parasitic diode.

In the example of FIG. 10, the control unit 10 controls on/off of the switching elements Q3 and Q4 according to the polarity of the power supply voltage, and controls on/off of the switching elements Q1 and Q2 according to the polarity of the power supply current. but not limited to this. The control unit 10 may control on/off of the switching elements Q1 and Q2 according to the polarity of the power supply voltage, and may control on/off of the switching elements Q3 and Q4 according to the polarity of the power supply current.

Also, the second current threshold Ith2 is, as described above, the current value when the voltage required to cause the current to flow through the parasitic diode and the switching element have the same value, but is not limited to this. The second current threshold Ith2 may be a value determined according to the characteristics of the voltage required to pass the current through the parasitic diode and the characteristics of the voltage required to pass the current through the switching element.

For example, the second current threshold Ith2 is set to a value larger than the current value when the voltage necessary for the current to flow through the parasitic diode and the switching element are the same, according to the switching loss generated in the switching element. can be a value. This makes it possible to determine the second current threshold Ith2 in consideration of the switching element generated when switching the switching element from on to off. In this case, even if the absolute value of the power supply current increases while the switching element is on, if the loss cannot be reduced by turning off the switching element, the control unit 10 keeps the switching element on. to As a result, the motor drive device 100 can further suppress a decrease in efficiency.

Further, the second current threshold Ith2 may be a value obtained by adding or subtracting a specified value to the current value when the voltage required to flow the current to the parasitic diode and the switching element have the same value. . This makes it possible to determine the second current threshold value Ith2 in consideration of the difference in characteristics due to variations in the components of each element. In this case, the control unit 10 improves the reduction of loss compared to the case where the second current threshold Ith2 is the current value when the voltage required to flow the current to the parasitic diode and the switching element are the same. It may not be possible. However, the control unit 10 can reduce the loss more than the case where the switching element is kept on even if the absolute value of the power supply current increases while the switching element is on.

FIG. 11 is a flow chart used for explaining the operation of the main part in Embodiment 1. FIG. FIG. 11 shows a processing flow in which the control unit 10 of the motor drive device 100 controls the switching elements Q1 and Q2 to turn on and off. Here, as an example, a case where the polarity of the power supply current is positive will be described.

The control unit 10 compares the absolute value |Is| of the detected value Is of the power supply current with the first current threshold (step S21). When the absolute value |Is| is smaller than the first current threshold (step S21, No), the control unit 10 prohibits turning on the switching element Q1 (step S22). When the absolute value |Is| is greater than or equal to the first current threshold (step S21, Yes), the controller 10 compares the absolute value |Is| with the second current threshold (step S23). When the absolute value |Is| is equal to or less than the second current threshold (step S23, No), the control unit 10 permits turning on of the switching element Q1 (step S24). When the absolute value |Is| is greater than the second current threshold (step S23, Yes), the control unit 10 prohibits turning on the switching element Q1 (step S22). After step S22 or step S24, the control unit 10 returns to step S21 and repeats the above process. When the polarity of the power supply current is negative, the control unit 10 performs the same processing as described above for the switching element Q2.

In step S21 above, the determination is "Yes" when the absolute value |Is| is equal to the first current threshold value, but the determination may be "No". That is, if the absolute value |Is| is equal to the first current threshold value, the determination may be either "Yes" or "No". Further, in step S23 above, the case where the absolute value |Is| is equal to the second current threshold is determined as "No", but it may be determined as "Yes". That is, if the absolute value |Is| is equal to the second current threshold value, the determination may be either "Yes" or "No".

By the way, as explained in the section [Problems to be Solved by the Invention], the efficiency of the synchronous rectification operation is improved by the on-voltage of the MOSFET, that is, the drain-source voltage when the MOSFET is turned on. It is assumed that the parasitic diode connected in parallel has a higher voltage drop. On the other hand, depending on the operating state of the MOSFET, the on-voltage of the MOSFET may be higher than the voltage drop of the parasitic diode connected in parallel. This point will be described below with reference to FIGS. 12 to 14. FIG. FIG. 12 is a diagram showing temperature characteristics of forward current of a parasitic diode in a general MOSFET. FIG. 13 is a diagram showing temperature characteristics of on-resistance of a general MOSFET. FIG. 14 is a diagram showing how the cross points shown in FIG. 9 vary with temperature. 14, the solid line K1 indicates the solid line K1 shown in FIG. 9, and the broken line K2 indicates the broken line K2 shown in FIG.

As shown in FIG. 12, the source-drain voltage V _SD of the parasitic diode in a general MOSFET decreases as the temperature T _a rises with respect to the same forward current I _S . Therefore, the solid line K1 shown in FIG. 14 moves downward and to the right in the paper surface like the solid line K1' when the temperature _Ta rises. Also, as shown in FIG. 13, the on-resistance _RDS of _a general MOSFET tends to increase as the temperature Ta increases with respect to the same drain current _ID . Therefore, the dashed line K2 shown in FIG. 14 moves to the upper left direction of the paper surface like the dashed line K2' when the temperature _Ta rises. Due to this characteristic, the second current threshold Ith2, which is the current value at the cross point, moves to the left side of the paper, that is, in the direction in which the current value decreases as the temperature rises. When the temperature is not a condition, in the flowchart of FIG. 11, synchronous rectification is performed between the second current threshold Ith2 and the second current threshold Ith2′ in FIG. 14, but synchronous rectification is not performed. less loss and better efficiency. Therefore, if the second current threshold value Ith2 in the flowchart of FIG. 11 is corrected based on the ambient temperature of the switching element or the ambient temperature of the device in which the switching element is mounted, the efficiency of the DC power supply 50 can be further improved. can be increased.

Next, the configuration of the switching element will be described. In the motor drive device 100, one method of increasing the switching speed of the switching element is to reduce the gate resistance of the switching element. As the gate resistance becomes smaller, the charging and discharging time of the gate input capacitance becomes shorter, and the turn-on period and the turn-off period become shorter, so that the switching speed becomes faster.

However, there is a limit to reducing the switching loss by reducing the gate resistance. Therefore, an example of configuring the switching element with a WBG semiconductor such as GaN or SiC will be described. By using a WBG semiconductor for the switching element, the loss per switching can be further suppressed, the efficiency is further improved, and high-frequency switching becomes possible. In addition, since high-frequency switching becomes possible, the size of the reactor 2 can be reduced, and the size and weight of the motor drive device 100 can be reduced. Also, by using a WBG semiconductor for the switching element, the switching speed is improved and the switching loss is suppressed. As a result, it is possible to simplify measures for heat dissipation so that the switching elements can continue to operate normally. Moreover, by using a WBG semiconductor for the switching element, the switching frequency can be set to a sufficiently high value, for example, 16 kHz or higher. Thereby, noise caused by switching can be suppressed.

In addition, in the GaN semiconductor, a two-dimensional electron gas is generated at the interface between the GaN layer and the aluminum gallium nitride layer, and this two-dimensional electron gas provides high carrier mobility. Therefore, a switching element using a GaN semiconductor can realize high-speed switching. Here, if the AC power supply 1 is a commercial power supply of 50 Hz or 60 Hz, the audible frequency ranges from 16 kHz to 20 kHz, that is, from 266 times to 400 times the frequency of the commercial power supply. GaN semiconductors are suitable for switching at frequencies above this audible range. When the switching elements Q1 to Q4 made of silicon (Si), which is the mainstream semiconductor material, are driven at a switching frequency of several tens of kHz or higher, the ratio of switching loss increases and heat dissipation measures are essential. On the other hand, the switching elements Q1 to Q4 made of GaN semiconductor have a very small switching loss even when driven at a switching frequency of several tens of kHz or higher, more specifically, at a switching frequency higher than 20 kHz. Therefore, heat dissipation measures are not required, or the size of a heat dissipation member used for heat dissipation measures can be reduced, and the motor drive device 100 can be made smaller and lighter. In addition, since high-frequency switching becomes possible, the size of the reactor 2 can be reduced. In order to prevent the primary component of the switching frequency from entering the measurement range of the noise terminal voltage standard, the switching frequency is preferably 150 kHz or less.

In addition, since the WBG semiconductor has a smaller electrostatic capacity than the Si semiconductor, less recovery current is generated due to switching, and loss and noise due to the recovery current can be suppressed. Therefore, WBG semiconductors are suitable for high frequency switching.

SiC semiconductors have a lower on-resistance than GaN semiconductors. For this reason, the first upper arm element 311 and the first lower arm element 312 of the first leg 31 having a higher switching frequency than the second leg 32 are made of GaN semiconductor, and the second leg 31 having a lower switching frequency is used. The second upper arm element 321 and the second lower arm element 322 of the leg 32 may be made of SiC semiconductor. This makes it possible to make the most of the respective characteristics of the SiC semiconductor and the GaN semiconductor. In addition, by using a SiC semiconductor for the second upper arm element 321 and the second lower arm element 322 of the second leg 32 having a smaller number of switching times than the first leg 31, the second upper arm Among the losses of the element 321 and the second lower arm element 322, the percentage of conduction loss increases, and turn-on loss and turn-off loss decrease. Therefore, an increase in heat generation due to switching of the second upper arm element 321 and the second lower arm element 322 is suppressed, and the second upper arm element 321 and the second lower arm element constituting the second leg 32 are suppressed. 322 chip area can be relatively small. As a result, the SiC semiconductor, which has a low yield in chip manufacturing, can be effectively used.

Also, the second upper arm element 321 and the second lower arm element 322 of the second leg 32 having a low switching frequency may be SJ-MOSFETs with a super junction structure. By using the SJ-MOSFET, it is possible to take advantage of the low on-resistance which is an advantage of the SJ-MOSFET, while suppressing the demerit of high electrostatic capacity and easy occurrence of recovery. Also, by using the SJ-MOSFET, the manufacturing cost of the second leg 32 can be reduced as compared with the case of using the WBG semiconductor.

Moreover, the WBG semiconductor has higher heat resistance than the Si semiconductor, and can operate even at high junction temperatures. Therefore, by using a WBG semiconductor, the first leg 31 and the second leg 32 can be configured with small chips having high thermal resistance. In particular, SiC semiconductors, which have a low yield during chip manufacturing, can be used for small chips to achieve cost reduction.

In addition, even when the WBG semiconductor is driven at a high frequency of about 100 kHz, the increase in loss generated in the switching element is suppressed, so the loss reduction effect due to the miniaturization of the reactor 2 is increased, and a wide output band, that is, a wide load range. Under these conditions, a highly efficient converter can be realized.

In addition, the WBG semiconductor has higher heat resistance than the Si semiconductor, and has a high allowable level of heat generation due to switching loss imbalance between the arms.

As described above, according to Embodiment 1, the voltage detection unit 7, which is the first physical quantity detection unit, detects the bus voltage, which is the first physical quantity representing the operating state of the output side of the converter 3, Voltage detector 5 , which is a second physical quantity detector, detects power supply voltage, which is a second physical quantity representing the operating state of the input side of converter 3 . The first and second physical quantities are input to the controller 10 . Control unit 10 controls the operation of converter 3 based on the first and second physical quantities. The control unit 10 further controls the conduction of each switching element of the converter 3 according to the ambient temperature of the switching element, the outside air temperature of the converter, or an operation command to the converter 3, thereby operating the converter 3 in different operation modes. It has multiple modes. This makes it possible to achieve further efficiency in applying synchronous rectification.

Further, according to Embodiment 1, when the absolute value of the power supply current is equal to or greater than the first current threshold and equal to or less than the second current threshold, the control unit 10 controls the switching element having a smaller loss than the parasitic diode within this range. turn on. Further, when the absolute value of the power supply current is larger than the second current threshold, the control unit 10 prohibits turning on the switching element whose loss is larger than that of the parasitic diode within this range. As a result, in the converter 3, the current can flow through an element with a small loss according to the current value. As a result, it is possible to obtain a highly efficient DC power supply device 50 with reduced efficiency drop and loss. In addition, in this control, it is preferable to correct the second current threshold value based on the ambient temperature of the switching element or the outside air temperature of the equipment in which the switching element is mounted. As a result, a DC power supply device 50 with even higher efficiency can be obtained.

Next, a hardware configuration for realizing the functions of the control unit 10 according to Embodiment 1 will be described with reference to FIGS. 15 and 16. FIG. FIG. 15 is a block diagram showing an example of a hardware configuration that implements the functions of the control section 10 according to Embodiment 1. As shown in FIG. FIG. 16 is a block diagram showing another example of the hardware configuration that implements the functions of the control section 10 according to the first embodiment.

When realizing the function of the control unit 10 in Embodiment 1, as shown in FIG. It can be configured to include an interface 304 for performing.

The processor 300 may be an arithmetic means such as an arithmetic unit, a microprocessor, a microcomputer, a CPU (Central Processing Unit), or a DSP (Digital Signal Processor). The memory 302 includes non-volatile or volatile semiconductor memories such as RAM (Random Access Memory), ROM (Read Only Memory), flash memory, EPROM (Erasable Programmable ROM), EEPROM (registered trademark) (Electrically EPROM), Examples include magnetic discs, flexible discs, optical discs, compact discs, mini discs, and DVDs (Digital Versatile Discs).

The memory 302 stores programs for executing the functions of the control unit 10 according to the first embodiment. Processor 300 performs the above-described processing by exchanging necessary information via interface 304, executing programs stored in memory 302, and referring to tables stored in memory 302 by processor 300. It can be carried out. Results of operations by processor 300 may be stored in memory 302 .

Further, when realizing the function of the control unit 10 in Embodiment 1, the processing circuit 305 shown in FIG. 16 can also be used. The processing circuit 305 corresponds to a single circuit, a composite circuit, an ASIC (Application Specific Integrated Circuit), an FPGA (Field-Programmable Gate Array), or a combination thereof. Information to be input to the processing circuit 305 and information to be output from the processing circuit 305 can be obtained via the interface 306 . Even in the configuration using the processing circuit 305, part of the processing in the control unit 10 may be performed by the processor 300 configured as shown in FIG.

Embodiment 2.
In the second embodiment, control when gate drive circuit 15 that drives each switching element of converter 3 includes a bootstrap circuit will be described.

FIG. 17 is a diagram showing a configuration example of the gate drive circuit 15 according to the second embodiment. In FIG. 17, the gate drive circuit 15 includes drive circuits 51 and 52 and a bootstrap circuit 54 . The drive circuit 51 is a drive circuit used when driving the first upper arm element 311 of the first leg 31 . The drive circuit 52 is a drive circuit used when driving the first lower arm element 312 of the first leg 31 . The second upper arm element 321 and the second lower arm element 322 of the second leg 32 are also driven by two similar drive circuits.

The bootstrap circuit 54 includes a resistor 54a, a diode 54b, and a capacitor 54c, which is a bootstrap capacitor. A driving voltage is applied to the capacitor 54c from the driving power supply 55 through a series circuit of the resistor 54a and the diode 54b. In the bootstrap circuit 54 configured as described above, when the lower arm switching elements Q2 and Q4 are turned on, a current flows through the resistor 54a, the diode 54b, the capacitor 54c, and the lower arm switching elements Q2 and Q4, and the capacitor 54c is turned on. charged. The charging voltage of the capacitor 54c becomes a gate drive voltage for driving the upper arm switching elements Q1 and Q3.

As in the example of FIG. 17, in the gate drive circuit 15 including the bootstrap circuit 54, the gate drive voltage for driving the upper arm switching elements Q1 and Q3 turns on the lower arm switching elements Q2 and Q4. obtained by letting With the control signals S311 to S322 as shown in FIG. 8, it is possible to reliably generate the gate drive voltage for driving the switching elements of the upper arm.

On the other hand, when the gate drive circuit 15 includes the bootstrap circuit 54, the power consumption of the drive circuit 51 that drives the upper arm switching elements Q1 and Q3 increases in proportion to the number of switching times. Therefore, in the second embodiment, the waveforms of some of the control signals S311 to S322 shown in FIG. 8 are changed to those shown in FIG. 18 suitable for the configuration including the bootstrap circuit . FIG. 18 is a diagram showing operation waveforms in the second embodiment corresponding to FIG.

FIG. 18 shows operation waveforms obtained by partially changing the control signals S311 to S322 in the operation mode shown in FIG. Differences between the operating waveforms shown in FIG. 18 and the operating waveforms shown in FIG. 8 are as follows.

First, in the rectification mode, the operating waveforms are the same because each switching element does not turn on. Also, the operating waveforms are the same in the synchronous rectification mode. In the synchronous rectification mode, there is no need to perform switching control to form a power supply short-circuit path, and the number of times of switching of the switching element in the upper arm is once per half cycle of the power supply cycle, so the effect of changing the switching pattern is small. Therefore, the switching pattern of FIG. 8B is used without change.

On the other hand, in slow switching mode, FIG. 8(c) is changed to FIG. 18(c). Also in FIG. 18(c), since the upper arm switching elements Q1 and Q3 do not operate, the loss in the bootstrap circuit 54 is reduced. As can be seen from the comparison between FIG. 8(d) and FIG. 18(d), the high-speed switching mode has the same operating waveforms. Referring to FIG. 8(d), it can be seen that the switching elements Q1 and Q3 of the upper arm are not operating, and the loss in the bootstrap circuit 54 is reduced by the original operating waveforms. Whether to select the switching pattern of FIG. 8 or the switching pattern of FIG. 18 is determined based on the load condition. Details of the load condition will be described in a third embodiment described later.

As described above, according to the second embodiment, gate drive circuit 15, which is a drive circuit for driving converter 3, drives upper-arm switching elements Q1 and Q3 connected to the positive side of smoothing capacitor 4. A bootstrap circuit 54 is provided as a drive power supply for the. The control unit 10 has a low-speed switching mode as an operation mode, and stops the operation of the switching elements connected to the bootstrap circuit 54 based on the load conditions when operating in the low-speed switching mode. Thereby, in the configuration including the bootstrap circuit 54, the DC power supply device 50 can be operated efficiently.

Further, according to the second embodiment, the gate drive circuit 15, which is a drive circuit for driving the converter 3, is a drive power source for driving the switching elements Q1 and Q3 of the upper arm connected to the positive side of the smoothing capacitor 4. A bootstrap circuit 54 is provided. The control unit 10 has a high-speed switching mode as an operation mode, and the switching element connected to the bootstrap circuit 54 does not turn on in the high-speed switching mode. Thereby, in the configuration including the bootstrap circuit 54, the DC power supply device 50 can be operated efficiently.

Embodiment 3.
Embodiment 3 describes an application example of motor drive device 100 described in Embodiment 1. FIG. FIG. 19 is a diagram showing the configuration of air conditioner 400 according to Embodiment 3. As shown in FIG. Motor drive device 100 described in Embodiments 1 and 2 can be applied to products such as fans, compressors, and air conditioners. In Embodiment 3, as an application example of the motor drive device 100 according to Embodiments 1 and 2, an example in which the motor drive device 100 is applied to an air conditioner 400 will be described.

In FIG. 19 , a motor 500 is connected to the output side of the motor driving device 100 and the motor 500 is connected to a compression element 504 . Compressor 505 comprises motor 500 and compression element 504 . The refrigerating cycle section 506 is configured to include a four-way valve 506a, an indoor heat exchanger 506b, an expansion valve 506c, and an outdoor heat exchanger 506d.

The flow path of the refrigerant circulating inside the air conditioner 400 passes from the compression element 504, the four-way valve 506a, the indoor heat exchanger 506b, the expansion valve 506c, the outdoor heat exchanger 506d, and again through the four-way valve 506a. , and return to the compression element 504 . The motor drive device 100 receives supply of AC power from the AC power supply 1 and rotates the motor 500 . The compression element 504 can compress the refrigerant by rotating the motor 500 and circulate the refrigerant inside the refrigeration cycle section 506 .

In the air conditioner 400, operation under an intermediate condition in which the output is less than half of the rated output, that is, under a low output condition is dominant throughout the year. Therefore, the degree of contribution to annual power consumption is high under intermediate conditions. In addition, in air conditioner 400, motor 500 rotates at a low speed, and the bus voltage required to drive motor 500 tends to be low. Therefore, it is effective in terms of system efficiency to operate the switching elements used in air conditioner 400 in a passive state. Therefore, the DC power supply 50 capable of reducing loss in a wide range of operation modes from passive state to high frequency switching state is useful for the air conditioner 400.

In addition, since the DC power supply 50 can suppress the switching loss, the temperature rise of the DC power supply 50 is suppressed. Cooling capacity can be secured. Therefore, the DC power supply device 50 is highly efficient and suitable for the air conditioner 400 with a high output of 4.0 kW or more.

Further, according to Embodiment 3, switching loss is reduced by high-frequency driving of the switching elements, the energy consumption rate is low, and highly efficient air conditioner 400 can be realized.

Furthermore, air conditioner 400 according to Embodiment 3 is configured to operate in the operation mode shown in FIG. FIG. 20 is a diagram showing features of operation modes in the third embodiment.

FIG. 20 shows operation modes corresponding to temperature conditions and load conditions when the air conditioner 400 is operated, and operations of switching elements of the upper arm when the air conditioner 400 is operated in the operation modes.

First, it is desirable to use the rectification mode because the air conditioner 400 does not operate stably in terms of control during start-up and in the event of an abnormality. An example of the temperature condition of the air conditioner is the ambient temperature of the device in which the switching element is mounted.

In air conditioner 400, the cooling operation is used when the outside air temperature is high. In this cooling operation, there are load conditions such as cooling intermediate, cooling rated, and cooling overload. When the outside air temperature is high, the ambient temperature around the switching element tends to be high.

The loss in the diode is obtained by multiplying the forward current by the forward voltage drop, which is the forward voltage drop due to the forward current. Also, the loss of the MOSFET is found by the square product of the current flowing through the MOSFET and the ON resistance of the MOSFET. Further, as described above, diodes tend to have a lower forward voltage drop as the temperature rises, and MOSFETs tend to have a higher on-resistance as the temperature rises. In other words, the loss of a diode improves as the temperature rises, and the loss of a MOSFET worsens as the temperature rises.

Also, in the air conditioner 400, the bootstrap circuit 54 described above is widely used. A loss occurs in the drive circuit 51 that drives the MOSFETs of the upper arm, and the loss of the drive circuit 51 increases in proportion to the number of times of switching the MOSFETs. Therefore, depending on the load conditions, the loss of the parasitic diode may be smaller than the loss including the loss of the upper arm MOSFET and the loss of the drive circuit 51 . In particular, when the ambient temperature of the converter 3 is high and the current is large, the effect is remarkable. For this reason, the switching operation of the upper arm MOSFET is stopped, and an operation mode considering control to positively conduct the parasitic diode is set.

Based on the above points, the following operation modes are constructed. First, when the air conditioner 400 is in cooling operation, the outside temperature is high, so the ambient temperature of the switching element is also high. In addition, since the temperature rises due to heat generation of the switching element itself, especially in the low-speed switching mode and high-speed switching mode where the current is large, the operation efficiency of the upper arm switching element is stopped and the diode is made conductive. to improve

Specifically, in the example of FIG. 20, the synchronous rectification mode in which the upper arm switching element is operated is selected in the middle of cooling with a light load. For a high-load cooling rating, a low-speed switching mode is selected in which the upper arm switching element is not operated. In a cooling overload, which is an overload, a high-speed switching mode is selected in which the upper arm switching element is not operated.

On the other hand, since the outside air temperature is low during the heating operation, the ambient temperature of the switching element is also lower than during the cooling operation. Therefore, during heating operation, synchronous rectification is positively performed by operating the switching element of the upper arm to improve the operation efficiency.

Specifically, in the example of FIG. 20, the synchronous rectification mode in which the switching element of the upper arm is operated is selected in the middle of heating with a light load. At the high load heating rating, the low speed switching mode with the upper arm switching element operated is selected. In a heating overload, which is an overload, a high-speed switching mode in which the upper arm switching element is operated is selected. In other words, the operation mode in which the switching element of the upper arm is operated is selected in any of the heating intermediate load which is light load, the heating rated load which is high load, and the heating overload which is overload.

As described above, the conduction of the switching element of the converter 3 is controlled according to the operation command to the converter 3 such as the cooling operation or the heating operation, more specifically, the intermediate cooling operation or the intermediate heating operation. In these controls, when the upper arm switching element is not driven, it is not necessary to drive the lower arm switching element in consideration of securing the drive voltage for the upper arm switching element. Thereby, control over the converter 3 can be simplified.

Further, as can be understood from the switching pattern of FIG. 18, there is no need to provide a dead time to prevent the upper arm MOSFET and the lower arm MOSFET from being turned on at the same time. The dead time is a non-overlapping time during which the upper arm MOSFET and the lower arm MOSFET are not turned on at the same time, and is also called short-circuit prevention time. If control is performed without dead time, the degree of matching between the control command value and the actual command value is enhanced. As a result, controllability is improved, and the air conditioner 400 with high efficiency and high control stability can be realized.

Note that the characteristics of the MOSFET shown in FIGS. 12 and 13 are current characteristics, and may be improved in the future. Therefore, the determination as to whether or not to operate the switching element of the upper arm may be made flexibly in accordance with the technical trends of MOSFETs, as shown in parentheses in FIG.

It should be noted that the configuration shown in the above embodiment shows an example of the contents of the present invention, and it is possible to combine it with another known technique, and the configuration can be changed without departing from the gist of the present invention. It is also possible to omit or change part of

1 AC power supply 2 Reactor 3 Converter 3a, 3b, 3c, 3d, 26a, 26b, 26c Connection point 4 Smoothing capacitor 5, 7 Voltage detection unit 6, 9 Current detection unit 10 Control unit 12 Load , 14 power supply circuit, 15, 17 gate drive circuit, 16a, 16b DC bus, 18 inverter, 18A, 18B, 18C leg, 18a transistor, 18b, 54b, D1, D2, D3, D4 diode, 18UN, 18VN, 18WN Bottom arm element, 18UP, 18VP, 18WP upper arm element, 31 first leg, 32 second leg, 50 DC power supply, 51, 52 drive circuit, 54 bootstrap circuit, 54a resistor, 54c capacitor, 55 drive power supply, REFERENCE SIGNS LIST 100 motor drive device, 300 processor, 302 memory, 304, 306 interface, 305 processing circuit, 311 first upper arm element, 312 first lower arm element, 321 second upper arm element, 322 second lower arm Element 400 Air conditioner 500 Motor 504 Compression element 505 Compressor 506 Refrigeration cycle unit 506a Four-way valve 506b Indoor heat exchanger 506c Expansion valve 506d Outdoor heat exchanger 600 Semiconductor substrate 601,603 Region, 602 oxide insulating film, 604 channel, D drain electrode, G gate electrode, Q1, Q2, Q3, Q4 switching element, S source electrode.

Claims (13)

a reactor;
A converter comprising four bridge-connected unidirectional elements, connected to an AC power supply via the reactor, and converting a power supply voltage, which is an AC voltage output from the AC power supply, into a DC voltage and applying it to a load. ,
a smoothing capacitor connected between output terminals of the converter;
a first physical quantity detector that detects a first physical quantity representing an operating state of the output side of the converter;
a second physical quantity detector that detects a second physical quantity representing an operating state of the input side of the converter;
a control unit to which the first and second physical quantities are input and which controls the operation of the converter;
with
Two of the four unidirectional elements in the converter are connected in series to form a first leg, and the remaining two unidirectional elements are connected in series. constitute the second leg,
At least two unidirectional elements in the first and second legs connected to the positive side of the smoothing capacitor, or two unidirectional elements in the first and second legs connected to the negative side of the smoothing capacitor. A switching element is connected in parallel to each of the unidirectional element, or the two unidirectional elements in the first leg, or the two unidirectional elements in the second leg,
As operation modes for operating the converter in different operation modes, the control unit controls a rectification mode in which the unidirectional element is energized and a channel of the corresponding switching element at the timing at which the unidirectional element is energized. and has a synchronous rectification mode that allows
A first switching mode in which the power supply voltage is short-circuited via the reactor at least once every half cycle of the power supply voltage, which is the voltage of the AC power supply; at least one of a second switching mode in which the power supply voltage is short-circuited multiple times as the operation mode;
The control unit further selects the operation mode based on an ambient temperature of the switching element or an outside air temperature of the converter, and a load condition when driving the load .
DC power supply. A drive circuit that drives the converter based on a control signal output from the control unit,
The drive circuit includes a bootstrap circuit, which is a power supply circuit for driving the switching element connected to the positive side of the smoothing capacitor,
Having the second switching mode as the operation mode,
The DC power supply device according to claim 1 , wherein in the second switching mode, the switching element connected to the bootstrap circuit does not turn on. A drive circuit that drives the converter based on a control signal output from the control unit,
The drive circuit includes a bootstrap circuit, which is a power supply circuit for driving the switching element connected to the positive side of the smoothing capacitor,
Having the first switching mode as the operation mode,
3. The DC power supply device according to claim 1 , wherein the operation of the switching element connected to the bootstrap circuit is stopped based on the load condition when operating in the first switching mode. The DC power supply device according to any one of claims 1 to 3, wherein the switching element is a metal oxide semiconductor field effect transistor made of a wide bandgap semiconductor. The DC power supply device according to claim 4 , wherein the wide bandgap semiconductor is silicon carbide, gallium nitride, gallium oxide, or diamond. The DC power supply device according to any one of claims 1 to 3, wherein the switching element is a metal oxide semiconductor field effect transistor with a super junction structure. The direct-current power supply device according to any one of claims 1 to 6 , wherein the unidirectional element is a parasitic diode of a metal oxide semiconductor field effect transistor. The DC power supply device according to any one of claims 1 to 7 , wherein the unidirectional element is a diode. A DC power supply device according to any one of claims 1 to 8 ;
an inverter that converts the output voltage of the DC power supply into an AC voltage and applies the AC voltage to a motor provided in the load;
A motor drive device with A blower comprising the motor drive device according to claim 9 . A compressor comprising the motor drive device according to claim 9 . An air conditioner comprising at least one of the blower according to claim 10 and the compressor according to claim 11 . 13. The air conditioner according to claim 12 , wherein the load condition is at least one of startup of the air conditioner, an abnormality, and cooling or heating operation of the air conditioner.

Images