JP2015149875A - Electric motor controller - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide an electric motor controller capable of controlling an electric motor to stably drive without using any position sensor in a range from a state a rotor is ceased to a state the rotor rotates at a high speed by precisely estimating the magnetic pole phase of the rotor.SOLUTION: The electric motor controller includes: a phase estimation section 6 that estimates the magnetic pole phase to output an estimated phase θs in a relatively low speed range; a magnetic flux estimator 8 estimates the magnetic pole phase to output an estimated phase θ0 in a relatively high speed range; and a phase switching section 9 that directly switches the phase from the estimated phase θs to the estimated phase θ0; or from the estimated phase θ0 to the estimated phase θs at predetermined preset speed to output a phase signal θm.

Description

  The present invention relates to an electric motor control device that drives and controls an electric motor in a wide speed range from a stopped state to a high speed state without using a position sensor.

In general, when an AC motor is driven using a power converter, a current sensor, a position sensor, or the like is used, and vector control is performed in which a command voltage is applied according to the magnetic pole phase of the rotor detected by the position sensor. The power converter is a device that is connected between an AC power system and a DC power system and converts the power into DC-AC, is supplied with DC power from a storage battery or the like, and outputs AC power to drive the motor. .
On the other hand, the position sensor is a device that detects the magnetic pole phase of the rotor of the electric motor, and there are various types such as an encoder and a resolver. In general, since a position sensor is expensive and inferior in durability, several methods for vector control of an electric motor without using a position sensor are known.

  For example, Patent Document 1 applies a predetermined voltage vector to each combination of multi-phase windings in a state where the driving current is not supplied to the windings of the electric motor, and according to the applied voltage vector An electrical angle detector that measures the behavior of the current flowing in each multi-phase winding and detects the magnetic pole phase of the rotor from the behavior of the current, and a synchronous motor driving device using the same are disclosed.

  Further, for example, Patent Document 2 estimates a current controller that outputs a voltage command from a current command and current on two rotation axes, and an angular frequency and an estimated rotor magnetic flux based on the current and voltage command on the two rotation shafts. Disclosed is a control device for a synchronous motor that includes an adaptive observer for calculating a rotation speed and controls the synchronous motor by an inverter.

Further, for example, in Patent Document 3, when the electric motor is rotating at a predetermined rotation speed or more, the first detection method using the induced voltage tries to detect the magnetic pole phase, and the predetermined rotation in which the position cannot be detected by the induced voltage. If the number is less than the number, a motor control device that controls the motor by detecting the magnetic pole phase by the second detection method of multiplexing the voltage signals for position detection is disclosed.
It is common to detect the rotation speed of the rotor of the electric motor and select the first and second detection methods according to the magnitude of the rotation speed. When switching the detection method, the magnetic pole phase estimation means Phase jump of the estimated magnetic pole phase may occur due to a transient state or computation delay, etc., so the phase amount corresponding to the current motor speed for a predetermined time until the estimated value of the magnetic pole phase estimation means after switching is established The estimated magnetic pole phase is compensated by the phase switching unit using Δθ.

Japanese Patent Laid-Open No. 7-177788 (see claims 1 and 5) Japanese Patent No. 4672236 (see claim 1) Japanese Patent Laid-Open No. 2002-315386 (refer to claim 14, FIGS. 7, 8, etc.)

In the conventional apparatus of Patent Document 1, even if a voltage vector for estimating the magnetic pole phase is applied and driving from the stopped state, the change in the magnetic pole phase within the time during which the voltage vector is applied increases as the rotational speed increases. There is a problem that the magnetic pole phase cannot be correctly estimated at high speeds because it cannot be ignored.
Further, the device of Patent Document 2 is a method that uses an induced voltage, and there is a problem that a voltage is not generated unless the rotor performs a rotation operation, and the magnetic pole phase cannot be detected in a stopped state.

On the other hand, the apparatus of Patent Document 3 is a first magnetic pole position estimating means that assumes a first detection method for estimating a magnetic pole phase using an induced voltage when the electric motor rotates at a predetermined rotational speed or more. And a second magnetic pole phase estimation means that assumes a second detection method for estimating the magnetic pole phase by multiplexing the voltage signals for position detection when the rotational speed is less than a predetermined rotational speed at which the position cannot be detected by the induced voltage. I have.
Further, for switching between both detection methods, a phase switching unit for compensating the estimated magnetic pole phase for a predetermined time until the estimated value of the magnetic pole phase estimating means after switching is established is provided separately from both estimating means, The estimated magnetic pole phase θ2 different from the estimated magnetic pole phases θ1 and θ3 by the both estimating means is generated using the phase amount Δθ corresponding to the speed of the motor (see paragraphs 0048 and 0049 of Patent Document 3 and FIG. 8).

  For this reason, in consideration of changes in rotational speed at the time of switching, it is necessary to set the switching time for compensating the estimated magnetic pole phase by a certain amount longer by the phase switching unit, and the phase error increases and the motor is continuously driven stably. There was a problem that it was difficult.

  The present invention was made to solve these conventional problems, and in an apparatus for driving and controlling an electric motor without using a position sensor, even when the rotor is rotating at high speed from a stopped state, An object of the present invention is to obtain an electric motor control device that can accurately estimate the magnetic pole phase of a rotor and drive the electric motor stably.

  An electric motor control device according to the present invention is composed of a plurality of switching elements, converts a DC voltage from a DC power source and outputs an AC voltage to be supplied to the electric motor, based on a command value input from the upper control system A voltage command generation unit that generates and outputs an AC voltage command value, a gate signal generation unit that generates and outputs a gate signal for controlling on / off of the switching element of the power conversion unit based on the AC voltage command value, and flows to each phase of the motor Current detector for detecting current, first phase estimation means for estimating the magnetic pole phase of the motor by calculation under the condition that the rotational speed of the motor is not more than a predetermined set speed, and outputting it as a first phase signal, rotational speed of the motor Is a second phase estimation means for estimating the magnetic pole phase of the electric motor by calculation under a condition that is equal to or higher than the set speed, and outputting it as a second phase signal; A phase switching unit that inputs a phase signal, selects either the first phase signal or the second phase signal based on the rotation speed of the motor or information corresponding to the rotation speed, and outputs the selected phase signal as a phase signal; The command generation unit converts the current value of each phase detected by the current detector based on the phase signal into a quadrature two-phase current detection value of quadrature two-phase coordinates and outputs the second coordinate conversion unit, orthogonal to the command value A quadrature two-phase voltage command generation unit that generates and outputs a quadrature two-phase voltage command value in quadrature two-phase coordinates based on the two-phase current detection value, and an quadrature two-phase voltage command value based on the phase signal. A first coordinate conversion unit for converting to a value and outputting the value is provided.

As described above, the phase switching unit in the electric motor control device according to the present invention changes from the first phase signal to the second phase signal and from the relatively high speed range to the low speed when shifting from the relatively low speed range to the high speed range. When shifting to the area, since the second phase signal is switched directly and continuously to the first phase signal, a phase signal different from the first and second phase signals is separately calculated and output during the transition operation transition. There is no need, and control at the time of transition is particularly simple, and phase estimation accuracy is also good.
Therefore, without using a position sensor, the motor is stably driven by accurately estimating the magnetic pole phase of the rotor in a wide speed range from a stopped state to a high speed state, with low cost and excellent durability. It is possible to obtain an electric motor control device that can be used.

It is a block diagram which shows the whole structure of the electric motor control apparatus by Embodiment 1 of this invention. It is a circuit diagram which shows the internal structure of the power converter 10 of FIG. It is a figure which shows the voltage vector which the pulse generation part 5 of FIG. 1 outputs. It is a figure which shows the electric current waveform which flows when the voltage vector of FIG. 3 is applied. It is a figure for demonstrating operation | movement of the gate switching part 7 of FIG. It is a block diagram which shows the internal structure of the phase estimation part 6 of FIG. It is a diagram which shows the relationship between the electric current deviation of each phase calculated by the phase calculating part 17 of FIG. 6, and a magnetic pole phase. It is a block diagram which shows the internal structure of the magnetic flux estimation device 8 of FIG. It is a figure for demonstrating operation | movement of the phase switching part 9 of FIG. It is a block diagram which shows the structure of the phase estimation part 6 of the electric motor control apparatus by Embodiment 2 of this invention. It is a figure for demonstrating operation | movement of the phase switching part 9 of the electric motor control apparatus by Embodiment 3 of this invention.

Embodiment 1 FIG.
1 is a diagram showing an overall configuration of an electric motor control apparatus according to Embodiment 1 of the present invention. The quadrature two-phase voltage command generator 1 receives a torque command value input from a higher control system (not shown in FIG. 1) and current information (orthogonal two-phase coordinates) converted into quadrature two-phase coordinates by a second coordinate converter 3 described later. The quadrature two-phase voltage command values vd * and vq * are calculated from the phase current detection values id and iq, and output to the first coordinate converter 2 described later.

  In the quadrature two-phase voltage command generation unit 1, for example, the deviation between the quadrature two-phase current command value id * and iq * obtained from the torque command value and the quadrature two-phase current detection value id and iq shown in Equation (1) is calculated. The quadrature two-phase voltage command values vd * and vq * can be calculated by calculating the PI control used.

In Expression (1), Kp is a proportional gain, and Ki is an integral gain.
The quadrature two-phase voltage command values vd * and vq * calculated by the equation (1) are input to the first coordinate conversion unit 2 and expressed in the three-phase AC coordinate system by the conversion equation shown in the equation (2). It is converted into a three-phase AC voltage command value vu *, vv *, vw *.

  In the equation (2), the angle θ represents the direction of the magnetic pole phase with the U-phase of the three-phase alternating current as 0 degree as a reference and the field magnetic flux direction of the rotor as the d-axis.

  The first coordinate conversion unit 2 performs an operation (θ = θm) shown in Expression (2) based on a phase signal θm output from the phase switching unit 9 described later, and performs quadrature two-phase voltage command values vd *, vq. * Is converted into a three-phase AC voltage command value vu *, vv *, vw * and output to the PWM control unit 4 as a drive control unit.

  The second coordinate conversion unit 3 converts the current from the three-phase AC coordinate system to the orthogonal two-phase coordinate system. That is, the second coordinate conversion unit 3 is based on the phase signal θm, and is detected by the current detectors 11, 12, and 13, and the three phases flowing in the respective phases of the motor, here, the permanent magnet embedded PM motor 14. The alternating currents iu, iv, iw are converted into quadrature two-phase current detection values id, iq by the conversion formula shown in formula (3) and output to the quadrature two-phase voltage command generation unit 1.

  The quadrature two-phase voltage command generation unit 1, the first coordinate conversion unit 2, and the second coordinate conversion unit 3 constitute a voltage command generation unit 50.

The PWM control unit 4 performs PWM (pulse width modulation) processing on the three-phase AC voltage command values vu *, vv *, and vw * to generate a gate signal (driving gate signal) for driving the power conversion unit 10 described later. It converts and outputs to the gate switching part 7 mentioned later.
In the PWM processing of the three-phase AC voltage command value, the PWM processing is generally performed by performing amplitude comparison processing with a triangular wave carrier signal.

  Here, the drive control unit is a PWM control unit that generates a PWM gate signal as a drive gate signal by PWM processing. However, for example, a PAM control unit that generates a drive gate signal by PAM (pulse amplitude modulation) processing It is good.

The gate switching unit 7 switches the drive gate signal from the PWM control unit 4 and a magnetic pole phase detection gate signal generated by the pulse generation unit 5 to be described later. Output to the converter 10.
The PWM control unit 4, the pulse generation unit 5, and the gate switching unit 7 constitute a gate signal generation unit 51.

  FIG. 2 is a circuit diagram showing an internal configuration of the power conversion unit 10 of FIG. Here, as an example, the configuration of a power converter corresponding to three-phase alternating current is shown. In FIG. 2, VDC represents the positive side of the DC power supply, and VGND represents the negative side of the DC power supply. In the power converter 10, as shown in the figure, six switching elements S1 to S6 are arranged and connected to each arm, and six diodes are arranged and connected in parallel with the switching elements. On-off control of these switching elements S1 to S6 based on the gate signals UP to WN generates a three-phase AC signal corresponding to the U phase, the V phase, and the W phase, and the desired operating frequency and voltage command value are set. The PM motor 14 is driven by outputting an AC signal controlled to a value.

  In FIG. 1, the power conversion unit 10 performs switching processing of each switching element based on the gate signal input from the gate switching unit 7, and outputs three-phase AC voltages vu, vv, vw to the PM motor 14. As an example, the PM motor 14 describes a synchronous motor using a permanent magnet as a rotor as a control target. Current detectors 11, 12, and 13 detect three-phase alternating currents iu, iv, and iw that flow between power conversion unit 10 and PM motor 14.

  Next, a phase estimation unit 6 as a first phase estimation unit for estimating a magnetic pole phase in a relatively low speed range including a stop state based on the magnetic pole phase detection gate signal generated by the pulse generation unit 5 and a phase estimation method thereof explain.

  FIG. 3 is a diagram showing a voltage vector output to the PM motor 14 by the magnetic pole phase detection gate signal from the pulse generator 5. P1 to P6 represent each voltage vector output for estimating the magnetic pole phase of the rotor. For example, in the voltage vector P1, the switching elements S1, S4, and S6 of the power conversion unit 10 illustrated in FIG. 2 are turned on. At this time, as gate control, the UP, VN, and WN gate signals are activated. In the state of the voltage vector P1, current flows from the switching element S1 to S4 and S6.

  Similarly, in voltage vector P2, switching elements S1, S3, and S6 of power converter 10 shown in FIG. 2 are turned on. At this time, as gate control, the UP, VP and WN gate signals are activated. In the state of the voltage vector P2, it flows from the switching elements S1 and S3 to S6. The voltage vectors P3 to P6 perform the same control operation, and are controlled so that currents flow in different directions.

  FIG. 4 is a diagram showing a current waveform when each voltage vector is applied. The pulse waveform in FIG. 4 indicates the timing of the pulse signal input to the switching element, and each phase current has a peak value when the pulse signal changes from on to off. Moreover, the pattern of the voltage applied to each phase by a series of pulse signals becomes a different pattern. By continuously outputting these six types of voltage vectors, a difference in current flow depending on the magnetic pole phase appears.

  In FIG. 4, in the state of the voltage vector P1, a positive current flows in the U phase, and a reverse current of approximately half the magnitude of the U phase flows in the V phase and the W phase. In the voltage vector P4 following the voltage vector P1, a negative current flows in the U phase, and a reverse current having a magnitude approximately half that of the U phase flows in the V phase and the W phase. In this way, six types of voltage vectors are continuously applied. Phase currents iu, iv, and iw that flow when each voltage vector is applied are defined as shown in FIG. By using the peak values iu1 to iu6, iv1 to iv6, and iw1 to iw6 of each phase current, it is possible to estimate the magnetic pole phase of the rotor.

  In managing the output timing of the voltage vectors in the pulse generation unit 5, six voltage vectors are output in the order shown in FIG. At this time, in the section between the respective voltage vectors, all the gate signals are turned off, and the pulsed current is converged to zero at one end. In the management of the current detection capturing timing in the phase estimation unit 6, control is performed so as to perform the calculation by capturing the peak value of each current value in FIG. The order shown in FIG. 4 is preferable for the order in which the voltage vectors are output. However, the order shown in FIG. 4 is not necessarily required. The pattern of the voltage applied to each phase by a series of pulse signals becomes a different pattern. You can do that.

FIG. 5 shows the operation of the gate switching unit 7, that is, the timing for switching between the PWM gate signal from the PWM control unit 4 and the magnetic pole phase detection gate signal from the pulse generation unit 5.
In the present invention, in the speed range from the stop state to a predetermined set speed set in advance, the phase estimation unit 6 is responsible for the magnetic pole phase estimation operation. During the period in which the phase estimation unit 6 operates, the gate switching unit 7 Thus, the PWM gate signal from the PWM control unit 4 and the magnetic pole phase detection gate signal from the pulse generation unit 5 are alternately output at a predetermined cycle determined in advance by time division.

As a result, in the PWM section in which the PWM gate signal is output, the voltage command based on the torque command value is output, the torque is output according to the command value, and the magnetic pole phase detection outputs the magnetic pole phase detection gate signal. In the section, the torque output is temporarily stopped, the pulse output for detecting the magnetic pole phase is performed, and the magnetic pole phase is estimated.
The magnetic pole phase detection interval is preferably as short as possible compared to the PWM interval, although it depends on the pulse width of the output pulse. For example, the PWM interval is about 15 ms, the repetition period of pulses output in the magnetic pole phase detection interval is about several hundred μs, and the entire magnetic pole phase detection interval is about 5 ms.

The phase estimation unit 6 takes in the peak value of the pulse current generated by each voltage vector shown in FIG. 3 and estimates the magnetic pole phase of the rotor.
FIG. 6 is a block diagram showing an internal configuration of the phase estimation unit 6. In FIG. 6, current values iu, iv, iw of the respective phases detected by the current detectors 11, 12, 13 in FIG. 1 are input to the input terminal 15. A timing management signal from the pulse generator 5 is input to the input terminal 16.

  The phase calculation unit 17 takes in the peak value of the current in synchronization with the gate signal of the voltage vector to be output, and estimates and outputs the magnetic pole phase θp by calculation. Hereinafter, an example of the calculation method of the magnetic pole phase θp in the phase calculation unit 17 will be described.

  First, current deviations ΔxU, ΔxV, and ΔxW are obtained by Equation (4) using peak values iu1 to iu6, iv1 to iv6, and iw1 to iw6 of the pulse current.

Here, current deviations ΔxU, ΔxV, and ΔxW indicate current deviations when the voltage vectors are applied in the U-phase, V-phase, and W-phase directions in the reverse direction of 180 degrees.
The magnetic pole phase can be estimated using three types of current deviations ΔxU, ΔxV, and ΔxW. For example, when the rotor d-axis is oriented in the same direction as the U phase (θ = 0 degree), the magnetic flux of the permanent magnet and the magnetic flux generated by the armature current are in the same direction, causing magnetic saturation. As a result, if a state of magnetic saturation occurs by flowing a large armature current, ΔxU shows a maximum value at 0 degrees.

  On the other hand, when the d-axis of the rotor faces the direction opposite to the U phase (θ = 180 degrees), on the other hand, it is in a state where magnetic saturation is hardly caused, and ΔxU indicates a minimum value. As a result, when the voltage vectors P1 to P6 shown in FIG. 3 are applied, the current deviations ΔxU, ΔxV, and ΔxW are sinusoidal waveforms having a phase difference of 120 degrees as shown in FIG. 7 according to the magnetic pole phase of the rotor. The characteristics of By using this characteristic, the magnetic pole phase θp can be estimated from the values of the current deviations ΔxU, ΔxV, ΔxW.

  As described above, the magnitudes and pulse widths of the voltage vectors P1 to P6 based on the magnetic pole phase detection gate signal cause magnetic saturation in a part of the iron core when these voltage vectors are applied to the PM motor 14. The condition, that is, the condition value where nonlinearity appears in the magnetic characteristics is set. When this condition is satisfied, the magnetic pole phase can be estimated based on this nonlinearity.

  In the calculation, first, the three-phase current deviations ΔxU, ΔxV, ΔxW of the U phase, the V phase, and the W phase, which are 120 degrees out of phase, are expressed by the equation (5) as the α axis and β axis of the orthogonal two-phase coordinate system. The current deviations Δxα and Δxβ above are converted.

At this time, the α axis is a coordinate axis taken in the same direction as the U phase, and the β axis is a coordinate axis orthogonal to the α axis.
From the equation (5), the phase θpt indicated by the current deviation can be calculated by the equation (6).

  Here, considering the interval from 0 to 2π with the positive and negative signs of Δxα and Δxβ, the condition of the magnetic pole phase θp of the rotor can be estimated in the entire phase region from 0 to 2π if the condition is determined according to Equation (7). I can do it.

  As described above, the phase calculation unit 17 obtains the magnetic pole phase θp of the rotor, and the voltage vector for detecting the magnetic pole phase θp is applied at regular intervals such as intervals of several hundred μs to detect a series of magnetic pole phases. Since the pulse voltage group for applying is applied at regular intervals such as intervals of several ms, as shown in the lower part of FIG. 5, the magnetic pole phase θp is similarly calculated at the same regular cycles. Therefore, in order to drive the PM motor 14 using the magnetic pole phase θp, it is necessary to convert it into a continuous phase signal θs.

  In FIG. 6, the speed calculator 18 calculates the rotational speed ω using the phase signal θp detected intermittently. The rotational speed ω can be obtained by differentiating the magnetic pole phase θp. The calculated rotation speed ω is input to a low-pass filter (LPF) 19 for removing noise and the like.

  The output (first speed signal) ωs of the LPF 19 is output from the output terminal 24 and input to the speed correction unit 20. In the speed correction unit 20, a correction speed ωs ′ for conversion into a phase is obtained by the subsequent integration calculation unit 21 using Expression (8).

  Here, Δωs is acceleration information, and is calculated here as the amount of change per unit time of the rotational speed ω calculated for each magnetic pole phase detection section. Ts adopts a sampling period corresponding to a certain period of time, for example, an update period of the voltage command value, a period in which the current detectors 11 to 13 detect the current instantaneous value, and the like.

  In this case, the acceleration information Δωs is obtained from the amount of change per unit time of the rotational speed ωs input from the LPF 19 as described above, but other than this method, for example, a quadrature two-phase voltage command generation unit Based on the torque command value Tref input to 1, it is also possible to calculate using the equation (9).

  Here, α is a coefficient depending on the moment of inertia of the entire rotating system including the PM motor 14, and by multiplying the coefficient α by the torque command value Tref, the acceleration information Δωs, which is a change in speed in the PWM section, is obtained. Can be estimated.

  The output ωs ′ of the speed correction unit 20 is input to the integration calculation unit 21. The integration calculation unit 21 integrates the correction speed ωs ′ and outputs the correction magnetic pole phase θi. The switching processing unit 22 switches between the magnetic pole phase θp output from the phase calculation unit 17 and the corrected magnetic pole phase θi output from the integration calculation unit 21, and outputs the first phase signal θs from the output terminal 23.

  That is, as shown in the lower part of FIG. 5, the switching processing unit 22 selects the magnetic pole phase θp calculated by the phase calculating unit 17 at the detection time point when the magnetic pole phase is calculated in the magnetic pole phase detection section, and the detection time point At the time point excluding, the correction magnetic pole phase θi is selected and output as the first phase signal θs.

As described above, during the period in which the PM motor 14 is being driven and during the PWM period in which the magnetic pole phase θp cannot be obtained from the phase calculation unit 17, the corrected magnetic pole phase θi obtained by integrating and adding the correction speed ωs ′ is output. As a result, a continuous phase signal θs is obtained.
With this correction, the PM motor 14 can be stably driven and controlled without detecting the magnetic pole position for several tens of milliseconds.

  Here, since the peak values iu1 to iu6, iv1 to iv6, and iw1 to iw6 of the pulse current shown in FIG. 4 can be detected even when the PM motor 14 is stopped, the magnetic pole phase can be estimated. Is possible. Further, when the PM motor 14 is stopped, the phase signal θp detected intermittently always has the same value, and the rotation speed ω is calculated as 0 by the speed calculation unit 18.

  When a speed sensor for detecting speed is provided outside the block shown in the control configuration of FIG. 1, the speed calculation unit 18 is omitted and speed information input from the outside is input to the speed correction unit 20. It is also possible to obtain the corrected magnetic pole phase θi from the integral calculation unit 21.

Furthermore, in the above embodiment, a three-phase synchronous motor has been described as an example of the PM motor. However, the present invention can also be applied to a three-phase or more multi-phase synchronous motor. The same applies to the following embodiments.
In the case of a three-phase synchronous motor, the phase can be detected with six types of pulses at 60 degrees, but in the case of a six-phase motor, each phase has a phase difference of 60 degrees, so that it is described in the present invention. The magnetic pole phase can be detected by applying a plurality of voltage vectors having different phases by 30 degrees with the same pulse. Thus, the present invention can also be applied to a multiphase synchronous motor having three or more phases.

  However, when the rotational speed of the PM motor 14 increases and reaches a speed at which the change in the magnetic pole phase cannot be ignored during the voltage vector application time for detecting the magnetic pole phase (corresponding to the predetermined set speed described above), Since a highly reliable magnetic pole phase cannot be obtained, a magnetic flux estimator 8 serving as a second phase estimating means, which will be described later, is used instead of the phase estimator 6 that estimates the magnetic pole phase based on the magnetic saturation characteristics of the PM motor. The magnetic pole phase is estimated using the induced voltage associated with.

  For this reason, the gate switching unit 7 always outputs only the PWM gate signal from the PWM control unit 4 instead of the operation in which the PWM gate signal and the magnetic pole phase detection gate signal have been output in a time division manner.

  FIG. 8 is a block diagram showing the internal configuration of the magnetic flux estimator 8. As described above, the magnetic flux estimator 8 estimates the magnetic pole phase using an induced voltage accompanying rotation, and is basically the same as that introduced in the above-mentioned Patent Document 2, An outline of the configuration operation will be described according to an application example of the present invention.

In FIG. 8, the third coordinate conversion unit 27 is on two stationary axes using the estimated phase θ0 output from the phase estimation unit 38 by using the voltage command values vu *, vv *, and vw * input to the input terminal 25. The voltage command vectors vαs * and vβs * on the α-β axis are converted.
The fourth coordinate conversion unit 28 converts the current detection values iu, iv, iw input to the input terminal 26 into current vectors iαs, iβs on the α-β axis using the estimated phase θ0 output from the phase estimation unit 38. To do. Here, the armature resistance of the PM motor is R, the armature inductance is L, the estimated speed is ωr0, and the matrices A, B, C1, C2, and H are defined as in Expression (10).

Here, the values of h11, h12, h21, h22, h31, h32, h41, and h42 of the matrix H are controlled to change the value of each amplification gain according to the rotation speed.
The gain matrix calculator 30 outputs a result obtained by multiplying the matrix B by the voltage command vector (vαs *, vβs *) T. T represents a transposed matrix. The adder / subtractor 29 adds and subtracts current vectors iαs and iβs and estimated currents iαs0 and iβs0 described later according to the sign shown in FIG. 8 to output a current deviation vector (eα, eβ) T. The gain matrix calculator 31 outputs a result obtained by multiplying the current deviation vector (eα, eβ) T by the matrix H as an amplified deviation vector (e1, e2, e3, e4) T. The adder / subtracter 32 outputs a vector obtained by adding / subtracting the output of the gain matrix calculator 30, the output of the gain matrix calculator 31 and the output of a gain matrix calculator 33 described later according to the sign shown in FIG.

  The integrator 34 integrates the vector output from the adder / subtractor 32 element by element, and calculates the estimated armature reaction vectors pαs0 and pβs0 on the α-β axis and the estimated magnetic flux vectors pαr0 and pβr0 on the α-β axis as vectors. (Pαs0, pβs0, pαr0, pβr0) is output as T. The gain matrix calculator 33 obtains the matrix A of Expression (10) based on the estimated speed ωr0, and outputs the result of multiplying the matrix A by the vector (pαs0, pβs0, pαr0, pβr0) T.

  In this series of operations, the input of the integrator 34 corresponds to the right side of each matrix of the equation (11). Further, since the left side of the equation (11) is a differentiation of the vector (pαs0, pβs0, pαr0, pβr0) T and is also an input of the integrator 34, the output of the integrator 34 is a vector (pαs0, pβs0, pαr0, pβr0). T.

  As described above, if the voltage command vector (vαs *, vβs *) T on the α-β axis and the amplified deviation vector (e1, e2, e3, e4) T are given, the estimated armature on the α-β axis Reaction vectors pαs0 and pβs0 and estimated magnetic flux vectors pαr0 and pβr0 on the α-β axis can be obtained.

  The speed estimator 35 calculates the formula (12) based on the input estimated magnetic flux vector (pαr0, pβr0) T and the current deviation vector (eα, eβ) T, and estimates the speed (second speed signal). ωr0 is output.

  Here, s is a Laplace operator (differential operator), kp is a proportional gain, and ki is an integral gain.

  The gain matrix calculator 36 performs matrix calculation of Expression (13) and outputs estimated currents (iαs0, iβs0) T. The gain matrix calculator 37 performs a matrix calculation of Expression (14) and outputs an estimated magnetic flux vector (pαr0, pβr0) T.

  The phase estimation unit 38 performs the calculation of Expression (15) based on the input estimated magnetic flux vector (pαr0, pβr0) T, and outputs the estimated phase (second phase signal) θ0.

As described above, the magnetic flux estimator 8 calculates and outputs the estimated phase θ0 and the estimated speed ωr0 of the PM motor 14.
If the estimated speed ωs calculated by the phase estimator 6 is set as the initial value in the PI calculation shown in Expression (12), the time until the operation of the magnetic flux estimator 8 converges and stabilizes after switching is shortened. The
As a phase estimation method, the estimated speed ωr0 can be simply obtained by integration. At this time, the phase θs obtained by the phase estimation unit 6 is input as an initial value, and the estimated speed ωr0 is integrated and added. Go.

  The magnetic flux estimator 8 can also estimate the speed based on the voltage equation of the electric motor shown in the equation (16).

  Based on the estimated speed (first speed signal) ωs from the phase estimator 6 or the estimated speed (second speed signal) ωr0 from the magnetic flux estimator 8, the phase switching unit 9 outputs the estimated phase ( The phase signal θm is output by switching between the first phase signal θs and the estimated phase (second phase signal) θ0 output from the magnetic flux estimator 8. Next, a method for switching between the estimated phase θs and the estimated phase θ0 will be described.

  FIG. 9 is a diagram illustrating the timing when switching from the phase estimation operation by the phase estimation unit 6 to the phase estimation operation by the magnetic flux estimator 8. In this case, when the estimated speed ωs by the phase estimation unit 6 reaches the predetermined set speed described above, the estimated phase θp is obtained in the final magnetic pole phase detection section before switching based on the arrival time (A) point Thus, the gate switching unit 7 shifts to an operation of outputting only the PWM gate signal and enters the PWM section. At the timing of the point (A), after the magnetic flux estimator 8 is operated with an estimated speed ωs as an initial value, a predetermined set time (magnetic flux estimator operation section) elapses and the magnetic flux estimator 8 The phase signal θm used for the control is switched from the estimated phase θs to the estimated phase θ0 at the timing of the point (B) when the operation is stable.

As described above with reference to FIG. 6, the phase estimation unit 6 outputs the continuous estimated phase θs, and the magnetic flux estimator 8 starts up the calculation operation, and the operation is stabilized and the estimated phase θ0 is highly accurate. In this stage, θs is directly switched to θ0, so that the control at the time of transition is particularly simple, the problem described in the conventional patent document 3 does not occur, the phase estimation accuracy is good, and the PM motor is stable. Can be controlled.
Since no new magnetic pole phase estimation is performed during this magnetic flux estimator operation period, it is necessary to keep the time of about several tens of milliseconds so that the PM motor does not step out.

  As an initial value when the operation of the magnetic flux estimator 8 is started at the point (A) in FIG. 9, the output voltage command values vu *, vv *, vw * and the detected current values iu, iv, It is also possible to set iw, and by providing both or one of them as an initial value of the input of the magnetic flux estimator 8, the operation of the magnetic flux estimator 8 is quickly stabilized and the magnetic flux estimator operation section is shortened. I can do it.

  When the phase switching unit 9 switches between the estimated phase θs output from the phase estimation unit 6 and the estimated phase θ0 output from the magnetic flux estimator 8, when switching from the former θs to the latter θ0, the estimation output from the phase estimation unit 6 When switching from the latter θ0 to the former θs with the speed ωs as a reference, it is common to use the estimated speed ωr0 output from the magnetic flux estimator 8 as a reference. It is also possible to switch based on the modulation factor m calculated by equation (17).

  Here, vd * and vq * are quadrature two-phase voltage command values, and vdc is the voltage of the DC power supply of the power conversion unit 10.

  When switching from the estimated phase θ0 output from the magnetic flux estimator 8 to the estimated phase θs output from the phase estimation unit 6, the phase estimation unit 6 takes time until the output is stabilized, as in the magnetic flux estimator 8. Since it is not necessary to ensure the above, it may be switched instantaneously when a predetermined set speed is reached.

  As described above, in the motor control device according to Embodiment 1 of the present invention, the phase estimation unit 6 that estimates the magnetic pole phase in the relatively low speed range and outputs the estimated phase θs and the magnetic pole phase in the relatively high speed range. And a magnetic flux estimator 8 that estimates and outputs the estimated phase θ0, and directly switches from the estimated phase θs to the estimated phase θ0 or from the estimated phase θ0 to the estimated phase θs at a predetermined setting speed. Therefore, the motor can be stably controlled by accurately estimating the magnetic pole phase of the rotor in a wide speed range from the stopped state to the high speed state at low cost with excellent durability. I can do it.

Embodiment 2. FIG.
FIG. 10 is a block diagram showing a configuration of phase estimating unit 6 of the motor control device according to Embodiment 2 of the present invention. Unlike the first embodiment, the motor control device according to the second embodiment has a saliency applied to the motor in addition to the function of estimating the magnetic pole phase using the magnetic saturation characteristics as a phase calculation method in the phase estimation unit 6. In other words, the magnetic pole phase of the rotor is estimated using the characteristic when the magnetic field has an inductance characteristic depending on the magnetic pole phase, and the magnetic pole phase can be estimated with a smaller current.
In the latter estimation of the magnetic pole phase, as will be described later, it is not necessary to use the magnetic saturation characteristic. Therefore, the magnitude of the applied voltage vector and the pulse width can be reduced to reduce the flowing current.

  FIG. 10 shows the block configuration of the phase estimation unit 6 as in the first embodiment. The same or corresponding parts as those in FIG. . 6 is different from FIG. 6 in that a second phase calculation unit 41 is provided in addition to the phase calculation unit 17 (referred to herein as the first phase calculation unit) 17 in FIG. explain.

  In FIG. 10, currents iu, iv, iw and timing signals of the respective phases flowing through the PM motor 14 are input to the second phase calculation unit 41, and the initial value θp of the magnetic pole phase is input from the first phase calculation unit 17. And the correction magnetic pole phase θi from the integration calculation unit 21 is input. The magnetic field phase θp ′ is estimated using the input current values iu, iv, iw.

  At this time, the first phase calculation unit 17 estimates the magnetic pole phase of the rotor using the peak value of the pulse current generated by each voltage vector, as in the first embodiment, but the second phase calculation unit The calculation method of the magnetic pole phase at 41 is different from that of the first embodiment.

  In the estimation calculation of the first phase calculation unit 17, the first magnetic pole phase detection gate signal having such a magnitude and pulse width that the voltage vector applied to the PM motor 14 generates magnetic saturation in a part of the iron core is the pulse generation unit. However, in the estimation calculation of the second phase calculation unit 41, since there is no need to generate magnetic saturation, the second magnetic pole phase detection gate signal having a small magnitude and pulse width of the first magnetic pole phase detection gate signal is obtained. Output from the pulse generator 5.

  Hereinafter, an example of the phase estimation method performed by the second phase calculation unit 41 will be described. First, average values ΔxU ′, ΔxV ′ of current absolute values are obtained by the following equation (18) using the peak values iu1 to iu6, iv1 to iv6, and iw1 to iw6 of the pulse currents flowing by the second magnetic pole phase detection gate signal ΔxW ′ is obtained.

  The absolute current average values ΔxU ′, ΔxV ′, and ΔxW ′ are obtained when the current generated by the voltage vector applied in each direction of the U phase, the V phase, and the W phase is further applied in the opposite direction by 180 degrees in each phase. The current average value is shown. The magnetic pole phase can be estimated using the three types of absolute current average values ΔxU ′, ΔxV ′, and ΔxW ′.

  In an electric motor having saliency characteristics, a difference in inductance occurs according to the magnetic pole phase of the rotor, and a difference appears in the magnitude of the armature current that flows even when the same voltage is applied. For example, when the rotor d-axis is oriented in the same direction as the U-phase (θ = 0 degrees), or the rotor d-axis is advanced (or delayed) 90 degrees from the U-phase (θ = 90 degrees), the inductance is different. As a result, when the voltage vectors P1 to P6 shown in FIG. 3 are applied, the absolute current average values ΔxU ′, ΔxV ′, ΔxW ′ A sinusoidal characteristic having a phase difference of 120 degrees may be exhibited according to the magnetic pole phase.

By using the inductance characteristics depending on the magnetic pole phase, the magnetic pole phase can be estimated from the values of the absolute current average values ΔxU ′, ΔxV ′, and ΔxW ′.
The magnitude and pulse width of the voltage vector applied at this time are not conditions under which nonlinearity appears in the magnetic characteristics, but may be smaller in magnitude and pulse width than conditions under which nonlinearity appears in the magnetic characteristics.

However, the difference in inductance according to the magnetic pole phase appears in a cycle of π [rad] (180 degrees). For this reason, a method for distinguishing between 0 to π [rad] and π to 2π [rad] is necessary.
When describing the measure, first, absolute average values ΔxU ′, ΔxV ′, ΔxW ′ on the three-phase AC axes of the U phase, the V phase, and the W phase, which are 120 degrees apart, are expressed by the equation (19). The values are converted into values Δxα ′ and Δxβ ′ on the α axis and β axis of the orthogonal two-phase coordinate system.

  Based on equation (19), θpp is defined by equation (20).

  Then, taking the sign into consideration and assuming that the magnetic pole phase calculated and output by the second phase calculation unit 41 is θp ′, the magnetic pole phase θp ′ can be obtained by the equation (21) using θpp.

  As described above, the magnetic pole phase θpp of the rotor in the region of 0 to π [rad] can be calculated by the equation (21).

As described above, the second embodiment further requires means for distinguishing the phases of 0 to π [rad] and π to 2π [rad].
For this reason, the magnetic pole phase θp detected by the first phase calculator 17 is used. In the second embodiment, the magnetic pole phase is calculated from the applied voltage vector by two methods, and the magnetic pole phase θp calculated by the first phase calculating unit 17 is given as an initial value of the second phase calculating unit 41.

  For example, in the initial state where the rotor of the PM motor 14 is stopped, the first phase calculation operation is performed, the first phase calculation unit 17 sets the first magnetic pole phase calculation value θp, and the second phase calculation unit 41 sets the first phase calculation operation. A two-pole phase calculation value θp ′ is obtained. Then, the calculated value θp ′ is corrected using the calculated value θp so that the two calculated values θp and θp ′ match.

Specifically, it is determined whether or not the calculated value θp is approached by adding π [rad] to the calculated value θp ′. If the determination is the former, that is, “approaching”, it is assumed that θp ′ + π≈θp. Therefore, the second phase calculation unit 41 sets a value θp ′ + π obtained by adding π to the internally calculated value θp ′. The phase signal θp ′ is output to the switching processing unit 22.
If the determination is the latter, that is, “No (distant)”, it is assumed that θp′≈θp. Therefore, the second phase calculation unit 41 uses the calculated value θp ′ obtained internally as it is as the phase signal θp ′. Is output to the switching processing unit 22 as follows.

  As described above, the phase estimation unit 6 has the magnetic pole phase θp ′ obtained by performing the correction processing based on the result of the first phase calculation described above, and the corrected magnetic pole phase for outputting to the PWM section based on this θp ′. The operation of switching θi by the switching processing unit 22 and outputting it as the first phase signal θs is continued.

  In this case, it is desirable to monitor this deviation and make the necessary corrections using equation (22) so that the deviation between the calculated values θp ′ and θi does not become abnormally large.

  That is, the change in inductance due to the saliency appears not in the 2π [rad] period of the magnetic pole phase but in the period of π [rad]. Therefore, if no processing is performed, the estimation error of the phase signal may exceed π [rad], and there is a possibility that control will be hindered. Therefore, the first phase signal θs calculated by the second phase calculator 41 does not deviate by π [rad] or more after being calibrated by the first phase calculator 17 with the calculated value θp calculated using the magnetic saturation phenomenon. That is why.

  In the calculation of the peak current shown in Expression (18), the same effect can be obtained even if the calculation is performed with one absolute current value without using the absolute average value of the two current values.

  As described above, in the phase estimation unit 6 of the electric motor control device according to the second embodiment of the present invention, the first phase calculation unit 17 that estimates the magnetic pole phase using the magnetic saturation characteristics of the PM motor 14 and the saliency And the second phase calculation unit 41 for estimating the magnetic pole phase. After the correction of the calculation value by the calculation values from the two calculation units, the calculation value of the second phase calculation unit 41 is Since the first phase signal θs is output based on this, the magnitude and pulse width of the voltage vector applied to the PM motor 14 are made smaller, so that the magnetic pole phase can be estimated with less current.

Embodiment 3 FIG.
FIG. 11 is a diagram showing the timing of the switching operation by phase switching unit 9 of the motor control device according to Embodiment 3 of the present invention.
Unlike the first embodiment, the motor control device according to the third embodiment is the second phase output from the magnetic flux estimator 8 from the first phase signal θs output from the phase estimation unit 6 as the switching method in the phase switching unit 9. When switching to the signal θ0, the operation of the magnetic flux estimator 8 is started in a state where a driving current is flowing as a PWM section, and the magnetic flux estimator 8 outputs a phase signal θm used for control after a predetermined time has elapsed. Switch to the second phase signal θ0.

  More specific description will be given with reference to FIG. When the estimated speed ωs by the phase estimator 6 reaches the predetermined set speed described above, the estimated phase θp is obtained at the point (A) where the estimated phase θp is obtained in the last magnetic pole phase detection section before switching based on the arrival time. The switching unit 7 shifts to an operation of outputting only the PWM gate signal and enters the PWM section. Then, after continuing the driving operation based on the predetermined first set time PWM gate signal from the point (A) to the point (D), the output voltage command of the driving state at the timing of the point (D) After setting the values vu *, vv *, vw * and the detected current values iu, iv, iw and the estimated speed ωs to the initial values and operating the magnetic flux estimator 8, a predetermined second set time set in advance The phase signal θm used for control is switched from the estimated phase θs to the estimated phase θ0 at the timing of (E) when the operation of the magnetic flux estimator 8 is stabilized after the passage of the magnetic flux estimator operation section.

  In this case, since the operation of the magnetic flux estimator 8 is started from a state in which the drive operation of the PM motor 14 has advanced by a certain amount, the magnetic flux estimator operation section from the point (D) to the point (E) is the first embodiment. It can be shortened than the case of.

  As described above, in the phase switching unit 9 of the motor control device according to the third embodiment of the present invention, after continuing the driving operation based on the first set time PWM gate signal, the operation of the magnetic flux estimator 8 is started. Thereafter, the phase signal θm used for control is switched from the estimated phase θs to the estimated phase θ0 at the timing when the second set time (the magnetic flux estimator operation section) has elapsed, so that it is ensured to stabilize the operation. The magnetic flux estimator operation section can be shortened.

  It should be noted that the present invention can be freely combined with each other within the scope of the invention, and each embodiment can be appropriately modified or omitted.

1 quadrature two-phase voltage command generator, 2 first coordinate converter, 3 second coordinate converter,
4 PWM control unit, 5 pulse generation unit, 6 phase estimation unit, 7 gate switching unit,
8 magnetic flux estimator, 9 phase switching unit, 10 power conversion unit, 11-13 current detector,
14 PM motor, 17 (first) phase calculation unit, 18 speed calculation unit, 20 speed correction unit, 21 integral calculation unit, 22 switching processing unit, 41 second phase calculation unit, 50 voltage command generation unit, 51 gate signal generation Department.

Claims (13)

A power converter that consists of multiple switching elements that converts the DC voltage from the DC power supply and outputs the AC voltage supplied to the motor. Generates and outputs an AC voltage command value based on the command value input from the upper control system. A voltage command generator for generating, a gate signal generator for generating and outputting a gate signal for controlling on / off of the switching element of the power converter based on the AC voltage command value, and a current detection for detecting a current flowing in each phase of the motor A first phase estimating means for estimating the magnetic pole phase of the motor by calculation under a condition that the rotational speed of the motor is equal to or lower than a predetermined set speed and outputting it as a first phase signal, and the rotational speed of the motor is the set Second phase estimation means for estimating a magnetic pole phase of the motor by calculation under a condition equal to or higher than a speed and outputting the second phase signal as a second phase signal; Phase switching for inputting a second phase signal and selecting either the first phase signal or the second phase signal based on the rotation speed of the motor or information corresponding to the rotation speed and outputting it as a phase signal Part
The voltage command generation unit converts a current value of each phase detected by the current detector based on the phase signal into a quadrature two-phase current detection value of a quadrature two-phase coordinate and outputs the second coordinate conversion unit A quadrature two-phase voltage command generation unit that generates and outputs a quadrature two-phase voltage command value of quadrature two-phase coordinates based on the command value and the quadrature two-phase current detection value; and the quadrature based on the phase signal An electric motor control device including a first coordinate conversion unit that converts a two-phase voltage command value into the AC voltage command value and outputs the same. The gate signal generation unit generates a gate signal for controlling on / off of the switching element to drive the electric motor based on the AC voltage command value generated by the voltage command generation unit, and outputs the gate signal as a drive gate signal A pulse for generating a gate signal for controlling on / off of the switching element for applying a series of voltage pulses to the electric motor for the purpose of estimating the magnetic pole phase of the electric motor by the first phase estimating means and outputting it as a magnetic pole phase detection gate signal In the period in which the generator, the drive gate signal and the magnetic pole phase detection gate signal are input and the first phase estimation means operates, the drive section and the magnetic pole phase detection section are divided at a predetermined period by time division. Set alternately, the drive gate signal in the drive section and the magnetic pole phase detection in the magnetic pole phase detection section Over preparative signals respectively output, in the second period in which the phase estimation means is operated, a gate switching unit to output only the driving gate signals,
The first phase estimating means calculates a magnetic pole phase of the electric motor based on a current value of each phase detected by the current detector when a series of voltage pulses are applied to the electric motor based on the magnetic pole phase detection gate signal. The motor control device according to claim 1, wherein the motor control device outputs the first phase signal. The pulse generator is configured to generate a magnetic saturation in a part of the iron core of the motor when a voltage pulse based on the magnetic pole phase detection gate signal is applied to the motor as the magnetic pole phase detection gate signal. Produces
The first phase estimating means calculates a magnetic pole phase of the electric motor based on a current value of each phase detected by the current detector when a series of voltage pulses are applied to the electric motor based on the magnetic pole phase detection gate signal. The motor control device according to claim 2, wherein the motor control device outputs the first phase signal. When the motor has an inductance characteristic depending on the magnetic pole phase,
The pulse generator is configured to generate a magnetic saturation in a part of the iron core of the motor when a voltage pulse based on the magnetic pole phase detection gate signal is applied to the motor as the magnetic pole phase detection gate signal. A first magnetic pole phase detection gate signal having a pulse width and a second magnetic pole phase detection gate signal having a magnitude and a pulse width that do not cause magnetic saturation in the iron core of the motor,
The first phase estimation means determines the magnetic pole phase of the electric motor based on the current value of each phase detected by the current detector when a series of voltage pulses are applied to the electric motor based on the first magnetic pole phase detection gate signal. A first phase calculation unit for calculating, and a magnetic pole phase of the motor based on a current value of each phase detected by the current detector when a series of voltage pulses are applied to the motor based on the second magnetic pole phase detection gate signal And a second phase calculation unit that outputs a magnetic pole phase obtained by correcting the calculation value based on a calculation value by the first phase calculation unit as the first phase signal. The motor control device described. With the electric motor stopped, the first phase calculation unit and the second phase calculation unit respectively perform a first phase calculation operation to obtain a first magnetic pole phase calculation value and a second magnetic pole phase calculation value, The phase calculation unit adds π (rad) to the second magnetic pole phase calculation value obtained in the first phase calculation operation, so that the added second magnetic pole phase calculation value becomes the first magnetic pole phase calculation value. In the former case, π (rad) is added to the second magnetic pole phase calculation value, and in the latter case, the correction is performed to adopt the second magnetic pole phase calculation value as it is. The motor control device according to claim 4, wherein the magnetic pole phase is output as the first phase signal. The first phase estimator calculates a rotation speed of the electric motor from a magnetic pole phase calculated for each magnetic pole phase detection section that outputs the magnetic pole phase detection gate signal or the second magnetic pole phase detection gate signal. A speed correction unit that outputs a correction speed by correcting the rotation speed calculated by the speed calculation unit in the drive section that outputs the drive gate signal based on the acceleration information of the motor; a correction magnetic pole that integrates the correction speed and corrects the magnetic pole An integral calculation unit for outputting a phase, and the calculated magnetic pole phase is selected at a detection time point when the magnetic pole phase is calculated in the magnetic pole phase detection section, and the corrected magnetic pole phase is selected at a time point other than the detection time point 6. The motor control device according to claim 2, further comprising a switching processing unit that outputs the signal as a single phase signal. The motor control device according to claim 6, wherein the acceleration information of the motor is a change amount per unit time of the rotation speed of the motor calculated for each magnetic pole phase detection section. When a torque command value is input as the command value, the acceleration information of the electric motor is a product of a coefficient dependent on the moment of inertia of the entire rotating system including the electric motor and the torque command value. The motor control device according to claim 6. The second phase estimating means is configured to apply the AC voltage command value and the current value of each phase detected by the current detector when an AC voltage based on the AC voltage command value is applied to the electric motor based on the drive gate signal. 9. The motor control device according to claim 2, wherein the magnetic pole phase of the motor is calculated and output as the second phase signal. The first phase estimating means calculates and outputs a first speed signal together with the first phase signal, and the second phase estimating means calculates and outputs a second speed signal together with the second phase signal. The phase switching unit switches from the first phase signal to the second phase signal based on a point in time when the first speed signal becomes equal to or higher than the set speed. The motor control device according to claim 9, wherein the switching to the one-phase signal is performed based on a time point when the second speed signal becomes equal to or lower than the set speed. As information corresponding to the rotation speed, a modulation factor in a DC / AC conversion operation is calculated from the DC voltage and the quadrature two-phase voltage command value, and the phase switching unit includes the first phase signal and the second phase signal. The motor control device according to claim 9, wherein the switching is performed based on the modulation rate. When switching from the first phase signal to the second phase signal, the second phase estimation means starts operation at the end of the magnetic pole phase detection section before switching, and the phase switching unit 12. The phase signal to be output is switched from the first phase signal to the second phase signal after elapse of a predetermined set time after the estimation means starts the operation. The electric motor control device according to any one of the above. When switching from the first phase signal to the second phase signal, the second phase estimation means starts operation after the elapse of a predetermined first set time from the end point of the magnetic pole phase detection section before switching. The phase switching unit changes the phase signal to be output from the first phase signal to the second phase signal after elapse of a predetermined second set time after the second phase estimation means starts the operation. The motor control device according to claim 9, wherein the motor control device is switched.

Images