JP2012254020A - Linear motor control apparatus - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a linear motor control apparatus capable of suppressing thrust ripples involved in an output voltage amplitude limitation when switching a power converter.SOLUTION: A linear motor control apparatus, supplying electric power to a plurality of stator windings to move a field mover for controlling a linear motor whose stator windings are segmented into a plurality of segments in the moving direction of the field mover by switching power converters provided for each segment on the basis of a field mover position detected by a position sensor, prevents an output voltage limitation or an output current response delay of the power converter when switching the power converters.

Description

  The present invention relates to a linear motor that controls a linear motor in which a stator winding is divided into a plurality of sections in the moving direction of the field mover so as to move the field mover by switching a power converter provided for each section. The present invention relates to a control device.

In the first conventional linear motor control device, a stator winding that is driven by an inverter whose frequency is independently controlled is arranged in each section divided in the moving direction of the moving element. Is arranged such that when the moving element passes the boundary of the adjacent section, the voltage phase and frequency of the stator winding in the subsequent section are controlled to be the same as those in the preceding section, and the stator winding is arranged. It is described that a moving element made of a permanent magnet moving along a traveling path alleviates a shock received when entering a different speed section (see, for example, Patent Document 1).
In addition, in the second conventional linear motor control device, it is described that a position / speed estimator performs induced voltage estimation in order to perform sensorless control (see, for example, Patent Document 2).

Japanese Unexamined Patent Publication No. 04-229092 (page 2-3, FIG. 1) JP 2006-087178 A (page 4-5, FIG. 1)

In general, a linear motor has a peculiar phenomenon in which both ends exist in the length direction (field mover moving direction) or an end effect due to a finite length occurs. Due to this end effect, when the field mover moves, a thrust ripple is generated due to an interphase voltage imbalance at the end of the field mover (particularly, it occurs greatly during high-speed operation where the influence of the induced voltage increases) There was a problem that high accuracy such as constant speed control could not be pursued.
In the first conventional linear motor control device, only a means for mitigating the switching shock when the field mover moves is described, and the phase voltage unbalance at the end of the field mover due to the end effect still remains. There was a problem that the resulting thrust ripple occurred.
In addition, the second conventional linear motor control device does not take into account the temperature change of the stator winding and the sensor measurement error due to energization. There is a problem that the induced voltage estimation accuracy may be greatly deteriorated when an error occurs in the value.
Furthermore, even if the technique described in the second conventional linear motor control apparatus is applied to the technique described in the first conventional linear motor control apparatus, the induced effect cannot be accurately estimated, so that the end effect The phase voltage imbalance at the end of the field mover due to cannot be sufficiently compensated, and as a result, the generation of thrust ripple cannot be solved.

  Still further, the first and second conventional linear motor control devices do not consider the voltage range that can be output from the power converter, for example, switching of the power converter at the time of high thrust command or high speed movement There was also a problem that the thrust ripple that sometimes occurs may not be prevented. That is, the voltage that can be output from the power converter is limited because it is limited depending on the DC bus voltage of the power converter (for example, a DC voltage obtained by rectifying the input three-phase AC power supply). If the output voltage amplitude of the power converter is limited in response to a sudden change in the current command, the response of the current output from the power converter may be delayed more than usual with respect to the current command. . When switching the power converter to be energized, if the response of the output current of each power converter is uneven due to such voltage limitation, the sum of output currents that should be constant before and after switching is constant. Otherwise, thrust ripple may occur. Especially during high thrust commands where a sudden change in the current command occurs, or during high-speed movement that requires a large voltage output due to induced voltage disturbance, such voltage limitation is likely to occur, so there is a possibility that thrust ripple will occur. high. When such thrust ripple occurs, there is a possibility that high accuracy such as constant speed control cannot be pursued in the operation pattern of the field mover including switching in the linear motor control device.

  The present invention has been made in view of such a problem. A linear motor in which a stator winding is divided into a plurality of sections in the moving direction of the field mover is switched to a field converter by switching a power converter provided for each section. Provided is a linear motor control device that controls to move a magnetic mover, and that can reduce thrust ripple associated with output voltage amplitude limitation at the time of switching.

  A representative invention in the present application is that a linear motor in which a stator winding is divided into a plurality of sections in the direction of movement of the field mover is provided with power conversion provided for each section based on the position of the field mover detected by the position sensor. A linear motor control device that supplies electric power to the stator winding so as to move the field mover by switching the voltage, and when switching the power converter to be energized, the voltage command output from the power converter and It is configured so that the output voltage limit of the power converter to be switched does not occur and the total sum of the currents output is kept constant by predicting the current and the initial value operation of the integral controller at the time of switching. It is.

  According to the representative invention in the present application, it is possible to compensate for the thrust ripple when the energized power converter is switched. In addition, switching can be completed quickly without inadvertently delaying the current response due to compensation. Furthermore, by using the prediction calculation, it can be realized with a simple configuration that does not require high-speed communication between power converters or between a host controller and a power converter.

Schematic which shows the structure of the linear motor and linear motor control apparatus in embodiment of this invention. The block diagram which shows the structure of the linear motor control apparatus which concerns on 1st Example of this invention. The block diagram which shows the structure of the electric current control part in the power converter which concerns on 1st Example of this invention. Time change of induced voltage of stator winding 103 Time change of induced voltage of stator winding 104 Time change of induced voltage of stator winding 105 Time change of induced voltage of stator winding 106 Time change of induced voltage of stator winding 107 Time change of induced voltage of stator winding 108 The figure which shows the field mover thrust before the induced voltage compensation which concerns on 1st Example of this invention The figure which shows the field mover thrust after the induced voltage compensation which concerns on 1st Example of this invention The block diagram which shows the structure of the linear motor control apparatus which concerns on 2nd Example of this invention. The block diagram which shows the structure of the current control part in the power converter which concerns on 2nd Example of this invention. The block diagram which shows the structure of the q-axis current controller and linear motor model which concern on 2nd Example of this invention. Time variation of current flowing through the stator winding 104 and the stator winding 108 after voltage compensation Time variation of the sum of currents flowing through the stator winding 104 and the stator winding 108 after voltage compensation Time change of current flowing in the stator winding 104 and the stator winding 108 before voltage compensation Time variation of the sum of currents flowing through the stator winding 104 and the stator winding 108 before voltage compensation

  Hereinafter, embodiments of the present invention will be described with reference to the drawings.

A first embodiment of the present invention will be described below.
FIG. 1 is a schematic diagram illustrating a configuration of a linear motor and a linear motor control device according to an embodiment of the present invention. In the figure, a linear motor 101 includes a field mover 102 and stator windings 103 to 108 arranged in a plurality of sections in the moving direction of the field mover 102. In addition, the linear motor control device in the present embodiment outputs a plurality of power converters 109 to 114 corresponding to the plurality of stator windings 103 to 108 and current commands and the like to the power converters 109 to 114, respectively. And a control unit 115.

The linear motor control device according to the present embodiment controls the linear motor 101 in which one field mover 102 moves with respect to six stator windings 103 to 108 that are continuously arranged in a plurality of divisions. is there. Further, according to the current position of the field mover 102, driving power is supplied from the six power converters 109 to 114 to the six stator windings 103 to 108.
For the sake of simplicity, in this embodiment, the control unit 115 switches the power converter so as to always generate driving force by supplying driving power to the four stator windings. When the stator 102 faces the position of the stator windings 103 to 106, the stator windings 103 to 106, and when it faces the position of the stator windings 104 to 107, the stator windings 104 to 107, the stator. When facing the positions of the windings 105 to 108, drive power is supplied to the stator windings 105 to 108.

FIG. 2 is a block diagram showing the configuration of the linear motor control device. In the figure, the control unit 115 in the linear motor control device has a speed command w * from a host device (not shown), a field mover position x from a position sensor 201 that detects the current position of the field mover 102, and each power. Based on the detected current values i _dn and i _qn from the converters 109 to 114, the induced voltage compensation amounts Δe _dn and Δe _qn and the q-axis current command value i _qn ^* are controlled and calculated to the power converters 109 to 114. Distribute output. The subscript n corresponds to the number of power converters provided in plurality, and in this embodiment, n = 1, 2, 3, 4, 5, 6. The subscript * represents a command value.
Although not shown, the control unit 115 includes a function of controlling operation of the ^* d-axis current command value _{i dn,} distributes outputs the d-axis current command value _{i dn} ^* Each power converter 109-114.

The speed calculator 202 inputs the field mover position x from the position sensor 201 and calculates the field mover speed w by, for example, subtracting the field mover position x from time. The speed controller 203 inputs the deviation between the speed command w * from the host device (not shown) and the field mover speed w, and calculates the thrust command F ^* by, for example, proportional integral control calculation.
The thrust distributor 210 receives the thrust command F ^* and the field mover position x, and distributes and outputs the q-axis current command value i _qn ^* to each of the power converters 109 to 114 by performing, for example, vector control calculation. . Here, in the distribution of the thrust command F ^* , it is assumed that the sum of the thrust commands does not change before and after the distribution, and for example, ¼ of the thrust command F ^* is distributed and output to four stator windings.
The induced voltage compensator 211 receives the field mover position x and the current detection values i _dn and i _qn from the respective power converters 109 to 114, and induces voltage compensation amounts Δe _dn and Δe by a calculation method described later. _qn is calculated and output to each power converter 109-114.

  FIG. 3 is a block diagram illustrating a configuration of a current control unit in the power converter. In the figure, a current control unit in each of the power converters 109 to 114 includes a d-axis current controller 302, a q-axis current controller 303, a two-phase three-phase converter 304, a three-phase two-phase converter 305, and a PWM modulator. 306, and other adder / subtractor. In this current control unit, so-called vector control calculation is performed based on each signal from the control unit 115, and power is supplied to each stator winding of the linear motor.

Hereinafter, a method of calculating the induced voltage compensation amounts Δe _dn and Δe _qn in the induced voltage compensator 211 of the present embodiment will be described. In this embodiment, it is assumed that the linear motor to be controlled is controlled by vector control, but the effect is the same when controlled by three-phase AC control.

First, in the state in which the field mover 102 moves, the induced voltages of all the stator windings across the field mover 102 are estimated (induced voltage estimated values _edn , _eqn ). If the state is handled in the dq coordinate system, the voltage equation of the linear motor is expressed by equation (1).

Here, v _κn ^* , i _κn ^, L _κ (subscript κ = d, q, subscript ^ is an estimated value) is a voltage command value for the d-axis or q-axis of the Nth power converter, Nth Of the d-axis or q-axis of the power converter of FIG. Ra is a stator winding resistance, omega is field mover speed, K _E is the induced voltage constant, p is the differential operator.
Incidentally, the stator winding inductance value is a motor parameter L _kappa, stator winding resistance Ra, the induced voltage constant K _E is a known value that can be known in advance.

In order to calculate the estimated current i _κn ^, first, the equation (1) is converted into a differential equation relating to the estimated current i _κn ^. Expression (2) to Expression (4) are modified with respect to Expression (1), and the resulting differential equation is shown in Expression (5).

  Further, if n is the number of samples, T is the sampling period, and the detected current value is discretized, it is expressed by Expression (6).

The equation (6) is applied to the equation (5), and further transformed through the equation (7) to be converted into a difference equation (8) relating to the command current. The d-axis and q-axis voltage command values v _κn ^* , the stator winding inductance value L _κ , the stator winding resistance value Ra, and the field mover speed ω of the Nth power converter used in the mathematical model are: When it is considered that it is equal to the actual value, the estimated current value i _κn ^ flowing in the d-axis or q-axis is theoretically obtained by the equation (8).

The difference value between the estimated current value i _κn ^ and the actually detected current detection value i _κn corresponds to the estimation error of the induced voltage estimated value. Therefore, the difference value Δi _κn of the current value is multiplied by the induced voltage estimation gain Ke _κ as a proportionality coefficient, and fed back to the induced voltage estimation values _edn and e _qn . That is, the above-described difference value Δi _κn of the current value is obtained by calculating Equation (9) and the induced voltage estimated values _edn and _eqn are calculated by Equation (10).
Since the induced voltage estimation formula of Expression (10) includes a proportional calculation with respect to the current difference value Δi _κn obtained by Expression (9), the induced voltage estimated value e is set so that the above-described current difference value Δi _κn is zero. _It has a function of adjusting _dn and e _qn . Since the induced voltage estimated values e _dn and e _qn when the current difference value Δi _κn becomes 0 are considered to be equal to the true value, the induced voltage estimated values e _dn and e _qn are automatically true according to the equation (10). Converges to a value.

It is desirable that the induced voltage estimation gain Ke _κ is selected so as not to oscillate and is set to converge to a true value quickly.
Here, when comparing the power converter that bears the end of the field mover that is strongly influenced by the end effect and the power converter in the center of the field mover that is not affected by the end effect, modeling of the linear motor Errors, sensor measurement errors, thrust disturbances, etc. are common. Therefore, it can be considered that the difference between the induced voltage estimated values e _dn and e _qn of each power converter obtained by the equation (10) depends only on the end effect of the power converter that bears the field mover end. it can.

Next, the induced voltage compensation amounts Δe _dn and Δe _qn to the power converter at the end of the field mover affected by the end effect are obtained based on the estimated voltage estimated values e _qn and e _dn of each power converter.
Using the field mover position x from the position sensor 201, the positional relationship between each power converter and the field mover is obtained. The power converter that carries the center of the field mover and supplies power to the stator winding in which the flux linkage is stable is the A-th power converter (subscript n = A. For example, in FIG. 111 or equivalent to the power converter 112), and each d, the induced voltage estimated value of q-axis _e dA _(n), and _e qA (n).
Also, the power converter that bears the end of the field mover and has unstable flux linkage is the Bth (subscript n = B, and corresponds to, for example, the power converter 110 and the power converter 113 in FIG. 1). _Assuming that the induced voltage estimated values of the d and q axes are e _dB (n) and e _qB (n).

Estimated induced voltages e _dA (n) and e _qA (n) generated in the power converter A are linear motor modeling errors and sensor measurements due to changes in the temperature of the stator winding in the current operating conditions. It can be handled as a model of normative induced voltage in a state including error and thrust disturbance.
If the estimated voltage estimate of all power converters that generate thrust to the field mover can be controlled to be the same as the power converter that plays the role of the center of the field mover, the optimum operating situation with end effect compensation Can be considered.

The induced voltage compensation amount Δe _dB (n), Δe for the B-th power converter that performs the calculation of the expression (11) in the induced voltage compensator 211 and supplies power to the stator winding that bears the end of the field mover. _qB (n) is determined.

The induced voltage compensator 211 performs the above-described series of calculations based on the field mover position x and the detected current values i _dn and i _qn, and the induced voltage compensation amount Δe _dB (n ), Δe _qB (n) are calculated at any time and are sequentially updated, so that induced voltage compensation is performed on the power converter that supplies power to the stator winding that bears the end of the field mover.
In other words, the induced voltage estimated value _edA (n), _eqA (n) generated in the power converter A is _treated as a model of the normative induced voltage, and this induced voltage estimated value _edA (n), e _{In order to set the difference between qA} (n) and the induced voltage estimated value e _dB (n), e _qB (n) generated in the power converter B as an induced voltage compensation amount for the power converter B, the induced voltage estimation Thus, it is possible to suppress the influence of the modeling error of the linear motor and the sensor measurement error due to the temperature change of the stator winding, and to reduce the thrust ripple caused by the unbalance of the interphase voltage.

Finally, in the power converter, as represented by Expression (12), the induced voltage compensation amounts Δe _dB (n) and Δe _qB (n) calculated by the induced voltage compensator 211 are used as the Nth power converter. Induced voltage compensation amounts Δe _dn and Δe _{qn to} the voltage command values v _dn ^* and v _qn ^* of the power converter that bears the field mover end, and new voltage command values v _dn ^* ′ and v _qn ^* Get ´.
For the power converter that the induced voltage compensator 211 has determined based on the field mover position x from the position sensor 202 that the field mover is not applied to the stator winding, the induced voltage compensation amount. Δe _κn is zero. In this case, the new voltage command values v _dn ^* ′ and v _qn ^* are expressed by Expression (13).

As described above, by controlling the linear motor using the new voltage command values v _dn ^* ′ and v _qn ^* ′, it is possible to reduce the thrust ripple caused by the phase voltage imbalance.
In addition, since the induced voltage estimation method shown in this embodiment does not include a delay element such as a filter, particularly when the current detection cycle is sufficiently short, the compensation effect is exhibited without any problem even when the field mover moves at high speed. Is done.
Furthermore, the present invention is applied to a method of switching a power converter in an ultra-precise device such as a semiconductor manufacturing apparatus having a high-accuracy specification and a linear motor for long-distance conveyance, in which the influence of the end effect of the linear motor is not allowed. By doing so, it is most effective from the viewpoint of the heat generation amount, power consumption amount and current resolution of the linear motor or linear motor control device.

  Here, experimental results when the linear motor is controlled using the induced voltage compensation amount in the present embodiment are shown below. The results of each experiment show that in the configuration of the linear motor in FIG. 1, the field mover 102 moves in a section from the left end of the stator winding 103 to the right end of the stator winding 108 by giving a constant thrust command. Shows the case.

  4 shows the time change of the induced voltage of the stator winding 103, FIG. 5 shows the time change of the induced voltage of the stator winding 104, FIG. 6 shows the time change of the induced voltage of the stator winding 105, and FIG. FIG. 8 shows the time change of the induced voltage of the stator winding 107, and FIG. 9 shows the time change of the induced voltage of the stator winding 108. FIG. In each figure, the vertical axis is the induced voltage amplitude axis, and the horizontal axis is the time axis, and each figure is for the case where the induced voltage compensation amount in this embodiment is not used. Moreover, in each figure, the time when the induced voltage amplitude is 0 means that the field mover 102 does not exist on the stator winding. Further, when the field mover 102 moves at a constant speed, the amplitude of the induced voltage of each phase in the three-phase linear motor becomes almost constant and a three-phase equilibrium state is obtained.

When the field mover 102 moves from the left end of the stator winding 103 to the right end of the stator winding 108, when looking at FIGS. 4 to 9 in order, the stator winding 103 to the stator winding 108. It can be seen that an imbalance of the induced voltage occurs sequentially.
For example, in the stator winding 103 (FIG. 4), since the end of the field mover 102 is taken from when the field mover 102 starts to move, the three-phase unbalanced state from the early stage of the time axis. The induced voltage is generated. In the stator winding 104 (FIG. 5) and the stator winding 105 (FIG. 6), since the field mover 102 starts to move from the start of movement, the center portion of the field mover 102 is assumed. At an early stage, an induced voltage in a three-phase equilibrium state is generated, and as the time axis becomes a late stage (as time progresses), the field mover 102 passes over each stator winding and the field mover. Since an end of 102 is assumed, an induced voltage in a three-phase unbalanced state is generated. In the stator winding 106 (FIG. 7), the stator winding 107 (FIG. 8), and the stator winding 108 (FIG. 9), there is no field mover on the stator winding at an early stage of the time axis. Although the induced voltage amplitude is 0, as the field movable element 102 passes over each stator winding (from the time zone that bears the end of the field mover 102, the time that bears the central part of the field mover 102) As the band changes, the generation of the induced voltage in the three-phase unbalanced state changes to the generation of the induced voltage in the three-phase balanced state.

FIG. 10 is a diagram showing the field mover thrust before the induced voltage compensation, and FIG. 11 is a diagram showing the field mover thrust after the induced voltage compensation of this embodiment. In each figure, the vertical axis is the thrust amplitude axis, and the horizontal axis is the time axis.
Referring to FIG. 10, when the field mover 102 moves from the left end of the stator winding 103 to the right end of the stator winding 108, an induced voltage in a three-phase unbalanced state that occurs in the order of FIGS. 4 to 9. It can be seen that thrust ripple occurs due to the above. On the other hand, it can be seen from FIG. 11 that the thrust ripple is reduced by the induced voltage compensation of the present embodiment.

  A second embodiment of the present invention will be described below. In the first embodiment of the present invention, the technology for reducing the thrust ripple caused by the interphase voltage imbalance at the end of the field mover due to the end effect in the linear motor has been described. In the second embodiment of the present invention, a technique for reducing the thrust ripple caused by the output current response delay of the power converter accompanying the output voltage limitation depending on the DC bus voltage of the power converter will be described.

FIG. 12 is a block diagram showing the configuration of the linear motor control apparatus according to the second embodiment of the present invention. In the figure, the control unit 115 in the linear motor control device is based on a speed command w * from a host device (not shown) and a field mover position x from a position sensor 201 that detects the current position of the field mover 102. The switching compensation selection signal S _n , the predicted voltage command E _q ^* , and the q-axis current command value i _qn ^* are controlled and calculated and distributed to the power converters 109 to 114. The subscript n corresponds to each of the plurality of power converters, and n = 1, 2, 3, 4, 5, 6 as in the first embodiment. The subscript * represents a command value. In addition, since the effect attached to the structure which attached | subjected the code | symbol same as drawing used for 1st Embodiment is the same as 1st Embodiment, the detailed description is abbreviate | omitted.

Switch compensation selector 401, calculates the switching compensation selection signal S _n on the basis of the calculation method described below the field mover position x as input and output to each power converters 109-114. The thrust fluctuation suppressor 402 receives the q-axis current command value i _qn ^* from the thrust distributor 210 as an input, calculates a predicted voltage command E _q ^* by a calculation method described later, and outputs the predicted voltage command E _q ^* to each of the power converters 109 to 114. .

FIG. 13 is a block diagram illustrating a configuration of a current control unit in each of the power converters 109 to 114. In the figure, the q-axis current controller 309 adds a switching compensation selection signal _Sn and a predicted voltage command E _q ^* from the control unit 115 to the q-axis current controller 303 in the first embodiment (FIG. 3). Entered.

FIG. 14 is a block diagram showing a configuration of a q-axis current controller and a linear motor model in each power converter. In the figure, a current controller 309 includes a predictive compensator 504 and other adder / subtracters in addition to a proportional controller 502 and an integral controller 503 for performing proportional-integral control on the motor model 501. Prediction compensator 504, q-axis current command value _{i qn} from control 115 ^*, calculates a voltage compensation value Delta] E _qn ^* based on the switching compensation selection signal _{S n} and the predicted voltage command _E ^{q *.}

Here, the principle of suppressing thrust ripple in the present embodiment will be described using Expressions (14) to (29).
Here, when the power converter is switched, the power converter that is switched to start supplying power is set to the Ath (subscript is n = A), and the power converter that is switched to stop power supply is Bth. (Subscript is n = B). For example, in FIG. 1, power is supplied from the power converters 110 to 113 to the four stator windings 104 to 107 to drive the field mover 102 in the right direction, based on the position of the field mover. When the power converter is switched and the power supply is switched from the power converters 111 to 114 to four of the stator windings 105 to 108, the Ath power conversion that is switched to start the power supply The power converter 108 corresponds to the power converter 108, and the B-th power converter that is switched to stop the power supply corresponds to the power converter 104.

  In FIG. 14, the transfer function Gc of the current controller 309 excluding the predictive compensator 504 is the expression (14), the transfer function Gm of the motor model 501 is the expression (15), and the transfer function Gi of the q-axis current combining them is the expression It is represented by (16). Here, Kp is a proportional control gain, Ti is an integration time constant, Ra is a stator winding resistance, and Lq is a q-axis stator winding inductance. Further, using the stator winding resistance value Ra and the stator winding inductance Lq, the proportional gain Kp = 2π × Kp ′ × L, the integration time constant Ti = L / R, and the q-axis current transfer function Gi is cut. It is regarded as a first-order lag system with an off frequency Kp ′ [Hz].

When the output voltage limitation depending on the DC bus voltage of the power converter does not occur in the Ath and Bth power converters, the sum i of the currents output from the two power converters (Ath and Bth) to be switched _{Since qA} + i _qB becomes a constant value i _qA ^* , thrust ripple does not occur at the time of switching.
On the other hand, when the output voltage limit occurs such as when the q-axis current command value i _qn ^* is large or when a large induced voltage disturbance occurs due to high speed running, the voltage command V _qn ^* is the maximum output voltage. Limited by value ± V _max . As a result, the current response of the power converter whose output voltage is limited is delayed, and the sum i _qA + i _{qB of the} currents output from the two power converters ( _{Ath and Bth} ) to be switched is not kept constant. Thrust ripple will occur.

Normally, the current value i _qn flowing through the stator winding is uniquely determined by an arbitrary q-axis current command value i _qn ^* from the equation (16). However, since the current controller 309 includes the integral controller 503, If the instantaneous integral output is set to an arbitrary value (hereinafter referred to as the initial setting of the integral controller), the current transient response after switching will naturally change.
Therefore, when switching the power converter to be energized, the output voltage limit can be reduced by predicting the voltage command and output current of the two power converters (Ath and Bth) and setting the initial value of the integral controller. It is possible to compensate so that the sum of the two power converter (A-th and B-th) currents that are not generated does not fluctuate.

In order to consider the influence of the output value of the integral controller on the voltage command V _qn ^* and the current value i _qn , the motor model 501 and the current controller 309 in FIG. Represented by It is assumed that the voltage disturbance _VLn such as the induced voltage can be regarded as constant with respect to the current control period in the current controller 309.

Here, the input _{u n} is ^* q-axis current command value _{i qn,} state variables _{x 1n} the q-axis current value _{i qn,} state variables _{x 2n} is integral controller output _{Vi qn} ^*, output _{y 1n} the q-axis current value i _qn and y _2n are q-axis voltage command values V _qn ^* .
Assuming that the time to switch the power converter is t = 0, and that the output current was sufficiently following the current command before that, the input un of the _Ath and Bth power converters and the state variable x _1n, the initial value of _{x 2n} is represented by the formula (19). Here, ΔE _qn ^* is a voltage compensation value, and Ts is a sampling time of the current controller 309.

  If the state equations of Equation (17) and Equation (18) are modified under the condition of Equation (19), Equation (20) relating to the q-axis current response is obtained.

Formula (21) is defined so that the sum i _qA + i _qB of the currents output from the two power converters ( _Ath and _Bth ) to be switched is always a constant value of the q-axis current command value i _qA ^* . Furthermore, if Formula (21) is solved using Formula (19) and Formula (20), Formula (22) will be obtained.

If the voltage compensation value ΔE _qn ^* applied when setting the initial value of the integration controller satisfies the equation (22), the sum i _qA + i of the currents output from the two power converters (Ath and _Bth ) to be switched _qB is maintained at a constant value of the q-axis current command value i _qA ^* . If the condition that the voltage command V _qn ^* does not exceed the maximum value ± V _max of the output voltage is given here, thrust ripple at the time of switching the power converter can be suppressed. Considering that the maximum value of the voltage command V _qn ^* is set to an arbitrary predicted voltage command E _q ^* by performing this voltage compensation, the voltage compensation value ΔE _qn ^* is set.
First, the maximum value y _2nMAX of y _2n (q-axis voltage command value V _qn ^* ) is obtained from the equation (15), and _these are _defined as equations (23) and (24).

In order to satisfy the equation (22) and set the maximum value y _2nMAX of the voltage command as an arbitrary predicted voltage command E _q ^* , the voltage compensation value ΔE _qn ^* is defined as in the equation (25). Here, _VLM is an output voltage model used in the calculation, and means voltage disturbance and voltage loss due to winding resistance. This output voltage model _VLM needs to satisfy Expression (26).

By adding the voltage compensation value ΔE _qn ^* defined in Equation (25) to the output of the integral controller 503 once at the moment of switching the power converter, and setting the initial value of the integral controller, the voltage command The maximum value y _2nMAX of the current can be suppressed to an arbitrary predicted voltage command E _q ^* , and the sum of the currents i _qA + i _qB output from the two power converters (A-th and B-th) to be switched is changed to the q-axis current command. The value i _qA ^* can be kept constant. That is, the thrust ripple at the time of switching a power converter can be suppressed.

Next, an arbitrary predicted voltage command E _q ^* is set. The predicted voltage command E _q ^* may be equal to or less than the output voltage maximum value ± V _max of the power converter. However, when the predicted voltage command E _q ^* is always set to the maximum value ± V _max , when the power converter is switched. A large overshoot or undershoot may occur in the response of the output current i _qn . Therefore, it is necessary to vary the predicted voltage command E _q ^* in accordance with the q-axis current command value i _qn ^* to improve the output current response.
Considering the improvement of the output current response when the output voltage limitation depending on the DC bus voltage of the power converter does not occur, the predicted voltage command E _q ^{* is set} so that the convergence of the output current response is accelerated and no overshoot or undershoot occurs ^. Set. For example, when the transfer function Gi of the q-axis current in the current controller 309 is set as shown in Expression (16), the function of the predicted voltage command E _q ^* is expressed as shown in Expression (27) using the proportionality coefficient α. Defined in

そして、ｑ軸電流応答に関する式（２０）から、Ａ番目およびＢ番目の出力電流応答が任意の極配置になるように比例係数αを定める。α＝Ｋｐとすれば、極配置が通常制御時と同等であり、オーバーシュートやアンダーシュートしない、式（２８）および式（２９）の電流応答式が得られる。つまり、電流応答を悪化させない、推力リプルを抑制するが可能となる。

Then, from the equation (20) regarding the q-axis current response, the proportionality coefficient α is determined so that the A-th and B-th output current responses have an arbitrary pole arrangement. If α = Kp, the current response equations of Equations (28) and (29) are obtained, in which the pole arrangement is the same as that in normal control and no overshoot or undershoot occurs. That is, thrust ripple can be suppressed without deteriorating the current response.

Thus, the principle which suppresses thrust ripple in this embodiment was demonstrated using Formula (14)-Formula (29).
Next, the switching compensation selection signal S _n output from the switching compensation selector 401 in FIG. 12, the predicted voltage command E _q ^* output from the thrust fluctuation suppressor 402, and the voltage compensation amount ΔE _qn output from the prediction compensator 504 in FIG. ^* Calculation and implementation procedures are described.

First, define the switching compensation selection signal _{S n} in the switching compensation selector 401. Switch compensation selection signal S _n is a signal for determining whether to switch the compensation for each power converter, one for the A-th and B-th power converter switch, the other power converter On the other hand, 0 is set. Further, the switching compensation selection signal _Sn is a signal that is output only one pulse at the moment of switching of the power converter, and is always 0 when switching is not performed.

For example, as described above, when switching from the power converters 104 to 107 in FIG. 1 to the power converters 105 to 108, the Ath power converter that is switched to start supplying power is the power converter. 108. Since the B-th power converter that is switched to stop the power supply corresponds to the power converter 104, the switching compensation selection signal _Sn is _expressed by Equation (30). Incidentally, the right side of the equation (30), the switching compensation selection signals S ₁ to the power converter 103 from left to _right, switching the compensation selection signal S ₂ to the power converter _104, switch compensation selection signal S to the power converter 105 _3, switching compensation selection signal _{S 4} to the power converter 106, switch compensation selection signal _{S 5} to the power converter 107, switch compensation selection signal _{S 6} to the power converter 108, a. The switching compensation selection signal S _n in accordance with the field mover position x based on the movement of the field mover 102, the state (1 or 0) is switched.

Next, a predicted voltage command E _q ^* is determined in the thrust fluctuation suppressor 402. Using the output voltage model V _LM , the q-axis current command value i _qA ^{* for} the A-th power converter, and the proportionality coefficient α = Kp, it is determined by equation (27). At this time, the output voltage model V _LM is determined so as to satisfy the equation (26), but the voltage disturbance value V _Ln in the equation (26) may be determined by an induced voltage estimation calculation, a voltage disturbance observer, or the like. A value obtained by subtracting the voltage loss due to the winding resistance from the output value of the integral controller of the current controller may be used. If the output voltage model V _LM satisfies the equation (26), the voltage disturbance value V _Ln used for the calculation may have some error, and when the voltage disturbance is estimated more than the actual value. Thus, a predicted voltage command having a margin with respect to the limit value is created.

Next, the predictive compensator 504 calculates the voltage compensation amount ΔE _qn ^* . The calculation is performed using Equation (31) in the A-th power converter that is switched to start the supply of power, and Equation (32) in the B-th power converter that is switched to stop the power supply. In Expressions (31) and (32), since the switching compensation selection signal S _n is multiplied, the switching compensation selection signal _Sn is 0 in the power converters other than the A-th and B-th, so _{that the} voltage compensation amount ΔE _qn ^* Is 0. Also in the A-th and B-th power converters, the voltage compensation amount ΔE _qn ^* other than the moment of switching of the power converters is zero. The thrust ripple compensation in this embodiment is achieved by adding the obtained ΔE _qn ^* to the output of the integration controller 503.

It should be noted that the thrust ripple compensation in this embodiment can be combined with the first embodiment. In that case, the output voltage maximum value V _max is estimated to be small in advance in consideration of the induced voltage compensation amount Δe _qn in the first embodiment, and the equation (27) is calculated as the equation (33) (induced voltage). When there are a plurality of compensation amounts Δe _qn , the largest one is selected and applied).

The experimental result at the time of using voltage compensation amount (DELTA) _Eqn ^* in the 2nd Embodiment of this invention is shown. Each of the experimental results shown in FIGS. 15 to 18 is based on the power converter from the power converters 110 to 113 to the stator windings 104 to 107 in the configuration of the linear motor and the linear motor control device in FIG. The case where it switches to the electric power supply from 111 to 114 to the stator windings 105-108 is shown.
FIG. 15 shows the time variation of the current flowing through the stator winding 104 and the stator winding 108 after voltage compensation. FIG. 16 shows the time of the sum of the current flowing through the stator winding 104 and the stator winding 108 after voltage compensation. FIG. 17 shows the time change of the current flowing through the stator winding 104 and the stator winding 108 before voltage compensation. FIG. 18 shows the sum of the current flowing through the stator winding 104 and the stator winding 108 before voltage compensation. The time change of is shown. 15 and 17, the current flowing through the stator winding 108 is indicated by a solid line, and the current flowing through the stator winding 104 is indicated by a broken line. In each figure, the vertical axis represents the current amplitude, and the horizontal axis represents the time axis.

  Before voltage compensation, when the power converter is switched (around the time axis of 0.1 s), a ripple is generated in the sum of the currents output from the two switched power converters 110 and 114 (see FIG. 18). On the other hand, after voltage compensation, no ripple is generated in the above-mentioned current sum (see FIG. 16). Therefore, it can be seen that the thrust ripple is reduced by the voltage compensation of the present embodiment. Also, comparing FIG. 17 (before voltage compensation) and FIG. 15 (after voltage compensation), there is no significant delay in the current response output by the power converters 110 and 114, and it corresponds to the output voltage limit. It can be seen that prompt switching of the power converter can be realized.

DESCRIPTION OF SYMBOLS 101 Linear motor 102 Field mover 103-108 Stator winding 109-114 Power converter 115 Control part 201 Position sensor 202 Speed calculator 203 Speed controller 210 Thrust distributor 211 Induced voltage compensator 302 d-axis current controller 303, 309 q-axis current controller 304 two-phase three-phase converter 305 three-phase two-phase converter 306 PWM modulator 307, 308 adder 401 switching compensation selector 402 thrust fluctuation suppressor 501 motor model 502 proportional controller 503 integral control 504 Predictive compensator 505, 506 Adder

Claims (4)

A linear motor in which a stator winding is divided into a plurality of sections in the moving direction of the field mover, and a power converter provided for each section is switched on the basis of the position of the field mover detected by the position sensor. A linear motor control device for supplying electric power to the stator winding so as to move a mover,
A control unit that performs control calculation of a current command according to a command from a host and outputs the command to each of the power converters;
A switching compensation selector that determines the power converter to be switched based on the field mover position and outputs a switching compensation selection signal to the power converter;
Based on the current command, a thrust fluctuation suppressor that calculates a predicted voltage command for preventing output voltage limitation or output current response delay of the power converter and outputs the predicted voltage command to the power converter;
Based on the current command, the switching compensation selection signal, and the predicted voltage command, the power converter that controls and calculates the voltage and current supplied to the linear motor;
A linear motor control device comprising: The power converter performs a current control calculation based on the current command and the detected current value, and calculates a voltage compensation value based on the current command, the switching compensation selection signal, and the predicted voltage command. A predictive compensator to output,
The voltage compensation value is added to the voltage command value that is the output of the current control calculation to calculate a new voltage command value, and the current supplied to the linear motor is detected to calculate the detected current value. The linear motor control device according to claim 1. The thrust fluctuation suppressor determines the predicted voltage command from a first value that is the maximum value of the output voltage of the power converter, or a value obtained by multiplying the current command by a proportional coefficient, depending on voltage disturbance and winding resistance. Set to the second value obtained by subtracting the voltage value according to the voltage loss,
The power converter performs a current control calculation based on the current command and the detected current value, and when the switching compensation selection signal indicates that switching compensation is performed, the current command is multiplied by a predetermined proportional gain. A value obtained by adding the value obtained by integrating the current command and a value obtained by integrating the current command with a predetermined integration time constant from the predicted voltage command, or a value obtained by adding a voltage value corresponding to the voltage loss to the subtraction result or The new voltage command value is calculated by adding the voltage compensation value to the voltage command value that is the output of the current control calculation, and using the value obtained by reversing positive and negative as the voltage compensation value. The linear motor control device described.   The linear motor control device according to claim 1, wherein the control unit and the power converter perform vector control or three-phase AC control.

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