JP2008295125A - Controller for voltage-type inverter - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a controller for voltage-type inverter which is applicable to V/f constant control and vector control and is capable of minimizing the output voltage distortion of an inverter without complicated circuit constitution or adjustment. <P>SOLUTION: This controller includes a means which calculates first output voltage command values v<SB>dn</SB>* and v<SB>qn</SB>* where counter electromotive force components and interferent species are excluded from the output current command value of an inverter 1, a current detecting means 6, a voltage strain estimating means 20 which estimates an output voltage from an output current detected value and estimates voltage strain components from the first output voltage command values v<SB>dn</SB>* and v<SB>qn</SB>* and an output voltage estimate value and gets voltage corrections values Δv<SB>d</SB>and Δv<SB>q</SB>from the voltage strain component estimate value, and a means which generates final output voltage command values v<SB>d</SB>* and v<SB>q</SB>* to the inverter 1 by correcting the second output voltage command values v<SB>dO</SB>* and v<SB>qO</SB>* where the counter electromotive force components and interferent species are added to the first output voltage command values v<SB>dn</SB>* and v<SB>qn</SB>*, by the voltage correction values Δv<SB>d</SB>and Δv<SB>q</SB>. <P>COPYRIGHT: (C)2009,JPO&INPIT

Description

  The present invention relates to a control apparatus for a voltage source inverter, and for example, when an AC motor is driven by a voltage source inverter that outputs an AC voltage of a desired magnitude and frequency by turning on and off a semiconductor switching element, the output voltage of the inverter The present invention relates to a control device for compensating for errors and distortions included in.

In a normal voltage source inverter, semiconductor switching elements constituting upper and lower arms are alternately conducted to generate an AC voltage having a desired magnitude and frequency. However, as is well known, the switching element has an operation delay at the time of turn-off, and thereby a short-circuit prevention period (dead time) is provided in the switching pattern in order to prevent an arm short circuit due to the upper and lower arms being simultaneously turned on.
However, due to this dead time, the output voltage of the voltage source inverter has an error with respect to the output voltage command, and distortion of the output voltage due to this causes torque ripple when the motor is driven.

As a conventional technique for compensating for the distortion of the output voltage due to the dead time described above, as described in Patent Document 1, the output voltage distortion is estimated based on the output voltage command value and the output current of the inverter, A method of adjusting the amplitude (compensation amount) of a compensation signal to be added to an output voltage command value based on this estimated value is known.
The prior art will be described below with reference to FIG.

In FIG. 5, reference numeral 101 denotes a voltage-type pulse width modulation (PWM) inverter that outputs a voltage proportional to the voltage command value, and reference numeral 102 denotes an AC motor driven by the inverter 101.
In the inverter 101, a block 101 a indicated by a broken line models that the output voltage distortion Δv due to the dead time described above occurs according to the polarity of the output current i detected by the current detection unit 110. .

Reference numeral 103 denotes a d-axis voltage command value v _d ^{* in} the direction of the magnetic flux axis of the electric motor 102 in the rotating magnetic field coordinate system, a q-axis voltage command value v _q ^* orthogonal thereto, and a phase reference signal θ ^* (= ω ^* t). Is inputted to the coordinate transformation means 109 for outputting the output voltage command value v ^* of the stator coordinate system, 109 is an addition means for adding the output voltage command value v ^* and a compensation signal Δv ^* from the compensation means 105 described later, 106 Is a coordinate conversion means for converting the output current i into a d-axis current i _d and a q-axis current i _q in the rotating magnetic field coordinate system, 107 is a d-axis voltage command value v _d ^* , a q-axis voltage command value v _q ^* , a d-axis Distortion component estimation means for estimating a voltage distortion component (compensation residual component) based on the current _id and the q-axis current _iq , and a compensation amount for adjusting the compensation amount based on the estimated voltage distortion component and the output current i Adjusting means 108a is polarity inversion of voltage distortion component Means 108b is a polarity detecting means for outputting a switching signal according to the polarity of the output current i, 108c is an integrating means for integrating the output of the distortion component estimating means 107 selected by the switching signal or its inverted output, and 105 is an output. Compensating means for outputting a compensation signal Δv ^* corresponding to the polarity of the current i to the adding means 109.

In the above configuration, the output voltage distortion Δv can be estimated by a model in which a distortion component acts on a path from the voltage command values v _d ^* , v _q ^{* of} each axis to the output voltage of the inverter 101, and distortion component estimation is performed. The means 107 estimates a voltage distortion component according to the principle of a disturbance observer using the d-axis voltage command value v _d ^* , the q-axis voltage command value v _q ^* , the d-axis current i _d and the q-axis current i _q . The compensation amount adjusting means 108 controls the compensation signal Δv ^* from the compensation means 105 so that the voltage distortion component approaches zero, and the addition means 109 adds this to the original output voltage command value v ^*. Thus, the output voltage distortion Δv is minimized.

That is, by adding the compensation signal Δv ^* to the original output voltage command value v ^* and performing feedforward control of the output voltage distortion Δv, the output voltage v is suppressed from being included and compensated for the output voltage distortion. is doing. The amplitude of the compensation signal (compensation amount) has the largest compensation effect when it matches the amplitude of the voltage distortion component, and a compensation residual component is generated even if it is larger or smaller than that. Whether the compensation amount is excessive or insufficient is determined by 108, and the compensation amount is controlled to the optimum value via the compensation means 105 based on the determination result.

However, in the above prior art, the voltage distortion component estimated by the distortion component estimation means 107 serving as a disturbance observer includes the back electromotive force (speed electromotive force) of the motor 102, so that the estimated distortion component is output as it is. The voltage command value cannot be compensated. For this reason, it is necessary to adjust the compensation amount using the compensation amount adjusting unit 108 and the compensation unit 105.
The compensation amount adjusting means 108 and the compensating means 105 determine overcompensation or undercompensation based on the waveform of the voltage distortion component by performing integration, a difference calculation of the integral value, etc., and compensating according to the discrimination result. Complicated processing of determining the amount is performed, which leads to complicated circuit configuration and increased calculation load.

In view of the above points, the applicant has already applied for a voltage source inverter control device capable of solving the above problems as Japanese Patent Application No. 2007-6687.
FIG. 6 is a block diagram showing the configuration of the invention of the prior application, wherein 1 is a voltage-type PWM inverter that outputs an AC voltage having a desired magnitude and frequency, and 2 is an induction connected to the output terminal of each phase of the inverter 1. motor, 3 a frequency setting device for setting the primary angular frequency command omega ₁ ^* for an induction motor 2, integrated and integrating means for outputting a phase reference signal theta ₁ ^* of the primary angular frequency command omega ₁ ^* 4, 5 rotate The d-axis voltage command value v _1d ^* and the q-axis voltage command value v _1q ^* in the magnetic field coordinate system are used as the phase voltage command values v _u ^* , v _v ^* , v in the stator coordinate system based on the phase reference signal θ ₁ ^*. Coordinate conversion means for converting to _w ^* .

Reference numeral 6 denotes current detection means for detecting the output currents i _u , i _v , i _w of the inverter 1, and 7 denotes the output currents i _u , i _v , i _w of the stator coordinate system based on the phase reference signal θ ₁ ^* . A coordinate conversion means for converting the d-axis current i _1d and the q-axis current i _1q in the rotating magnetic field coordinate system, and V / f conversion means 8 for outputting the q-axis voltage command value v _1q ^* from the primary angular frequency command ω ₁ ^*. , 9 is a high-speed disturbance estimation observer as a first estimating means for estimating and outputting the disturbance voltage v _dis from the q-axis voltage command value v _1q ^* and the q-axis current i _1q , and 10 is the q-axis voltage command value v _1q ^*. And a low-speed disturbance estimation observer as second estimation means for estimating and outputting the back electromotive force component v _emf from the q-axis current i _1q , 11 is a d-axis current i _1d , a primary angular frequency command ω ₁ ^*, and a leakage inductance L d-axis interference component v using _σ d-axis interference component calculation means as third estimation means for calculating _cmp , 12 is a current control for outputting a d-axis voltage command value v _1d ^* with the d-axis current command value i _1d ^* and the d-axis current i _1d as inputs. means, 13 subtraction means for subtracting the d-axis interference component _{v cmp} from the counter electromotive force component _{v emf,} 14 is subtracted for outputting a value obtained by subtracting the output of the subtraction means 13 from the disturbance voltage _{v dis} as a compensation voltage _{v c} means, 15 is the original of the q-axis voltage command value v _1q ^* and the compensation voltage v _c and q-axis voltage command value v _1q ^* output adding means corrected by adding.

Next, the operation of FIG. 6 will be described. A well-known V / f constant control is used for the control method in the prior invention.
First, the primary angular frequency command ω ₁ ^* set by the frequency setting unit 3 is input to the V / f conversion unit 8 and a primary voltage command value (output voltage command value) corresponding to the primary angular frequency command ω ₁ ^* is obtained. Is output. Here, since the d-axis component of the voltage command value does not directly affect the torque generated by the induction motor 2, the V / f conversion means 8 outputs the q-axis voltage command value v corresponding to the primary angular frequency command ω ₁ ^*. _1q ^* is output as the primary voltage command value.
Hereinafter, q-axis voltage command value _{v 1q} ^* of the primary voltage command value _v ^1q *, the primary voltage of the q-axis voltage _{v 1q v 1q,} also referred to as primary current _{i q} and q-axis current _{i q.}

The primary voltage command value v _1q ^* is input to the coordinate conversion means 5, and output voltage command values v _u ^* , v _v ^* , v _w of sine waves are obtained by coordinate conversion using the phase reference signal θ ₁ ^* from the integration means 4. ^* Is generated. By turning on / off the semiconductor switching element of the inverter 1 in accordance with the PWM signal obtained by comparing these voltage command values v _u ^* , v _v ^* , v _w ^* with the carrier wave signal inside the inverter 1, an output voltage is obtained. v _u , v _v and v _w are controlled and supplied to the induction motor 2.

On the other hand, the output voltage distortion caused by the dead time of the inverter 1 is compensated by the disturbance voltage v _dis estimated by the high-speed disturbance estimation observer 9 in the rotating magnetic field coordinate system.
Here, when the relationship between the primary current i _1q and the primary voltage v _1q of the induction motor 2 is obtained from the voltage equation of the induction motor expressed in the rotating magnetic field coordinate system given by the formula 1, the formula 2 is obtained.

In these equations, φ _2d , φ _2q : d-axis motor flux and q-axis motor flux, v _1d , i _1d : d-axis voltage and d-axis current on the primary side of the induction motor, v _1q , i _1q : of the induction motor Q-axis voltage and q-axis current on the primary side, ω ₁ : primary angular frequency, ω _m : rotational angular frequency (electrical angular velocity), R ₁ : primary resistance value, R ₂ : secondary resistance value, L _σ : leakage inductance, L _m : excitation inductance, p: differential operator.

In Equation 2, the second term on the right side is the interference component due to the d-axis, and the fourth term is the back electromotive force component. The third term on the right side is considered to be zero because there is no effect at low speed.
When the sum of the error voltage due to the dead time of the inverter 1 and the d-axis interference component and the back electromotive force component is defined as the disturbance voltage v _dis , Equations 2 to 3 can be obtained.

Here, it is considered that the disturbance voltage v _dis is estimated by a disturbance observer. That is, based on the relationship between the primary current i _1q and the primary voltage v _1q shown in Formula 3, the output voltage (= (R ₁ + R ₂ + pL _σ ) i _1q ) of the inverter 1 applied to the induction motor 2 is primary. Estimated from current i _1q .
Then, the high-speed disturbance estimation observer 9 takes the difference between the estimated output voltage (= (R ₁ + R ₂ + pL _σ ) i _1q ) and the primary voltage command value v _1q ^* as shown in the following equation (4). multiplied by a gain K through T ₁ of the low-pass filter, and outputs as a disturbance voltage estimation value _{^ v dis.} In Equation 4, any value with the symbol “^” is an estimated value.

Here, as the disturbance component due to the dead time, when the primary output frequency is f (= ω ₁ / 2π), an AC ripple component having a frequency six times that appears on the q axis. Accordingly, the time constant T ₁ in Equation 4 is set to the range shown in Equation 5, for example, when the disturbance voltage is estimated in a time sufficiently faster than 6 times the output frequency.

Further, as described above, the disturbance voltage estimated value ^ _vdis includes the back electromotive force component of the induction motor 2 and the interference component due to the d axis in addition to the error voltage due to the dead time. The back electromotive force component is also compensated, and the voltage becomes excessive during open loop control such as V / f control, which causes a problem in stability.
Accordingly, the counter electromotive force component v _emf of the fourth term on the right side in Equation 2 is calculated by the low-speed disturbance estimation observer 10, and the d-axis interference component v _cmp (= ω) of the second term on the right side is calculated by the d-axis interference component calculation unit 11. ₁ L _σ i _1d ) is calculated and inputted to the subtracting means 13 and 14 with the sign shown in the figure, thereby compensating the disturbance voltage estimated value ^ v _dis which is the output of the high-speed disturbance estimation observer 9 and compensating voltage v _c. Is generated.

Slow disturbance estimation observer 10 constitutes a disturbance estimation observer for outputting the estimated back EMF ^ v _emf shown has the same structure as equation 4, the time constant in the formula 6 and T _2.

Here, the back electromotive force v _emf depends on the rotational angular frequency ω ₁ from Equation 2, and the change is very slow with respect to the disturbance component due to the dead time. Therefore, the time constant T ₂ in Equation 6 can be estimated by setting the back electromotive force v _emf to an extent that the disturbance component due to the dead time can be neglected. For example, the time constant T ₂ is set in the range shown in Equation 7.

The estimated back electromotive force ^ v _emf output from the low-speed disturbance estimation observer 10 includes the d-axis interference component v _{cmp of} the second term on the right side of Equation 2, but it is moderate if it is estimated to v _cmp and compensated as a disturbance component. Since stability deteriorates in the high-speed region, this is calculated using the primary angular frequency ω ₁ , the d-axis current i _1d and the leakage inductance L _σ as shown in Formula 8, and is compensated forward.

The d-axis interference component v _cmp calculated as described above is compensated by subtracting it from the estimated counter electromotive force ^ v _emf by the subtracting means 13. The output of the subtraction means 13 compensates for the disturbance voltage estimation value ^ v _dis is input to the next stage of the subtraction means 14, and outputs the result as a compensation voltage v _c.

Compensation voltage v _c since added to the primary voltage command value v _1q ^* in addition means 15, the result, estimates only error voltage due to dead time, excluding the counter electromotive force corresponding amount from the disturbance voltage as a voltage distortion component A system for compensating the primary voltage command value v _1q ^* can be configured. Thereby, the output voltage distortion of the inverter 1 can always be minimized.

Japanese Patent No. 3536114 (paragraphs [0007] to [0012], FIG. 1 etc.)

The prior invention shown in FIG. 6 proposes a disturbance observer applicable to V / f constant control. However, a high-speed disturbance estimation observer 9 with a fast time constant and a low-speed disturbance estimation observer 10 with a slow time constant are combined. If the time constants of these two observers 9 and 10 are not set optimally, The voltage distortion component may increase. Since adjustment of the time constant requires empirical knowledge, the prior invention has a problem that the processing becomes complicated.
Further, the invention of the prior application is intended for simple V / f constant control, and it is difficult to apply to vector control or a method of controlling the torque by calculating the voltage command value forward from the current command value. .

  Therefore, the problem to be solved by the present invention can minimize the output voltage distortion of the inverter without requiring a complicated circuit configuration or adjustment, and can be applied not only to constant V / f control but also to various control methods such as vector control. It is an object of the present invention to provide a control device for a voltage source inverter that can be used.

In order to solve the above-mentioned problem, in claim 1, the output voltage command value is calculated from the output current command value of the inverter, and the difference between the output voltage command value obtained from the output current detection value and the output voltage command value is calculated. The voltage distortion component is estimated, and the output voltage command value is corrected using the estimated voltage distortion component value.
That is, the invention according to claim 1 is a control device for a voltage source inverter that outputs an alternating voltage of a desired magnitude and frequency by turning on and off a semiconductor switching element.
Means for calculating a first output voltage command value obtained by removing the back electromotive force component and the interference component from the output current command value of the inverter;
Current detection means for detecting the output current of the inverter;
Means for estimating the output voltage of the inverter from the output current detection value of the inverter;
Means for estimating a voltage distortion component from the first output voltage command value and the output voltage estimated value of the inverter;
Means for obtaining a voltage correction value from the estimated voltage distortion component value;
The final output voltage command value for the inverter is generated by correcting the second output voltage command value obtained by adding the back electromotive force component and the interference component to the first output voltage command value with the voltage correction value. Means.

The voltage distortion obtained from the output voltage command value and the estimated output voltage value by calculating the output voltage command value from the output current command value that is a forward feedforward component in vector control or the like having current adjusting means. The output voltage command value is corrected using the component estimated value.
That is, the invention according to claim 2 is characterized in that, in claim 1, the current adjusting means that operates so as to eliminate the deviation between the output current command value of the inverter and the detected output current, and the output of the current adjusting means is the second output. Means for adding to the output voltage command value.

4. The voltage distortion obtained from the output voltage command value and the estimated output voltage value by calculating the output voltage command value from the output current command value which is a forward feedforward component in vector control or the like having a current adjusting means. The effect of the current adjusting means is improved by correcting the output current command value using the component estimated value.
That is, the invention according to claim 3
In a control device for a voltage source inverter that outputs an alternating voltage of a desired magnitude and frequency by turning on and off a semiconductor switching element,
Means for calculating a first output voltage command value obtained by removing the back electromotive force component and the interference component from the output current command value of the inverter;
Current detection means for detecting the output current of the inverter;
Means for estimating an output voltage of the inverter from an output current detection value of the inverter;
Means for estimating a voltage distortion component from a first output voltage command value and an output voltage estimated value of the inverter;
Means for obtaining a current command correction value from this voltage distortion component estimated value;
Means for correcting the output current command value by the current command correction value;
Current adjusting means that operates so as to eliminate a deviation between the output current command value corrected by this means and the output current detection value;
By correcting the second output voltage command value obtained by adding the back electromotive force component and the interference component to the first output voltage command value by the output of the current adjusting means, the final output voltage command value for the inverter is obtained. Generating means.

  According to a fourth aspect of the present invention, in the first to third aspects, the means for calculating the first output voltage command value includes an output current command value on the rotation coordinates of the inverter, a resistance value and an inductance value of a load such as an AC motor. Is used to calculate the output voltage command value.

  According to a fifth aspect of the present invention, in the first to fourth aspects, the means for estimating the output voltage of the inverter uses an output current detection value on the rotation coordinate of the inverter, a resistance value and an inductance value of a load such as an AC motor. The output voltage is estimated.

  According to the present invention, a voltage distortion component due to an error between the output voltage command value of the inverter and the estimated output voltage value is compensated, and for example, torque ripple and rotation unevenness when a motor is used as a load are improved to achieve high efficiency and high accuracy. A control device can be provided. The present invention can be realized with a simple configuration, and can be applied to a wide range of control such as vector control as well as V / f control without performing complicated adjustment.

Hereinafter, embodiments of the present invention will be described with reference to the drawings.
First, the present invention differs from the prior invention in that the voltage distortion component is estimated using the output voltage command value calculated from the output current command value of the inverter and the output voltage estimated value calculated from the output current detection value. The output voltage command value is corrected using the correction value obtained from the estimated voltage distortion component value.

FIG. 1 is a block diagram showing a first embodiment of the present invention, which corresponds to claims 1, 4, and 5.
In this embodiment, as shown in FIG. 1, the d-axis voltage command value v _d0 ^{* is} determined from the d-axis current command value i _d ^* and the q-axis current command value i _q ^* according to the voltage equation of the induction motor 2 shown in Formula 9 ^. The q-axis voltage command value v _q0 ^* is calculated. Here, the d-axis voltage command value v _d0 ^* and the q-axis voltage command value v _q0 ^* are referred to as a second output voltage command value for convenience.
This Equation 9 is the same as Equation 1 described above except that the d-axis and q-axis voltage command values v _d0 ^* and v _q0 ^* are calculated using the d-axis and q-axis current command values i _d ^* and i _q ^*. It is the same.

However, in Formula 9, φ _2d , φ _2q : secondary magnetic flux, R ₁ , R ₂ : primary and secondary resistance, L _σ : leakage inductance, L _m : mutual inductance (all values converted from a T-type equivalent circuit) ), Ω ₁ : electrical angular frequency, ω _m : mechanical angular frequency, p: differential operator.
The parameter may be a detected value (actual value) or an estimated value.

In FIG. 1, as an example, a configuration in which φ _2q = 0 in Formula 9 is shown, and the present invention can be applied to a configuration in which Formula 9 is simplified in order to simplify the control.
In FIG. 1, the interference term is a component by ± ω ₁ L _σ (set value (estimated value) of a voltage drop component by reactance), and the back electromotive force is a back electromotive force component of the load by ω _m φ _2d . Reference numerals 21 to 25 denote addition / subtraction means.

In FIG. 1, the calculation is based on the voltage equation of the induction motor. However, even when the synchronous motor is a target, it can be applied by changing the voltage equation. In general, there are many means for calculating the current command value based on the control target such as speed control and torque control. However, the present invention does not particularly focus on the method for calculating the current command value, and the d The axis current command value i _d ^* and the q axis current command value i _q ^* may be given.

The voltage correction values Δv _d and Δv _q calculated by the voltage distortion estimator 20 are added to v _d0 ^* and v _q0 ^* calculated as described above, respectively, so that final d-axis voltage command values v _d ^* and q The shaft voltage command value v _q ^* is obtained. The voltage command values v _d ^* and v _q ^* are substantially the same as the voltage command values v _1d ^* and v _1q ^* in FIG.
These voltage command values v _d ^* and v _q ^* are converted into a stationary coordinate system by the coordinate conversion means 5 to obtain an output voltage command value corresponding to the angular frequency ω ₁ . Thereafter, the PWM generator in the voltage-type PWM inverter 1 calculates the pulse pattern of the inverter 1 and supplies the desired output voltage to the induction motor 2 by turning on and off the semiconductor switching element according to the pattern. To do.

On the other hand, the current detection means 6 detects the current flowing through the electric motor 2 based on the output voltage of the inverter 1 and converts it into a rotating coordinate system by the coordinate conversion means 7 to obtain i _d and i _q . The detected current values i _d and iv _q are substantially the same as the detected current values i _1d and i _1q in FIG.
These current detection values i _d and i _q are input to the voltage distortion estimation means 20, which estimates the output voltage of the inverter 1 and estimates the voltage distortion component from the estimated voltage value and the voltage command value. Further, voltage correction values Δv _d and Δv _q are calculated.

Next, the structure of the voltage distortion estimation means 20 which is the principal part of this embodiment is demonstrated.
In the voltage distortion estimating means 20, first, the output voltage estimated values {circumflex over (v) _} and {circumflex over (v ₎ } _q are calculated from the d-axis and q-axis current detection values i _d and i _q on the rotation coordinates by the following equation (10).

Where τ _{2 is the} motor secondary time constant (= R ₂ / L _m ).

In Equation 10, the output voltage estimated value of q-axis side ^ v _q is the same as the prior application 2 of the first term formulas in the invention, also calculates an output voltage estimated value ^ v _d for the d-axis side here . The resistance values R ₁ and R ₂ and the inductance values L _σ and L _m of the electric motor 2 are the same as those used for the calculation of the voltage command value described above.

On the other hand, the d-axis and q-axis voltage command values v _dn ^* and v _qn ^* excluding the back electromotive force component and the components that interfere with each other on the d-axis and q-axis are calculated by Equation 11, and these voltage command values are converted into voltage distortions. Input to the estimation means 20. Here, the d-axis voltage command value v _dn ^* and the q-axis voltage command value v _qn ^* are referred to as a first output voltage command value for convenience.

Equations 10 and 11 are exactly the same when there is no voltage distortion. If voltage distortion exists, the voltage distortion causes the current detection values i _d and i _{q to be} between current detection values i _d ^* and i _q ^*. An error occurs.
Taking the difference between Equation 10 and Equation 11 as the voltage distortion component estimated values ^ v _drip and ^ v _qrip , these are expressed by Equation 12.

The d-axis and q-axis voltage correction values Δv _d and Δv _q are obtained by Equation 13 as a disturbance observer through the low-pass filter (1 / (1 + sτ)) and the gain K to the voltage distortion component obtained by Equation 12.

FIG. 4A is a block diagram showing the configuration of the voltage distortion estimation means 20. In FIG. 4 (a), 20a is a block for calculating the block, 20b, 20c is _{^ v drip, ^} v subtraction means for calculating a _qrip, 20d is a formula 13 for calculating the formula 10.

In the present embodiment, the d-axis and q-axis voltage correction values Δv _d and Δv _q obtained by Expression 13 are used as second output voltage command values (back electromotive force component and interference component) by the addition / subtraction means 23 and 25 in FIG. output voltage command _value) ^v d0 * _{including, v q0} ^* to sum respectively.
Here, in the invention of the prior application shown in FIG. 6, since the back electromotive force component is superimposed on the voltage command value, two observers 9 and 10 of high speed and low speed are prepared, the time constant is optimally selected, and the voltage distortion Only the components must be extracted. On the other hand, in the present embodiment, the back electromotive force component is not originally included in the expression 11 related to the first output voltage command value, and the observer time constant τ is simply set higher than the frequency of the voltage distortion component. Because it is good, no complicated adjustment is required.
Therefore, according to the present embodiment, it is possible to easily compensate for the voltage distortion component with a simple configuration as compared with the prior invention.

  Next, FIG. 2 is a block diagram showing a second embodiment of the present invention, which corresponds to claims 2, 4 and 5. In this embodiment, the present invention is applied to a case where a current adjusting means for correcting a deviation between a current command value and a current detection value is used in vector control or the like.

In FIG. 2, the difference from the first embodiment shown in FIG. 1 is as follows.
That is, deviations between the d-axis and q-axis current command values i _d ^* and i _q ^* and the d-axis and q-axis current detection values i _d and i _q are obtained by the addition / subtraction means 31 and 32, respectively. Input to means 30. The current adjusting means 30 performs an adjusting operation so as to eliminate each of the current deviations, and the output is added to the second output voltage command values v _d0 ^* and v _q0 ^* by the adding and subtracting means 26 and 27, respectively. The subsequent configuration and operation are the same as those in the first embodiment, and the configuration of the voltage distortion estimation means 20 is also as shown in FIG.

Hereinafter, the reason why the second embodiment can be applied to vector control will be described.
The current adjusting means normally calculates the voltage command value so that the deviation between the current command value and the current detection value becomes zero using PI control or PID control. Therefore, when the performance of the current adjusting means is sufficiently high, the voltage distortion component can be corrected by the current adjusting means. However, in reality, the current adjusting means has a limited response due to the calculation time of the control circuit and the like, and compensation of the voltage distortion component becomes insufficient at a high frequency.

On the other hand, in the present embodiment, only the voltage distortion component that cannot be compensated for by the current adjusting means 30 can be supplementarily compensated by the voltage correction values Δv _d and Δv _q from the voltage distortion estimating means 20 described above. . This is because the low-pass filter (1 / (1 + sτ)) and the gain K in Equation 13 are designed to have a higher response than that of the current adjustment unit 30, and equivalently a high response in parallel with the current adjustment unit 30. This is because accurate feedback control is configured.
As described above, this embodiment is preferably applied to control such as vector control using the current adjusting means.

FIG. 3 is a block diagram showing a third embodiment of the present invention. This embodiment corresponds to claims 3, 4 and 5.
In this embodiment, the voltage distortion estimation means 20 _calculates current command correction values Δi _d ^* and Δi _q ^* from the voltage distortion component estimated values ^ v _drip and ^ v _qrip shown in Equation 12, and these current command corrections are performed. value Δi _d ^*, _d-axis which is the input of the current regulating means 30 by Δi _q ^*, _q-axis current command value _i ^d _*, corrects _i ^{q *,} respectively. With such a configuration, the control performance of the current adjusting means 30 is improved and the voltage distortion component is effectively removed.

That is, the voltage distortion estimation unit 20, the calculation of the following equation 14, the voltage distortion component estimation value _^ v _{drip, ^ v} from _qrip by the high-pass filter _{(sτ a / (1 + sτ} a)) to extract the high frequency components, further gains current command correction value by multiplying a K _a Δi _d ^*, seek .DELTA.i _q ^*.
Here, FIG. 4 (b) is a block diagram showing a structure of a voltage distortion estimation unit 20, 20a is a block which calculates a formula 10, 20b, 20c is _^ v _drip, addition and subtraction means for calculating a _{^ v qrip} , 20e are blocks for calculating Formula 14.
As shown in FIG. 3, the current command correction values Δi _d ^* and Δi _q ^* are added to the original current command value in the addition / subtraction means 33 and 34 and input to the current adjustment means 30 through the addition / subtraction means 31 and 32.

As described above, according to the present embodiment, a disturbance component such as a dead time is extracted from the voltage distortion component, and the output current command value is corrected so as to cancel it, thereby reducing the output voltage distortion and the electric motor 2. Generation of torque ripple or the like can be prevented.
The current command correction values Δi _d ^* and Δi _q ^* are obtained from the voltage distortion component, the high-pass filter, and the gain as described above, and the distortion components included in the actual current detection values i _d and i _q are extracted. It may be obtained by multiplying the gain.

  Each of the above embodiments is directed to the case where the induction motor 2 is driven by the voltage-type PWM inverter 1, but the present invention can also be applied to a drive system of another motor such as a synchronous motor.

1 is a block diagram showing a first embodiment of the present invention. It is a block diagram which shows 2nd Embodiment of this invention. It is a block diagram which shows 3rd Embodiment of this invention. It is a block diagram which shows the structure of the voltage distortion estimation means in each embodiment. It is a block diagram which shows the prior art described in patent document 1. It is a block diagram which shows prior invention.

Explanation of symbols

1: Voltage type PWM inverter 2: Induction motor 5, 7: Coordinate conversion means 6: Current detection means 20: Voltage distortion estimation means 21-27, 31-34: Addition / subtraction means 30: Current adjustment means

Claims (5)

In a control device for a voltage source inverter that outputs an alternating voltage of a desired magnitude and frequency by turning on and off a semiconductor switching element,
Means for calculating a first output voltage command value obtained by removing a back electromotive force component and an interference component from the output current command value of the inverter;
Current detection means for detecting an output current of the inverter;
Means for estimating the output voltage of the inverter from the output current detection value of the inverter;
Means for estimating a voltage distortion component from a first output voltage command value and an estimated output voltage value of the inverter;
Means for obtaining a voltage correction value from the estimated voltage distortion component value;
A final output voltage command value for the inverter is generated by correcting a second output voltage command value obtained by adding a back electromotive force component and an interference component to the first output voltage command value with the voltage correction value. Means to
A control device for a voltage source inverter. In the control apparatus for the voltage source inverter according to claim 1,
Current adjusting means that operates to eliminate a deviation between the output current command value of the inverter and the output current detection value;
Means for adding the output of the current adjusting means to the second output voltage command value;
A control device for a voltage source inverter. In a control device for a voltage source inverter that outputs an alternating voltage of a desired magnitude and frequency by turning on and off a semiconductor switching element,
Means for calculating a first output voltage command value obtained by removing a back electromotive force component and an interference component from the output current command value of the inverter;
Current detection means for detecting an output current of the inverter;
Means for estimating the output voltage of the inverter from the output current detection value of the inverter;
Means for estimating a voltage distortion component from a first output voltage command value and an estimated output voltage value of the inverter;
Means for obtaining a current command correction value from this voltage distortion component estimated value;
Means for correcting the output current command value by the current command correction value;
Current adjusting means that operates so as to eliminate a deviation between the output current command value corrected by this means and the output current detection value;
A final output voltage command value for the inverter is corrected by correcting a second output voltage command value obtained by adding a back electromotive force component and an interference component to the first output voltage command value by the output of the current adjusting means. Means for generating
A control device for a voltage source inverter. In the control apparatus of the voltage type inverter as described in any one of Claims 1-3,
The means for calculating the first output voltage command value is:
A control apparatus for a voltage source inverter, wherein an output voltage command value is calculated using an output current command value, a load resistance value, and an inductance value on a rotation coordinate of the inverter. In the control apparatus for the voltage source inverter according to any one of claims 1 to 4,
The means for estimating the output voltage of the inverter is:
A control apparatus for a voltage source inverter, wherein an output voltage is estimated using an output current detection value, a load resistance value, and an inductance value on a rotation coordinate of the inverter.

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