JP2012239271A - Controller of rotary machine - Google Patents

Abstract

PROBLEM TO BE SOLVED: To solve such a problem that the estimation accuracy of electrical angle lowers as the high frequency voltage signal is made small.SOLUTION: A high frequency voltage signal setting unit 50 sets a high frequency voltage command signal. An operation signal generation unit 32 sets the operation signal g*#(*=u,v,w;#=n,p) of an inverter based on the high frequency voltage command signal. Meanwhile, a highpass filter 62 extracts high frequency current signals idh, iqh from currents id, iq flowing through a motor generator. An outer product operation unit 64 calculates the outer product of the high frequency voltage command signal and the high frequency current signal. Rotation angle θ is operated so that the outer product becomes zero. Input of a highpass filter 64 is generated using only those having a large absolution value out of the detection values of phase currents iu, iv, iw.

Description

  The present invention controls a control amount of the rotating machine by operating a power conversion circuit including a switching element that selectively opens and closes between a terminal of the rotating machine having saliency and a voltage applying unit having a voltage value different from each other. In this case, superimposing means for superimposing a high frequency voltage signal having a frequency higher than the electrical angular frequency of the rotating machine on the output voltage of the power conversion circuit, and a high frequency flowing in the rotating machine in accordance with the superimposed high frequency voltage signal. And a estimator that estimates a rotation angle of the rotating machine based on a current signal.

  As this type of control device, for example, as shown in Patent Document 1 below, when a high-frequency voltage signal that vibrates in the positive and negative directions of the estimated d-axis of a three-phase motor is applied, the high-frequency that actually propagates to the motor Some have also been proposed that estimate the electrical angle of an electric motor based on a current signal.

Japanese Patent No. 3312472

  By the way, since the frequency of the high-frequency voltage signal is usually within an audible frequency band, noise perceived by a person when estimating the electrical angle may occur. In order to reduce this noise, it is effective to reduce the high-frequency voltage signal. However, in this case, the inventors have found that the estimation accuracy of the electrical angle is lowered.

  The present invention has been made in the process of solving the above-mentioned problems, and its object is to detect by superimposing a high-frequency voltage signal having a frequency higher than the electrical angular frequency of the rotating machine on the output voltage of the power conversion circuit. Another object of the present invention is to provide a new control device for a rotating machine that estimates the rotation angle of the rotating machine based on the high-frequency current signal.

  Hereinafter, means for solving the above-described problems and the operation and effect thereof will be described.

  According to the first aspect of the present invention, the operation of the rotating machine is performed by operating a power conversion circuit including a switching element that selectively opens and closes between a terminal of the rotating machine having saliency and a voltage applying unit having a voltage value different from each other. When controlling the control amount, superimposing means for superimposing a high-frequency voltage signal having a frequency higher than the electrical angular frequency of the rotating machine on the output voltage of the power conversion circuit, and the rotation according to the superimposed high-frequency voltage signal And an estimation means for estimating a rotation angle of the rotating machine based on a high-frequency current signal flowing through the machine, wherein the rotating machine includes a plurality of windings connected to each other, The high-frequency current signal is acquired based on a detection result of a detection unit that detects a current flowing through each of the windings, and the estimation unit estimates the rotation angle. Regarding the contribution of the detected value of the current of the winding to the high-frequency current signal that is reasonably used, the contribution of the absolute value exceeding the specified value over the contribution of the detected value whose absolute value is less than or equal to the specified value It is characterized by comprising a contribution varying means for making the degree smaller than the degree.

  When the current flowing through the rotating machine is small, the detection accuracy of the detection means tends to be lowered. In particular, when the high-frequency voltage signal is reduced under a condition where the current flowing through the winding of the rotating machine is small, the high-frequency current signal flowing through the winding also becomes small, and the detection accuracy tends to be lowered. In the above invention, in view of this point, the degree to which the absolute value of the detection value of the current flowing through the winding contributes to the high-frequency current signal used for the estimation process is reduced. Thereby, the influence of the detection value with low accuracy can be reduced, and the estimation accuracy of the rotation angle can be improved.

  According to a second aspect of the present invention, in the first aspect of the invention, the variable contribution means prohibits the use of a detected current value of the winding having an absolute current value equal to or less than a specified value. Prohibiting means is provided, and the high-frequency current signal is obtained based on a detected value of current for the winding whose absolute value exceeds a specified value.

  In the above invention, since the detection value with low accuracy is not used for the estimation processing, the estimation accuracy of the rotation angle can be further improved.

  According to a third aspect of the present invention, in the first or second aspect of the present invention, the specified value is set to a value equal to or greater than a minimum resolution of the detection means.

  According to a fourth aspect of the present invention, in the first or second aspect of the present invention, the windings of the rotating machine are arranged at equal electrical angles with respect to the rotation direction of the rotating machine. The contribution varying means switches the detection value for decreasing the contribution for each “1 / N” degree of electrical angle, where N is the number of windings.

  In the above invention, the currents flowing through the windings are out of phase with each other by “2π / N”, and each of the windings is at the timing of zero crossing every “π / N”. And the process which makes a contribution degree small can be simply performed by switching the detection value which makes a contribution degree small using the periodicity of the electric current which flows through such a coil | winding.

  The invention according to claim 5 is the invention according to any one of claims 1 to 4, wherein the estimating means calculates a component of a two-dimensional coordinate system of the high-frequency current signal from the detection result, The rotation angle is estimated by using a selective transmission means for selectively transmitting the frequency component of the high-frequency voltage signal from the detection result of the detection means and inputting the frequency component to the estimation means. .

  In the above invention, by including the selective transmission means, it is possible to suitably suppress the noise from being mixed into the parameters used for calculating the components of the two-dimensional coordinate system.

  According to a sixth aspect of the present invention, in the first aspect of the present invention, the frequency component of the high-frequency voltage signal is selectively amplified from the detection result of the detection means and the estimation means is used. It comprises a selective amplification means for inputting.

  In the above invention, by including the selective amplification means, the current detection accuracy can be improved even when the amplitude of the current flowing through the winding is small.

  The invention according to claim 7 is the invention according to any one of claims 1 to 6, wherein the detection means detects a magnetic field around a wiring connected to each winding of the rotating machine. And a means for detecting an electric current, and a shield means for shielding an electric field at a portion of the wiring facing the detection means on an outer periphery thereof.

  When electric current is generated in the detection means due to the electric field effect due to the current flowing through the wiring, the detection accuracy of the detection means decreases. In this regard, in the above invention, by providing the shielding means, it is possible to suitably suppress the phenomenon caused by the electric field effect.

  The invention according to claim 8 is the invention according to any one of claims 1 to 7, wherein the power conversion circuit selectively connects a terminal of the rotating machine to each of a positive electrode and a negative electrode of a DC voltage source. And a switching element connected to the positive electrode and a switching element connected to the negative electrode are turned on and off, respectively. When switching from one state to the other, the dead time period during which both of the other and the other are turned off is provided. In the dead time period, the winding of the rotating machine is wound by the DC / AC converter circuit. The high frequency voltage signal that is actually superimposed due to the dead time voltage applied to the end of the wire. And further comprising an error reducing means for reducing the error generated.

  When the DC / AC converter circuit is used, the voltage applied to the terminal (end of the winding) of the rotating machine during the dead time period depends on the polarity of the current flowing through the terminal. The voltage applied to the terminal of the rotating machine during this time can be an error with respect to the high frequency voltage signal intended to be superimposed by the superimposing means. The ratio of the error voltage to the high frequency voltage signal increases as the high frequency voltage signal is reduced. For this reason, the smaller the high-frequency voltage signal, the greater the error that the high-frequency voltage signal actually superimposed has on the intended one. This is a factor that lowers the estimation accuracy of the rotation angle and is a factor different from the detection error of the detection means. In view of this point, the above invention includes error reduction means.

1 is a system configuration diagram according to a first embodiment. FIG. The block diagram which shows the estimation process of the rotation angle concerning the embodiment. The time chart which shows the PWM process concerning the embodiment. The time chart which shows the dead time compensation process concerning the embodiment. The time chart explaining the error which arises in a high frequency voltage signal due to dead time. The vector diagram which shows the error which arises in a high frequency voltage signal due to dead time. The figure which shows the setting of the high frequency voltage command signal concerning the said embodiment. The time chart explaining the error which arises in a high frequency voltage signal due to dead time. The figure which shows the suppression principle of the error resulting from the dead time concerning the said embodiment. The time chart which shows the error which arises in the detected value of a high frequency current signal resulting from a current sensor. The time chart which shows the switching process of use of the current sensor concerning the said embodiment. The figure which shows the gain specification of the band pass filter concerning the embodiment. The circuit diagram which shows the circuit structure of the band pass filter concerning the embodiment. The time chart which shows the effect of the embodiment. The time chart which shows the switching process of use of the current sensor concerning 2nd Embodiment. The system block diagram concerning 3rd Embodiment.

<First Embodiment>
Hereinafter, a first embodiment in which a control device for a rotating machine according to the present invention is applied to a control device for a rotating machine as an in-vehicle main machine will be described with reference to the drawings.

  FIG. 1 shows a system configuration diagram according to the present embodiment.

  The motor generator 10 is a three-phase permanent magnet synchronous motor. The motor generator 10 is a rotating machine (saliency pole machine) having saliency. Specifically, the motor generator 10 is an embedded magnet synchronous motor (IPMSM).

  Motor generator 10 is connected to high voltage battery 12 via inverter IV and converter CV. Here, inverter IV has three sets of series connection bodies of switching elements S * p, S * n (* = u, v, w), and the connection point of each series connection body is the U of motor generator 10. , V and W phases. In the present embodiment, insulated gate bipolar transistors (IGBTs) are used as the switching elements S * p and S * n. In addition, diodes D * p and D * n are connected in antiparallel to these. On the other hand, the high voltage battery 12 is a secondary battery having a terminal voltage of, for example, 100 V or more.

  In this embodiment, the following is provided as detection means for detecting the state of the motor generator 10 and the inverter IV. First, current sensors 16, 17 and 18 for detecting currents iw, iv and iu flowing through the respective phases of the motor generator 10 are provided. Moreover, the voltage sensor 19 which detects the input voltage (power supply voltage VDC) of the inverter IV is provided. Here, the current sensors 16, 17, and 18 are configured to include Hall elements, and detect the magnetic flux generated in the circumferential direction according to the current flowing through each phase as a parameter having a correlation with the current. Means.

  The detection values of the various sensors are taken into the control device 14 constituting the low pressure system via the interface 13. The control device 14 generates and outputs an operation signal for operating the inverter IV and the converter CV based on the detection values of these various sensors. Here, the signals for operating the switching elements S * p and S * n of the inverter IV are the operation signals g * p and g * n. The signals for operating the converter CV are the operation signals gcp and gcn.

  FIG. 2 shows processing performed by the control device 14. In the following, first, “control amount control” will be described, and then “rotation angle estimation processing” will be described.

`` Control amount control ''
The command current setting unit 20 sets a command current idr on the d axis and a command current iqr on the q axis, which are current command values of the rotating two-phase coordinate system, based on the required torque Tr. On the other hand, the currents iu, iv, iw flowing through the phases of the motor generator 10 are, in the dq conversion unit 22, an actual current id on the d axis and an actual current iq on the q axis, which are actual currents in the rotating two-phase coordinate system. Is converted to The deviation calculator 24 calculates the difference between the d-axis command current idr and the actual current id, and the deviation calculator 26 calculates the difference between the q-axis command current iqr and the actual current iq. The current control unit 28 feedback-controls the command voltage vdr on the d axis as an operation amount for feedback control of the actual current id on the d axis to the command current idr, and the actual current iq on the q axis to the command current iqr. A command voltage vqr on the q-axis is calculated as an operation amount for this. These manipulated variables may be the sum of the output of the proportional element and the output of the integral element with the corresponding difference as an input.

  The three-phase conversion unit 30 converts the command voltages vdr and vqr in the rotating two-phase coordinate system into three-phase command voltages vur, vvr and vwr, and normalizes them with the power supply voltage VDC to thereby convert the duty signal Du. , Dv, Dw are calculated. The dead time compensation unit 34 calculates dead time correction amounts Δvu, Δvv, Δvw for feedforward correction of the duty signals Du, Dv, Dw based on the corresponding phase currents iu, iv, iw. Then, each of the correction units 36, 38, and 40 corrects the duty signals Du, Dv, and Dw based on the dead time correction amounts Δvu, Δvv, and Δvw, respectively. The operation signal generation unit 32 generates the operation signal g * # by PWM processing based on the magnitude comparison between the duty signals Du, Dv, Dw and the carrier.

  FIG. 3 shows details of processing by the operation signal generation unit 32. In the present embodiment, based on the magnitude comparison between the triangular wave carrier CS in which the gradual increase speed and the gradual decrease speed are the same and the gradual increase period and the gradual decrease period are the same, and the duty signals Du, Dv, Dw of each phase, PWM signals gu, gv, and gw are generated. Based on the PWM signal g * (* = u, v, w), an upper arm operation signal g * p and a lower arm operation signal g * n are generated. At this time, by performing dead time generation processing, the operation signal g * # (* = u, v, w; # = p, n) is delayed in the rising timing by the dead time DT with respect to the PWM signal g *. Will be. Note that the update cycle of the duty signals Du, Dv, Dw (command voltages vur, vvr, vwr) is made to coincide with the update cycle of the carrier CS. More specifically, in the present embodiment, the duty signals Du, Dv, Dw are updated at the timing when the carrier CS reaches a peak.

  FIG. 4 shows details of the processing of the dead time compensation unit 34.

  As shown in FIG. 4A, when the phase current i * (* = u, v, w) is positive, current flows through the lower arm diode D * n during the dead time period. The ON period of the operation signal g * p is shorter than the ON period of the PWM signal g * by the dead time DT, and the rising edge is delayed by the dead time DT. For this reason, the dead time compensation unit 34 corrects the duty signal D * to be increased by the dead time correction amount Δv *, so that both the rising edge and the falling edge of the PWM signal g * are “½ of the dead time DT. Correct by one by one. Thereby, the ON period of the operation signal g * p can be matched with the ON period of the PWM signal g * before correction, and the delay amount of the rising edge can be halved.

  As shown in FIG. 4B, when the phase current i * (* = u, v, w) is negative, current flows through the upper arm diode D * p during the dead time period. The on period of the operation signal g * p is longer by the dead time DT than the on period of the PWM signal g *. Therefore, the dead time compensation unit 34 corrects the duty signal D * to be decreased by the dead time correction amount Δv *, so that both the rising edge and the falling edge of the PWM signal g * are “½ of the dead time DT. Correct by one by one. Thereby, the ON period of the operation signal g * p can be matched with the ON period of the PWM signal g * before correction. However, at this time, the rising edge of the operation signal g * p is delayed by “½” of the dead time DT with respect to the rising edge of the PWM signal g * before correction.

  As shown in FIG. 4C, when the phase current i * (* = u, v, w) is inverted from negative to positive in the period from the rising edge to the falling edge of the PWM signal g *, the dead corresponding to the rising edge During the time period, current flows through the upper arm diode D * p, and during the dead time period corresponding to the falling edge, current flows through the lower arm diode D * n. For this reason, the ON period of the operation signal g * p coincides with the ON period of the PWM signal g *. Therefore, in this case, the dead time correction amount Δv * is set to zero.

"Rotation angle estimation process"
The high frequency voltage signal setting unit 50 shown in FIG. 2 sets the high frequency voltage command signal Vhr = (vdhr, vqhr). Here, in this embodiment, vqhr = 0 and vdhr is a signal that reverses its polarity every half cycle of the PWM processing. The superimposing unit 52 corrects the d-axis command voltage vdr output from the current control unit 28 with the d-axis component vdhr of the high-frequency voltage command signal and outputs the corrected signal to the three-phase conversion unit 30.

  On the other hand, the high-pass filter 62 extracts harmonic components (high-frequency current signals idh, iqh) from actual currents id, iq that have been processed later. Here, the high frequency component is a component having a higher frequency than the fundamental wave component. In particular, here, the same frequency component as the high-frequency voltage command signal Vhr is extracted. As the high-pass filter 62, for example, a means for outputting a difference between values before and after a half cycle of the PWM signal for the actual currents id and iq may be used.

  The outer product calculation unit 64 calculates the outer product value of the high-frequency voltage command signal Vhr and the high-frequency current signals idh and iqh. This outer product value has a correlation with the angle between the vectors of the high-frequency voltage signal and the high-frequency current signals idh and iqh, and is a parameter (angle correlation amount) having a correlation with the rotation angle of the motor generator 10. In particular, in this embodiment, the error correlation amount has a correlation with the error of the rotation angle θ. The outer product value as the error correlation amount is input to the speed calculation unit 66. The speed calculation unit 66 calculates the electrical angular speed ω as the sum of a proportional element and an integral element having the outer product value as an input. Then, the angle calculation unit 68 calculates the rotation angle θ as a time integral value of the electrical angular velocity ω. As a result, the rotation angle θ becomes an operation amount for feedback control of the outer product value to its target value of zero.

  The target value of the outer product value is zero because the motor generator 10 is IPMSM, and therefore the d-axis inductance Ld is smaller than the q-axis inductance Lq. That is, in this case, if the high-frequency voltage in the d-axis direction is superimposed on the voltage for controlling the control amount as the output voltage of the inverter IV, the high-frequency current signal is also in the d-axis direction, and the outer product value becomes zero. When the outer product value is not zero, the rotation angle θ is manipulated so that the outer product value becomes zero, and the rotation angle θ matches the correct angle.

However, the estimation accuracy of the rotation angle θ is lowered by reducing the high-frequency voltage signal. This is due to a voltage error caused by the dead time DT and detection errors of the current sensors 16, 17, and 18. Below, each of these pairs of causes and those solution means concerning this embodiment are explained in order.
"Dead time error compensation"
When the ratio of the change amount of the ON time and OFF time of the operation signal g * # due to the superposition is reduced by decreasing the high frequency voltage signal, the high frequency voltage actually superimposed is reduced. The error due to the signal dead time DT becomes large, and as a result, the accuracy of estimating the rotation angle θ decreases. Such an error is due to the provision of the dead time compensation unit 34 so that the phase current i * (* = u, v, w) is inverted from negative to positive during the period from the rising edge to the falling edge of the PWM signal g *. Can be avoided. This is because, as shown in FIG. 4, the ON period of the operation signal g * # is defined by the PWM signal g * by the compensation by the dead time compensation unit 34, and the phase is only “DT / 2”. This is because the line voltage matches that specified by the PWM signal g * before correction because of the delay. In other words, this case is equivalent to the case where the phase of the carrier CS is delayed by “DT / 2” and PWM processing is performed, and no error occurs in the line voltage.

  However, when the phase current i * (* = u, v, w) is inverted from negative to positive during the period from the rising edge to the falling edge of the PWM signal g *, the phase of the operation signal g * # for that phase is Since there is no delay, this is equivalent to advancing only that phase by “DT / 2” compared to the other phases. For this reason, in this case, the line voltage deviates from that defined by the PWM signal g * before correction, and an error occurs in the high-frequency voltage signal. FIG. 5 shows the PWM signal g * after processing by the dead time compensation unit 34. In the example shown in the figure, the U phase has a zero-cross period, and in this case, only the U phase is in the same state as when the voltage is advanced by “DT / 2”. In other words, when the PWM process is performed by retarding the phase of the carrier CS by “DT / 2”, it is equivalent to the voltage of only the U phase being advanced by “DT / 2”. As a result, as indicated by the alternate long and short dash line in the upper part of the figure, when the high-frequency voltage signal (vdh) is superimposed on the signals in the positive and negative directions of the U axis for each half cycle of the PWM, the lower part of the figure As shown by a two-dot chain line, the amplitude of the actually superposed high-frequency voltage signal increases. Note that what is indicated by a one-dot chain line in FIG. 5 is that the d-axis component vdhr of the high-frequency voltage command signal is normalized by the power supply voltage VDC.

  For this reason, as shown in FIG. 6, when the phase current zero-crosses, the high frequency voltage signal Vh that is actually superimposed on the high frequency voltage command signal Vhr has an error.

  Therefore, in this embodiment, as shown in FIG. 7, the error voltage vector due to the dead time and the high-frequency voltage command signal Vhr are set in parallel. Specifically, the command value Ir = (command current idr, iqr) for controlling the control amount is defined as the q-axis direction. Furthermore, a period in which the high-frequency voltage command signal Vhr is superimposed in the d-axis positive direction is the first half of the PWM process, and a period in which the high-frequency voltage command signal Vhr is superimposed in the d-axis negative direction is the second half of the PWM process; Is switched to the second half of the PWM process and the period in which the period superimposed in the negative d-axis direction is the first half of the PWM process. This is a setting for avoiding that the error voltage and the high-frequency voltage command signal due to the dead time are in opposite directions. FIG. 8A shows the relationship between the error indicated by the outer product value and the electrical angle for the case where the rotation angle θ has the error “± Δ” and the case where the rotation angle θ does not have the error “± Δ”.

  Here, in the example shown in FIG. 8B, since the error voltage caused by the dead time and the high frequency voltage command signal are in the same direction, the actually superposed high frequency voltage signal has an amplitude value higher than that of the high frequency voltage command signal. Only grows. On the other hand, in the example shown in FIG. 8C, since the error voltage due to the dead time and the high frequency voltage command signal are in opposite directions, the actually superimposed high frequency voltage signal is more than the high frequency voltage command signal. The amplitude value becomes smaller. In particular, when the amount of reduction of the amplitude becomes large, the actually superposed high-frequency voltage signal becomes a voltage signal in the opposite direction to the high-frequency voltage command signal. When such a phenomenon occurs, the outer product value as the angle correlation amount may not be a highly accurate representation of the rotation angle θ information. Incidentally, even if the main current is orthogonal to the high-frequency voltage signal when the error is “± Δ”, the outer product value fluctuates largely near the zero cross and deviates from the correct value because the direction in which the high-frequency voltage command signal is superimposed is d This is because the d-axis component tends to be smaller than the error voltage due to the dead time because it is offset from the axis.

Therefore, in the present embodiment, as shown in FIG. 9, when the main current I (id, iq) moves in the six regions A to F defined by the voltage vector defined by the stator of the motor generator 10, the high frequency The d-axis component vdhr of the voltage command signal is alternately switched between positive and negative in the first half of the PWM processing cycle Tc. This is because a pair of phenomena of a phenomenon in which the positive direction of the d-axis and the positive direction of each phase coincide with each other and a phenomenon in which the positive direction of the d-axis and the negative direction of each phase coincide with each other as the direction of the main current I rotates. In view of the fact that both of these occur. That is, when the positive direction of the d-axis is the positive direction of a specific phase, the d-axis component vdhr of the high-frequency voltage command signal is made positive in the first half of the PWM processing cycle Tc, so that FIG. ). On the other hand, when the positive direction of the d-axis is a negative direction of a specific phase, the d-axis component vdhr of the high-frequency voltage command signal is made positive in the first half of the PWM processing cycle Tc, so that FIG. This is the case shown in (c). Therefore, in this case, the d-axis component vdhr of the high-frequency voltage command signal is set to be negative in the first half of the PWM processing cycle Tc, so that the case shown in FIG.
"Compensation for detection accuracy of current sensors 16, 17, 18"
When there is a zero crossing in the phase currents iu, iv, iw of the motor generator 10, there is a possibility that distortion occurs in the high-frequency current signals idh, iqh calculated from the detection values by the current sensors 16-18. In particular, as shown in FIG. 10, if the effective value of the high-frequency voltage command signal Vhr is reduced, distortion is likely to occur. In the figure, a “real waveform” indicated by a one-dot chain line is an actual d-axis high-frequency current, and a “current sensor value” indicated by a solid line is a high-frequency current signal calculated from detection values by the current sensors 16 to 18. idh.

  Therefore, in the present embodiment, the output signals of the current sensors 16 to 18 are taken into the zero cross current calculation unit 70 as shown in FIG. The zero cross current calculation unit 70 calculates the current value of this phase based on the current values of the remaining two phases when the absolute value of the phase currents iu, iv, iw is less than the specified value. Here, the specified value is set to be equal to or higher than the minimum resolution of the current sensors 16-18. In particular, in this embodiment, the minimum resolution is set to 10 times or more. FIG. 11 shows a use area of detection values of the current sensors 16 to 18 according to the present embodiment.

  The output signal of the zero cross current calculation unit 70 is input to the band pass filter 80. Here, as shown in FIG. 12, the bandpass filter 80 is an analog filter that selectively transmits and amplifies the frequency band A of the high-frequency voltage command signal Vhr. FIG. 13 illustrates the configuration of the bandpass filter 80. As shown in the figure, the band pass filter 80 includes an operational amplifier 81. The positive input terminal of the operational amplifier 81 is connected to a connection point of a series connection body of a capacitor 82 and a resistor constituting a differentiation circuit. The end of the series connection body on the capacitor 82 side is connected to the input terminal of the band pass filter 80, and the end of the resistor 83 side is grounded. The negative input terminal of the operational amplifier 81 is grounded via a resistor 84 and is connected to the output terminal of the operational amplifier 81 via a parallel connection body of a resistor 85 and a capacitor 86 constituting an integrating circuit. The output terminal of the operational amplifier 81 is the output terminal of the band pass filter 80.

  The three-phase current values output from the bandpass filter 80 are taken into the dq conversion unit 60 shown in FIG. 2 and are dq-converted here, so that the actual current id, iq is calculated.

  FIG. 14A shows the effect of this embodiment. FIG. 14A shows the relationship between the error and the electrical angle determined from the outer product value when the rotation angle θ is correct in the present embodiment. On the other hand, the comparative example shown in FIG. 14B is a case where neither the processing by the bandpass filter 80 nor the processing of the zero cross current calculation unit 70 is performed. In this comparative example, the error is very large in the zero cross period. This is due to the influence of noise and the like in addition to the influence of the distortion shown in FIG. On the other hand, in this embodiment, not only the detection value in the zero cross period is not used, but also the influence of noise can be removed by the frequency selection function of the bandpass filter 80. That is, in the present embodiment, in order to reduce the effective value of the high-frequency voltage command signal Vhr, the high-frequency current signals idh and iqh associated therewith become small, and as a result, changes in the fundamental wave components of the phase currents iu, iv and iw are regarded as errors. May have a major impact. Further, even when noise having a frequency higher than that of the high-frequency voltage command signal Vhr is mixed, there is a possibility that the influence is increased. For these reasons, signals outside the frequency band to be used were attenuated. Further, the amplification function of the band pass filter 80 reduces the influence of calculation errors in the process of estimating the rotation angle θ by amplifying the phase currents iu, iv, iw even when the amplitudes are small. Can do.

  According to the embodiment described in detail above, the following effects can be obtained.

  (1) The high-frequency current signals idh and iqh were acquired based only on the detected current value for the current flowing through each phase of the motor generator 10 (current of the stator winding) exceeding the specified value. Thereby, since the detection value with low accuracy by the current sensors 16 to 18 is not used for the estimation process, the estimation accuracy of the rotation angle θ can be improved.

  (2) The phase currents iu, iv, iw used for calculating the high-frequency current signals idh, iqh are filtered by the band-pass filter 80. As a result, it is possible to remove the influence of noise having a lower frequency than the high-frequency voltage command signal Vhr (fundamental wave components of the phase currents iu, iv, iw) and high-frequency noise. Further, by amplifying the frequency band of the high-frequency voltage command signal Vhr, it is possible to reduce calculation errors in the calculation process of the rotation angle θ even when the amplitudes of the phase currents iu, iv, iw are small.

  (3) A straight line having the direction of the error voltage due to the dead time and a straight line having the direction of the high-frequency voltage command signal were set in parallel. Thereby, the angle formed by the straight line having the direction of the high-frequency voltage command signal and the straight line having the direction of the high-frequency voltage signal actually superimposed can be made zero.

  (4) The polarity of the d-axis component vdhr of the high-frequency voltage command signal is switched so that the vector direction of the error voltage caused by the dead time matches the vector direction of the high-frequency voltage command signal. Thereby, the situation where the high frequency voltage command signal is canceled by the error voltage due to the dead time can be preferably avoided.

(5) A dead time compensation unit 34 is provided. As a result, when there is no zero crossing in the phase current, it is possible to avoid the occurrence of an error in the high frequency voltage signal due to the dead time.
<Second Embodiment>
Hereinafter, the second embodiment will be described with reference to the drawings with a focus on differences from the first embodiment.

In the zero cross current calculation unit 70 according to the first embodiment, the current value of the phase corresponding to the one having a small absolute value of the current is calculated based on the detected values of the remaining two phases. In contrast, in the present embodiment, as shown in FIG. 15, the phase for calculating the current value based on the detected values of the remaining two-phase currents according to the rotation angle θ of the motor generator 10 is “60 °”. Switch periodically every time. Here, the period during which any one of the detection values of the current sensors 16 to 18 is not used is “60 °”. In this embodiment, one period of the electrical angle of the rotor is equally divided. This is related to the placement of the stator. In this case, the phase currents iu, iv, and iw are sine waves whose phases are shifted from each other by “120 °”, so that any phase is zero-crossed by an electrical angle of “60 °”. For this reason, by setting the “60 °” region including the advance side and the retard side of this region as the unused period, the unused period can be easily set.
<Third Embodiment>
Hereinafter, the third embodiment will be described with reference to the drawings with a focus on differences from the first embodiment.

  FIG. 16 shows a system configuration according to the present embodiment. Note that, in FIG. 16, the same step numbers are assigned for convenience to those corresponding to the members shown in FIG.

As shown in the figure, in the present embodiment, in the wiring connected to each phase of the motor generator 10, the portion where the current sensors 16, 17, 18 are arranged in the direction orthogonal to the current flow direction is shielded. It is constituted by lines 92, 94, 96. These shield lines 92, 94, and 96 have a function of shielding an electric field. As a result, it is possible to avoid the generation of electric charges in the current sensors 16, 17, and 18 due to the electric field generated when the current flows through the wiring, thereby further reducing the current detection accuracy of the current sensors 16, 17, and 18. Can be suppressed.
<Other embodiments>
Each of the above embodiments may be modified as follows.

“Prohibited measures”
The current value of the phase in which the absolute value of the current is equal to or less than the specified value is not limited to that calculated by Kirchhoff's law from the detected values of the other two phases whose absolute value exceeds the specified value. For example, in consideration of the fact that the current value of two phases out of three phases only needs to be input as dq conversion, another two-phase detection value whose absolute value exceeds a specified value is converted into a detection value on the dq axis. It may be. In this case, the conversion process differs depending on which phase current is not used.

"Contributing variable means"
Of the detection values of the current sensors 16, 17, and 18, the detection values are not limited to those that do not use an absolute value equal to or less than a specified value (prohibiting means). For example, a detection value of a phase whose absolute value is not more than a specified value, and a weighted average processing value of a current value of a phase not more than a specified value calculated from the other two phases exceeding the specified value, and detection of the current of the other phase The high-frequency current signals idh and iqh may be calculated using the values.

"Selective transmission means"
The selective transmission means is not limited to the bandpass filter 80 described above. For example, it may be a low-pass filter that cuts off components higher than the frequency of the high-frequency current signals idh and iqh, and may have a function of amplifying a signal in the transmission frequency band. In this case, although changes in the fundamental wave components of the actual currents iu, iv, and iw are transmitted, the high frequency noise can be sufficiently removed.

  Further, the filter is not limited to an analog filter, and may be a digital filter.

  Further, the frequency band to be transmitted may be variably settable.

  Note that the selective transmission means may not have a function of amplifying a signal in a frequency band to be transmitted.

"Selective amplification means"
For example, a band pass filter that transmits a specific frequency but does not have an amplification function and an amplifier that amplifies the output of the band pass filter may be used. Further, for example, a low-pass filter that cuts off components higher than the frequencies of the high-frequency current signals idh and iqh and does not have an amplification function, and an amplifier that amplifies the output signal of the low-pass filter. May be.

  The fact that this is not necessary is as described in the section “Selective transmission means”.

About detection means
It is not restricted to what is equipped with a Hall element. In short, the application of the present invention is effective if the absolute value of the current decreases and the detection accuracy decreases.

"About estimation means"
Feedback control of the phase difference (vector direction difference) between the high-frequency voltage signal and the high-frequency current signal to zero is not limited to feedback control of the outer product value to zero. For example, the phase difference between the high-frequency current signal and the high-frequency voltage signal is directly calculated based on the angle formed by the vector direction of the high-frequency current signals idh and iqh and the positive d-axis direction, and this is feedback-controlled to zero. Good.

  Further, the angle correlation amount is not limited to the one using the phase difference between the high frequency current signal and the high frequency voltage signal. For example, as described in Japanese Patent Application Laid-Open No. 2008-220289, it is a product of vector norms of each high-frequency current signal when the high-frequency voltage command signal is a d-axis direction voltage and a q-axis direction voltage, respectively. May be. In this case, the rotation angle θ is operated as an operation amount for controlling the product as the angle correlation amount to a target value other than zero.

  The means for operating the rotation angle θ to feedback control the angle correlation amount to the target value is not limited to first operating the electrical angular velocity ω to control the angle correlation amount to the target value. θ may be directly manipulated.

"Error reduction measures"
It is not restricted to what was illustrated in said each embodiment. For example, the dead time correction amount Δv * (* = u, v, w) in the dead time compensation unit 34 may be changed as follows for each half cycle of the carrier of the PWM processing. That is, in the gradual increase period of the carrier, if the phase current i * is positive, the dead time correction amount Δ * is doubled as compared to the first embodiment, whereas if the phase current i * is negative, no correction is performed. In the gradual decrease period of the carrier, if the phase current i * is negative, the dead time correction amount Δ * is doubled as compared to the first embodiment, while if the phase current i * is positive, no correction is performed. By doing so, the actual PWM signal with compensation shown in FIG. 4 and the PWM signal g * can be always matched.

Moreover, it is not restricted to what reduces an error by operation of the output voltage of the inverter IV. For example, an accurate high-frequency voltage signal may be calculated based on the voltage during the dead time period and used for the estimation process of the rotation angle θ.
When the error reducing means is as described above, the phases of the command currents idr and iqr can be variably set. In this case, the period of “60 °” in the second embodiment may be variably set according to the phases of the command currents idr and iqr. If variably set in this way, the phase at which any phase crosses zero can be set to the center of the unused period.

"Dead time compensation function"
The dead time compensation means is not limited to the one that feedforward corrects the command voltage (Duty signal) based on the polarity of the phase current. For example, the detected value of the output voltage of each phase of the inverter may be feedback controlled to a command value. Even in this case, if correction is performed such that the start point and end point of the on-operation command period are shifted by the same time, it is possible to preferably avoid occurrence of an error in the high-frequency voltage signal outside the zero-cross period.

  Further, even if there is no dead time compensation means for controlling the error of the line average voltage due to the dead time as a direct control amount to zero, the current feedback control shown in FIG. If there is no current that crosses zero, the line voltage shift due to dead time is fully compensated. For this reason, even in this case, except for the period until the feedback control follows, an error occurs in the actually superimposed high-frequency voltage only when there is a zero crossing. Incidentally, when there is a zero-crossing phase, the error due to the dead time in that phase is, as can be seen from the description of FIG. 5, the error of the average voltage in one cycle Tc of PWM processing although it has a high frequency component. Does not contribute. For this reason, an error occurs in the high-frequency voltage signal in the zero cross period even by the dead time compensation function by feedback control.

“About Carrier CS”
The carrier CS is not limited to a triangular wave, and may be any one as long as the gradual increase period and the gradual decrease period are symmetrical by setting the gradual increase speed and the gradual decrease speed to be equal to each other and the gradual increase period and the gradual decrease period being equal to each other. . In this case, the dead time compensation function makes it easy to set the correction to shift the start point and the end point of the ON operation command period of the operation signal by the same time.

  However, the present invention is not limited to this, and for example, a sawtooth wave may be used. In this case, even if the control amount is feedback controlled, an error due to the dead time may occur in the high-frequency voltage signal even outside the zero-cross period, so the error due to the dead time is calculated as described in the “Error reduction means” column. Therefore, it is desirable to use for estimation processing.

"About rotating machines"
The rotating machine is not limited to a three-phase rotating machine having three stator windings connected to each other, and may be, for example, a five-phase rotating machine having five stator windings connected to each other. However, in this case, when applying the method shown in the second embodiment, a current sensor that is not used is “(360 ° ÷ 5) / n = 72 / n °: n = 1, It is desirable to invert every 2, 3,.

"Power conversion circuit"
The power conversion circuit including a switching element that selectively opens and closes between a terminal of the rotating machine and a voltage applying unit having a voltage value different from each other is not limited to the inverter IV. For example, it may be provided with a switching element that selectively opens and closes between a voltage applying unit that applies three or more different voltages to each phase of the multiphase rotating machine and a terminal of the rotating machine. . An example of a power conversion circuit for applying three or more different voltages to a terminal of a rotating machine is exemplified in Japanese Patent Application Laid-Open No. 2006-174697.

"others"
The final control amount of the motor generator 10 is not limited to torque, and may be, for example, a rotational speed. Further, the present invention is not limited to performing current vector control, and for example, torque feedback control may be performed. At this time, if the command voltage is set as the operation amount for controlling the control amount, and the operation signal is set based on the magnitude comparison between the carrier having symmetry and the command voltage, the dead time is set by feedback control of the control amount. A time compensation function can be provided.

  The rotating machine having saliency is not limited to the motor generator 10 because of its structure. For example, a synchronous reluctance motor (SynRM) may be used.

  The rotating machine is not limited to the in-vehicle main machine. For example, an electric motor mounted on a vehicle-mounted power steering may be used.

  DESCRIPTION OF SYMBOLS 10 ... Motor generator, 14 ... Control apparatus, 50 ... High frequency voltage signal setting part, 60 ... Outer product calculating part.

Claims (8)

When controlling the control amount of the rotating machine by operating a power conversion circuit including a switching element that selectively opens and closes between a terminal of the rotating machine having saliency and a voltage application unit having a voltage value different from each other, Superimposing means for superimposing a high-frequency voltage signal having a frequency higher than the electrical angular frequency of the rotating machine on the output voltage of the power conversion circuit, and based on a high-frequency current signal flowing through the rotating machine in accordance with the superimposed high-frequency voltage signal In a control device for a rotating machine comprising: an estimation means for estimating a rotation angle of the rotating machine;
The rotating machine includes a plurality of windings connected to each other,
The high-frequency current signal is acquired based on a detection result of a detection unit that detects a current flowing through each of the windings,
The estimation means determines the contribution of the detected value of the winding current to the high-frequency current signal used for the estimation process of the rotation angle of the detected value whose absolute value is less than a specified value. A control device for a rotating machine, comprising: a contribution variable means for making the absolute value smaller than a contribution of the absolute value exceeding a specified value.   The contribution varying means includes prohibiting means for prohibiting the use of a detected current value of the winding having an absolute current value equal to or less than a specified value, and the absolute value of the current of the winding is specified. 2. The control device for a rotating machine according to claim 1, wherein the high-frequency current signal is acquired based on a detected value of a current exceeding a value.   3. The control device for a rotating machine according to claim 1, wherein the specified value is set to a value equal to or higher than a minimum resolution of the detection means. The windings of the rotating machine are arranged at equal electrical angles with respect to the rotating direction of the rotating machine,
The said contribution degree variable means switches the detection value which makes the said contribution degree small for every "1 / N" degree of an electrical angle, when the number of the said windings is set to N. 2. A control device for a rotating machine according to 2. The estimation means calculates a component of a two-dimensional coordinate system of the high-frequency current signal from the detection result, and estimates the rotation angle using the component.
5. The selective transmission unit according to claim 1, further comprising a selective transmission unit that selectively transmits a frequency component of the high-frequency voltage signal from the detection result of the detection unit and inputs the frequency component to the estimation unit. Rotating machine control device.   6. The method according to claim 1, further comprising selective amplification means for selectively amplifying a frequency component of the high-frequency voltage signal from the detection result of the detection means and inputting the amplified frequency component to the estimation means. Rotating machine control device. The detection means is means for detecting a current by detecting a magnetic field around a wiring connected to each winding of the rotating machine,
The rotating machine control device according to any one of claims 1 to 6, further comprising shielding means for shielding an electric field of a portion of the wiring that is opposed to the detection means on an outer periphery thereof. The power conversion circuit includes a switching element that selectively connects a terminal of the rotating machine to each of a positive electrode and a negative electrode of a DC voltage source, and a DC / AC conversion circuit including a diode connected in reverse parallel to the switching element,
When switching from the state where either one of the switching element connected to the positive electrode and the switching element connected to the negative electrode and the other are turned on and off to the state where one and the other are turned off and on, respectively, There is a dead time period to turn off,
Error reduction means for reducing an error that occurs in a high-frequency voltage signal that is actually superimposed due to a dead time voltage applied to the end of the winding of the rotating machine by the DC / AC converter circuit during the dead time period. The rotating machine control device according to claim 1, further comprising:

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