JP2012175756A - Inverter control device - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a control device by which the inverter control does not affect a switching frequency in a PWM region and a current response with a small current deviation amount can be obtained independently of an operational frequency in a 1-pulse region.SOLUTION: In an embodiment, an inverter control device uses a current follow-up type PWM control circuit directly generating a switching control signal of an inverter driving an electric motor based on comparison between a PWM correction current reference and an electric motor current, and making automatic transition of a control mode to PWM control in middle and low speed regions of the electric motor and to 1-pulse control in a high speed region. The inverter control device has: a stationary deviation correction circuit amplifying a deviation between a current reference obtained by a vector operation and the electric motor current to control a magnetic flux and a torque of the electric motor; an adding circuit adding an output signal of the stationary deviation correction circuit to the current reference to calculate the PWM correction current reference; and a switching circuit, when the current follow-up control type PWM control circuit operates in the PWM control mode, setting the control in the stationary deviation correction circuit to integration control, and when the current follow-up control type PWM control circuit operates in the 1-pulse mode, switching over to proportional integration control.

Description

  The present invention relates to a voltage source inverter control device widely used in various fields such as electric power, industry, and transportation.

  FIG. 3 shows an example of the configuration of an induction motor speed control device.

  In FIG. 3, 1 is a DC power supply that supplies driving power to the motor, 2 is a smoothing capacitor, 3 is an inverter that converts DC power into three-phase AC power, 4 is an induction motor, 5U, 5V, and 5W detects motor current. The rotation sensor 6 is a hall CT that detects the position of the rotor of the electric motor.

  The rotation detection circuit 7 calculates and outputs an electrical angle signal θr and a speed ωr corresponding to the position of the rotor from the output signal of the rotation sensor 6. The subtracter 8 calculates the difference between the speed reference ωr * and the speed detection value ωr. This speed reference ωr * is a value corresponding to, for example, a speed command at the time of ATC or ATO or a speed command by a driver of the cab. The speed control circuit 9 performs PI control so that the speed deviation output from the subtracter 8 becomes zero, and generates a torque command Trq * so that the speed ωr follows the speed reference ωr *.

The vector calculation circuit 10 converts the magnetic flux reference Φ * and the torque reference Trq * into the torque component current reference i _q * and the magnetic flux component current reference i _d *, and also the slip angle based on the magnetic flux reference Φ * and the torque reference Trq *. Calculate θ _S * and output it. The adder 11 adds the rotor position signal θr and the slip angle θ _S * and outputs a magnetic flux position signal θ ₀ . This magnetic flux position signal θ ₀ is a d-axis electrical angle reference of the electric motor. The current detection circuit 12 scales the output signal of Hall CT (5U, 5V, 5W) to a value that can be used in this control circuit, and outputs the result as current detection values i _u , i _v , i _w . The coordinate conversion circuit 13 uses the magnetic flux position signal θ ₀ to detect the magnetic flux component current detection value i _d , the torque component current detection value on the dq axis coordinates in which the current detection values i _u , i _v , i _w are synchronized with the magnetic flux of the motor. i Convert to _q . Here, the d axis is an axis of magnetic flux generated in the rotor, and the q axis is an axis perpendicular to the d axis. Further, the current detection values i _d and i _q are DC values and are values that do not change much compared to the current detection values i _u , i _v , and i _w (sine waves).

The subtractors 14d and 14q calculate the difference between the current references i _d * and i _q * and the current detection values i _d and i _q for the d axis and the q axis, respectively. The steady deviation correction circuits 15d and 15q amplify the current deviation output from the subtracters 14d and 14q. The limiter 16 limits the correction signal i _qc * output from the correction circuit 15 q on the q-axis side. The adder 17d adds the correction signal i _dc * output from the correction circuit 15d and the current reference i _d *, and outputs a current tracking PWM current reference i _d **. The adder 17q adds the limiter 16 output and i _q *, and outputs a current tracking type PWM current reference i _q **. The subtractor 18 calculates the difference between the q-axis side correction circuit 15q output i _qc * and the weakening control start level i _qCLIM *.

  The magnetic flux weakening control circuit 19 amplifies (PI control) the output of the subtractor 18. The limiter 20 limits the lower limit of the output of the flux weakening control circuit to zero. The subtracter 21 subtracts the magnetic flux weakening control circuit output from the strong magnetic flux command value Φ **, and outputs a magnetic flux command Φ * according to the state of the motor. Here, the strong magnetic flux command value Φ ** may be a value that changes according to the rotation speed of the electric motor or a constant value that is appropriately set in advance.

The coordinate conversion circuit 22 converts the current tracking type PWM current references i _d ** and i _q ** into three-phase current references i _U *, i _V *, i of the stator stationary coordinates based on the magnetic flux position θ _0. Convert to _W *. Subtractor 23U, 23V, 23W are 3-phase current reference _i U _{*, i} V _*, it takes respective differences between _{i W} * and 3-phase current detection value _{i U, i V, i W} , current tracking type PWM control This is applied to the circuit 24.

The current follow-up type PWM control circuit 24 generates a PWM signal in which the detected current values i _U , i _V , i _W follow the current references i _U *, i _V *, i _W *, and is based on this PWM signal. Gate signals Gu, Gv, Gw, Gx, Gy, and Gz are output. The switching elements constituting the inverter 3 are on / off controlled by gate signals Gu, Gv, Gw, Gx, Gy, Gz.

The switching frequency control circuit 25 inputs the gate signals Gu, Gv, Gw, Gx, Gy, Gz, the speed ωr, and the torque component current command i _q * to increase the allowable error area of the current tracking type PWM control circuit 24. The determined hysteresis Hys is output. In a battery driving device or a train where the inverter DC voltage Vdc changes over a wide range, it is better to adjust the hysteresis Hys also by the DC voltage Vdc.

  This current tracking type PWM method directly generates a PWM signal such that the motor current follows the command value without using a triangular wave carrier wave, so that the current response is extremely fast. Further, in the normal triangular wave PWM method, when the motor rotation speed increases, the arithmetic control is not in time, and it is necessary to switch from the triangular wave PWM method to the synchronous PWM method. At this time, special control is required to perform switching smoothly. However, in the current follow-up type PWM method, the PWM waveform is automatically and continuously switched according to the motor operating frequency, so that it is not necessary to intentionally switch control to the synchronous PWM method or the like. Further, the current tracking type PWM method does not fall out of control in the high speed region unlike the control method based on the combination of the triangular wave comparison PWM and the dq axis current control, and one pulse method (including the synchronous PWM method) from the PWM region. The current control can be continuously performed up to the region.

  However, the following current error exists in principle in the current tracking type PWM. That is, the current tracking type PWM compares the current reference on the stator stationary coordinate and the current detection value by the comparator, and generates the PWM signal only by the magnitude relationship, so the proportional gain is infinite. If operated as it is, the switching frequency becomes too high. For this reason, a dead zone due to hysteresis and a delay time due to a timer are provided to limit the switching frequency. A steady-state error occurs in this dead zone and delay time.

  The wider the hysteresis width (dead zone), the lower the switching frequency of the inverter 3 and the larger the steady-state error (difference between the actual motor current average value and the current reference). That is, the lower the switching frequency, the larger the steady-state error, which may cause a reduction in the control performance of the motor.

  The current tracking type PWM has a great merit that a high-speed response can be obtained regardless of the switching frequency. Applying current tracking PWM to large industrial drives that use high-current switching elements, train main machine drives, etc. can speed up the current response without increasing the switching frequency, which can dramatically improve performance. . It is conceivable to actively lower the switching frequency to improve both performance and efficiency. However, in order to apply the current tracking type PWM to these applications, it is necessary to realize current control without a steady deviation. The steady deviation correction circuits 15d and 15q are for this purpose.

If the current detection values i _d and i _q are smaller than the current references i _d * and i _q *, the correction circuits 15 d and 15 _q increase their outputs i _dc * and i _qc *. As a result, the correction current references i _d ** and i _q ** are increased, so that the motor currents i _d and i _q are increased by the current tracking type PWM, and the current detection values i _d and i _q and the current reference i _d * are increased. , I _q * and the difference becomes small. If the correction circuits 15d and 15q have an integral element, even if the deviation output by the subtracters 14d and 14q is very small, it is integrated to obtain the correction current references i _d ** and i _q **. Since the correction is made, the steady deviation can be made zero for both the d-axis and the q-axis.

Since the induced voltage of the induction motor is small in the middle / low speed region of the motor, the current command i * and the detected current value i are substantially equal. Accordingly, since the correction signal i _qc * output from the correction circuit 15q is minute, the output signal of the weak control circuit 19 becomes negative and is limited to the lower limit value 0 by the limiter 20. Therefore, the magnetic flux command Φ * given to the vector calculation circuit 10 is equal to the strong magnetic flux command Φ **.

When the motor rotates at a high speed, the induced voltage increases and current, particularly q-axis current, cannot flow into the motor. This is because the induced voltage (ωΦ) is generated in the phase advanced by 90 ° relative to the magnetic flux Φ in the rotor d-axis direction, that is, in the q-axis direction (at this time, the d-axis direction magnetic flux Φ hardly changes. ). As a result, the current detection value iq decreases, and the output signal of the steady deviation correction circuit 15q on the q-axis side increases. When the 15q output signal exceeds the weakening start level i _qCLIM *, the output of the subtractor 18 turns positive, and the output of the weakening control circuit 19 starts to increase. As a result, the magnetic flux command Φ * input to the vector arithmetic circuit 10 becomes a value obtained by subtracting the output of the weakening control circuit 19 from the strong magnetic flux command Φ **, and starts to weaken. The i _d * output from the vector arithmetic circuit 10 becomes smaller and the motor magnetic flux Φ becomes smaller. As a result, the increase of the induced voltage is limited, and the value of the output signal of the correction circuit 15q is controlled to the limit value i _qCLIM ^* .

The q-axis side correction circuit 15q output i _qc * is added to the current reference I _q * by the adder 17q via the limiter 16. However, if the limiter 16 is not provided, the flux weakening may not be in time and the control may be impossible. is there. For example, when a large-amplitude power running torque command is given, i _qc * becomes a large signal transiently, so that flux weakening operates. Consequently, d-axis side current reference id * is narrowed, it becomes smaller than the current detection value i _d. As a result, i _dc * swings to the negative side. On the other hand, i _qc * swings to the positive side due to a large torque command. When the torque command is a large amplitude step change, the current deviation on the q-axis side is large and i _qc * increases rapidly. In one pulse region, q-axis current cannot flow unless the magnetic flux is weakened. For this reason, if there is no limiter 16, i _qc * increases in the positive direction. On the other hand, when the magnetic flux current reference i _dc * is reduced, i _dc * also increases to the negative side. Even if current follow-up type PWM control tries to control current without steady deviation on both the d-axis side and the q-axis side based on the current references i _d ** and i _q ** with this correction, the d-axis Both the side and the q-axis side can no longer pass the standard current.

The limiter 16 controls the q axis side while allowing a deviation transiently. On the other hand, since the d-axis side is not _limited , the magnetic flux is _continuously weakened while i _qc * is weakened and exceeds the control start level. Eventually, the magnetic flux is sufficiently weakened and i _qc * is controlled to a constant value of the weakening control start level. Since the steady deviation correction circuits 15d and 15q have integral elements, i _qc * is controlled to a constant value, i _q is controlled to be equal to i _q *. On the other hand, since the d-axis side current is always controlled so as not to have a deviation, both the currents i _d and i _q can be controlled according to the reference value even in one pulse region.

  When the switching frequency is limited by hysteresis in the current tracking control type PWM, the switching frequency varies depending on various factors such as the inductance of the motor, the inverter DC voltage, the motor rotation speed, and the current magnitude. If the switching frequency of the large-capacity switching element tends to fluctuate to the higher side, a large cooling capacity is required and the cooling fins must be enlarged. When trying to reduce the inverter loss at a switching frequency lower than that of the prior art by utilizing the high-speed tracking capability of the present method, the fluctuation of the switching frequency toward the higher side is not preferable. The switching frequency control circuit 25 is for this purpose.

  An example of the switching frequency control circuit is shown in FIG. In FIG. 4, 30 is a switching frequency detection circuit, 31 is a subtractor, 32 is an integrator, 33 is an inverse circuit, 34 is a limiter, 35 is a look-up table circuit, and 36 is an adder.

  In FIG. 4, the switching frequency detection circuit 30 measures the number of rising and falling times in a predetermined time for each gate signal of Gu, Gv, Gw, Gx, Gy, and Gz, and outputs the sum of them as a switching frequency detection value fsw. To do. The switching frequency can be detected from the three-phase PWM signal in the previous stage of the gate signal instead of the 6-arm gate signal, but it is preferable to detect the switching frequency from the gate signal. Since the gate signal is a signal after a dead time for preventing the upper and lower arms from being short-circuited, the gate signal eliminates the count of pulses removed during the dead time of the PWM signal. In addition, since the pulse with a very narrow width included in the PWM signal damages the element, the minimum width of the pulse is secured in the gate signal. The narrow pulse may be expanded to the minimum width, but may be removed due to the dead time, and the number of pulses differs between the PWM signal and the gate signal. For this reason, it is more accurate to count the switching frequency by using the gate signal actually applied to the element. In the case of a gate signal, there is an advantage that an accurate detection circuit can be easily made because there is no counting error if counting is performed with clock pulses having a period equal to or less than the minimum width.

  The subtractor 31 calculates a difference (fsw * −fsw) between the switching frequency detection value fsw and the switching frequency command value fsw *, and the deviation (fsw * −fsw) is integrated by the integrator 32. Since the hysteresis must be widened to lower the switching frequency, the reciprocal circuit 33 takes the reciprocal of the integral value. The limiter 34 limits the lower limit value of the reciprocal circuit output signal to 0 to obtain a switching frequency control signal HysI by feedback control. The control signal HysI becomes a larger value as the switching frequency command value fsw * is smaller (lower).

The look-up table circuit 35 has a built-in hysteresis table that shows the relationship between the hysteresis at which the switching frequency becomes a predetermined value and the operating frequency ωr for each current value of i _q *, and is appropriately interpolated and given ωr. And a switching frequency control signal HysFF corresponding to i _q *. The two switching frequency control signals HysI and HysFF are added by the adder 36, and are given as the hysteresis Hys to the current tracking type PWM control circuit 24 of FIG.

  Unlike the conventional combination method of dq axis current control and triangular wave comparison PWM control, current tracking type PWM control has a feature that the speed of current response is not affected by the switching frequency. Therefore, if the current tracking type PWM control is used, the switching frequency can be lowered without lowering the control performance, the heat generation of the element can be suppressed, and the cooling device can be downsized. However, the design of a specific cooling device requires not only that the switching frequency can be reduced more than before, but also a clear guarantee that the switching frequency will not rise above the predetermined frequency. Will be needed.

  If the control of the hysteresis width that influences the switching frequency of the current tracking type PWM control is performed only by integral control, an overshoot of the switching frequency is unavoidable when the current reference step changes, the operating frequency suddenly changes, or the like. That is, the change in the width of the hysteresis Hys occurs with a delay from the change in the operating frequency (speed reference ω *). Therefore, even if the current reference or the operating frequency suddenly changes and becomes a constant value, the switching frequency (the width of the hysteresis Hys) continues to change, and a phenomenon in which the switching frequency changes too much occurs. In particular, when the switching frequency must be controlled to be low, the time required for one switching frequency calculation process becomes long, so the response is slow, and the period during which the switching frequency is equal to or higher than a predetermined value is also long. In the case of only integral control, the control device attempts to control the switching frequency according to the command value up to the vicinity of one pulse region. However, since the operation frequency and the switching frequency are equal in the one-pulse region, the switching frequency control becomes unstable near the switching point.

  On the other hand, the hysteresis necessary for maintaining the switching frequency at a predetermined value fluctuates due to various factors. Therefore, feedforward control that can cope with all factors is extremely difficult.

  Accordingly, as in the switching frequency control circuit of FIG. 4, feedforward (path through the hysteresis table 35) and feedback (path including the circuits 30 to 34) are combined. In other words, the approximate value is given by feedforward and precisely adjusted by feedback control to satisfy both the transient response of the switching frequency and the control accuracy.

  As a result, highly accurate current control can be performed using the current tracking type PWM having excellent current response, and high-performance vector control excellent in both accuracy and response can be realized.

Also, in the conventional method combining the PI control type dq-axis current control and the triangular wave comparison PWM control, the q-axis current control output (voltage reference) is advanced so as not to exceed the q-axis voltage that can actually be output. I had to weaken it. However, in the current tracking PWM method, when the voltage for current control is insufficient, that is, when the induced voltage of the motor exceeds the maximum output voltage of the inverter (the output i _qc * of the correction circuit 15q is greater than or _equal to i _qCLIM *). When the magnetic flux weakening circuit 19 operates, the magnetic flux is weakened. This makes it possible to control the flux weakening with a complete one-pulse voltage, and increase the output capacity and improve the efficiency in the weakened region.

Japanese Patent No. 3267524 JP 2003-235270 A

In the method described above, the low speed range of the motor is PWM control with a low switching frequency, and the high speed range is 1 pulse control. Regardless of the PWM range / 1 pulse range, the current reference i _d * output from the vector arithmetic circuit, It is controlled to flow currents i _d and i _q faithfully to i _q *.

  Here, the difference in current deviation depending on the operating frequency of the motor is considered.

The low speed range in which performs PWM control, the current-following PWM, the voltage that can be regarded as average in the sine wave is applied to the motor, substantially current reference i _{u *,} i v _*, a sine wave is equal to _i w _* Currents i _u , i _v and i _w flow. Since the current deviations diu (= i _u * −i _u ), div (= i _v * −i _v ), and diw (= i _w * −i _w ) are small, the dq axis current deviation did (= i _d * −i _d ) and diq (= iq * -iq) are also small. In the methods of Patent Documents 2 and 3, most of the steady-state deviation appears in the q-axis current. Further, the methods of Patent Documents 2 and 3 indicate that “the inverter output voltage vector is a non-zero voltage vector and the current deviation is close to zero. When the current deviation is sufficiently close to zero, the inverter is switched to the zero voltage vector. The current deviation increases due to the change in the induced voltage, and when the magnitude of the current deviation reaches Hys, the inverter output voltage vector is switched to the non-zero vector again to reduce the current deviation. . As described above, since the induced voltage is generated on the q-axis, the q-axis portion dI _q of the current deviation is controlled to be approximately between zero and Hys. Therefore, the q-axis steady-state deviation is approximately 0 and the average value of Hys (Hys / 2). The steady deviation for the d-axis is almost zero. In both cases, the amount of steady-state deviation is almost constant.

  The appearance is different in one pulse region (also called mode) during high-speed rotation.

  FIG. 5 shows an output voltage waveform of each phase (here, u phase) of the inverter in the 1 pulse mode. Thus, in the 1-pulse mode, one voltage pulse is output in synchronization with the inverter period (θ = 0 to 2π), and the amplitude of this pulse matches the maximum output voltage of the inverter (the output voltage of the DC power supply 1). . A method of outputting one or more pulse voltages in synchronization with the inverter cycle is referred to as a synchronous PWM method. In the figure, the sine wave is the ideal waveform of the output voltage, and the dotted waveform is the waveform in the 3-pulse mode as another example of the synchronous PWM system. In the 3-pulse mode, 3 pulses P1, P2, and P3 are output in one cycle of the output voltage.

  In the 1-pulse mode, the electric motor is operated at high speed and maximum power. In the current tracking type PWM control method, the current deviation and the hysteresis Hys width increase as the rotational frequency increases, and the rotational frequency approaches the frequency of each output signal (gate signal switching frequency) of the subtractors 23U, 23V, and 23W. And shift to the 1-pulse mode. As described above, in this method, when the rotation frequency increases, the control mode automatically shifts from the PWM mode to the minority pulse mode and the one-pulse mode.

  The induced voltage is a sine wave, but the inverter output voltage is a square wave. For this reason, the current waveform becomes a non-sinusoidal wave, and the current deviations diu, div, diw are very large and change due to the electrical angle is large. Moreover, even if the operating frequency changes in one pulse region, the output voltage is constant and the magnitude of the induced voltage is almost constant. However, since the direction change of the induced voltage becomes faster as the operating frequency becomes higher, the control becomes higher as the operating frequency becomes higher. Gain is required. For simplicity, when trying to achieve a constant control gain over the entire range, a high gain that can be stably controlled at the maximum operating frequency must be employed.

However, the high gains of the correction circuits 15d, 15q, etc. have a large current deviation in the region where the operation frequency is low even in one pulse region, and the correction amounts i _dc *, i _qc * become too large and the control becomes unstable. , Produce the evil that.

Thus, the high gain causes inconveniences such as a large current ripple in the PWM region or a problem in control stability. If the amount of change in the current references i _d ** and i _q ** corrected by the steady deviation correction circuit due to the high gain increases, the current response of the current follow-up type PWM has a high speed, so that there is a probability that switching will occur immediately. Get higher. This widens the hysteresis to keep the switching frequency fsw constant, resulting in a large current ripple. As current ripple increases, motor harmonic loss increases. If the motor loss increases at the same switching frequency, it is caused by the high gain of the steady deviation correction circuit.

  In the first place, the role of the steady deviation correction circuit is completely different between the PWM region and the one-pulse region. In the PWM region, current control is carried out by current tracking type PWM control, and the steady deviation correction circuit only needs to eliminate the steady deviation caused by the current tracking type PWM control being proportional control by a correction circuit having only an integral term. On the other hand, in one pulse region, the current tracking type PWM control functions only as a PWM controller that merely generates a square wave, and it is the steady deviation correction circuit that actively controls the current.

  The object of the present invention is to switch the gain of the steady-state deviation correction circuit according to the pulse mode and the operating frequency, so that the switching frequency is not adversely affected in the PWM region, and the current deviation amount is small in one pulse region regardless of the operating frequency. An object of the present invention is to provide an inverter control device capable of obtaining a current response.

  One embodiment of the present invention directly generates a switching control signal for an inverter that drives an electric motor based on a comparison between the PWM correction current reference and the electric motor current. In the inverter control device using a current follow-up type PWM control circuit that automatically shifts to 1-pulse control, the deviation between the current reference obtained by vector calculation to control the magnetic flux and torque of the motor and the motor current is amplified. A steady-state deviation correction circuit that adds the steady-state deviation correction circuit output signal to the current reference to calculate a PWM correction current reference, and a current tracking control type PWM control circuit that operates in the PWM control mode. Switching circuit that sets the control in the steady deviation correction circuit to integral control and switches to proportional integral control when operating in the 1 pulse mode. Comprising a.

The structure of embodiment of the inverter control apparatus by this invention is shown. 1 shows a detailed configuration of a gain switching circuit according to an embodiment. The control structure of the motor control apparatus using the conventional current tracking type PWM control circuit is shown. The detailed structure of a switching frequency control circuit is shown. The output voltage waveform of each phase (here u phase) in the inverter in 1 pulse mode is shown.

  Embodiments of an AC motor control device according to the present invention will be described below with reference to the drawings.

[Constitution]
FIG. 1 is a diagram showing a configuration of an embodiment of an inverter control device according to the present invention. This inverter control device is an inverter control device using a current follow-up type PWM control circuit, and the same components as those in FIG. 3 are denoted by the same reference numerals, and detailed description thereof is omitted.

  In FIG. 1, the hysteresis Hys output from the switching frequency control circuit 25 is halved by the multiplier 26. The adder 27 adds the output signal of the adder 17q and the output signal of the multiplier 26.

The gain switching circuit 28 is a main part of this embodiment. FIG. 2 shows a detailed configuration of the gain switching circuit 28. The gain switching determination unit 40 generates a gain switching signal based on the flux-weakening control start level i _qCLIM * and the q-axis side steady deviation correction signal i _qc *, and _provides the gain switching signal to the integral gain controller 41 and the proportional gain controller 42. .

The gain switching determination unit 40 includes comparators 50 and 52, a zero level setting unit 51, and an RS flip-flop 53, and determines whether or not the operation of the current tracking type PWM control unit 24 is in one pulse region. Comparator 50 receives weak flux control start level i _qCLIM * and q-axis side steady deviation correction signal i _qc *, and q-axis side steady deviation correction signal i _qc * is greater than weak flux control start level i _qCLIM *. 1 and 0 otherwise. The comparator 52 generates 1 when the q-axis side steady deviation correction signal i _qc * is smaller than zero.

The output of the comparator 50 is connected to the set input of the RS flip-flop 53, and the output of the comparator 52 is connected to the reset input of the RS flip-flop 53. Therefore, the RS flip-flop 53 is set when the q-axis side steady deviation correction signal i _qC * exceeds the flux-weakening control start level i _qCLIM *, and is reset when it is less than zero (negative). As a result, the RS flip-flop 53 generates a logic signal of 1 when the operation of the current tracking type PWM control unit 24 is in the 1 pulse region. The output of the RS flip-flop 53 is given to the integral gain adjuster 41 and the proportional gain adjuster 42 as a gain switching signal.

The integral gain adjuster 41 includes a PWM domain integral gain setter 54, a single pulse domain integral gain setter 55, an integral gain reference frequency setter 56, a divider 57, a multiplier 58, and a selector 59. The integral gain (constant) Ki is provided to 15d and 15q. The set value K _i2 of the PWM region integral gain setter 54 is set to a value equal to or smaller than the set value K _i1 of the one-pulse region integral gain setter 55. The set value omega _r1 integral gain reference frequency setter 56, when the operating frequency omega _r of the motor is equal to the set value omega _r1, the multiplier 58 output is equal to the set value K _i1. This is merely for explanation. If K _i1 / ω _r1 is set from the beginning and inputted to the multiplier 58, only one constant setting unit is required, and the divider 57 is unnecessary. If the set value ω _r1 is set to a value that is approximately close to the weaker control start frequency, and the gain K _i2 in the PWM region and the gain K _{i1 in} the one-pulse region are the same, the gain when switching from the PWM region to the one-pulse region Are equal and it is easy to set the standard for adjustment, and there is no instability when switching. There is no need to set the gain in one pulse region lower than the gain in the PWM region.

  The proportional gain adjuster 42 includes a proportional gain setter 60 for PWM region, a proportional gain setter 61 for 1 pulse region, and a selector 62, and provides a proportional gain (constant) Kp to the steady deviation correction circuits 15d and 15q.

[Action]
A half value of the hysteresis Hys output from the switching frequency control circuit 25 in FIG. 1 is added to the output of the adder 17 q by the multiplier 26 and the adder 27. In the methods of Patent Documents 1 to 3 which are the premise of the present invention, the q-axis steady-state deviation in the PWM region is (Hys / 2), and is added to the current reference in advance. Even if this is not added, i _qc * bears “Hys / 2” due to steady deviation correction on the q-axis side. Therefore, it is not necessary to add, but in this case, the configuration of the gain switching circuit 28 is different. . In the embodiment, by adding “Hys / 2”, in the PWM region, the output of the steady deviation correction circuit may be substantially zero for both the d-axis and the q-axis. In the case of not adding, the value to be set by the setting device 51 connected to the A input of the comparator 52 of the gain switching circuit 28 may be set not to 0 (Hys / 2).

In FIG. 2, the output (gain switching signal) of the RS flip-flop 53 is set to 0 in the initial state. Since the induced voltage is small in the PWM region and the output i _qc * of the q-axis side steady deviation correction circuit 15q in FIG. 1 is almost 0, the output of the comparator 50 is 0 and the output of the RS flip-flop 53 remains zero. It is. As a result, the integral gain adjuster 41 outputs the set value K _i2 of the PWM region integral gain setter 54. On the other hand, the proportional gain adjuster 42 outputs the set value 0 of the proportional gain setting unit 60 for the PWM region. Thus, the steady deviation correction operation in the PWM region is a gentle integration control based on the integration gain K _i2 . Therefore, the ripples included in the correction signals output from the steady deviation correction circuits 15d and 15q in FIG. 1 are small, and the switching of the inverter 3 due to the output ripple of the steady deviation correction circuit can be suppressed.

When the rotational speed of the motor increases and approaches the one-pulse region, the q-axis current becomes difficult to flow due to the induced voltage increase, so that the output i _qc * of the steady deviation correction circuit 15q increases rapidly. By providing the current tracking type PWM control circuit 24 with a current reference i _q ** that is much larger than the value of i _q that can actually flow into the motor, the voltage shifts from a PWM waveform to a one-pulse waveform. At this time, even if the current reference i _q ** is large, there is no overcurrent. The current reference is i _q * output from the vector calculation circuit 10 to the last, and a current i _q equal to the command value i _q * cannot be flown into the motor. This is because a large induced voltage is generated due to high rotation. Although i _qc * output from the correction circuit 15q increases and i _q ** becomes larger, no overcurrent flows.

Here, if the magnetic flux weakening control start level i _qCLIM * is set to zero, the magnetic flux weakening control is always performed, and even if the rotation becomes high, the PWM does not shift to one pulse. As the magnetic flux weakening control start level i _qCLIM ^* is increased, the magnetic flux weakening control is started on the high rotation side, so that it becomes easier to shift to one pulse. Further, the frequency range in which minority pulses such as 3 pulses and 5 pulses immediately before shifting to 1 pulse appears becomes narrower. _i qCLIM ^* of the current command vector by a value _{(i d **, i q **} ) phase relationship between the current response vector _(i d, _{i q)} may vary. That _is, the compensation value to be added to the current command _{i q} * by _{i QCLIM} is the _limit, will also change the output value of the adder 17q _{when i QCLIM} is limit is worked if Kaware, resulting in the current command vector ( i _d **, i _q **) also changes, and the phase relationship with the current responses i _d , i _q changes. In the embodiment, by changing i _d ** by the correction signal i _dc * output from the d-axis side correction circuit 15d, a current having a phase equal to the original i _q * and i _d * output from the vector arithmetic circuit 10 is obtained. Keep flowing.

Modes such as PWM, a small number of pulses, and one pulse transition according to the acceleration / deceleration of the motor. For example, when the motor accelerates, the transition proceeds from PWM → few pulses → one pulse, and the transition timing corresponds to the rotational speed of the motor. When the magnetic flux weakening control start level i _qCLIM is increased, the pulse immediately shifts to one pulse even at a low rotational speed. Therefore, the value of the magnetic flux weakening control start level i _qCLIM ^* is selected mainly to set the appearance frequency region of the minority pulse when switching between PWM and one pulse.

In FIG. 2, when the correction signal i _qc * reaches the magnetic flux weakening control start level i _qCLIM *, the output of the comparator 50 becomes 1 and the RS flip-flop 53 is set. That is, it is determined that the inverter control mode is the 1 pulse mode. Thereby, the integral gain controller 41 outputs the gain K _i1 · (ω _r / ω _r1 ), and the proportional gain controller 42 outputs the gain K _p1 . Since the proportional gain K _p1 and the integral gain K _i1 · (ω _r / ω _r1 ) are set, the correction circuits 15d and 15q perform proportional integral control instead of integral control. Proportional integral control _allows changes in i _dc * and i _qc * to be performed at higher speeds, thus enabling higher-speed control than in the PWM mode. The integral gain K _i1 * (ω _r / ω _r1 ) needs a gain sufficient to obtain a response necessary for current control. In the 1-pulse mode, the inverter output voltage is a constant value (maximum value), but the induced voltage changes according to the rotation frequency. The integration coefficient is varied according to this change. That is, K _i1 is increased as the rotational frequency is higher.

On the other hand, the integral gain K _i2 in the PWM mode may be small. Since the switching frequency in the PWM mode is determined from the switching element power capacity to be used and its cooling capacity, when the high power switching element is used in a small cooler, the switching frequency must be considerably low. When calculating the switching frequency, the number of switchings generated during a predetermined time is counted and converted to the switching frequency. This predetermined time is set as a sampling time. Further, when calculating the switching frequency, sufficient accuracy cannot be obtained unless the number of switching samples is large to some extent. Accordingly, when the switching frequency is low, the sampling time becomes long, so that the switching frequency control is necessarily delayed. Integral gain _{K 12} of the PWM region from this point may be smaller than the integral gain _{K i1} of one pulse region.

If the switching frequency is fast, the values of K _i2 and K _i1 may be equal, and there is no necessity that Ki2> Ki1. According to the present embodiment, the speed of the steady deviation correction in the PWM region can be set irrespective of the speed of the current control in one pulse region, so that the steady deviation correction amount does not adversely affect the switching frequency control in the PWM region. Can be made smaller.

If the operating frequency further increases and the induced voltage increases, the q-axis current hardly flows, and i _qc * _tends to increase. When i _qc * exceeds i _qCLIM *, the magnetic flux weakening control circuit 19 weakens the magnetic flux, so that the induced voltage is controlled to a constant value and an increase in i _qc * is suppressed.

As the operating frequency increases, the rate of change of the current references i _u *, i _v *, i _w * and the induced voltage increases. Since the current tracking type PWM control used in this embodiment is based on a three-phase instantaneous value, a higher response is required as the operating frequency increases, but the integral gain is proportional to K _i1 · (ω _r / ω _r1 ). Therefore, even if the operating frequency is increased, current control with good followability becomes possible.

When the operating frequency decreases, the induced voltage decreases if the magnetic flux is constant. Therefore, since the current easily flows, the steady deviation correction circuit output i _qc * becomes small. As a result, the output of the magnetic flux weakening control circuit 19 decreases, and the magnetic flux command Φn * is gradually strengthened. As a result, even if the operating frequency is lowered, the induced voltage is kept constant, and the steady deviation correction circuit output i _qc * is also kept constant. When the operating frequency is further lowered and the output of the magnetic flux weakening control circuit 19 is lowered to zero, the magnetic flux is not further strengthened and becomes constant. At operating frequencies below this, the induced voltage begins to decrease with the operating frequency, and current tends to flow. The steady deviation correction circuit output i _qc * gradually decreases and reaches zero. Since there is always an overshoot and i _qc * is negative, the comparator 52 in FIG. 2 outputs 1 and the RS flip-flop 53 is reset.

As a result, the integral gain is set to the low gain K _i2 for the PWM region, and the proportional gain is set to zero. In the PWM region, the switching frequency control circuit 25 of FIG. 1 for adjusting the hysteresis Hys starts to operate in order to control the switching frequency. In this case, since the control response of the steady deviation correction circuit is suppressed to a low value of K _i2 , the steady deviation correction does not adversely affect the switching frequency control. In the PWM region, high-speed current control is performed by current tracking type PWM, and the steady-state deviation correction circuit provides a substantially constant steady-state deviation on both the d-axis and q-axis sides, so that the response need not be fast.

[effect]
An increase in switching frequency is directly linked to an increase in the size and weight of the element cooling device. For this reason, it is extremely important to suppress the switching frequency in a drive device using a large semiconductor element having a large switching loss.

  The current follow-up type PWM control as in this embodiment has a feature that the speed of the current response is not influenced by the switching frequency, unlike the combination method of the conventional dq axis current control and the triangular wave comparison PWM control. . Therefore, if current tracking type PWM control is used, the cooling frequency can be reduced by reducing the switching frequency and suppressing the heat generation of the element without lowering the control performance.

  However, in order to design a specific cooling device, it is necessary not only to reduce the switching frequency compared to the prior art, but also to have a clear guarantee that the switching frequency does not rise above a predetermined frequency. Management becomes necessary. Conventionally, steady-state deviation correction is adversely affected to control the switching frequency, resulting in an increase in current ripple despite the same switching frequency, resulting in a deterioration in motor efficiency.

  The present embodiment has been made to avoid this. In the PWM region, the current follow-up type PWM control is responsible for high-speed response, and the steady-state deviation is almost constant in the entire PWM region. For this reason, in this embodiment, steady-state deviation correction in the PWM region is set as low gain integral control. This can avoid adverse effects on the switching frequency control. Since only a constant constant deviation is corrected in the entire PWM region, the response of the steady deviation correction may be slow. The current follow-up type PWM control circuit responds to a step change in the current command, a current change due to a change on the load side, and the like.

  In the one-pulse region, the current follow-up type PWM control circuit operates as a simple square wave generator, and it is the steady-state deviation correction circuit that actively controls the current. Therefore, the steady-state deviation correction must be fast. In the present embodiment, the steady-state deviation correction in one pulse region is set to high gain proportional integral control. The integral gain is proportional to frequency. This enables high-speed current control in one pulse region.

  As described above, an increase in current ripple due to interference between steady-state deviation correction and switching frequency control, and hence deterioration in motor efficiency can be avoided.

[Other Embodiments]
In the case of an electric motor control device in which the switching speed of an element to be used is high and a switching frequency control circuit is unnecessary, the steady deviation correction circuit avoids an increase in switching frequency and a wide one pulse region. The present invention is effective for realizing current control with good followability over the operating frequency range.

In the above embodiment, the discrimination between the PWM mode and the one-pulse mode is performed by the gain switching determination unit 40 of FIG. 2 based on the magnitude of i _q *. However, in the 1-pulse mode, the switching frequency and the operating frequency coincide with each other, and in the PWM mode, the switching frequency is higher than the operating frequency. Therefore, the integration gain of FIG. You may provide to the regulator 41 and the proportional gain adjustment 42. FIG. In this case, when the comparison result is 1, an integral gain and a proportional gain for one pulse are provided to the correction circuits 15d and 15q.

  The above description is an embodiment of the present invention, and does not limit the apparatus and method of the present invention, and various modifications can be easily implemented. For example, various inventions can be configured by appropriately combining a plurality of constituent elements disclosed in the embodiment.

  DESCRIPTION OF SYMBOLS 1 ... DC power supply, 2 ... Capacitor, 3 ... Inverter, 4 ... Motor, 5U, 5V, 5W ... Hall CT, 6 ... Rotation sensor, 7 ... Rotation detection circuit, 8 ... Subtractor, 9 ... Speed control circuit, 10 ... Magnetic flux weakening function generator, 10 ... vector operation circuit, 11 ... adder, 12 ... current detection valuer, 13 ... coordinate conversion circuit, 14d, 14q ... subtractor, 15d, 15q ... steady deviation correction circuit 16 ... limiter, 17d , 17q ... adder, 18 ... subtractor, 19 ... magnetic flux weakening control circuit, 20 ... limiter, 21 ... subtractor, 22 ... coordinate conversion circuit, 23U, 23V, 23W ... subtractor, 24 ... current tracking type PWM control circuit , 25 ... switching frequency control circuit, 26 ... multiplier, 27 ... adder, 28 ... gain switching circuit, 30 ... switching frequency detection circuit, 31 ... subtractor, 32 ... integrator, 33 ... inverse circuit 34 ... Limiter, 35 ... Look-up table circuit, 36 ... Adder, 40 ... Gain switching determination unit, 41 ... Integral gain adjuster, 42 ... Proportional gain adjuster, 50, 51 ... Comparator, 52 ... Zero level setter 53... RS flip-flop, 54... PWM region integral gain setter 54, 55... One pulse region integral gain setter, 56... Integral gain reference frequency setter, 57. Selector, 60... PWM region proportional gain setter, 61... 1 pulse region proportional gain setter, 62.

Claims (5)

Based on the comparison between the PWM correction current reference and the motor current, it directly generates a switching control signal for the inverter that drives the motor. The control mode is PWM control in the middle and low speed range of the motor, and 1 pulse control and automatic transition in the high speed range. In an inverter control device using a current tracking type PWM control circuit that
A current reference obtained by vector calculation to control the magnetic flux and torque of the motor, and a steady deviation correction circuit that amplifies the deviation from the motor current;
An adding circuit for calculating the PWM correction current reference by adding the steady deviation correction circuit output signal to the current reference;
A switching circuit for setting the control in the steady deviation correction circuit to integral control when the current tracking control type PWM control circuit is operating in the PWM control mode, and switching to proportional integral control when operating in the one-pulse mode; ,
An inverter control device comprising:   2. The switching circuit determines whether the current tracking type PWM control circuit is in a PWM control mode or a one-pulse control mode based on the steady deviation correction circuit output signal. Inverter control device.   The switching circuit determines whether the current follow-up type PWM control circuit is in a PWM control mode or a one-pulse control mode based on a comparison result between a frequency of a switching control signal and a motor operation frequency. The inverter control device according to claim 1.   4. The inverter control device according to claim 1, wherein the switching circuit changes an integral gain of a steady deviation correction circuit in a one-pulse mode in proportion to an operation frequency. Based on the comparison between the PWM correction current reference and the motor current, it directly generates a switching control signal for the inverter that drives the motor. The control mode is PWM control in the middle and low speed range of the motor, and 1 pulse control and automatic transition in the high speed range. A control method in an inverter control device using current tracking type PWM control means,
In order to control the magnetic flux and torque of the motor, the deviation between the current reference iq * obtained by vector calculation and the motor current iq is amplified by the steady deviation correcting means,
Calculate the PWM correction current reference by adding the steady deviation correction means output signal to the current reference,
When the current tracking control type PWM control means is operating in the PWM control mode, the control by the steady deviation correcting means is set to integral control, and when operating in the one-pulse mode, switching to proportional integral control is provided. Inverter control method.

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