JP2007166690A - Motor control device - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To reduce the input current of a motor caused by periodical torque disturbance. <P>SOLUTION: In this motor control device provided with an inverter that drives the motor connected to a load, and a controller that drives the inverter by pulse-width modulation control, the controller extracts current variation amount related to an actual value of a motor current that flows to the motor, and is provided with a current variation compensation means that corrects the current command of the pulse-width modulation control by generating a correction signal that suppresses the current variation amount, thus reducing the input current due to the periodical torque disturbance. <P>COPYRIGHT: (C)2007,JPO&INPIT

Description

  The present invention relates to an electric motor control device, and more particularly to a technique for suppressing a periodic torque disturbance generated by a synchronous motor and a load.

  As a method for suppressing a periodic torque disturbance in which a load of a synchronous motor is generated, a technique described in Patent Document 1 is known. According to this, the pulsation component included in the detected speed value of the electric motor is extracted, and the inverter output voltage is corrected so as to cancel it out. In particular, if the speed fluctuation of the motor driving the rotary compressor as a load is smaller than the average speed fluctuation for several revolutions of the compressor, the speed fluctuation is not compensated, and if it is larger than the average speed fluctuation, the primary component of the speed fluctuation Compensation corresponding to is performed. As a result, the peak value of the motor current can be reduced as compared with the case where control is always performed in which the speed fluctuation is zero. Patent Documents 2 and 3 are known as conventional techniques related to motor control.

Japanese Patent Laid-Open No. 10-174488 JP 2002-272194 A JP-A-8-19263

  According to the control described in Patent Document 1, it is possible to suppress the periodic torque disturbance, but this document gives sufficient consideration to reducing the input current and the inverter loss. Not.

  An object of the present invention is to reduce the input current of an electric motor caused by periodic torque disturbance.

  In order to solve the above-described problem, the present invention provides an electric motor control device including an inverter that drives an electric motor connected to a load, and a controller that drives the inverter by pulse width modulation control. Current fluctuation compensation means for extracting a current fluctuation amount related to an effective value of the motor current flowing through the electric motor, generating a correction signal for suppressing the current fluctuation quantity, and correcting the current command of the pulse width modulation control is provided. Features.

  In other words, the current fluctuation compensation means extracts the current fluctuation relating to the effective value of the motor current, that is, the current component having a frequency component other than the fundamental wave of the motor current, and the current of the pulse width modulation control so as to suppress the current fluctuation. Since the command is corrected, for example, an increase in the effective value of the motor current due to load torque fluctuation can be suppressed. As a result, the input current can be reduced and the inverter loss can be reduced. In particular, even if the load torque fluctuation pattern varies depending on the type of compressor, operating pressure conditions, etc., it is not necessary to adjust the amount of correction to suppress the current fluctuation with a limiter, so the input current can be easily reduced. it can.

  In addition, the motor control device of the present invention detects a current fluctuation amount related to an effective value of the motor current flowing through the motor, generates a first correction signal that suppresses the current fluctuation amount, and generates a current for the pulse width modulation control. Current fluctuation compensation means for correcting the command, and vibration for detecting the load torque fluctuation amount of the motor, generating a second correction signal for suppressing the load torque fluctuation amount, and correcting the current command of the pulse width modulation control It can comprise a suppressing means, a switching means for switching the current fluctuation compensating means and the vibration suppressing means.

  According to this, it is possible to properly reduce the input current and appropriately suppress the torque disturbance according to the operating condition of the load by properly using the current fluctuation compensating means and the vibration suppressing means. For example, when the load is a compressor, the vibration and noise are controlled by controlling the vibration during the low rotation speed, and when the load is high and the load is low, the current fluctuation compensation means is switched to the inverter by low input operation. Loss can be reduced.

  Further, the current fluctuation compensating means can be configured to include limiter means for limiting a correction amount by the correction signal. As a result, control divergence due to detection error or the like can be prevented.

  ADVANTAGE OF THE INVENTION According to this invention, the input current of the electric motor resulting from a periodic torque disturbance can be reduced.

  Hereinafter, with reference to FIGS. 1-8, one Embodiment of the electric motor control apparatus of this invention is described. The present embodiment is a control device for a synchronous motor that drives a compressor of an air conditioner, and is an example in which a compressor is driven by a permanent magnet type synchronous motor (hereinafter abbreviated as PM motor). However, the electric motor targeted by the present invention is not limited to the PM motor, and can be similarly applied to other synchronous motors such as a winding synchronous motor and a reluctance motor. Moreover, a scroll compressor, a rotary compressor, and a reciprocating compressor can be used for a compressor.

  FIG. 1 is a block diagram of a PM motor control apparatus according to this embodiment. As shown in the figure, a DC power supply 4, an inverter 3 having a three-phase bridge configuration that converts a DC output from the DC power supply 4 into an AC having a desired frequency and voltage, a controller 2 that controls the inverter 3, and a controller 3 is provided with a rotation speed command generator 1 for giving a rotation speed command ωr * of the electric motor. The inverter 3 is driven in accordance with a pulse width modulated wave signal (PWM signal) output from the controller 2, applies an AC voltage having a desired frequency and voltage to the PM motor 5, and causes the compressor 6 that is a load of the PM motor 5 to operate. The drive is controlled. A current detector 7 is provided in a DC circuit connecting the DC power supply 4 and the inverter 3. The direct current power source 4 includes an alternating current power source 41, a diode bridge 42 that rectifies alternating current, and a smoothing capacitor 43 that suppresses pulsation contained in the direct current power source.

  The controller 2 has a function described below. The rotational speed command ωr * input from the rotational speed command generator 1 is input to the conversion gain 22. The conversion gain 22 converts the input rotational speed command ωr * into an angular frequency command ω1 * using the number of poles P of the PM motor 5, and outputs it to the integrator 23. The integrator 23 integrates the input electrical angular frequency command ω1 * and converts it into a magnet magnetic flux position θdc corresponding to the actual magnet magnetic flux position (phase angle) θd of the PM motor 5. The magnet magnetic flux position θdc is a magnet magnetic flux position of the PM motor 5 assumed inside the controller 2, as will be described later. In this specification, the subscript “*” of each symbol means a command, and “c” means an estimated value in the controller 2 or the like.

  On the other hand, the direct current I 0 detected by the current detector 7 is input to the current reproducer 8. The current reproducer 8 reproduces the three-phase alternating currents Iu, Iv, and Iw flowing through the PM motor 5 based on the direct current I0 by calculation and outputs them to the dq coordinate converter 9. The dq coordinate converter 9 coordinates the input three-phase alternating currents Iuc, Ivc, and Iwc into current components Idc and Iqc on the axes d and q based on the magnet magnetic flux position θdc output from the integrator 23. Convert. The converted d-axis current component Idc is input to the period Id * generator 10 described later. Further, the converted q-axis current component Iqc is input to the Iq * generator 14.

  A d-axis current command Id * and a q-axis current command Iq * are respectively output from the Id * generator 13 and the Iq * generator 14 to the voltage command calculator 15. Based on the d-axis current command Id *, the q-axis current command Iq *, and the angular frequency command ω1 * output from the conversion gain 22, the voltage command calculator 15 generates a d-axis voltage command Vdc * and a q-axis voltage command Vqc *. Is output to the dq inverse converter 16. The dq inverse converter 16 converts the d-axis voltage command Vdc * and the q-axis voltage command Vqc * into the three-phase AC voltage commands Vu *, Vv *, and Vw * based on the magnet magnetic flux position θdc output from the integrator 23. The data is converted and output to the PWM pulse generator 17. The PWM pulse generator 17 generates a pulse width modulation signal (PWM signal) for controlling the switch element of the inverter 3 based on the input three-phase AC voltage commands Vu *, Vv *, Vw *. As a result, the inverter 3 generates an AC voltage having a frequency and voltage according to the three-phase AC voltage commands Vu *, Vv *, and Vw * and supplies the AC voltage to the PM motor 5.

  Further, the Δθ estimator 18 calculates an axis error estimated value Δθdc corresponding to an error between the actual magnet magnetic flux position θd of the PM motor 5 and the magnet magnetic flux position θdc assumed in the controller 2, and performs an adder / subtracter. 19 output. The adder / subtracter 19 subtracts the axis error estimated value Δθdc from the reference value = 0 output from the zero generator 20 that generates the reference of the axis error estimated value Δθdc, and outputs the result to the proportional compensator 21. The proportional compensator 21 calculates an angular frequency correction value Δω1 for controlling Δθdc to zero, corrects the angular frequency command ω1 * via the adder / subtractor 19, and generates a magnet magnetic flux position θdc. It has become. Thus, coordinate conversion is performed in the dq coordinate converter 9 and the dq inverse converter 16 based on the magnet magnetic flux position θdc in phase with the magnet magnetic flux position θd.

The ΔTm estimator 24 calculates a torque pulsation amount ΔTmc of the PM motor 5 based on the axis error estimated value Δθdc estimated by the Δθ estimator 18 and outputs the torque pulsation amount ΔTmc to the torque control generator 25. The torque control generator 25 calculates a q-axis current correction command Iq _SIN * so as to cancel the shaft error Δθ caused by the torque pulsation component ΔTmc, and the q-axis current command via the switch 12 and the adder / subtractor 19. Iq * is corrected.

The period Id * generator 10 calculates the fluctuation amount of the d-axis current component Idc converted by the dq coordinate converter 9, obtains a d-axis current correction command Id _SIN * so as to cancel the fluctuation amount, and the adder / subtractor The d-axis current command Id * input to the voltage command calculator 15 via 19 is corrected. The period Iq * generator 11 calculates the variation of the q-axis current component Iqc transformed by dq coordinate converter 9, obtains a correction command Iq _{SIN *} of the q-axis current so as to cancel the fluctuation, The q-axis current command Iq * input to the voltage command calculator 15 is corrected via the switch 12 and the adder / subtractor 19. Further, the switch 12 switches between using the torque control generator 25 and the cycle Iq * generator 11 as a correction value for correcting the q-axis current command Iq * input to the voltage command calculator 15. It has become.

  The detailed configuration of the embodiment configured as described above will be described together with the operation. The rotational speed command ωr * input from the rotational speed command generator 1 is converted into the angular frequency ω1 * of the PM motor 5 by the conversion gain 22, and the phase θdc of the AC voltage applied to the PM motor 5 is obtained by the integrator 23. It is done. On the other hand, the current reproducer 8 calculates and reproduces the three-phase AC currents Iuc, Ivc, and Iwc flowing through the PM motor 5 based on the DC current I0 detected by the current detector 7. This reproduction calculation can be performed by, for example, a known method described in Patent Document 3 or the like. In short, the timing at which the switching elements of the respective phases connected to the three-phase bridge that are controlled by switching change from on to off and from off to on is detected based on the PWM signal, and the on / off change timings of the switching elements of each phase are detected. The difference between the DC currents I0 before and after is obtained, and the difference currents are defined as AC currents Iuc, Ivc, and Iwc of each phase.

  The three-phase AC currents Iuc, Ivc, and Iwc reproduced in this way are the d-axis current component Idc and the q-axis current component on the rotation orthogonal coordinate axis (dq axis) rotating at the angular frequency ω1c in the dq coordinate converter 9. Converted to Iqc. The q-axis current component Iqc is processed by the Iq * generator 14 to become a q-axis current command Iq *. On the other hand, the Id * generator 13 generates a d-axis current command Id *. However, in the PM motor 5 of a non-salient pole type rotor, Id * = 0 normally. When the d-axis current command Id * and the q-axis current command Iq * are input to the voltage command calculator 15, the voltage command calculator 15 receives the PM motor based on the angular frequency command ω1 * input from the conversion gain 22. 5 calculates a d-axis voltage command Vdc * and a q-axis voltage command Vqc * of the voltage applied to 5.

  The d-axis voltage command Vdc * and the q-axis voltage command Vqc * obtained by the voltage command calculator 15 are input to the dq inverse converter 16 and the three-phase AC voltage according to the phase θdc of the AC voltage output from the integrator 13. It is converted into commands Vu *, Vv *, Vw *. The PWM signal generator 17 generates a PWM signal having a pulse width corresponding to the amplitude and polarity of the three-phase AC voltage commands Vu *, Vv *, and Vw * of the AC amount. Although this PWM signal can be generated by calculation or the like, it is conceptually generated by modulating AC voltage commands Vu *, Vv *, Vw * with a carrier wave such as a triangular wave. By this PWM signal, the switching elements of the upper and lower arms configured in a three-phase bridge of the inverter 3 are turned on / off, and the PM motor 5 is driven and controlled by voltages according to the three-phase AC voltage commands Vu *, Vv *, Vw *. Is done.

  The Δθ estimator 18 calculates an error Δθdc between the magnet magnetic flux position θd of the PM motor 5 and the magnet magnetic flux position θdc in the controller 2. Here, the error Δθdc is defined by the vector diagram shown in FIG. The position (phase) of the actual magnet magnetic flux Φ inside the PM motor 5 is defined as the d-axis, and the axis orthogonal thereto is defined as the q-axis. On the other hand, the dq axis assumed in the controller 2 is defined as the dc-qc axis, and the deviation between the two corresponds to the axis error Δθdc. Therefore, if Δθdc can be obtained, the estimated value θdc is corrected so that the dq axis related to the actual magnet magnetic flux of the PM motor 5 matches the dc-qc axis assumed in the controller 2. Therefore, so-called magnetic pole position sensorless control can be realized.

  Specifically, for example, as shown in FIG. 3, the estimation calculation of Δθdc can be obtained by multiplying the difference between the q-axis current command Iq * and the q-axis current component Iqc by a proportional gain K0 to obtain an estimated value Δθdc. That is, since the q-axis current component Iqc varies according to the deviation between θd and θdc caused by load variation or the like, Δθ can be estimated on the contrary based on the variation of the q-axis current component Iqc. However, in the configuration of FIG. 3, it is difficult to obtain Δθdc with high accuracy. In order to improve accuracy, for example, as described in Patent Document 2, it is preferable to obtain Δθdc by the following equation (1). In the equation, R is the winding resistance of the PM motor 5, L is an inductance, and L = Ld = Lq in the case of a non-salient pole type.

Δθdc = tan ⁻¹ (Vdc * −R · Idc−ω1c · L · Iqc) /
(Vqc * -R · Iqc-ω1c · L · Idc) (1)
Based on the axial error estimated value Δθdc obtained in this way, the difference between the reference “0” output from the zero generator 20 and Δθdc is obtained by the adder / subtractor 19 so that this becomes zero, and the difference is added via the proportional compensator 21. Feedback control for correcting the angular frequency ω1 *. As shown in the vector diagram of FIG. 2, when Δθdc is positive, the dc-qc axis is more advanced than the dq axis. Therefore, Δθdc can be reduced by lowering the angular frequency ω1 * by the correction amount Δω1, and conversely, when Δθdc is negative, the angular frequency ω1 * is increased by the correction amount Δω1 to obtain the dq axis. Match the dc-qc axes. By controlling in this way, the magnet magnetic flux position θdc in the controller can be matched with the actual magnet magnetic flux position θd in the PM motor 5 without using the magnetic pole position sensor of the PM motor 5.

  An estimation calculation method of the torque fluctuation component ΔTm in the ΔTm estimator 24 will be described with reference to FIG. This estimation calculation is described in Patent Document 2. First, the relationship between the estimated axis error value Δθdc and the torque fluctuation component ΔTm is the relationship shown in FIGS. In the figure, symbol J is inertia of PM motor 5 and load 6, R is winding resistance of PM motor 5, L is inductance L of PM motor 5, P is the number of poles, Ke is a power generation constant (magnet magnetic flux), and s is The differential operator, Δωr, is the change in angular velocity at the mechanical angle.

  Here, when the torque fluctuation occurs about ΔTm when the motor is rotating at the angular velocity ωr, the angular velocity fluctuation amount Δωr can be expressed using the inertia J of the motor, and this can be expressed using the number of magnetic poles P. FIG. 4A is established by converting to the axis error Δθdc. If this is modified so that it can be inversely calculated, it becomes (d) of FIG. 4, and the torque fluctuation ΔTm can be extracted by substituting s = jωr * into this equation. Therefore, as shown in FIG. 4 (d), the shaft error estimated value Δθdc is used as an input, the torque fluctuation component ΔTm is calculated, and this is solved by Laplace transform.

  FIG. 5 shows a configuration of the ΔTm estimator 21 that embodies FIG. That is, the ΔTm estimator 24 includes a proportional gain 241 that increases Δθdc by 2J / P times and two multipliers 242, and performs the calculation of FIG.

  The torque control generator 25 is a vibration suppressing means, and as shown in FIG. 6 (1), a single-phase-dq coordinate converter 251, a first-order lag filter 252, an integral controller 253, a dq-single-phase inverse converter 254, The adder / subtractor 19 and the zero generator 20 are also included. Then, the output ΔTmc of the ΔTm estimator 24 is decomposed into a sin component ΔTqs and a cos component ΔTds by the single phase-dq coordinate converter 251. The conversion formula of the single phase-dq coordinate converter 251 is the following formula (2).

ΔTds = ∫ΔTmc × cos (ωr · t) dt
ΔTqs = ∫ΔTmc × sin (ωr · t) dt (2)
The expression (2) is not only the extraction of the ΔTmc fundamental wave and the harmonic component with the fundamental frequency ωr in the Fourier series expansion, but if the frequency component of the mechanical angular velocity ωr is included in ΔTmc, ΔTds, ΔTqs depending on the amount thereof. The average value of is a non-zero value, and the harmonic component appears as an AC amount of ΔTds and ΔTqs. Using this property, the same frequency component as ωr of the torque fluctuation component ΔTmc of the motor rotating at the average mechanical angular velocity ωr becomes the average value of the above equation (2). In order to extract this average value, the first-order lag filter 252 deletes the AC component. I _ATR is a first-order lag time constant, and an input error is compensated by adjusting this value. As a result, ΔTds and ΔTqs become a pulsation cos component and a sin component included in ΔTmc. Next, in order to make each component zero, a reference command “zero” signal is given from the zero generator 17, and a deviation from the reference command is calculated by the adder / subtractor 16. Based on these deviations, the integral controller 253 performs integral compensation and controls the pulsation to zero. K _iATR is a proportional gain, and control responsiveness can be changed by changing this value. The limiter unit 255 limits the correction current value calculated by the integration controller 253 and outputs the correction current values Ids and Iqs. As a result, control divergence due to detection error or the like can be prevented. Corrected current value Ids, Iqs outputted from the limiter unit 255, dq-by a single-phase converter 254 are converted back to a single-phase signal, it is outputted to the switch 12 as a correction command _{Iq SIN} * of the q-axis current component. That is, the correction command Iq _SIN * is added to the q-axis current command Iqc * input to the voltage command calculator 15 via the switch 12 via the adder / subtractor 19. The inverse transformation in the dq-single phase converter 254 is calculated according to the following equation (3).

Iq _SIN * = Iqsind + Iqsinq
Iqsind = ∫Ids × cos (ωr · t) dt
Iqsinq = ∫Iqs × sin (ωr · t) dt (3)
As described above, in the torque control generator 25, the torque pulsation amount ΔTmc becomes a direct current amount after the coordinate conversion is performed by the equation (2), and thus the deviation can be eliminated by the integration controller 253. That is, when viewed from the outside, the torque control generator 25 is equivalent to a compensation element whose gain is infinite at the angular frequency ωr.

  By the way, it is known that the input power of the compressor is minimized when the crest values of the motor current are aligned. In addition, in order to reduce vibration by suppressing load torque that fluctuates periodically, a correction signal that suppresses torque fluctuation is added to the inverter, and the correction amount for torque suppression is limited by a limiter so that the AC effective value can be reduced. Reduction has been done. However, the fluctuation pattern of the load torque of the compressor varies depending on the type of compressor, the operating pressure condition, and the like, and the correction amount for aligning the peak values of the motor current changes. Therefore, it is very difficult to always set an optimum value for the limiter.

Therefore, in the present embodiment, the period Id * generator 10 and the period Iq * generator 11 are provided to effectively reduce the input current of the PM motor 5. 6 (2) and 6 (3) show block configurations of the period Id * generator 10 and the period Iq * generator 11. As can be seen from the figure, the block configuration itself is the same as that of the torque control generator 25 shown in FIG. The period Id * generator 10 takes in the d-axis current component Idc output from the dq coordinate converter 9 and decomposes the variation ΔIdc into a sin component ΔIdds and a cos component ΔIdqs by the single-phase-dq coordinate converter 251. To do. Next, the AC component is deleted by the first-order lag filter 252, and the pulsation components of the sin component ΔIdds and the cos component ΔIdqs are extracted. In order to make these pulsations zero, the adder / subtracter 19 calculates a deviation from zero given from the zero generator 20. Based on these deviations, the integral controller 253 performs integral compensation and controls the pulsation to zero. K _IIdACR is a proportional gain, it can change the response of the control by changing this value. The limiter unit 255 limits the correction current value calculated by the integration controller 253 and outputs the correction current values Ids and Iqs. The correction current values Ids and Iqs output from the integration controller 253 are inversely converted into single-phase signals by the dq-single phase converter 254 and output as a d-axis current component correction command Id _SIN *. The correction command Id _SIN * is added to the d-axis current command Idc * input to the voltage command calculator 15 via the adder / subtractor 19.

Similarly, the period Iq * generator 11 takes in the q-axis current component Iqc output from the dq coordinate converter 9 and decomposes the variation ΔIqc into a sin component ΔIqds and a cos component ΔIqqs. Next, in order to extract those pulsations and make these pulsations zero, the adder / subtracter 19 obtains a deviation from zero given from the zero generator 20, performs integral compensation by the integral controller 253, and pulsations Control minutes to zero. K _iIqACR is a proportional gain, and the control responsiveness can be changed by changing this value. The limiter unit 255 limits the correction current value output from the integration controller 253 and outputs correction current values Ids and Iqs. Corrected current value Ids, Iqs outputted from the limiter unit 255, dq-by a single-phase converter 254 are converted back to a single-phase signal, it is outputted as a correction command _{Iq SIN} * of the q-axis current component. The correction command _{Iq SIN} * via the switch 12, are added through the q-axis current command Iqc * to the adder-subtracter 19 is input to the voltage command calculator 15.

  Thus, the calculation formulas themselves of the period Id * generator 10 and the period Iq * generator 11 are the same as those used in the torque control generator 25, and only the difference between ΔTmc, ΔIqc, and ΔIdc is input. It is.

As described above, according to the period Id * generator 10 and the period Iq * generator 11 of the present embodiment, the correction current command Id _SIN *, which cancels the pulsation of each frequency component in each axis of the d and q axes. Iq _SIN * can be generated. These correction current commands are corrected by adding to the d-axis current command Id * and q-axis current command Iq *, respectively, and by adjusting the command voltages Vdc * and Vqc * for driving the inverter 3, The crest values can be aligned.

Next, the function of the switch 12 will be described with reference to FIG. 7, the vertical axis represents the average load torque of the PM motor 5 (N · m), the horizontal axis represents the rotational speed of the compressor 6 ^{(min -1).} The average load torque was calculated from the DC current I 0 supplied from the DC power source 4 to the inverter 3. Further, as shown in the figure, for example, the rotational speeds N1 to N2 and N2 to N3 and the average load torques T1, T2 and T3 are set and divided into regions 1, 2, and 3 and a hysteresis region. In the range of the rotational speeds N1 to N2, torque fluctuation suppression control is performed as a torque control (ATR) region in the entire range of the average load torque. In the range of the rotational speeds N2 to N3, the cycle Iq * generator 11 is selected up to the average load torque T1, and current fluctuation suppression control is performed as a current fluctuation suppression region. Further, torque fluctuation suppression control is performed as a torque control (ATR) region in the region where the average load torque is T2 to T3 or more in the range of the rotational speeds N2 to N3. Further, a hysteresis region is set between the current fluctuation suppression region and the torque control (ATR) region, and a difference is given to the average load torque for switching the control according to the increase or decrease of the average load torque. In this embodiment, both the region 1 and the region 3 are torque control, but hysteresis is used intentionally so that this control method can cope with the case where the current fluctuation suppression control is to be used for the region 1 due to the characteristics of the load device.

  That is, as shown in the figure, in an overload where the average load torque is large (region 3) and in a region where the load fluctuation is large (region 1) such as during low rotation, the torque control generator 25 is selected to perform torque control (ATR). I do. On the other hand, in the other normal operation region (region 2), the cycle Iq * generator 11 is selected and input current reduction control is performed. Further, regarding the switching between the region 1 and the region 2, since the control of the present embodiment is a frequency control method, an error in the rotational speed is unlikely to occur, and as described above, the torque control generator 25 and the cycle Iq * generator 11 are Since both are integral control, stable control can be performed even during the transition period at the time of switching. However, when this control is used for the current control method, it is necessary to use hysteresis in consideration of the reading error of the rotational speed.

  FIG. 8 shows a comparison of input currents in the case of torque control which is vibration suppression control by the torque control generator 25 and in the case of current fluctuation compensation control (IqACR + IdACR) by the period Id * generator 10 and the period Iq * generator 11. Show. As can be seen from the figure, in the case of IqACR + IdACR, the motor current waveform can be made uniform, and the input can be reduced to 98.8% of about 1.2% with respect to 100% of the input in the case of torque control. It was.

  As described above, the compensation control by the period Id * generator 10 and the period Iq * generator 11 of the present embodiment is aimed directly at equalizing the crest values of the motor current, and the limiter adjustment is performed as in the prior art. Therefore, it is possible to easily reduce the input current. In other words, the variation pattern of the load torque of the compressor varies depending on the type of compressor, operating pressure conditions, etc., so the correction amount for aligning the peak value of the motor current changes, so the correction amount is set in advance. However, the conventional technology cannot cope with this. In this regard, according to the present embodiment, fluctuations of d and q-axis current components Idc and Iqc correlated with the detected value of the motor current are obtained, and d and q-axis current command Id *, so as to reduce the fluctuations. Since Iq * is corrected, the input current can be reduced without adjusting the limiter.

  Further, according to the present embodiment, in addition to being able to suppress periodic torque disturbance, the input current can be reduced, so that vibration and noise can be suppressed, and the loss of the inverter can be reduced, so that the operation of the highly efficient compressor can be reduced. It is possible to realize an air conditioner that makes it possible. In addition, an air conditioner using a compressor that could not be operated at low rotation due to vibration and noise can be operated at low rotation and high efficiency. As a result, a low-speed and high-efficiency motor can be used, and significant energy savings can be expected.

  In the above-described embodiment, the description has been made using specific numerical values. However, these numerical values are examples, and it goes without saying that other numerical values may be used as long as they conform to the control concept of the present invention. .

  In addition, as described with reference to FIG. 7, the example in which the control by the torque control generator 25 and the control by the period Iq * generator 11 are switched according to the average load torque and the load rotational speed has been shown. Not limited to this, the area selection parameter is increased depending on the load condition, the current flowing in each phase of the motor winding is detected by a shunt resistor or direct Hall CT instead of the DC current I0, or the AC power supply 41 and the diode bridge 42 are detected. It goes without saying that the same effect can be obtained even if a current detected by a current detecting means using a transformer or the like is used.

  Further, in FIG. 1, by using the fact that the internal configurations of the cycle Iq * generator 11 and the torque control generator 25 are the same, the switcher 12 determines whether the input information is ΔTm or ΔIqc as shown in FIG. It can be done in one process by switching using. As a result, the program capacity can be reduced, and when the torque control and the input reduction control are switched, the already accumulated integral term can be used as it is, so that the control can be stably performed.

It is a block block diagram of the electric motor control apparatus of one Embodiment of this invention. It is a figure explaining the relationship between PM motor and the dq axis coordinate in a controller. It is a block block diagram of one Embodiment of an axial error ((DELTA) (theta)) estimator. It is a figure explaining the computing equation which calculates | requires a torque pulsation part with a torque pulsation part ((DELTA) Tm) estimator. It is a block block diagram of one Embodiment of (DELTA) Tm estimator. It is a block block diagram of one Embodiment of a torque control generator, a period Id * generator, and a period Iq * generator. It is a figure explaining the system which switches and controls a torque control generator and a period Iq * generator with a switch. It is explanatory drawing which compares the reduction effect of the input current by a torque control generator and a period Iq * generator. It is a block block diagram of one Embodiment for performing a torque control generator and a period Iq * generator by one process.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 DC power supply 2 Inverter 3 Controller 4 Rotation speed command generator 5 PM motor 6 Compressor 7 Current detector 8 Current reproduction device 9 dq coordinate converter 10 Period Id * generator 11 Period Iq * generator 12 Switcher 13 Iq * Generator 14 Id * Generator 15 Voltage command calculator 16 dq inverse converter 17 PWM pulse generator 18 Δθ estimator 21 Proportional compensator 22 Conversion gain 23 Integrator 24 ΔTm estimator 25 Torque control generator 26 Torque control / Period Iq generator

Claims (6)

In an electric motor control device comprising an inverter that drives an electric motor connected to a load, and a controller that drives the inverter by pulse width modulation control, the controller includes a current fluctuation related to an effective value of a motor current flowing through the electric motor. An electric motor control device comprising current fluctuation compensation means for extracting a current and generating a correction signal for suppressing the current fluctuation and correcting the current command of the pulse width modulation control. In an electric motor control device comprising an inverter that drives an electric motor connected to a load, and a controller that drives the inverter by pulse width modulation control, the controller includes a current fluctuation related to an effective value of a motor current flowing through the electric motor. Detecting a minute amount, generating a first correction signal for suppressing the current variation amount and correcting a current command of the pulse width modulation control, and detecting a load torque variation amount of the motor, A vibration suppression unit that generates a second correction signal that suppresses load torque fluctuation and corrects the current command of the pulse width modulation control; and a switching unit that switches between the current fluctuation compensation unit and the vibration suppression unit. An electric motor control device. In the motor control device according to claim 1 or 2,
The electric motor control device according to claim 1, wherein the current fluctuation is a current fluctuation due to a periodic fluctuation of a load torque included in the motor current. In the motor control device according to claim 1 or 2,
The motor control apparatus according to claim 1, wherein the current fluctuation compensation means includes limiter means for limiting a correction amount by the correction signal. An inverter that drives a motor connected to a load by pulse width modulation control, and a detected value of a motor current flowing through the motor is converted into a d-axis current component and a q-axis current component corresponding to the rotating coordinate system of the magnetic flux axis of the motor. Dq coordinate conversion means for generating, d-axis current command generating means for generating a d-axis current command, q-axis current command generating means for generating a q-axis current command based on the q-axis current component, and the d-axis current command And a voltage command calculation means for generating a d-axis voltage command and a q-axis voltage command based on the q-axis current command and the angular frequency command of the motor, and the d-axis voltage command and the q-axis voltage command as an AC voltage command. In an electric motor control device comprising: an inverse conversion means for converting to a PWM signal generation means for generating a pulse width modulation signal based on the AC voltage command and driving the inverter;
Based on the d-axis current component converted by the dq coordinate conversion means, a fluctuation amount of the d-axis current component is detected, and a d-axis current correction command for suppressing the fluctuation amount is generated, and the d-axis current component is generated. D-axis current correction command generating means for correcting the command;
Based on the q-axis current component converted by the dq coordinate conversion means, a variation in the q-axis current component is detected, a q-axis current correction command for suppressing the variation is generated, and the q-axis current is generated. An electric motor control device comprising q-axis current correction command generation means for correcting the command. In the motor control device according to claim 5,
A torque fluctuation calculating means for obtaining a load torque fluctuation of the electric motor; a vibration suppressing means for generating a q-axis current correction command for suppressing the load torque fluctuation and correcting the q-axis current command; and the q-axis An electric motor control device comprising: a current correction command generation unit; and a switching unit that switches the q-axis current correction command of the vibration suppression unit to correct the q-axis current command.

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