JP2012050215A - Single-phase signal input device and system interconnection device - Google Patents

Abstract

PROBLEM TO BE SOLVED: To estimate a phase (instantaneous phase) of an AC single-phase signal with a low operation load.SOLUTION: A filter 11 generates a filter signal vby delaying the phase of a target single-phase signal v(vs) only by θ(0<θ<π/2). A vector component calculation part 12 can convert a signal vand a signal vinto a rotation vector on a fixed coordinate system using the phase difference θbetween the signal vand signal vand can also convert the signal vand signal vinto a vector on a rotating coordinate system using θand θ. θ represents a phase of the rotating coordinate system viewed from the fixed coordinate system. A phase difference calculation part 13 calculates a phase difference Δθ between θ and the phase of the target single-phase signal vbased on two orthogonal components vand vof the vector on the rotating coordinate system. An addition part 14 calculates an estimate θof the phase of the target single-phase signal vby adding θ to Δθ.

Description

  The present invention relates to a single-phase signal input device that estimates a phase state or the like of a single-phase signal. The present invention also relates to a grid interconnection device that performs control related to grid interconnection.

  Various methods for estimating the instantaneous phase of an alternating single-phase signal have been proposed. Among these, a method for detecting the phase and frequency of a single-phase signal from the zero-cross timing of the single-phase signal is widely known. However, the method using zero cross timing is vulnerable to noise and harmonic components because the zero cross timing easily changes depending on noise and harmonic components included in the signal. In view of this, various methods have been proposed that contribute to estimation of the phase state of a single-phase signal.

  In the first conventional method, a complex vector (rotation vector) is obtained from a single-phase signal using Hilbert transform (see, for example, Patent Document 1 below).

  In the second conventional method, a signal having a phase difference of π / 2 with respect to a single-phase signal is generated using a filter obtained by adding integral or differential characteristics to the band-pass filter, and the phase of the single-phase signal is estimated using the generated signal. (For example, see Patent Documents 2 and 3 below).

  In the third conventional method, a pseudo three-phase signal is generated from a single-phase signal, and the pseudo three-phase signal is three-phase / dq converted (see Non-Patent Document 1 below).

  The fourth conventional method proposes a single-phase voltage type AC / DC converter using a phase-delayed single-phase AC generator (for example, see Patent Document 4 below).

JP 2003-143860 A JP 2006-129681 A JP 2008-141935 A JP 2009-219263 A

Nagaishi, et al., “High-speed voltage detection and PLL of single-phase circuit using pseudo three-phase conversion”, Journal of Power Electronics Society, March, 2006, Vol. 31, JISE-31-18, p. 136-142

  However, in the first and second conventional methods, a high-order filter is required, so that the calculation load is heavy.

  Further, in the third conventional method, the calculation load becomes heavy because the generation of the pseudo three-phase signal is required.

Further, the phase delay of the filter processing performed by the fourth conventional method is π / 2 (90 degrees). In order to realize a phase delay of π / 2, a high-order filter is required, so that the calculation load is heavy. Note that Patent Document 4 discloses that an approximate solution can be obtained even when the phase delay amount deviates from π / 2 (when ω _s and ω _co in Patent Document 4 are not equal). Since the method described in Patent Document 4 is based on the premise that the phase delay amount is π / 2, an error due to approximation occurs when the phase delay amount deviates from π / 2, which causes a problem in accuracy. Occurs.

  Accordingly, an object of the present invention is to provide a single-phase signal input device that makes it possible to estimate a state (for example, a phase state) of a single-phase signal with a simple calculation. It is another object of the present invention to provide a grid interconnection device using such a single-phase signal input device.

  The first single-phase signal input device according to the present invention generates, as a filter signal, a signal whose phase difference from the target single-phase signal is smaller than π / 2 or an inverted signal of the target single-phase signal. And a signal derivation unit for deriving a derivation target using the target single-phase signal and the filter signal, and the derivation target has a phase difference of π / 2 with respect to the target single-phase signal. It includes a component of a vector signal on a rotating coordinate system synchronized with the frequency of the π / 2 phase difference signal or the target single phase signal.

  Thereby, a π / 2 phase difference signal or the like that is a derivation source of phase information or the like of the target single-phase signal can be derived using a filter signal whose phase difference from the target single-phase signal is less than π / 2. Since a filter signal having a phase difference from the target single-phase signal of less than π / 2 can be generated by a low-order filter, the calculation load is small.

  Specifically, for example, in the first single-phase signal input device, the signal derivation unit includes a trigonometric function using a phase difference between the target single-phase signal and the filter signal, the target single-phase signal, and the The derivation target is derived based on the amplitude ratio between the filter signals.

  The second single-phase signal input device according to the present invention includes a phase lag signal that is delayed in phase from the target single-phase signal and a phase advance that is advanced in phase from the target single-phase signal. A filter unit that generates a signal, and a signal derivation unit that derives a derivation target using the phase lag signal and the phase advance signal, and the derivation target has a phase difference of π / from the target single-phase signal. 2 or a π / 2 phase difference signal which is 2, or a vector signal component on a rotating coordinate system synchronized with the frequency of the target single-phase signal.

  Since the phase delay signal and the phase advance signal for deriving the π / 2 phase difference signal and the like can be generated by a low-order filter, the calculation load is small.

  Specifically, for example, in the second single-phase signal input device, the signal derivation unit uses a trigonometric function that uses a phase difference between the target single-phase signal and the phase delay signal or the phase advance signal. Based on this, the derivation target is derived.

  In addition, for example, a single-phase signal information estimation unit that estimates phase information of the target single-phase signal using a derivation result of the signal derivation unit may be further provided in the first or second single-phase signal input device. Also good.

  Further, for example, in the first or second single-phase signal input device, the single-phase signal information estimation unit uses at least a frequency and an amplitude of the target single-phase signal by using a derivation result of the signal derivation unit. One may be further estimated.

  The third single-phase signal input device according to the present invention generates, as a filter signal, a signal having a phase difference of π / 4 from the target single-phase signal or an inverted signal of the target single-phase signal. And a single phase signal information estimation unit configured to estimate phase information of the target single phase signal using the target single phase signal and the filter signal.

  Since the above filter signal can be generated by a low-order filter, the calculation load is small. In addition, by setting the phase difference between the target single-phase signal and the filter signal to π / 4, it is possible to further reduce the calculation for estimating the phase information, compared to setting the phase difference to other than π / 4. is there.

  Further, for example, in the third single-phase signal input device, the single-phase signal information estimation unit may further estimate the frequency of the target single-phase signal using the estimated phase information.

  A grid interconnection apparatus according to the present invention includes a power conversion circuit connected to a power system, and a control circuit that controls the power conversion circuit, the control circuit being between the power system and the power conversion circuit. The single-phase signal input device according to any one of the above that receives an alternating single-phase signal as a target single-phase signal, and an isolated operation detection unit that detects an isolated operation state of the power conversion unit using the single-phase signal input device And characterized by having

  According to the present invention, it is possible to estimate a state of a single-phase signal (for example, a phase state) with a simple calculation.

It is an internal block diagram of the phase difference estimation unit which concerns on 1st Embodiment of this invention. FIG. 4A is a diagram illustrating a relationship between an αβ coordinate system and a PQ coordinate system, and FIG. 4B is a diagram illustrating a locus of a target single-phase signal and a filter signal on the αβ coordinate system. It is a figure which shows the example of an internal structure of the vector component calculation part of FIG. It is a figure which shows the 1st example of the waveform of each signal in the phase difference estimation unit of FIG. It is a figure which shows the 2nd example of the waveform of each signal in the phase difference estimation unit of FIG. It is an internal block diagram of the phase estimation unit which concerns on 1st Embodiment of this invention. It is an internal block diagram of the phase / frequency / amplitude estimation unit according to the first embodiment of the present invention. It is a figure which shows the example of a waveform of the input-output signal and intermediate generation signal in the phase / frequency / amplitude estimation unit of FIG. It is an internal block diagram of the phase / frequency estimation unit which concerns on 1st Embodiment of this invention. It is an internal block diagram of the phase / frequency estimation unit which concerns on 1st Embodiment of this invention. It is an internal block diagram of the phase difference estimation unit which concerns on 2nd Embodiment of this invention. It is a figure which shows the example of an internal structure of the vector component calculation part of FIG. It is a figure which shows the 1st example of the waveform of each signal in the phase difference estimation unit of FIG. It is a figure which shows the 2nd example of the waveform of each signal in the phase difference estimation unit of FIG. It is an internal block diagram of the phase estimation unit which concerns on 2nd Embodiment of this invention. It is an internal block diagram of the phase / frequency / amplitude estimation unit according to the second embodiment of the present invention. It is an internal block diagram of the phase / frequency estimation unit which concerns on 2nd Embodiment of this invention. It is an internal block diagram of the phase / frequency estimation unit which concerns on 2nd Embodiment of this invention. It is a block diagram of the configuration of the grid interconnection system according to the third embodiment of the present invention. FIG. 20 is an internal block diagram of the coordinate conversion unit in FIG. 19. It is a figure which concerns on 3rd Embodiment of this invention and shows the transient response simulation result at the time of a single operation. It is a figure which concerns on 3rd Embodiment of this invention and shows the transient response simulation result at the time of a single operation. It is a block diagram of the configuration of a power supply system including a single-phase active filter according to a fourth embodiment of the present invention. FIG. 24 is an internal block diagram of a coordinate conversion unit in FIG. 23. It is a block diagram of the configuration of a motor drive system according to a fifth embodiment of the present invention. FIG. 26 is an internal block diagram of a coordinate conversion unit in FIG. 25. FIG. 26 is an internal block diagram of a rotation speed variation estimation unit that can be included in the voltage calculation unit of FIG. 25. It is a block diagram of the configuration of an actuator system according to a sixth embodiment of the present invention. It is an internal block diagram of the estimation unit of FIG.

Hereinafter, an example of an embodiment of the present invention will be specifically described with reference to the drawings. In each of the drawings to be referred to, the same part is denoted by the same reference numeral, and redundant description regarding the same part is omitted in principle. In addition, in this specification, for simplification of description, a symbol that represents a physical quantity or a symbol that refers to a component or the like is added, and a name corresponding to the symbol or the symbol may be omitted or abbreviated. For example, when referring to the filter signal by the symbol v _2, the filter signal v _2, for example it may be referred to as a signal v _2.

<< First Embodiment >>
FIG. 1 shows an internal block diagram of a phase difference estimation unit 1 (hereinafter sometimes abbreviated as estimation unit 1) according to the first embodiment of the present invention. The estimation unit 1 includes a filter 11, a vector component calculation unit 12, and a phase difference calculation unit 13.

The symbol vs represents an alternating current and single phase signal from which phase information and the like are to be derived. Hereinafter, the signal vs is also referred to as a target single phase signal. The type of physical quantity (voltage, current, speed, etc.) represented by the target single-phase signal vs is arbitrary. Symbols v ₁ and v ₂ are also symbols representing signals. Unless otherwise stated, the signal v ₁ is the same as the target single-phase signal vs. Therefore, hereinafter, not only vs but also v ₁ may be referred to as a target single-phase signal. The signal v ₂ is a signal obtained by performing a filtering process on the signal v ₁ , and the signal v ₂ is particularly called a filter signal.

  FIG. 2A shows the relationship between the αβ coordinate system and the PQ coordinate system. The αβ coordinate system is a two-dimensional fixed coordinate system having α and β axes as fixed axes as coordinate axes. The α and β axes are orthogonal to each other, and the β axis is advanced from the α axis by 90 degrees in electrical angle. In the figure showing an arbitrary coordinate system including FIG. 2A, the counterclockwise direction corresponds to the phase advance direction. The PQ coordinate system is a two-dimensional rotational coordinate system having the rotation axes P and Q as coordinate axes. The P and Q axes are orthogonal to each other, and the Q axis is advanced from the P axis by an electrical angle of 90 degrees. The phase of the P axis viewed from the α axis is represented by θ. θ represents the advance angle of the phase of the P axis with respect to the α axis.

In FIG. 2A, reference numeral 800 represents a vector on an arbitrary coordinate system. Vector 800 represents the phase and amplitude state of any signal. Therefore, a signal whose phase and amplitude state are represented by the vector 800 can also be called a vector signal. Assume that the vector 800 is a vector that rotates in synchronization with the rotation of the PQ coordinate system. Then, since the vector 800 rotates on the αβ coordinate system, the vector 800 on the αβ coordinate system can also be called a rotation vector. The α axis, β axis, P axis, and Q axis components of the vector 800 are represented by v _α , v _β , v _p, and v _q, respectively. Each of v _α , v _β , v _p and v _q is a kind of signal. For example, v _α is a signal having only an α-axis component (the same applies to v _β , v _p and v _q ). v _α and v _β are two orthogonal components (two components orthogonal to each other) of the rotation vector 800 on the αβ coordinate system, and v _p and v _q are two orthogonal components of the vector 800 on the PQ coordinate system. .

A known frequency given as the frequency of the target single-phase signal v ₁ is represented by ω _n . ω _n corresponds to the angular frequency of the fundamental wave component of the signal v ₁ , ignoring the harmonic component. The angular frequency in the rotation of the PQ coordinate system can be set to ω _n . Therefore, it can be said that the PQ coordinate system is a rotating coordinate system that rotates in synchronization with the frequency ω _n of the target single-phase signal v ₁ .

Signals v ₁ and v ₂ are
v ₁ = AMP _v1 · cos (θ _v1 ),
v ₂ = AMP _v2 · cos (θ _v2 ),
Represented by AMP _v1 and _{AMP v2,} respectively represent the amplitude of the signal _{v 1} and _{v 2,} theta _v1 and theta _v2 are respectively represents the phase of the signal _{v 1} and _{v 2.} The phase difference between the signals v ₁ and v _{2 is} represented by θ ₁ . “Θ ₁ = θ _v1 −θ _v2 ”. Therefore, the signal v ₂ is
v ₂ = AMP _v2 · cos (θ _v1 −θ ₁ ),
It is also expressed. When θ ₁ is positive, the signal v ₂ is delayed in phase by θ ₁ relative to the signal v ₁ .

In FIG. 2B, solid lines 802 and 803 represent the trajectories of the signals v ₁ and v ₂ on the αβ coordinate system, respectively. As can be seen from FIG. 2B, the α axis is set so that the signal v ₁ has only the α axis component. Therefore, v ₁ = v _α and the phase difference between the signal v _β and the target single-phase signal v ₁ is π / 2. Therefore, when considering the target single-phase signal v ₁ as a reference, the signal v _β can also be called a π / 2 phase difference signal. In this specification, π represents a pi, unless otherwise specified, the phase or angle is a phase or angle in electrical angle, and the unit of phase or angle expressed using π is radians.

When the matrix C ₁ (θ ₁ ) of the following equation (A-1) is used, the signals v ₁ and v ₂ can be converted into orthogonal two components v _α and v _β of the rotation vector 800 on the αβ coordinate system. . That is, Expression (A-1a) is established using the matrix C ₁ (θ ₁ ) of Expression (A-1). here,
R _AMP = AMP _v1 / AMP _v2 ,
It is. That is, R _AMP is the amplitude ratio between the signals v ₁ and v ₂ . Also,
θ ₂ = π / 2−θ ₁
It is. As apparent from the equation (A-1) and (A-1a), a signal v _beta as [pi / 2 phase difference signal can be represented by the formula (A-1b) or (A-1c) . Since the matrix C ₁ (θ ₁ ) can also be said to depend on θ ₂ , “C ₁ (θ ₁ )” can also be expressed as “C ₁ (θ ₂ )”.

The matrix C ₁ (θ ₁ ) is represented by, for example, an expression (A-2) when θ ₁ is π / 4, and is represented by an expression (A-2a) when θ ₁ is π / 3.

The vector 800 on the PQ coordinate system can be obtained by rotating the rotation vector 800 on the αβ coordinate system by the angle θ. A matrix C ₂ (θ) for rotating the rotation vector 800 on the αβ coordinate system by an angle θ is represented by the following equation (A-3), and is a product of the matrices C ₁ (θ ₁ ) and C ₂ (θ). A certain matrix C (θ) is represented by the following formula (A-4). Using the matrix C (θ), the signals v ₁ and v ₂ can be converted into orthogonal two components v _p and v _q of the vector 800 on the PQ coordinate system. That is, the equation (A-4a) is established between v ₁ and v ₂ and v _p and v _q .

Filter 11 of Figure 1 generates a filter signal v ₂ by performing filtering processing for delaying or advancing the phase of the filtered target signal to the signal v ₁ of the filtering target signal. The transfer function of the filter 11 is represented by F (s). In the filter 11, the gain of the signal _{v 1} at frequency omega _n is _{1 / R AMP.} Therefore, by passing through the filter 11, the amplitude of the signal component of the frequency ω _n in the signal v ₁ becomes 1 / R _AMP .

The vector component calculation unit 12 calculates the signals v _p and v _q from the signals v ₁ and v ₂ by performing coordinate transformation according to the above equations (A-4) and (A-4a) using θ ₁ and θ. To do. As the vector component calculation unit 12, the vector component calculation unit 12A in FIG. 3 can be used. The vector component calculation unit 12A is provided with coordinate conversion units 14 and 15. The coordinate conversion unit 14 calculates the signals v _α and v _β from the signals v ₁ and v ₂ according to the above formulas (A-1) and (A-1a) based on θ ₁ . The coordinate conversion unit 15 calculates signals v _p and v _q by multiplying a matrix having elements of the signals v _α and v _β calculated by the coordinate conversion unit 14 by a matrix C ₂ (θ). That is, by rotating the rotation vector 800 on the αβ coordinate system having the signals v _α and v _β as two orthogonal components by θ using the matrix C ₂ (θ), the signals v _p and v _q are converted into two orthogonal signals. A vector 800 on the PQ coordinate system is calculated as a component.

The phase difference calculation unit 13 obtains a phase difference Δθ between θ and the phase θ _v1 of the target single-phase signal v ₁ from v _p and v _q calculated by the vector component calculation unit 12. A formula for deriving the phase difference Δθ and a method of using the phase difference Δθ will be described later.

In this embodiment and other embodiments described later, each part that performs a calculation operation (including filter processing and the like) performs a calculation to be performed by itself using the latest signal value obtained at each time point. The signal value to be calculated and output by itself is updated at a predetermined update period. Therefore, the value of the derivation target derived at any part where the calculation operation is performed represents the instantaneous value of the derivation target. For example, the value of the filter signal v ₂ derived by the filter 11 at a certain time t is an instantaneous value of the filter signal v _{2 at} the time t, and the phase difference Δθ derived by the phase difference calculation unit 13 at a certain time t. The value is an instantaneous value of the phase difference Δθ at time t. Similarly, for example, θ _est derived by a phase estimation unit 2 (see FIG. 6) described later at time t is an instantaneous value (instantaneous phase) of the phase of the target single-phase signal v ₁ at time t.

Assuming that the phase difference θ ₁ between the signals v ₁ and v ₂ is π / 4 or π / 3, specific calculation operations of the filter 11 and the vector component calculation unit 12 will be described. However, as long as “0 <| θ ₁ | <π / 2” is satisfied, the value of θ ₁ may be other than π / 4 and π / 3.

[Specific example when θ ₁ = π / 4]
A specific calculation operation of the filter 11 and the vector component calculation unit 12 when the phase difference θ ₁ between the signals v ₁ and v ₂ is π / 4 will be described. When θ ₁ = π / 4, the filter 11 applies the filter signal according to the following equation (A-5) that causes the transfer function “F (s) = 2ω _n / (s + ω _n )” of the filter 11 to act on the signal v _1. v to generate the _2. Filter signal of the formula (A-5) _{v 2} is a phase delay signal whose phase is delayed by [pi / 4 from the signal _{v 1.} An operation symbol “s” appearing in the expression (A-5) and some expressions described later is a Laplace operator.

The signals v ₁ and v ₂ are converted into orthogonal two components v _α and v _β of the rotation vector 800 on the αβ coordinate system by the following equation (A-6) based on the above equations (A-1a) and (A-2). can do. Here, as the amplitude of the signal v ₂ is √2 times the amplitude of the signal v _1, the gain of the filter 11 corresponding to the formula (A-5) is set. That is, R _AMP = 1 / √2. “√2” is a positive square root of 2.

V _α and v _β obtained by the equation (A-6) can be converted into orthogonal two components v _p and v _q of the vector 800 on the PQ coordinate system by the following equation (A-7).

Rather than calculating v _α and v _β once using equation (A-6) and then calculating v _p and v _q using equation (A-7), v v and v _q are calculated using equation (A-8) below. It is also possible to obtain v _p and v _q directly from ₁ and v ₂ .

Furthermore, when the phase difference θ ₁ is π / 4 as in this example, v _p and v _q can be directly obtained from v ₁ and v ₂ using the following equation (A-9).

As described above, a single-phase vector signal (vector signal having two orthogonal components v _p and v _q ) on the rotating coordinate system can be obtained by a relatively simple calculation using a primary filter. FIG. 4 shows a waveform example of each signal in the estimation unit 1 when θ ₁ is π / 4. In FIG. 4, curves 810v ₁ , 810v ₂ , 810v _α , 810v _β , 810v _p and 810v _q represent waveform examples of signals v ₁ , v ₂ , v _α , v _β , v _p and v _q , respectively. Yes. The horizontal axis in the graph of FIG. 4 corresponds to time, and the unit of numerical values on the horizontal axis is seconds (the same applies to FIGS. 5, 8, 13, 14, 21, and 22 described later). From FIG. 4, it can be seen that the vector on the PQ coordinate system is accurately (or substantially accurately) obtained in about a half cycle of the target single-phase signal v ₁ even during the transient response.

The formula (A-5) filter the signal v ₂ obtained according to the the signal v by _one for [pi / 4 phase is delayed, filter a signal is advanced the phase by [pi / 4 with respect to the signal v ₁ The signal v ₂ may be generated. In this case, the filter signal v ₂ may be generated according to the following equation (A-5a). Filter signal _{v 2} of the formula (A-5a) is a phase advance signal is advanced in phase by [pi / 4 from the signal _{v 1,} when generating a filter signal _{v 2} by a formula (A-5a), "θ ₁ = −π / 4 ″.

Further, the filter 11 may be generating an inverted signal of the signal represented by the right-hand side of the formula (A-5) as a filtered signal v _2. An inversion signal of an arbitrary signal of interest refers to a signal obtained by inverting the sign of the signal of interest. That is, for example, the inverted signal of the signal v ₁ is (−v ₁ ). Therefore, the inverted signal of the signal represented by the right side of the equation (A-5) is represented by “− (2ω _n v ₁ / (s + ω _n ))”. Inverted signal of the signal represented by the right-hand side of the formula (A-5) is 3 [pi] / 4 by signal advanced in phase with respect to the signal v _1. Thus, the primary filter 11 can advance the phase of the signal by π / 2 or more. Similarly, in the filter 11, the inverted signal of the signal represented by the right-hand side of the formula (A-5a) (-. (2sv 1 / (s + ω n)) may also be generated as a filter signal _{v 2} Equation The inverted signal of the signal represented on the right side of (A-5a) is a signal delayed in phase by 3π / 4 with respect to the signal v _{1. In} this way, the phase of the signal is changed by the primary filter 11. It can also be delayed by π / 2 or more.

[Specific example when θ ₁ = π / 3]
A specific calculation operation of the filter 11 and the vector component calculation unit 12 when the phase difference θ ₁ between the signals v ₁ and v ₂ is π / 3 will be described. When θ ₁ = π / 3, the filter 11 generates the filter signal v ₂ according to the following equation (A-10) that causes the transfer function F (s) of the filter 11 to act on the signal v ₁ . Filter signal of the formula (A-10) _{v 2} is a phase delay signal whose phase is delayed by [pi / 3 from the signal _{v 1.}

The signals v ₁ and v ₂ are converted into orthogonal two components v _α and v _β of the rotation vector 800 on the αβ coordinate system by the following equation (A-11) based on the above equations (A-1a) and (A-2a). can do. Here, as the amplitude of the signal v ₂ is the same as the amplitude of the signal v _1, the gain of the filter 11 corresponding to the formula (A-10) is set. That is, R _AMP = 1.

V _α and v _β obtained by the equation (A-11) can be converted into orthogonal two components v _p and v _q of the vector 800 on the PQ coordinate system by the following equation (A-12).

Rather than calculating v _α and v _β once using equation (A-11) and then calculating v _p and v _q using equation (A-12), it is possible to calculate v using equation (A-13) below. It is also possible to obtain v _p and v _q directly from ₁ and v ₂ .

As described above, a single-phase vector signal (vector signal having two orthogonal components v _p and v _q ) on the rotating coordinate system can be obtained by a relatively simple calculation using a primary filter. FIG. 5 shows a waveform example of each signal in the estimation unit 1 when θ ₁ is π / 3. In FIG. 5, curves 815v ₁ , 815v ₂ , 815v _α , 815v _β , 815v _p and 815v _q represent waveform examples of the signals v ₁ , v ₂ , v _α , v _β , v _p and v _q , respectively. Yes. From FIG. 5, it can be seen that the vector on the PQ coordinate system is obtained accurately (or substantially accurately) in about a half cycle of the target single-phase signal v ₁ even during the transient response.

The formula (A-10) filter signal obtained according to v ₂ is only the signal v ₁ against [pi / 3 phase is delayed, filter a signal is advanced the phase by [pi / 3 with respect to the signal v ₁ The signal v ₂ may be generated. In this case, the filter signal v ₂ may be generated according to the following equation (A-10a). Filter signal _{v 2} of the formula (A-10a) is a phase advance signal is advanced in phase by [pi / 3 from the signal _{v 1,} when generating a filter signal _{v 2} by a formula (A-10a), "θ ₁ = −π / 3 ″.

Further, the filter 11 may be generating an inverted signal of the signal represented by the right side of the equation (A-10) as a filtered signal v _2. Inverted signal of the signal represented by the right side of the equation (A-10) is a 2 [pi / 3 only signal advanced in phase with respect to the signal v _1. Thus, the primary filter 11 can advance the phase of the signal by π / 2 or more. Similarly, the filter 11 may be generating an inverted signal of the signal represented by the right-hand side of the formula (A-10a) as a filtered signal v _2. The inverted signal of the signal represented by the right side of the equation (A-10a) is a signal delayed in phase by 2π / 3 with respect to the signal v ₁ . Thus, the primary filter 11 can delay the phase of the signal by π / 2 or more.

[Estimation method of phase, frequency and amplitude]
The phase difference Δθ between the angle θ used in the vector component calculation unit 12 and the phase θ _v1 of the signal v ₁ can be obtained by the following equation (A-14). The phase difference Δθ can be calculated by the phase difference calculation unit 13 using the equation (A-14) (see FIG. 1). Since “Δθ = θ _v1 −θ”, the phase θ _v1 of the target single-phase signal v ₁ can also be obtained according to the equation (A-15). In order to realize this, a phase estimation unit 2 (hereinafter sometimes abbreviated as estimation unit 2) as shown in FIG. 6 may be formed. The estimation unit 2 is obtained by adding an adding unit 14 to the estimation unit 1 of FIG. The adder 14 calculates the phase θ _est by adding the angle θ to the phase difference Δθ calculated by the phase difference calculator 13. The phase θ _est calculated by the adding unit 14 represents the value of the phase θ _v1 estimated by the phase estimation unit 2.

Alternatively, a phase / frequency / amplitude estimation unit 3 (hereinafter sometimes abbreviated as estimation unit 3) as shown in FIG. 7 may be formed. The estimation unit 3 is obtained by adding a PI control unit 16, an addition unit 17, and an integration unit 18 to the estimation unit 1 of FIG. In the estimation unit 3, θ _est is matched with θ _v1 by configuring a PLL (Phase Locked Loop) according to the following equations (A-16) and (A-17) for allowing Δθ to converge to zero. K _P and K _i are the proportional coefficient and integral coefficient of the proportional-integral control executed by the PI control unit 16, respectively.

Specifically, based on Δθ calculated by the phase difference calculation unit 13 and ω _n given as a known angular frequency, the PI control unit 16 and the addition unit 17 perform frequency according to the equation (A-16). ω _est is calculated. The integrator 18 calculates the phase θ _est by integrating the calculated frequency ω _est according to the equation (A-17). In the vector component calculation unit 12 of the estimation unit 3, the phase θ _est calculated by the integration unit 18 is used as the phase θ in the coordinate conversion of the vector component calculation unit 12. In the estimation unit 3, since the PLL is configured such that the phase difference Δθ between the phase θ (= θ _est ) in the coordinate conversion and the phase θ _v1 of the signal v ₁ converges to zero, the phase θ _est is It functions as an estimated value of the phase θ _v1 . Therefore, the frequency ω _{est that} is the _basis of the estimated value θ _est of the phase θ _v1 corresponds to the estimated value of the angular frequency of the target single-phase signal v ₁ . For example, the frequency of the target single-phase signal v ₁ is given to the system as a known nominal frequency ω _n (eg 60 × 2π), but the actual frequency of the target single-phase signal v ₁ (eg 59.5 × 2π). ) Often deviates from the nominal frequency ω _n . According to estimation unit 3 can estimate the true frequency of interest single-phase signal v ₁ as a frequency omega _est.

Also, if Δθ is zero, PQ coordinate system target single-phase signal v ₁ with rotated fully synchronized with the frequency of interest single-phase signal v ₁ has no Q-axis component. That is, when Δθ is zero, the target single-phase signal v ₁ is a signal on the P axis having only the P axis component. Therefore, the value of the signal v _p calculated by the vector component calculation unit 12 of the estimation unit 3 represents the amplitude of the target single-phase signal v ₁ . In the estimation unit 3, the value of the signal v _p calculated by the vector component calculation unit 12 is output as the amplitude AMP _est . As is clear from the above description, the amplitude AMP _est represents an estimated value of the amplitude of the target single-phase signal v ₁ (that is, the amplitude AMP _v1 of the signal v ₁ ).

FIG. 8 shows a waveform example of the input / output signal and the intermediate generation signal of the estimation unit 3 in FIG. In FIG. 8, curves 820v ₁ , 820v ₂ , 820v _p and 820v _q represent waveform examples of signals v ₁ , v ₂ , v _p and v _{q in} the estimation unit 3, respectively, and a curve 821 represents the estimation unit. 3 represents an example of a waveform of a signal obtained by dividing the frequency ω _est calculated in 3 by 2π, and a sawtooth line 822 represents a signal of the phase θ _est calculated in the estimation unit 3 (signal value is −π˜ An example of a waveform of vibration between π is shown. At the time of derivation of each waveform in FIG. 8, ω _n / 2π = 60 was set, and the above equations (A-5) and (A-9) were used. 8, even during the transient response, the target single-phase signal v ₁ of the target single-phase signal v ₁ of the phase in about half cycle, we can be seen that the frequency and amplitude are estimated exactly (or, generally accurate).

The configuration of the estimation unit 3 in FIG. 7 may be simplified to form a phase / frequency estimation unit 4 (hereinafter sometimes abbreviated as the estimation unit 4) as shown in FIG. The estimation unit 4 includes a filter 11, a vector component calculation unit 12, a PI control unit 19, and an integration unit 20. In the estimation unit 4, θ _est is matched with θ _v1 by configuring the PLL so that v _q converges to zero. Thus, the vector component calculator 12 of the estimation unit 4, of the two orthogonal components _{v p} and _{v q} vector 800 on PQ coordinate system, sufficient if derive _{v q} only. The method for deriving v _q is as described above. However, in the vector component calculation unit 12 of the estimation unit 4, the phase θ _est calculated by the integration unit 20 is used as the phase θ in the coordinate conversion of the vector component calculation unit 12. The PI control unit 19 calculates the frequency ω _est that is the source of the phase θ _est by proportional-integral control so that v _q calculated by the vector component calculation unit 12 converges to zero, and the integration unit 20 The phase θ _est is calculated by integrating the frequency ω _est calculated by the unit 19. v If caused to converge _q to zero, the target single-phase signal v ₁ becomes a signal on P axis with only P-axis component, this state Δθ corresponds to the state of zero. Accordingly, similarly to the estimation unit 3 of FIG. 7, the phase θ _est and the frequency ω _est calculated by the estimation unit 4 of FIG. 9 also function as estimated values of the phase and frequency (angular frequency) of the target single-phase signal v _1. To do.

Further, as shown in FIG. 10, the estimation unit 4 may execute the filter processing in the filter 11 using the frequency ω _est instead of the frequency ω _n . That is, the filter 11 may generate the filter signal v ₂ from the signal v ₁ after using the frequency ω _est calculated by the PI control unit 19 as the frequency ω _n . This is also applicable to the estimation unit 3 in FIG. That is, the filter 11 of the estimation unit 3 may generate the filter signal v ₂ from the signal v ₁ after using the frequency ω _est calculated by the PI control unit 16 and the addition unit 17 as the frequency ω _n. good. By performing the filter processing using the frequency ω _est , it is possible to appropriately cope with frequency fluctuations.

As described above, the estimation unit according to the present embodiment (1, 2, 3 or 4) estimates the phase information of the target single-phase signal v ₁ and the like. As the phase information of the target single-phase signal v ₁ , Δθ or θ _est is derived in each estimation unit (1, 2, 3, or 4) according to the present embodiment. In each estimation unit (1, 2, 3, or 4) according to the present embodiment, the filter 11 performs a filtering process on the target single-phase signal v ₁ to thereby obtain a phase difference from the target single-phase signal v _1. There generates an inverted signal of the smaller signal than the phase difference is [pi / 2 between the small signal or target single-phase signal v ₁ than [pi / 2 as a filter signal v _2.

The vector component calculation unit 12 in each estimation unit (1, 2, 3, or 4) has a function as a signal derivation unit. The signal derivation unit derives a derivation target using the target single-phase signal v ₁ and the filter signal v ₂ . The deriving target, for example, the target single-phase signal v ₁ phase difference [pi / 2 and is [pi / 2 phase difference signal v _beta with or rotating coordinate system that is synchronized with the frequency of the target single-phase signal v ₁ (PQ It is a vector signal component (v _p and v _q , or v _q only) on the coordinate system). Each estimator unit (1, 2, 3 or 4), be considered as a single-phase signal information estimation unit for estimating the phase information of the target single-phase signal v ₁ using the derived result of the signal derivation unit is provided Can do. For example, the part including the phase difference calculation unit 13 in FIG. 1, the part including the phase difference calculation unit 13 and the addition unit 14 in FIG. 6, the phase difference calculation unit 13, the PI control unit 16, the addition unit 17 and the integration unit in FIG. 18 or the part including the PI control unit 19 and the integration unit 20 in FIG. 9 (or FIG. 10) can be considered to function as a single-phase signal information estimation unit. Some single-phase signal information estimation units may further estimate at least one of the frequency and amplitude of the target single-phase signal v ₁ using the derivation result of the signal derivation unit.

  A single-phase signal input device having a filter unit as the filter 11, the signal derivation unit, and the single-phase signal information estimation unit is included in each estimation unit (1, 2, 3, or 4) according to the present embodiment. It can also be considered that each estimation unit (1, 2, 3 or 4) corresponds to a single-phase signal input device.

Additional signal derivation unit includes a trigonometric function using the phase difference theta ₁ between the target single-phase signal v ₁ and the filter signal v ₂ (sine or cosine function) between the target single-phase signal v ₁ and the filter signal v ₂ of The derivation target is derived using an arithmetic process depending on the amplitude ratio R _AMP . This calculation process is, for example, a calculation process of v _{β by} the above formula (A-1b) or (A-1c) when the derivation target is a π / 2 phase difference signal v _β , and the derivation target is on the rotating coordinate system. In the case of the vector signal component in, the component is calculated by, for example, the above formula (A-4) and formula (A-4a).

<< Second Embodiment >>
A second embodiment of the present invention will be described. In the second embodiment, with respect to matters that are not particularly described, the description in the first embodiment can be applied to the second embodiment.

FIG. 11 shows an internal block diagram of a phase difference estimation unit 31 (hereinafter sometimes abbreviated as estimation unit 31) according to the second embodiment of the present invention. The estimation unit 31 includes filters 41 _A and 41 _B , a vector component calculation unit 42, and a phase difference calculation unit 43. Similarly to the signal v _2, the signal v ₃ is for a signal obtained by performing filter processing on the signal v _1, can also signal v ₃ is referred to as a filtered signal.
Signals v _{1 to} v ₃ are
v ₁ = AMP _v1 · cos (θ _v1 ),
v ₂ = AMP _v2 · cos (θ _v1 −θ ₁ ),
v ₃ = AMP _v3 · cos (θ _v1 + θ ₁ ),
Represented by AMP _v1 , AMP _v2 and AMP _v3 represent the amplitudes of the signals v ₁ , v ₂ and v ₃ , respectively, and θ _v1 , (θ _v1 −θ ₁ ) and (θ _v1 + θ ₁ ) are the signals v ₁ , v, respectively. ₂ and v ₃ phases are represented. In this embodiment, the theta ₁ is positive. Therefore, the signal v ₂ is the phase lag signal whose phase is delayed by theta ₁ with respect to the signal v _1, the signal v ₃ is the phase advance signal advanced in phase by theta ₁ with respect to the signal v _1.

Using the matrix C _A (θ ₁ ) of the following equation (B-1), the signals v ₂ and v ₃ can be converted into orthogonal two components v _α and v _β of the rotation vector 800 on the αβ coordinate system. . That is, Expression (B-1a) is established using the matrix C _A (θ ₁ ) of Expression (B-1). Moreover, as is clear from the formula (B-1) and (B-1a), a signal v _beta as [pi / 2 phase difference signal can be represented by the formula (B-1b).

The matrix C _A (θ ₁ ) is represented by, for example, an expression (B-2) when θ ₁ is π / 4, and is represented by an expression (B-2a) when θ ₁ is π / 3.

The vector 800 on the PQ coordinate system can be obtained by rotating the rotation vector 800 on the αβ coordinate system by the angle θ. A matrix C _B (θ) for rotating the rotation vector 800 on the αβ coordinate system by an angle θ is expressed by the following equation (B-3). _A matrix C ′ (θ), which is the product of the matrices C _A (θ ₁ ) and C _B (θ), is expressed by the following equation (B-4). Using the matrix C ′ (θ), the signals v ₂ and v ₃ can be converted into orthogonal two components v _p and v _q of the vector 800 on the PQ coordinate system. That is, the equation (B-4a) is established between v ₂ and v ₃ and v _p and v _q .

Filter 41 _A in FIG. 11, to generate a phase lag signal v ₂ by performing filter processing to delay the phase of the filtered target signal to the signal v ₁ of the filtering target signal. That is, the filter 41 _B generates a phase advance signal v ₃ by applying a filter process for advancing the phase of the filtering target signal to the signal v ₁ as the filtering target signal. The transfer functions of the filters 41 _A and 41 _B are represented by F _A (s) and F _B (s), respectively.

The vector component calculation unit 42 calculates the signals v _p and v _q from the signals v ₂ and v ₃ by performing coordinate transformation according to the above formulas (B-4) and (B-4a) using θ ₁ and θ. To do. As the vector component calculation unit 42, the vector component calculation unit 42A of FIG. 12 may be used. The vector component calculation unit 42A is provided with coordinate conversion units 44 and 45. The coordinate conversion unit 44 calculates the signals v _α and v _β from the signals v ₂ and v ₃ according to the above formulas (B-1) and (B-1a) based on θ ₁ . The coordinate conversion unit 45 calculates signals v _p and v _q by multiplying a matrix having elements of the signals v _α and v _β calculated by the coordinate conversion unit 44 by a matrix C _B (θ). That is, by rotating the rotation vector 800 on the αβ coordinate system having the signals v _α and v _β as orthogonal two components by θ using the matrix C _B (θ), the signals v _p and v _q are converted into two orthogonal signals. A vector 800 on the PQ coordinate system is calculated as a component.

The phase difference calculation unit 43 obtains a phase difference Δθ between θ and the phase θ _v1 of the target single-phase signal v ₁ from v _p and v _q calculated by the vector component calculation unit 42.

[Specific example when θ ₁ = π / 4]
_A specific calculation operation of the filters 41 _A and 41 _B and the vector component calculation unit 42 when the phase difference θ ₁ is π / 4 will be described. When a θ ₁ = π / 4, the filter 41 _A and 41 _B, the transfer function of the filter 41 _A to the signal _{v 1 "F A (s)} = ω n / (s + ω n)" and the transfer function of the filter 41 _B according _{"F B (s) = s} / (s + ω n)" formula for applying a (B-5), the signal _{v 1} from [pi / 4 by a phase lag signal phase is delayed _{v 2} and the signal _{v 1} from [pi / A phase advance signal v ₃ whose phase is advanced by 4 is generated.

The signals v ₂ and v ₃ are converted into orthogonal two components v _α and v _β of the rotation vector 800 on the αβ coordinate system by the following equation (B-6) based on the above equations (B-1a) and (B-2). can do. Here, the gains of the filters 41 _A and 41 _B corresponding to the equation (B-5) are set so that the amplitudes of the signals v ₂ and v ₃ are 1 / √2 times the amplitude of the signal v _1. Yes. Therefore, it should be noted that the matrix included in the formula (B-6) is different from the matrix of the formula (B-2) by the coefficient “1 / √2”. Since v _α = v ₁ , the signal v ₁ itself may be obtained as the signal v _α without using the signals v ₂ and v ₃ .

V _α and v _β obtained by the equation (B-6) can be converted into orthogonal two components v _p and v _q of the vector 800 on the PQ coordinate system by the following equation (B-7).

Rather than calculating v _α and v _β using equation (B-6) and then calculating v _p and v _q using equation (B-7), the above equations (B-4) and (B− It is also possible to determine v _p and v _q directly from v ₂ and v ₃ using 4a).

As described above, a single-phase vector signal (vector signal having two orthogonal components v _p and v _q ) on the rotating coordinate system can be obtained by a relatively simple calculation using a primary filter. FIG. 13 shows a waveform example of each signal in the estimation unit 31 when θ ₁ is π / 4. In FIG. 13, curves 830v ₁ , 830v ₂ , 830v ₃ , 830v _α , 830v _β , 830v _p, and 830v _q are signals v ₁ , v ₂ , v ₃ , v _α , v _β , v _p, and v _{q, respectively.} This shows an example waveform. From FIG. 13, it can be seen that the vector on the PQ coordinate system is obtained accurately (or substantially accurately) in the half cycle of the target single-phase signal v ₁ even during the transient response.

[Specific example when θ ₁ = π / 3]
_A specific calculation operation of the filters 41 _A and 41 _B and the vector component calculation unit 42 when the phase difference θ ₁ is π / 3 will be described. When θ ₁ = π / 3, the filters 41 _A and 41 _B cause the transfer functions F _A (s) and F _B (s) of the filters 41 _A and 41 _B to act on the signal v ₁ (B− according 8) to generate a phase lag signal v ₂ and the signal v ₁ from [pi / 3 by the phase-lead signal v ₃ advanced phase with the signal v ₁ [pi / 3 by a phase is delayed.

The signals v ₂ and v ₃ are converted into orthogonal two components v _α and v _β of the rotation vector 800 on the αβ coordinate system by the following equation (B-9) based on the above equations (B-1a) and (B-2a). can do. Here, the gains of the filters 41 _A and 41 _B corresponding to the equation (B-8) are set so that the amplitudes of the signals v ₂ and v ₃ are the same as the amplitude of the signal v ₁ . For this reason, the matrix included in Formula (B-9) is the same as the matrix of Formula (B-2a). Since v _α = v ₁ , the signal v ₁ itself may be obtained as the signal v _α without using the signals v ₂ and v ₃ .

V _α and v _β obtained by the equation (B-9) can be converted into orthogonal two components v _p and v _q of the vector 800 on the PQ coordinate system by the following equation (B-10).

Rather than calculating v _α and v _β using equation (B-9) and then calculating v _p and v _q using equation (B-10), the above equations (B-4) and (B− It is also possible to determine v _p and v _q directly from v ₂ and v ₃ using 4a).

As described above, a single-phase vector signal (vector signal having two orthogonal components v _p and v _q ) on the rotating coordinate system can be obtained by a relatively simple calculation using a primary filter. FIG. 14 shows a waveform example of each signal in the estimation unit 31 when θ ₁ is π / 3. In FIG. 14, curves 835v ₁ , 835v ₂ , 835v ₃ , 835v _α , 835v _β , 835v _p, and 835v _q are signals v ₁ , v ₂ , v ₃ , v _α , v _β , v _p, and v _{q, respectively.} This shows an example waveform. From FIG. 14, it can be seen that the vector on the PQ coordinate system is accurately (or substantially accurately) obtained in about a half cycle of the target single-phase signal v ₁ even during the transient response.

[Estimation method of phase, frequency and amplitude]
The phase difference Δθ can be calculated using v _p and v _q obtained by the vector component calculation unit 42 and the estimated values θ _est and ω _est of the phase, frequency and amplitude of the target single-phase signal v ₁ can be calculated. And AMP _est can be calculated, and the configuration and method described in the first embodiment can be adopted as the configuration and method for calculating them.

That is, for example, the phase difference calculation unit 43 can calculate the phase difference Δθ according to the above equation (A-14) based on v _p and v _q obtained by the vector component calculation unit 42. Since “Δθ = θ _v1 −θ”, the phase θ _v1 of the target single-phase signal v ₁ can also be obtained according to the above equation (A-15). In order to realize this, a phase estimation unit 32 (hereinafter sometimes abbreviated as estimation unit 32) as shown in FIG. 15 may be formed. The estimation unit 32 is obtained by adding an addition unit 44 to the estimation unit 31 of FIG. The adding unit 44 calculates the phase θ _est by adding the angle θ to the phase difference Δθ calculated by the phase difference calculating unit 43. The phase θ _est calculated by the adding unit 44 represents the value of the phase θ _v1 estimated by the estimation unit 32.

  Alternatively, a phase / frequency / amplitude estimation unit 33 (hereinafter sometimes abbreviated as estimation unit 33) as shown in FIG. 16 may be formed. The estimation unit 33 is obtained by adding a PI control unit 46, an addition unit 47, and an integration unit 48 to the estimation unit 31 of FIG. The PI control unit 46, the addition unit 47, and the integration unit 48 are the same as the PI control unit 16, the addition unit 17 and the integration unit 18 in FIG. 7, respectively, and the description of the estimation unit 3 in FIG. Also applies. In this application, “estimation unit 3, vector component calculation unit 12, phase difference calculation unit 13, PI control unit 16, addition unit 17 and integration unit 18” in the explanatory text of the first embodiment are referred to as “estimation unit 33, What is necessary is just to read as the vector component calculation part 42, the phase difference calculation part 43, the PI control part 46, the addition part 47, and the integration part 48 ''.

Further, the configuration of the estimation unit 33 in FIG. 16 may be simplified to form a phase / frequency estimation unit 34 (hereinafter, abbreviated as the estimation unit 34) as shown in FIG. The estimation unit 34 includes filters 41 _A and 41 _B , a vector component calculation unit 42, a PI control unit 49, and an integration unit 50. The PI control unit 49 and the integration unit 50 are the same as the PI control unit 19 and the integration unit 20 in FIG. 9, respectively, and the description for the estimation unit 4 in FIG. 9 is also applied to the estimation unit 34. In this application, “estimation unit 4, vector component calculation unit 12, PI control unit 19 and integration unit 20” in the explanatory text of the first embodiment are referred to as “estimation unit 34, vector component calculation unit 42, PI control unit 49. In addition, the integration unit 50 ″ may be read.

Further, as shown in FIG. 18, the estimation unit 34 may execute the filter processing in the filters 41 _A and 41 _B using the frequency ω _est instead of the frequency ω _n . That is, the filters 41 _A and 41 _B may generate the signals v ₂ and v ₃ from the signal v ₁ after using the frequency ω _est calculated by the PI control unit 49 as the frequency ω _n. . This is also applicable to the estimation unit 33 in FIG. That is, the filters 41 _A and 41 _B of the estimation unit 33 use the frequency ω _est calculated by the PI control unit 46 and the addition unit 47 as the frequency ω _n , and then use the signals v ₁ to v ₂ and v ₃ . You may make it produce | generate.

As described above, the estimation unit according to this embodiment (31, 32, 33 or 34) estimates the phase information of the target single-phase signal _{v 1} and the like. As the phase information of the target single-phase signal v ₁ , Δθ or θ _est is derived in each estimation unit (31, 32, 33, or 34) according to the present embodiment. In each estimation unit (31, 32, 33, or 34) according to the present embodiment, the filter unit composed of the filters 41 _A and 41 _B performs the filter process on the target single-phase signal v ₁ , thereby performing the target unit. The phase delay signal v ₂ whose phase is delayed from the phase signal v ₁ and the phase advance signal v ₃ whose phase is advanced from the target single phase signal v ₁ are generated.

The vector component calculation unit 42 in each estimation unit (31, 32, 33, or 34) has a function as a signal deriving unit. The signal derivation unit derives a derivation target using the signals v ₂ and v ₃ . The deriving target, for example, the target single-phase signal v ₁ phase difference [pi / 2 and is [pi / 2 phase difference signal v _beta with or rotating coordinate system that is synchronized with the frequency of the target single-phase signal v ₁ (PQ It is a vector signal component (v _p and v _q , or v _q only) on the coordinate system). Each estimator unit (31, 32, 33 or 34), be considered as a single-phase signal information estimation unit for estimating the phase information of the target single-phase signal v ₁ using the derived result of the signal derivation unit is provided Can do. For example, the part including the phase difference calculation unit 43 in FIG. 11, the part including the phase difference calculation unit 43 and the addition unit 44 in FIG. 15, the phase difference calculation unit 43, the PI control unit 46, the addition unit 47, and the integration unit in FIG. It can be considered that the part including 48 or the part including the PI control unit 49 and the integration unit 50 in FIG. 17 (or FIG. 18) functions as a single-phase signal information estimation unit. Some single-phase signal information estimation units may further estimate at least one of the frequency and amplitude of the target single-phase signal v ₁ using the derivation result of the signal derivation unit.

  The single-phase signal input device including the filter unit, the signal derivation unit, and the single-phase signal information estimation unit is considered to be included in each estimation unit (31, 32, 33, or 34) according to the present embodiment. It can also be considered that each estimation unit (31, 32, 33 or 34) corresponds to a single-phase signal input device.

The signal deriving unit uses the arithmetic processing that depends on the trigonometric function (sine or cosine function) using the phase difference θ ₁ between the target single-phase signal v ₁ and the signal v ₂ (or v ₃ ). Deriving the derivation target. This calculation process is, for example, a calculation process of v _β by the above formula (B-1b) when the derivation target is a π / 2 phase difference signal v _β , and the derivation target is a component of a vector signal on the rotating coordinate system. In some cases, for example, the component is calculated by the above formula (B-4) and formula (B-4a).

<< Third Embodiment >>
A third embodiment of the present invention will be described. The technique described in the first or second embodiment can be applied to the third embodiment and the embodiments described later. In the specific examples of the various systems given in the third embodiment and each of the embodiments described later, the technique described in the first embodiment is mainly used. However, in the second embodiment, those specific examples are used. The described technique may be used.

  FIG. 19 is a configuration block diagram of a grid interconnection system according to the third embodiment. The grid interconnection system of FIG. 19 includes a solar cell 201 that is a DC power source, a single-phase inverter 202 that is a power conversion circuit, a control device (control circuit) 203 that controls the single-phase inverter 202, a single-phase inverter And an electric power system 204 that outputs AC power. The control device 203 is provided with each part referred to by reference numerals 221 to 229. A device including only the control device 203 or a device including the single-phase inverter 202 and the control device 203 can be regarded as a grid interconnection device. Arbitrary parts (for example, current detection sensor 205 and voltage detection sensors 206 and 207) shown in FIG. 19 other than the single-phase inverter 202 and the control device 203 may be regarded as being included in the components of the grid interconnection device. Is possible.

FIG. 19 shows an equivalent circuit of the solar cell 201. The solar cell 201 generates power based on solar energy and generates a DC voltage. DC voltage holding unit C _d holds the DC voltage by storing charge corresponding to a DC voltage solar cell 201 occurs. The voltage detection sensor 207 detects the voltage value of the DC voltage E _d held by the DC voltage holding unit C _d and sends a signal E _d having the detected voltage value as a signal value to the DC voltage control unit 221.

Single phase inverter 202 having a plurality of switching elements by pulse width modulation using a switching operation of the switching elements and converts a DC voltage from the DC voltage holding unit C _d to an AC voltage of a single phase. The AC voltage obtained by the single-phase inverter 202 is output from the terminals 210 and 211 with the potential of the terminal 210 as a reference. The power system 204 has an AC voltage source, and outputs a single-phase AC voltage (for example, so-called commercial AC voltage) from the terminals 212 and 213 with the potential of the terminal 212 as a reference.

  An interconnection point 208 is provided on the wiring connecting the terminals 210 and 212, and an interconnection point 209 is provided on the wiring connecting the terminals 211 and 213, and an AC voltage between the interconnection points 208 and 209 is provided. Is connected between the connection points 208 and 209. As described above, in the grid interconnection system of FIG. 19, the load 214 is driven using the output AC power of the single-phase inverter 202 and the output AC power of the power system 204.

  Lc represents the reactance component of the indoor wiring between the terminal 210 and the connection point 208 and the reactance component of the indoor wiring between the terminal 211 and the connection point 209, and Rc represents the resistance component of each of the indoor wirings. . Ls represents the reactance component of the outdoor wiring connecting the terminal 212 and the interconnection point 208 and the reactance component of the outdoor wiring connecting the terminal 213 and the interconnection point 209, and Rs represents the resistance component of each outdoor wiring.

Current sensor 205 detects a current value of the output current of the single-phase inverter 202, sends a signal i _z having a detected current value as the signal value to the coordinate converter 225. The voltage detection sensor 206 detects the voltage value of the output voltage of the single-phase inverter 202 (that is, the voltage between the terminals 210 and 211 with the potential of the terminal 210 as a reference), and a signal v _z having the detected voltage value as a signal value. Is sent to a band pass filter (BPF) 227. In the present embodiment, ω _n is a nominal frequency determined on the power system 204 side, and is, for example, 60 × 2π or 50 × 2π. BPF227 attenuates the harmonic components included in the signal _{v z} (high-order components of omega _n), and outputs the done signal _{v z} of this attenuation as the signal _{v z} '. For example, it is possible to generate a signal _{v z} 'from the signal _{v z} according to the following formula (C-1). ζ is a predetermined attenuation coefficient. In a system where the presence of harmonic components does not matter, BPF 227 can be omitted.

The estimation unit 228 receives the signal v _z ′ as the target single-phase signal vs, and estimates the phase, frequency, and amplitude of the signal v _z ′ as the target single-phase signal vs, θ _est , ω _est, and AMP _{est, respectively.} Output as. For example, the estimation unit 3 in FIG. 7 or the estimation unit 33 in FIG. 16 can be used as the estimation unit 228. Note that if it is not necessary to calculate all of θ _est , ω _est, and AMP _est , the estimation unit 2, 4, 32, or 34 can be used as the estimation unit 228 (see FIG. 6 and the like).

The output current of the single phase inverter 202 is formed from an effective current and a reactive current. Coordinate conversion unit 225, the active current forms the output current of the single-phase inverter 202 by converting a signal i _z into two orthogonal components of the vector on the rotating coordinate system based on the phase theta _est from estimation unit 228 and disabled current is calculated, to generate a reactive current signal i _y having a current value of the active current signal i _x and reactive current having a current value of the active current as the signal value as the signal value.

FIG. 20 shows an example of an internal block diagram of the coordinate conversion unit 225. The coordinate conversion unit 225 includes a filter 241 and a vector component calculation unit 242. The filter 241 and the vector component calculation unit 242 are the same as the filter 11 and the vector component calculation unit 12 in FIG. However, in the coordinate conversion unit 225, the signal _iz is handled as the target single-phase signal vs, the signals i _x and i _y are handled as the signals v _p and v _q (see also FIG. 1), and the phase θ The phase θ _est is used.

In addition to the signal E _d , a DC voltage command value E _d ^* is given to the DC voltage control unit 221.
DC voltage command value E _d ^* matches the value of signal E _d for obtaining maximum power from solar cell 201 (in other words, signal E _d for maximizing the output power of single-phase inverter 202). The DC voltage control unit 221 calculates and outputs an active current command value i _x ^* so that the value of the signal E _d matches the command value E _d ^* by proportional integral control or the like. The reactive current command value i _y ^* is set to zero. Active current controller 222 and the reactive current controller 223, respectively, so that the value of the value and the reactive current signal i _y of the active current signal i _x, such as by the proportional integral control is coincident with the command value i _x ^* and i _y ^* The voltage specified values v _x ^* and v _y ^* are calculated, and the coordinate conversion unit 224 converts the specified values v _x ^* and v _y ^* into voltage command values v _z ^* based on the phase θ _est from the estimation unit 228. The voltage command value v _z ^* is a target value of the output voltage of the single-phase inverter 202. Therefore, the PWM circuit 226 controls each switching element in the single-phase inverter 2 so that the output voltage value of the single-phase inverter 202 matches the voltage command value v _z ^* .

The following formulas (C-2) to (C-5) are calculation formulas of designated values i _x ^* , v _x ^* , v _y ^*, and v _z ^* that can be used in the control device 203. K _pe and K _p are proportional coefficients in the proportional-integral control, and K _ee and K _i are integral coefficients in the proportional-integral control.

The frequency ω _est and the amplitude AMP _est estimated by the estimation unit 228 can be used for detecting an isolated operation. Independent operation refers to a state in which power is supplied to the load 214 only by the single-phase inverter 202. An isolated operation can occur when a power failure occurs on the power system 204 side. For the purpose of ensuring safety or the like, when an isolated operation occurs, it is required to quickly detect this and stop the output of the single-phase inverter 202.

The isolated operation detection unit 229 can detect the isolated operation state of the single-phase inverter 202 based on the frequency ω _est or the amplitude AMP _est estimated by the estimation unit 228 (that is, whether or not an isolated operation has occurred). Can be detected). When the single operation occurs, the frequency of the output voltage of the single-phase inverter 202 can greatly deviate from the nominal frequency ω _n determined on the power system 204 side. Therefore, for example, the frequency ω _est and the nominal frequency ω _n are compared, and if the difference absolute value | ω _est −ω _n | exceeds a predetermined determination frequency ω _TH , it is determined that the isolated operation has occurred, and the difference If the absolute value | ω _est −ω _n | is equal to or lower than the determination frequency ω _TH , it can be determined that no isolated operation has occurred. In addition, when the single operation occurs, the amplitude of the output voltage of the single-phase inverter 202 can greatly deviate from the nominal voltage amplitude AMP _n determined on the power system 204 side. Therefore, for example, the amplitude AMP _est is compared with the nominal voltage amplitude AMP _n , and if the difference absolute value | AMP _est −AMP _n | exceeds a predetermined determination amplitude AMP _TH , it is determined that the isolated operation has occurred. If the difference absolute value | AMP _est −AMP _n | is equal to or less than the determination amplitude AMP _TH , it can be determined that no isolated operation has occurred.

  There are also methods for detecting an isolated operation by detecting a zero-cross timing or a peak timing of the output voltage of the inverter. However, these methods require a target single-phase signal for several cycles for detecting the isolated operation. On the other hand, the isolated operation detection unit 229 can detect the isolated operation at a higher speed than the method using the zero cross timing or the like.

FIG. 21 shows a transient response simulation result when the frequency of the target single-phase signal v ₁ as the signal v _z (or v _z ′) is abruptly shifted from the nominal frequency by 5 Hz (Hertz) by single operation. The frequency suddenly shifts at the time of 0.06 seconds). In FIG. 21, curves 840v ₁ , 840v ₂ , 840v _p and 840v _q represent waveform examples of signals v ₁ , v ₂ , v _p and v _{q in} the estimation unit 228, respectively, and a curve 841 represents the estimation unit. 8 shows an example of a waveform of a signal obtained by dividing the frequency ω _est calculated in 228 by 2π, and a sawtooth line 842 indicates a signal of the phase θ _est calculated in the estimation unit 228 (signal value is −π˜ An example of a waveform of vibration between π is shown. In the simulation corresponding to FIG. 21, the nominal frequency ω _n was set to (60 × 2π). FIG. 21 shows that the shifted frequency is estimated only about 0.01 seconds after the frequency of the target single-phase signal v ₁ is shifted.

FIG. 22 shows a transient response simulation result when the amplitude of the target single-phase signal v ₁ as the signal v _z (or v _z ′) is abruptly shifted from the nominal voltage amplitude by 10% by single operation. .. Amplitude suddenly shifts at time of 06 seconds). In FIG. 22, curves 845v ₁ , 845v _2, and 845v _p represent examples of waveforms of the signals v ₁ , v _2, and v _{p in} the estimation unit 228, respectively, and the curve 846 is calculated by the estimation unit 228. An example of a waveform of a signal obtained by dividing the frequency ω _est by 2π is shown, and a sawtooth line 847 is a signal of the phase θ _est calculated by the estimation unit 228 (signal value oscillates between −π to π). An example of a waveform is shown. In the simulation corresponding to FIG. 22, the nominal frequency ω _n was set to (60 × 2π). FIG. 22 shows that the signal v _p representing the amplitude after the shift is correctly estimated without waiting for about 0.01 second after the shift of the amplitude of the target single-phase signal v ₁ .

<< Fourth Embodiment >>
A fourth embodiment of the present invention will be described. FIG. 23 is a configuration block diagram of a power supply system including a single-phase active filter according to the fourth embodiment. The power supply system of FIG. 23 includes an AC power source 301 that outputs single-phase AC power, a single-phase active filter 302, a rectifier bridge 303, and a load 304. The single-phase active filter 302 includes portions that are referred to by reference numerals 311 to 315 and Cd. However, any part (for example, rectification bridge 303) shown in FIG. 23 other than each part referenced by reference numerals 311 to 315 and Cd can also be regarded as being included in the components of the single-phase active filter 302. It can also be considered that some of the parts referenced by reference numerals 311 to 315 and Cd are not components of the single-phase active filter 302. The harmonic current detection unit 311 includes portions that are referred to by reference numerals 321 to 326.

The AC voltage from the AC power source 301 is rectified to a DC voltage by the rectifier bridge 303 and then supplied to the load 304. The equivalent circuit of the load 304 here is, for example, a parallel circuit of a capacitance component and a resistance component. The current output from the AC power source 301 is represented by i _sr . When the output of the rectifying bridge 303 is supplied to the load 304, even if the output voltage of the AC power source 301 is a sine wave voltage, the current waveform of the output current i _sr of the AC power source 301 does not become a sine wave, and distortion is caused. Including. In consideration of harmonic regulation, it is preferable that such distortion is removed, and the single-phase active filter 302 removes the distortion.

The DC voltage is maintained at a DC voltage holding unit C _d expressed by v _dc. The DC voltage v _dc is converted into an AC voltage by a single-phase inverter 315 equivalent to the single-phase inverter 202 of FIG. The wiring to which the AC voltage from the single-phase inverter 315 is applied and the wiring to which the AC voltage from the AC power source 301 is applied are connected at a connection point 305. The current i _sr is a current that flows from the AC power source 301 to the connection point 305. Represents current flowing from the single-phase inverter 315 to the connection point 305 at _{i Cr,} represent a combined current of the currents _{i sr} and _{i Cr,} the current flowing from the connection point 305 to the rectifier bridge 303 at _{i Lr.} In FIG. 23, Ls and Rs respectively represent the reactance component and the resistance component of the wiring connecting the AC power source 301 and the connection point 305. A reactance component is also inserted between the connection point 305 and the rectifier bridge 303 and between the connection point 305 and the single-phase inverter 315 as necessary.

In the description of the present specification including the fourth embodiment, a symbol representing an arbitrary physical quantity (such as i _sr , i _Cr, and i _Lr ) is a symbol representing a corresponding physical quantity value, or a corresponding physical quantity value. Can also be used as a symbol of a signal having as a signal value. Therefore, for example, i _Lr can be used as a symbol of a current flowing from the connection point 305 to the rectifying bridge 303, a symbol of a current value of the current, or a symbol of a signal (current signal) having the current value as a signal value.

A current value of the current i _Lr is detected by a current detection sensor (not shown), and a signal i _Lr having the detected current value as a signal value is input to the harmonic current detection unit 311. The estimation unit 321 receives the signal i _Lr as the target single-phase signal vs, and outputs an estimated value of the phase of the signal i _Lr as the target single-phase signal vs as θ _est . For example, any estimation unit (2, 3, 4, 32, 33, or 34) described in the first or second embodiment can be used as the estimation unit 321.

The coordinate conversion unit 322 converts the signal i _Lr into vector orthogonal two components i _Ld and i _Lq on the rotating coordinate system based on the phase θ _est from the estimation unit 321. i _Ld is a signal component in phase with the fundamental wave component of the output current i _sr of the AC power source 301. FIG. 24 shows an example of an internal block diagram of the coordinate conversion unit 322. The coordinate conversion unit 322 includes a filter 341 and a vector component calculation unit 342. The filter 341 and the vector component calculation unit 342 are the same as the filter 11 and the vector component calculation unit 12 in FIG. However, in the coordinate transformation unit 322, the signal i _Lr is handled as the target single-phase signal vs, the signals i _Ld and i _Lq are handled as the signals v _p and v _q (see also FIG. 1), and the phase θ The phase θ _est is used.

In the present embodiment, ω _n is a nominal frequency determined on the AC power source 301 side, and is, for example, 60 × 2π or 50 × 2π. The frequency component of the signal i _Lr includes a fundamental wave component that matches the frequency ω _n of the output AC voltage of the AC power source 301 and a harmonic component corresponding to the fundamental wave component. Therefore, each of the signals i _Ld and i _Lq based on the signal i _Lr includes a fundamental wave component and a harmonic component. LPFs (low-pass filters) 323 and 324 execute low-pass filter processing for reducing harmonic components contained in the signals i _Ld and i _Lq , respectively. The signals i _Ld and i _Lq after the low-pass filter processing are represented by i _Ld ′ and i _Lq ′, respectively.

In order to appropriately remove current distortion due to harmonics by the single-phase active filter 302, it is necessary to control the DC voltage value v _dc to be a desired voltage value (for example, to maintain a constant voltage). The direct-current voltage control unit 312 uses proportional integral control or the like for the signal i _Ld ′ to make the difference between the voltage command value v _dc ^* representing the desired voltage value and the direct-current voltage value v _dc zero. A correction amount Δi _Ld is generated. The coordinate conversion unit 325 synchronizes the vector signal on the rotating coordinate system having the signal (i _Ld ′ + Δi _Ld ) and i _Lq ′ as orthogonal two components with the signal i _Lr based on the phase θ _est from the estimation unit 321. Converted to the single-phase signal i _Lr1 . The signal i _Lr1 corresponds to a fundamental wave component included in the signal i _Lr . Subtraction unit 326 subtracts the signal _{i Lr} from the signal _{i Lr1,} to generate a current command value _i ^Cr * corresponds to the target value of the current _{i Cr (= i Lr1 -i Lr} ). _iCr ^* corresponds to a harmonic component (distortion component) included in the signal _iLr . The current control unit 313 performs simple integration via the PWM circuit 314 so that the current value _iCr detected by a current detection sensor (not shown) matches the current command value _iCr ^* while using proportional-integral control or the like. The phase inverter 315 is controlled.

  In the example of FIG. 23, the current control is performed on the alternating current on the fixed coordinate system, but various other current controls can be performed. For example, the current control may be performed on the rotating coordinate system.

<< Fifth Embodiment >>
A fifth embodiment of the present invention will be described. FIG. 25 is a configuration block diagram of a motor drive system according to the fifth embodiment. The motor drive system of FIG. 25 includes a three-phase permanent magnet synchronization having a rotor (not shown) with permanent magnets and a stator (not shown) with U-phase, V-phase and W-phase armature windings. Motor 401 (hereinafter abbreviated as motor 401), inverter 402 that drives motor 401 by supplying three-phase AC voltage obtained by applying pulse width modulation to the DC voltage, and control of inverter 402 A motor control device 403 that controls the rotation of the motor 401 via a current detection sensor 404 is provided. The motor control device 403 is provided with respective parts referred to by reference numerals 411 to 414.

  The three-phase AC voltage supplied to the motor 401 by the inverter 402 is composed of U-phase, V-phase, and W-phase voltages representing voltages applied to the U-phase, V-phase, and W-phase armature windings. The total applied voltage to the motor 401, which is a combined voltage of the U-phase, V-phase, and W-phase voltages, is referred to as a motor voltage. The U-phase component, the V-phase component and the W-phase component of the current supplied from the inverter 402 to the motor 401 by the application of the motor voltage, that is, the currents flowing through the U-phase, V-phase and W-phase armature windings, respectively, Called phase, V phase and W phase currents. The total supply current to the motor 401, which is a combined current of the U-phase, V-phase, and W-phase currents, is referred to as a motor current.

The current detection sensor 404 detects the current value of the current for one phase among the U-phase, V-phase, and W-phase currents. In the example of FIG. 25, the W-phase current value _iw is detected by the current detection sensor 404, but a U-phase or V-phase current value may be detected instead.

Coordinate conversion unit 411 converts the W-phase current value i _w on the basis of the estimated phase theta _est at a position / speed estimator 412 in two orthogonal components i _d and i _q of the vector signal on the rotating coordinate system. The i _d and i _q obtained here are a d-axis current and a q-axis current corresponding to the d-axis component and the q-axis component of the motor current, respectively. In a rotating coordinate system that rotates at the same speed as the rotational speed of the magnetic flux generated by the permanent magnet of the motor 401, the axis along the direction of the magnetic flux generated by the permanent magnet is the d-axis, and the phase is 90 degrees in electrical angle from the d-axis. The advanced axis is the q axis.

FIG. 26 shows an example of an internal block diagram of the coordinate conversion unit 411. The coordinate conversion unit 411 includes a filter 441 and a vector component calculation unit 442. The filter 441 and the vector component calculation unit 442 are the same as the filter 11 and the vector component calculation unit 12 in FIG. However, in the coordinate conversion unit 411, the signal i _w is handled as the target single-phase signal vs, the signals i _d and i _q are handled as the signals v _p and v _q (see also FIG. 1), and the phase θ The phase θ _est is used. Further, the filter 441, the position / estimated by the speed estimating section 412 omega _est or given to the voltage calculation unit 413 omega ^* is used as the omega _n.

The position / speed estimation unit 412 estimates the rotor position and rotation speed of the motor 401 based on all or part of i _d , i _q , v _d ^*, and v _q ^* , and the estimated rotor position and estimated rotation. The speed is output as θ _est and ω _est , respectively. Any known method can be used as a method for estimating the rotor position and the rotational speed. The estimated rotor position θ _est corresponds to an estimated value of the d-axis phase viewed from the U-phase armature winding fixed axis, and the estimated rotational speed ω _est corresponds to an estimated value of angular frequency in the d-axis rotation. .

Voltage calculation unit 413, based on the current value i _d and i _q from the estimated rotation speed omega _est and coordinate conversion unit 411 from the position / speed estimator 412 utilizes such proportional-integral control, the estimated rotation speed omega _The voltage command values v _d ^* and v _q ^* are calculated so that _est matches the desired rotation speed designated value ω ^* . The voltage command values v _d ^* and v _q ^* are target values for the d-axis component and the q-axis component of the motor voltage, respectively. Based on the estimated rotor position θ _est , the coordinate conversion unit 414 converts the voltage command values v _d ^* and v _q ^* on the rotating coordinate system into voltage command values v _u ^* and v _v on a three-phase fixed coordinate system. Convert to ^* and v _w ^* . The voltage command values v _u ^* , v _v ^*, and v _w ^* are target values for the U-phase, V-phase, and W-phase voltages, respectively. The inverter 402 uses a pulse width modulation so that the voltage values of the U-phase, V-phase, and W-phase voltages that are actually applied to the motor 401 coincide with v _u ^* , v _v ^*, and v _w ^* , respectively. A three-phase AC voltage for 401 is generated, and the generated three-phase AC voltage is applied to the motor 401.

As described above, when the method of the present invention for estimating the phase of the target single-phase signal is used, the current vector (current vector having i _d and i _q as two orthogonal components) from the current information for one phase in the multiphase motor. Information can be obtained. Although the motor control device 403 of FIG. 25 has a control configuration for vector control, the method of the present invention can be applied to other control configurations. For example, in a control configuration that adjusts the phase of the voltage applied to the motor based on the phase difference between the voltage vector that is the vector of the applied motor voltage and the current vector that is the detected motor current vector, The method of the present invention is applicable.

[Estimation of periodic rotational speed fluctuation]
In the voltage calculation unit 413 of FIG. 25, a rotation speed fluctuation estimation unit 450 (hereinafter sometimes abbreviated as the estimation unit 450) as shown in FIG. 27 may be provided. The estimation part 450 is provided with each site | part referred with the codes | symbols 451-456. When the load state of the motor 401 fluctuates periodically, the rotation speed of the motor 401 may fluctuate periodically even though the specified rotation speed value ω ^* is constant. The estimation unit 450 can estimate the phase and amplitude of such fluctuation, that is, the phase and amplitude of the signal (ω ^* −ω _est ).

The filter 451, vector component calculation unit 452, and phase difference calculation unit 453 are the same as the filter 11, vector component calculation unit 12, and phase difference calculation unit 13 of FIG. However, in the estimation unit 450, the signal (ω ^* −ω _est ) representing the difference between ω ^* and ω _est is handled as the target single-phase signal vs, and the angular frequency ω _r of the fluctuation of the signal (ω ^* −ω _est ). Is handled as ω _n in the filter 451, and the phase obtained by integrating the angular frequency ω _r is handled as θ in the vector component calculation unit 452. Integral unit 454, to calculate the phase θ by integrating the angular frequency ω _r. The adder 455 adds the phase difference Δθ and the phase θ calculated by the phase difference calculator 453 and the integrator 454 to obtain the estimated value θ _r of the phase of the signal (ω ^* −ω _est ). The amplitude calculation unit 456 obtains an estimated value Δω of the amplitude of the signal (ω ^* −ω _est ) based on v _p and v _q from the vector component calculation unit 452. θ _r = Δθ + θ and Δω = √ (v _p ² + v _q ² ). It should be noted that the value of ω _r is often known. For example, when the load of the motor 401 is a drum of a washing machine, the value of ω _r is determined from the mechanical rotation speed of the drum.

By generating a torque command according to the variation of the load state of the motor 401 based on the phase and the estimated value of the amplitude theta _r and [Delta] [omega, it is possible to achieve the suppression of vibration due to the fluctuation. The torque refers to the torque of the motor 401, and the torque command refers to the target value of the torque of the motor 401.

Alternatively, when the motor 401 is a motor for a washing machine, it can also be utilized an estimate theta _r and Δω to the determination of the load balancing state. The load balance state can be determined by determining whether the clothes are evenly distributed in the drum of the washing machine, or if the clothes are not evenly distributed in the drum of the washing machine. This includes determining whether clothing is concentrated on the part. Δω in a state where clothing is evenly dispersed in the drum is smaller than Δω in a state where clothing is not evenly dispersed in the drum. Therefore, for example, based on Δω, it can be determined whether clothing is evenly distributed in the drum. Further, when clothing is not evenly distributed in the drum of the washing machine, it is possible to infer which part of the drum the clothing is concentrated by referring to θ _r . Using this analogy result, it is possible to perform motor control for uniform clothing distribution.

In addition, the method for estimating the phase and amplitude in the fluctuation of the rotation speed can be applied to the fluctuation of the torque. In this case, by passing the torque or q-axis current value i _q was estimated based on the known torque equation to the band-pass filter extracts only the fluctuation component of the torque. Then, the extracted fluctuation component signal may be input to the estimation unit 450 in FIG. 27 instead of the signal (ω ^* −ω _est ). Then, as in the case where the signal (ω ^* −ω _est ) is input to the estimation unit 450, the phase and amplitude in the torque variation can be estimated. The load balance state can also be determined based on these estimation results regarding torque.

<< Sixth Embodiment >>
A sixth embodiment of the present invention will be described. FIG. 28 is a configuration block diagram of an actuator system according to the sixth embodiment. The actuator system shown in FIG. 28 includes an actuator 501 that is a linear vibration actuator and a single-phase AC voltage obtained by applying pulse width modulation to a DC voltage. An inverter 502 to be driven, an actuator control device 503 for controlling the vibration state of the actuator 501 through the control of the inverter 502, and a current detection sensor 504 are provided. The actuator control device 503 is provided with an estimation unit 511 and a single-phase voltage command unit 512. The actuator 501 is provided with a stator and a mover that vibrates with reference to the position of the stator. For example, an armature winding for flowing a current from the inverter 502 can be provided in the stator of the actuator 501, and a permanent magnet can be provided in the mover of the actuator 501.

Current sensor 504 detects a current value of the current supplied from the inverter 502 to the actuator 501 by applying an AC voltage of a single phase to the actuator 501, the output signal i _s with the detected current value as the signal value To do.

FIG. 29 shows an example of an internal block diagram of the estimation unit 511. The estimation unit 511 includes a filter 521, a vector component calculation unit 522, and a phase difference calculation unit 523 having the same functions as the filter 11, the vector component calculation unit 12, and the phase difference calculation unit 13 in FIG. However, in the estimation unit 511, the angular frequency of vibration in the mover of the actuator 501 is handled as ω _n , and θ _est from the single-phase voltage command unit 512 is used as θ in the vector component calculation unit 522. Then, the current signal component of the orientation and phase of the induced voltage vectors induced by the permanent magnet of the mover, and outputted as the signal v _q from the vector component calculation unit 522.

The single-phase voltage command unit 512 calculates a single-phase voltage command value v _LOA ^* and a phase θ _est based on the phase difference Δθ calculated by the phase difference calculation unit 523. As a method for calculating the phase θ _est based on the phase difference Δθ, the methods of the other embodiments described above can be used. The inverter 502 generates a single-phase AC voltage using pulse width modulation so that the voltage value of the voltage applied to the armature winding of the actuator 501 matches the voltage command value v _LOA ^*, and the generated AC voltage Is supplied to the actuator 501.

In the actuator 501 which is a linear vibration actuator, an operation using resonance is performed. Assuming that the induced voltage due to the inductance of the armature winding of the actuator 501 is negligibly small compared to the induced voltage (back electromotive force) due to the permanent magnet of the actuator 501, it is applied from the inverter 502 to the actuator 501 at the time of resonance. The phase of the voltage vector coincides with the phase of the vector of the current supplied from the inverter 502 to the actuator 501. Therefore, the resonance operation can be performed by adjusting the frequency of v _LOA ^* so that Δθ becomes zero. That is, the single-phase voltage command unit 512 may adjust the frequency of v _LOA ^* so that Δθ from the estimation unit 511 converges to zero while using proportional-integral control or the like. When the induced voltage due to the inductance is not negligible, considering the difference between the phase of the voltage vector applied from the inverter 502 to the actuator 501 and the phase of the current vector supplied from the inverter 502 to the actuator 501, The frequency of v _LOA ^* may be adjusted so that Δθ becomes a predetermined value.

In each of the embodiments described above, a filter signal (phase lag signal or phase advance signal) having a phase difference of less than π / 2 with respect to the target single phase signal is generated, and from the target single phase signal and filter signal or from the phase lag signal and phase. From the advance signal, a π / 2 phase difference signal (v _β ), which is a derivation source of phase information of the target single-phase signal, and a vector signal component on the rotating coordinate system are obtained. Since a filter signal having a phase difference from the target single-phase signal of less than π / 2 can be generated by a low-order filter, the calculation load is small. In addition, a π / 2 phase difference signal or the like can be generated with high accuracy regardless of the phase difference. Further, it is not necessary to generate a pseudo three-phase signal as in the fourth conventional method.

Further, if the vector signal on the rotating coordinate system is generated without the generation of the π / 2 phase difference signal (v _β ) (that is, in the above formulas (A-6) and (A-7), for example) If v _p and v _q are calculated by the above formula (A-8) or (A-9), the calculation load required for conversion to the rotating coordinate system can be reduced.

By setting the phase difference between the target single-phase signal and the filter signal to π / 4, the calculation load is further reduced. For example, as can be seen from the comparison between the above formula (A-6) and formula (A-11), in order to generate the π / 2 phase difference signal v _β when the phase difference θ ₁ is different from π / 4. Although it is necessary to multiply the target single-phase signal v ₁ and the filter signal v ₂ by different gains, if the phase difference θ ₁ is π / 4, the difference between the target single-phase signal v ₁ and the filter signal v ₂ This is because the π / 2 phase difference signal v _β can be generated simply by taking.

  If the estimation technique for the phase state and the like according to the present invention is applied to a grid interconnection system, a single-phase active filter, or the like, it is expected to reduce a calculation load in a control system of those systems.

<< Deformation, etc. >>
The embodiment of the present invention can be appropriately modified in various ways within the scope of the technical idea shown in the claims. The above embodiment is merely an example of the embodiment of the present invention, and the meaning of the term of the present invention or each constituent element is not limited to that described in the above embodiment. The specific numerical values shown in the above description are merely examples, and as a matter of course, they can be changed to various numerical values. As annotations applicable to the above-described embodiment, notes 1 to 3 are described below. The contents described in each comment can be arbitrarily combined as long as there is no contradiction.

[Note 1]
In each embodiment, a method for deriving all values (v ₂ , v _p, etc.) to be derived is arbitrary. That is, for example, they may be derived by calculation, or may be derived from preset table data.

[Note 2]
Arbitrary portions for performing the arithmetic operation in each embodiment can be formed by software, hardware, or a combination of software and hardware.

[Note 3]
The following points should be noted in this specification and the drawings. The Greek letters (including α and β) expressed as so-called subscripts in the description or drawing of the formula (formula (A-1b) etc.) in the black brackets expressed as the above numbers are those black marks. Outside of the parentheses, it can be written as a standard character that is not a subscript. Such differences between subscripts and standard characters in Greek letters are caused by font conversion performed by the electronic application software, and should be ignored as appropriate when reading this specification. .

DESCRIPTION OF SYMBOLS 1, 31 Phase difference estimation unit 2, 32 Phase estimation unit 3, 33 Phase / frequency / amplitude estimation unit 4, 34 Phase / frequency estimation unit 11 Filter 12 Vector component calculation part 13 Phase difference calculation part 41 _A , 41 _B filter 42 Vector component calculator 43 Phase difference calculator

Claims (9)

A filter unit that generates, as a filter signal, a signal whose phase difference from the target single-phase signal of AC is smaller than π / 2 or an inverted signal of the signal, from the target single-phase signal;
A signal derivation unit for deriving a derivation target using the target single-phase signal and the filter signal,
The derivation target includes a π / 2 phase difference signal having a phase difference of π / 2 with respect to the target single phase signal, or a vector signal component on a rotating coordinate system synchronized with the frequency of the target single phase signal. A single-phase signal input device. The signal derivation unit derives the derivation target based on a trigonometric function using a phase difference between the target single-phase signal and the filter signal and an amplitude ratio between the target single-phase signal and the filter signal. The single-phase signal input device according to claim 1. A filter unit that generates a phase lag signal that is delayed in phase from the target single phase signal and a phase advance signal that is advanced in phase from the target single phase signal, from an alternating target single phase signal;
A signal derivation unit for deriving a derivation target using the phase lag signal and the phase advance signal, and
The derivation target includes a π / 2 phase difference signal having a phase difference of π / 2 with respect to the target single phase signal, or a vector signal component on a rotating coordinate system synchronized with the frequency of the target single phase signal. A single-phase signal input device. The signal derivation unit derives the derivation target based on a trigonometric function using a phase difference between the target single phase signal and the phase lag signal or the phase advance signal. Single-phase signal input device as described. The single-phase signal information estimation part which estimates the phase information of the said object single phase signal using the derivation | leading-out result of the said signal derivation | leading-out part was further provided, The single in any one of Claims 1-4 characterized by the above-mentioned. Phase signal input device. The said single phase signal information estimation part further estimates at least 1 among the frequency and amplitude of the said object single phase signal using the derivation | leading-out result of the said signal derivation | leading-out part, The Claim 6 characterized by the above-mentioned. Single-phase signal input device. A filter unit that generates, as a filter signal, a signal having a phase difference of π / 4 from the target single-phase signal of AC or an inverted signal of the signal;
A single-phase signal input device, comprising: a single-phase signal information estimation unit that estimates phase information of the target single-phase signal using the target single-phase signal and the filter signal. The single-phase signal input device according to claim 7, wherein the single-phase signal information estimation unit further estimates a frequency of the target single-phase signal using the estimated phase information. A power conversion circuit connected to the power system;
In a grid interconnection device comprising a control circuit for controlling the power conversion circuit,
The single-phase signal input device according to any one of claims 1 to 8, wherein the control circuit receives an alternating single-phase signal between the power system and the power conversion circuit as a target single-phase signal, and the single-phase signal. An independent operation detection unit that detects an isolated operation state of the power conversion unit using a signal input device.

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