JP2012049019A - Induction heating cooker - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide an induction heating cooker which can efficiently cook regardless of a material of a heated object.SOLUTION: An induction heating cooker of the present embodiment comprises a DC power supply circuit, a resonance circuit including one induction heating coil and a resonance capacitor, an inverter supplying a high-frequency voltage to the resonance circuit, and control means controlling switching operation of the inverter. The control means performs power control to make heating power coincide with preset heating power by varying a drive frequency of a drive signal in a range that a frequency of a high-frequency current flowing in the resonance circuit becomes the maximum frequency or less and the minimum frequency or more. On performing the power control, the control means performs a thinning control alternately performing a signal change operation for changing the drive signal every period of the drive signal to change an on period to an off period and a normal operation without changing the drive signal when the heating power does not coincide with the preset heating power even if the drive frequency of the drive signal varies to the maximum value.

Description

  Embodiments of the present invention relate to an induction heating cooker.

  The induction heating cooker has advantages such as safety and temperature control function without using fire, and is rapidly spreading as an IH cooking heater incorporated in a system kitchen. By the way, since the cooking pot etc. which are heating objects are made with various materials, it is calculated | required that an induction heating cooking appliance can heat all those materials efficiently. For this reason, in addition to heating a high-permeability pan such as an iron pan, an induction heating cooker has been devised that can also heat a low-permeability pan such as an aluminum pan (for example, Patent Document 1, 2).

  As an example of such an induction heating cooker, two types of induction heating coils having different numbers of turns are provided, and each induction heating coil is switched according to the material of the cooked utensil, and a tap is provided on the induction heating coil. In order to increase the output voltage of the inverter when heating a high-permeability pan, it is possible to switch the number of turns by providing a single induction heating coil with the number of turns suitable for a low-permeability pan. And so on to obtain a desired output.

JP 2006-140088 A Japanese Patent Laid-Open No. 05-251172

  Then, the induction heating cooking appliance which can be heated effectively irrespective of the material of a to-be-heated material is provided.

  According to the induction heating cooker of the present embodiment, it includes a DC power supply circuit, a resonance circuit, an inverter, and a control means. The resonance circuit includes one induction heating coil and a resonance capacitor for induction heating an object to be heated based on set heating power. The inverter includes a plurality of switching elements, converts a DC voltage output from the DC power supply circuit into a high frequency voltage, and supplies the high frequency voltage to the resonance circuit. The control means includes a drive signal generation circuit that generates a plurality of drive signals for driving the plurality of switching elements, respectively, and controls the switching operation of the inverter. The control means performs power control to match the heating power with the set heating power by changing the driving frequency of the driving signal within a range in which the frequency of the high-frequency current flowing in the resonance circuit is equal to or lower than the maximum frequency and equal to or higher than the minimum frequency. . When controlling the power, when the heating power at that time does not match the set heating power even though the drive frequency of the drive signal is changed to the maximum value, the control means turns on the switching element. Thinning control is performed in which a signal change operation for changing the drive signal in units of one cycle of the drive signal and a normal operation in which the drive signal is not changed are alternately executed so that the period becomes an off period in which the switching element is turned off.

Electrical configuration diagram of induction heating cooker showing this embodiment Block diagram showing the configuration of the inverter drive pulse generator Block diagram showing one configuration example of the multiplier circuit The figure which shows the output signal waveform of the waveform generation circuit and the thinning circuit The figure which shows the example of a period setting of normal operation and signal change operation in thinning-out control The figure which shows one structural example of a dead time variable circuit The figure which shows the waveform of each part of the dead time variable circuit Fig. 7 equivalent when the dead time length is changed Flow chart showing contents of power control by input power control unit Diagram showing transistor collector-emitter voltage and gate voltage

Hereinafter, an embodiment of the present invention will be described with reference to the drawings.
FIG. 1 schematically shows the electrical configuration of an induction heating cooker. An induction heating cooker 1 shown in FIG. 1 includes a DC power supply circuit 2, an inverter 3, a resonance circuit 4, a control circuit 5, a gate control circuit 6, and the like. The full-wave rectifier circuit 7 constitutes a DC power supply circuit 2 together with a smoothing capacitor 8, and its AC input terminal is connected to an AC power supply 11 via AC power supply lines L 1 and L 2 and a noise filter 10. Yes. The AC power source 11 is, for example, a single-phase 200V commercial AC power source.

  The positive side DC output terminal of the full-wave rectifier circuit 7 is connected to one terminal of the capacitor 8 via the reactor 12 and the diode 13. The negative DC output terminal of the full-wave rectifier circuit 7 is connected to the other terminal of the capacitor 8. The NPN transistor 14 forms a boost converter 15 together with the reactor 12 and the diode 13. The collector of the transistor 14 is connected to the interconnection point of the reactor 12 and the diode 13, and the emitter is connected to the negative DC output terminal of the full-wave rectifier circuit 7. DC power supply lines 16 and 17 are connected to both terminals of the capacitor 8.

  The inverter 3 converts the DC voltage VDC output from the DC power supply circuit 2 into a high frequency voltage by the switching operation, and supplies it to the resonance circuit 4. The inverter 3 includes a first half bridge circuit (corresponding to an arm) in which the first and second transistors 18 and 19 are connected in series between the DC power supply lines 16 and 17, and the third and fourth transistors 20 and 21. Is a full bridge type inverter circuit composed of a second half bridge circuit (corresponding to an arm) connected in series.

  The first to fourth transistors 18 to 21 correspond to switching elements, and are, for example, IGBTs. In these first to fourth transistors 18 to 21, free wheel diodes 18 a to 21 a are connected between the collector and the emitter, respectively. The resonance circuit 4 is connected between a node N1 which is an interconnection point of the first transistor 18 and the second transistor 19 and a node N2 which is an interconnection point of the third transistor 20 and the fourth transistor 21. It is connected. Further, snubber capacitors 22 and 23 for reducing switching loss are connected between the nodes N1 and N2 and the DC power supply line 17.

  The first and second transistors 18 and 19 operate alternately so that when one is on, the other is off. The third and fourth transistors 20 and 21 also operate alternately so that when one is on, the other is off. The on / off operation of each of the transistors 18 to 21 is controlled by the gate control circuit 6. The gate control circuit 6 includes an inverter drive pulse generator 25 and a driver 26.

  The resonance circuit 4 includes an induction heating coil 27, first and second resonance capacitors 28 and 29, and a relay switch 30. For example, an induction heating coil 27 for heating a heated object 31 that is a pan and a first resonance capacitor 28 are connected in series between nodes N1 and N2. A series circuit of a second resonance capacitor 29 and a relay switch 30 is connected in parallel to the first resonance capacitor 28. The relay switch 30 is provided to switch the resonance frequency by switching the connection state of the resonance capacitors 28 and 29. In the present embodiment, the number of turns of the induction heating coil 27 is set to 60 turns so that the object to be heated 31 made of an aluminum-based material can be induction-heated.

  When the relay switch 30 is off, the induction heating coil 27 and the first resonance capacitor 28 constitute a series resonance circuit. The capacity of the first resonance capacitor 28 is set so that the resonance frequency when heating the aluminum pan (object to be heated) in this state is about 80 kHz. On the other hand, when the relay switch 30 is on, the resonance capacitors 28 and 29 are connected in parallel to each other, and the capacitance and the induction heating coil 27 constitute a series resonance circuit. The parallel capacitances of the resonant capacitors 28 and 29 are set so that the resonant frequency when heating the iron pan (object to be heated) in that state is about 20 kHz. In this case, the capacity of the first resonant capacitor 28 is determined so as to satisfy the condition when the relay switch 30 is in the off state first, and the second condition is satisfied so that the condition when the relay switch 30 is in the on state is satisfied later. The capacity of the resonant capacitor 29 is determined.

The control circuit 5 is constituted by, for example, a 32-bit microcomputer (reference clock = 40 MHz). The control circuit 5 includes an input power control unit 32, a load determination unit 33, and a base clock generation unit 34 as functions realized by the software.
The input voltage detection circuit 35 detects an AC input voltage between the AC power supply lines L 1 and L 2 and gives the detected value to the input power control unit 32. The input current detection circuit 36 detects an AC input current via the current transformer 37 and supplies the detected value to the input power control unit 32 and the load determination unit 33. The inverter current detection circuit 38 detects the current flowing through the inverter 3 via the current transformer 39 and gives the detected value to the load determination unit 33.

  Based on the detection values of the input voltage detection circuit 35 and the input current detection circuit 36, the power factor improvement unit 40 makes the waveform of the AC input current follow the waveform of the AC input voltage and A base signal for driving the transistor 14 is generated so that the output voltage is boosted to a set value. The base signal is given to the base of the transistor 14 through the driver 41. Thus, boost converter 15 applies set DC voltage VDC between DC power supply lines 16 and 17 and acts as a power factor correction chopper. In the present embodiment, the boost converter 15, the power factor improving unit 40, and the driver 41 constitute a power factor improving circuit 71 that performs a power factor improving operation accompanied by a boosting operation. The power factor correction circuit 71 performs a power factor improvement operation during a period when the input power control unit 32 does not execute a thinning control (described later), and does not perform a power factor correction operation during a period when the thinning control is executed. ing.

  The load determination unit 33 (corresponding to the material determination unit) determines the material of the object to be heated 31 from the AC input current detected by the input current detection circuit 36 and the inverter current IQ detected by the inverter current detection circuit 38. . The load determination unit 33 gives a determination signal indicating the determination result of the material to the base clock generation unit 34. The load determination unit 33 turns off the relay switch 30 via the relay switching circuit 42 when determining that the material of the article to be heated 31 is aluminum-based (a material having low resistance and low magnetic permeability). The load determination unit 33 turns on the relay switch 30 via the relay switching circuit 42 when determining that the material of the object to be heated 31 is iron-based (a material having high resistance and high magnetic permeability). .

  The power (heating power) consumed by the induction heating coil 27 is controlled by the input power control unit 32. The input power control unit 32 controls the power consumed by the induction heating coil 27 to match the target power value (set heating power) instructed from the outside. Since the inverter current IQ flowing through the induction heating coil 27 is a high-frequency current, it is not easy to detect the heating power. Therefore, in this embodiment, instead of directly controlling the power consumed by the induction heating coil 27, the power for matching the input power of the DC power supply circuit 2 to the input power target value calculated based on the target power value. Take control. As a result of the power control, the power consumption (heating power) of the induction heating coil 27 is made to coincide with the target power value.

  That is, the input power control unit 32 calculates the input power from the AC input voltage detected by the input voltage detection circuit 35 and the AC input current detected by the input current detection circuit 36. The input power control unit 32 provides the base clock generation unit 34 with a frequency command that instructs the target value of the heating frequency (frequency of the inverter current IQ) so that the calculated input power becomes equal to the input power target value. The target value of the heating frequency is set to a value in the range from the maximum frequency to the minimum frequency. The maximum frequency and the minimum frequency in this embodiment are both values generally used in induction heating cookers, and are 100 kHz and 20 kHz, respectively.

  When the load determining unit 33 determines that the material of the object to be heated 31 is a high-resistance and high-permeability material such as iron or stainless steel, the input power control unit 32 can perform the thinning control. Become. When executing the thinning control, the input power control unit 32 outputs a thinning command Sa to the inverter drive pulse generation unit 25. Although details will be described later, the thinning control is executed when the input power cannot be reduced to the minimum value (100 W in the present embodiment) even though the target value of the heating frequency is set to the maximum frequency. It has become.

  The base clock generation unit 34 includes a timer, for example. The base clock generation unit 34 is given a reference clock (for example, 40 MHz) of the microcomputer. The base clock generator 34 counts this reference clock with a timer, and generates a base clock signal having a frequency corresponding to the count value. That is, the base clock generator 34 generates a base clock signal by dividing the reference clock. The base clock generation unit 34 determines the frequency of the base clock signal based on the frequency command given from the input power control unit 32 and taking into account the determination result of the load determination unit 33. The base clock signal is output from, for example, a general-purpose I / O port of a microcomputer and is supplied to the inverter drive pulse generator 25.

  In the gate control circuit 6, the inverter drive pulse generator 25 uses the base clock signal supplied from the control circuit 5 to drive the drive pulses signals VG 1 to VG 4 (drive signals) for driving the transistors 18 to 21 of the inverter 3. Equivalent). The drive pulse signals VG1 to VG4 are supplied to the gates of the transistors 18 to 21 of the inverter 3 via the driver 26, respectively. In the present embodiment, the control circuit 9 is constituted by the control circuit 5 and the gate control circuit 6.

  FIG. 2 is a block diagram illustrating a configuration of the inverter drive pulse generation unit 25. The inverter drive pulse generation unit 25 includes a multiplication circuit 43, a waveform generation circuit 44, a thinning circuit 72, and a dead time variable circuit 45. The multiplier circuit 43 generates a multiplied clock signal obtained by multiplying the base clock signal supplied from the control circuit 5 (m times, m is an integer). The waveform generation circuit 44 (corresponding to the drive signal generation circuit) is configured by a digital circuit. The waveform generation circuit 44 generates drive pulse signals VG1 'to VG4' based on the multiplied clock signal supplied from the multiplication circuit 43.

  The thinning circuit 72 is configured by a digital circuit. When the thinning command Sa is not given, the thinning circuit 72 does not change the driving pulse signals VG1 ′ to VG4 ′ generated by the waveform generation circuit 44 and does not change the driving pulse signals VG1 ″ to VG4 ″. Output as. Further, when the thinning command Sa is given, the thinning circuit 72 applies predetermined changes (details will be described later) to the driving pulse signals VG1 ′ to VG4 ′, and then drives the driving pulse signals VG1 ″ to Output as VG4 ″. The dead time variable circuit 45 changes the rising timing of the drive pulse signals VG1 ″ to VG4 ″ supplied from the thinning circuit 72, and serves as drive pulse signals VG1 to VG4 for turning on / off the transistors 18 to 21 of the inverter 3. Output.

  FIG. 3 shows a configuration example of the multiplier circuit 43. The multiplication circuit 43 shown in FIG. 3 includes a phase comparator 46, an LPF 47, a VCO 48, an inverter 49, and a frequency divider 50. The phase comparator 46 outputs a voltage signal corresponding to the phase difference between the base clock signal and the output signal of the frequency divider 50. The output of the phase comparator 46 is smoothed through an LPF 47 (low-pass filter) and then supplied to the VCO 48.

  The VCO 48 (voltage controlled oscillator) outputs a signal having a frequency corresponding to the voltage value of a given voltage signal. The output of the VCO 48 is supplied to the frequency divider 50 via the inverter 49 and is output as a multiplied clock signal. The frequency divider 50 divides the multiplied clock signal by 1 / m and outputs it. With such a configuration, the multiplier circuit 43 generates a multiplied clock signal that is synchronized with the base clock signal and has a frequency m times that of the base clock signal.

  FIG. 4 shows an output signal waveform of the waveform generation circuit and an output signal waveform of the thinning circuit when a thinning command is given. Although not shown, the waveform generation circuit 44 includes a counter that counts the multiplied clock signal and a plurality of logic circuits that receive the output signal of the counter. With such a configuration, the waveform generation circuit 44 generates the drive pulse signals VG1 'to VG4' based on the multiplied clock signal. As shown in FIGS. 4A and 4B, the drive pulse signals VG1 'to VG4' have waveforms that turn on / off the transistors 18 to 21 with a duty of 50%.

As shown in FIGS. 4C and 4D, when the thinning-out command Sa is given, the thinning-out circuit 72 performs the normal operation that does not change the driving pulse signals (VG1 ′ to VG4 ′) and the driving pulse signal (VG1). The signal changing operation for changing '˜VG4') is executed alternately. The signal changing operation is an operation of changing the H level period (on period in which the transistors 18 to 21 are turned on) of each drive pulse signal into the L level period (off period in which the transistors 18 to 21 are turned off). . The normal operation and the signal changing operation are executed in units of one cycle of the drive pulse signal. In the present embodiment, both the normal operation and the signal changing operation are executed for a period corresponding to two cycles of the drive pulse signal. That is, the repetition cycle (thinning control cycle) of the normal operation and the signal changing operation corresponds to four cycles of the drive pulse signal. For this reason, the frequency fa of the thinning control is expressed by the following equation (1) when the drive frequency of the drive pulse signal is fb.
fa = (1/4) · fb (1)
When such thinning-out control is executed, the on-period of each drive pulse signal is halved compared to the case where the decimating control is not executed, so that the heating power is also halved.

  In the normal operation period Ta and the signal change operation period Tb, the frequency fa of the thinning control when the drive frequency fb is the maximum value (the value when the frequency of the inverter current IQ becomes the maximum frequency (100 kHz)) is Any change can be made as long as it is within the range of the minimum frequency (20 kHz) of the current IQ.

  Specifically, as shown in FIG. 5, the period Ta of the normal operation and the period Tb of the signal changing operation may be set so that the cycle of the thinning control is within 5 cycles of the drive pulse signal. That is, when the period Ta is one cycle of the drive pulse signal, the period Tb can be set to any one of 1 to 4 cycles of the drive pulse signal. When the period Ta is set to two cycles of the drive pulse signal, the period Tb can be set to any one of 1 to 3 cycles of the drive pulse signal. Further, when the period Ta is set to 3 cycles of the drive pulse signal, the period Tb can be set to 1 cycle or 2 cycles of the drive pulse signal. Further, when the period Ta is set to four cycles of the drive pulse signal, the period Tb can be set to one cycle of the drive pulse signal. Either one of the periods Ta and T2 cannot be set to 5 cycles or more of the drive pulse signal. This is because the condition of the above-described expression (1) cannot be satisfied.

  The normal operation period Ta in which the transistors 18 to 21 are turned on must satisfy the conditions shown in FIG. However, when the high-frequency current does not become a resonance waveform within the period Ta, the resonance state is determined by free resonance in the period Tb of the signal changing operation. In that case, in the normal operation performed after the signal changing operation, there is a possibility of being turned on at an abnormal timing before the resonance state ends, so the period Tb is a period sufficient for the free resonance to converge. It is good to set. In consideration of this, in this embodiment, both the period Ta and the period Tb are set to a period corresponding to two cycles of the drive pulse signal.

  FIG. 6 shows an example of the configuration of the dead time variable circuit 45, and FIGS. 7 and 8 show signal waveforms of respective parts of the dead time variable circuit 45. The dead time variable circuit 45 shown in FIG. 6 changes so as to delay the rising timing of the input drive pulse signal VG1 ″ and outputs it as the drive pulse signal VG1. Although not shown, the same configuration as this circuit is provided corresponding to the drive pulse signals VG2 ″ to VG4 ″.

  The dead time variable circuit 45 shown in FIG. 6 includes resistors 59 and 60, capacitors 61 and 62, a diode 63, Schmitt trigger type inverters 64 and 65, and a transistor 66. The input terminal of the drive pulse signal VG 1 ″ is connected to the ground line 67 through the resistor 59 and the capacitor 61. A series circuit of a diode 63 and a resistor 60 is connected in parallel with the resistor 59. The diode 63 is provided such that the cathode is on the input terminal side. A capacitor 62 and a transistor 66 are connected in series between a node N3, which is an interconnection point between the resistor 59 and the capacitor 61, and the ground line 67.

  The transistor 66 includes a base-emitter resistance and a base resistance, and a switching signal output from the control circuit 5 is supplied to the base. The control circuit 5 switches the switching signal to an H level (a level at which the transistor 66 can be sufficiently turned on) or an L level (a level at which the transistor 66 can be sufficiently turned off). The signal at node N3 is output as drive pulse signal VG1 through inverters 64 and 65.

  Since the transistor 66 is turned off when the switching signal is at the L level, only the capacitor 61 is connected between the node N3 and the ground line 67. On the other hand, since the transistor 66 is turned on when the switching signal is at the H level, the capacitors 61 and 62 are connected in parallel between the node N3 and the ground line 67. In the present embodiment, the capacity of the capacitor 62 is set to about twice the capacity of the capacitor 61. Further, the resistance value of the resistor 59 is set to about 10 times the resistance value of the resistor 60.

  With such a configuration, when the switching signal is at the L level, the capacitor 61 is connected via the resistor 59 when the drive pulse signal VG1 ″ (see FIG. 7A) rises (time t0). Since the battery is charged, the rising edge of the signal at the node N3 becomes gradual (see FIG. 7B). On the other hand, when the drive pulse signal VG1 ″ falls (time t1), the charge of the capacitor 61 is discharged through the resistor 60 and the diode 63 having a resistance value sufficiently smaller than that of the resistor 59. The signal falls steeply (see FIG. 7B). By passing the slowly rising signal generated in this way through the inverters 64 and 65, the driving pulse signal VG1 (see FIG. 7C) with the rising timing delayed by the time T1 is generated.

  When the switching signal is at the H level, the capacitors 61 and 62 are charged via the resistor 59 when the drive pulse signal VG1 ″ (see FIG. 8A) rises (time t0). The rising edge of the signal at the node N3 is very gradual (see FIG. 8B). On the other hand, when the drive pulse signal VG1 ″ falls (time t1), the fall of the signal at the node N3 becomes steep as in the case where the switching signal is at the L level (FIG. 8). (See (b)). The drive pulse signal VG1 in which the rising timing is delayed by a time T2 longer than the time T1 by passing the very slowly rising signal thus generated through the inverters 64 and 65 (see FIG. 8C). Is generated.

  In this way, by delaying the rising timing of the drive pulse signals VG1 to VG4, the dead time when the first and second transistors 18 and 19 are both turned off, and the third and fourth transistors 20 and 21 are In any case, a dead time for turning off is provided. The length of the dead time can be switched in two steps according to the level of the switching signal given from the control circuit 5. The control circuit 5 outputs an L level switching signal when the input power is greater than or equal to a predetermined value, and outputs an H level switching signal when the input power is less than the predetermined value. Therefore, the length of the dead time is shortened if the input power is equal to or greater than a predetermined value, and is increased if the input power is less than the predetermined value.

Next, the content of the power control in the induction heating cooker 1 having the above configuration will be described.
When a signal (not shown) indicating the start of heating is input from the outside, the load determination unit 33 determines the type (material) of the object to be heated 31. In making this determination, the load determination unit 33 turns on the relay switch 30. The input power control unit 32 instructs the base clock generation unit 34 to set, for example, 25 kHz as a target value for the heating frequency of the inverter 3. In this state, the detection value of the inverter current detection circuit 38 when the inverter 3 is driven is read. When the material of the object to be heated 31 is aluminum, the inductance of the induction heating coil 27 becomes a small value because of the low magnetic permeability. For this reason, the resonance frequency of the resonance circuit 4 is a value greatly deviated from 25 kHz (for example, 50 kHz). In this case, only a small current flows through the induction heating coil 27.

  On the other hand, when the material of the article to be heated 31 is iron-based, the resonance frequency is about 20 kHz. Since the heating frequency 25 kHz is very close to the value, a larger current flows through the induction heating coil 27 than in the case of an aluminum-based material. Therefore, it can be determined from the magnitude of the current flowing through the induction heating coil 27 whether the material of the article to be heated 31 is iron-based or aluminum-based. In addition, instead of the magnitude of the current flowing through the induction heating coil 27, it can be determined from the magnitude of the input power obtained by the input power control unit 32. In this case, the value of the input power is small in the aluminum system and larger in the iron material.

  After determining the material of the object to be heated 31, the relay switch 30 of the resonance circuit 4 is switched according to the material. When the material is aluminum, the relay switch 30 is turned off to set the resonance frequency to about 80 kHz. If the material is iron, the relay switch 30 is turned on to set the resonance frequency to about 20 kHz.

  The input power control unit 32 gives the target value of the heating frequency to the base clock generation unit 34 so that the calculated input power becomes equal to the input power target value calculated based on the set heating power set by the user. The base clock generation unit 34 outputs a base clock signal having a frequency corresponding to the instructed heating frequency to the inverter drive pulse generation unit 25. The inverter drive pulse generator 25 generates drive pulse signals VG1 to VG4 for turning on / off the transistors 18 to 21 so that a high-frequency current that matches the instructed heating frequency flows through the induction heating coil 27. These drive pulse signals VG1 to VG4 are applied to the gates of the transistors 18 to 21 through the driver 26, respectively.

  When the transistors 18 to 21 are driven by the drive pulse signals VG1 to VG4, a high-frequency current flows through the induction heating coil 27 and the object to be heated 31 is induction heated. As described above, the drive pulse signals VG1 to VG4 of this embodiment have waveforms that turn on / off the transistors 18 to 21 with a 50% duty. For this reason, the frequency of the drive pulse signals VG1 to VG4 (drive frequency) and the frequency of the high-frequency current flowing through the induction heating coil 27 (resonance circuit 4) are the same. That is, in this embodiment, a driving method of the inverter 3 is adopted in which the driving frequency is the same as the frequency of the high-frequency current flowing in the resonance circuit 4.

  Power control (heating power control) for matching the input power to the input power target value is performed by varying the switching frequency of the inverter 3 (drive frequency of the drive pulse signal). In general, since the series resonance circuit in the induction heating cooker is used as inductive, the drive frequency does not fall below the resonance frequency. In the induction heating cooker, the drive frequency of the inverter 3 is increased (changed in a direction away from the resonance frequency) to decrease the input power and the drive frequency is decreased (changed in a direction approaching the resonance frequency). As a result, the input power is increased.

  Now, when using one (one kind) induction heating coil 27 like the induction heating cooker 1 of this embodiment, the induction heating coil 27 is usually made of a low permeability material such as an aluminum pan. It is designed to have a combined number of turns (about twice the number of turns according to the iron-based material). Then, in order to obtain a desired maximum power (for example, 3 kW) when heating a material having a high magnetic permeability such as an iron pan, the inverter voltage is set high. By doing in this way, regardless of the material of the to-be-heated material 31, heating cooking can be performed with a desired maximum heating power. However, in such a configuration, since the inverter voltage is set high, when the object to be heated 31 is an iron pan or the like, the minimum value of the heating power is reduced to a certain value (for example, 200 W to 150 W). It can only be lowered. On the other hand, for example, there is a demand to reduce the minimum value to a smaller value for convenience of control specifications. Therefore, in the present embodiment, by enabling the following thinning control, the minimum heating power is reduced when the iron-based material to be heated 31 is heated.

  That is, when performing such power control, the input power control unit 32 can perform the thinning control when it is determined that the material of the article to be heated 31 is iron based on the load determination result. . FIG. 9 is a flowchart showing the contents of power control when it is determined that the material of the article to be heated 31 is iron-based. In the first step S1, set power (input power target value) is set. First, in this step S1, a case will be described in which the set power is set to such a value that power control can be performed without performing thinning control.

  In step S2, the input power at that time is obtained by calculation. In step S3, it is determined whether or not the obtained input power is smaller than the set power. When it is determined that the input power is smaller than the set power (“YES” in step S3), the process proceeds to step S4. In step S4, the drive frequency is lowered by a predetermined value in order to increase the input power. Then, the process proceeds to step S5, and it is determined whether or not the thinning control is being executed. In this case, since the thinning control is not executed, it is determined as “NO”, and the process proceeds to Step S6. In step S6, it is determined whether or not the drive frequency is less than the minimum value fmin. The minimum value fmin is a driving frequency when the frequency of the high-frequency current flowing through the resonance circuit 4 is the minimum frequency (20 kHz), and is 20 kHz in the present embodiment. If the drive frequency is 20 kHz or higher (NO), the process returns to step S2. And when a drive frequency is less than 20 kHz (YES), it returns to step S2 after performing step S7. In step S7, the drive frequency is set to the minimum value fmin.

  On the other hand, if it is determined that the input power is greater than the set power (“NO” in step S3), the process proceeds to step S8. In step S8, the drive frequency is increased by a predetermined value in order to reduce the input power. In step S9, it is determined whether the drive frequency is greater than the maximum value fmax. The maximum value fmax is a driving frequency when the frequency of the high-frequency current flowing through the resonance circuit 4 is the maximum frequency (100 kHz), and is 100 kHz in this embodiment. In this case, since the drive frequency is 100 kHz or less (NO), the process directly returns to step S2. Normally, the above steps are repeated so that the input power is controlled to match the set power. In the present embodiment, it is assumed that the input power is 150 W when the driving frequency is set to 100 kHz without executing the thinning control.

  Next, a case where a low power value (for example, 100 W) that cannot be reached only by the normal power control is set as the set power in step S1 will be described. In this case, the drive frequency is increased to 100 kHz by executing the above-described normal control. However, the input power value at that time has not yet decreased to the target power value of 100 W, but is 150 W. Therefore, “NO” is determined in step S3, and the process proceeds to step S8. In step S8, the drive frequency is further increased by a predetermined value from 100 kHz. Therefore, in step S9, “YES” is determined, and the process proceeds to step S10.

  In step S10, the drive frequency is set to the maximum value fmax (100 kHz). And it progresses to step S11 and execution of thinning-out control is started (thinning-out control setting). Thus, the process returns to step S2 in a state where the input power is reduced to 75W, which is 1/2 of 150W. That is, since the input power becomes lower than the target power value, “YES” is determined in the subsequent step S3, and the process proceeds to step S4. After the drive frequency is lowered by a predetermined value in step S4, the process proceeds to step S5. In this case, since the thinning control is being executed, “YES” is determined in the step S5, and the process proceeds to the step S12.

In step S12, it is determined whether or not the drive frequency is smaller than the thinning-out completion determination value fth. The thinning end determination value fth is determined as follows. That is, during execution of thinning control, the high-frequency current flowing through the resonance circuit 4 has a frequency component for thinning control. For this reason, when the frequency fa of the thinning control becomes smaller than 20 kHz (= the minimum value fmin of the driving frequency fb) which is the minimum frequency of the high frequency current, the frequency component contained in the high frequency current becomes an audible frequency, and abnormal noise is generated from the pan. In this case, the user may feel uncomfortable. Therefore, in the present embodiment, during the execution of the thinning control, the thinning control is canceled when the frequency fa of the thinning control becomes less than the minimum value fmin of the driving frequency fb (minimum frequency of the high-frequency current = 20 kHz). ing. The thinning end determination value fth is expressed by the following equation (2).
fth = fmin · (fb / fa) (2)

  According to the above equation (2), the decimation end determination value fth in this embodiment is 80 kHz. If the drive frequency is equal to or higher than the thinning end determination value fth, “YES” is determined in the step S12, and the process returns to the step S2. On the other hand, if the drive frequency is less than the thinning end determination value fth, “NO” is determined in the step S12, and the process proceeds to a step S13. In step S13, the drive frequency is set to the maximum value fmax. And it progresses to step S14 and execution of thinning-out control is stopped (thinning-out control cancellation | release). As a result, the process returns to step S2 in a state where the input power is 150 W, which is the lowest value when the thinning control is not executed.

  As described above, when the thinning control is executed, the power control is executed so that the input power matches the set power in the state in which the driving frequency is equal to or greater than the thinning end determination value fth while the thinning control is executed. The When the drive frequency is changed to the thinning end determination value fth but the input power at that time does not match the set power, the execution of the thinning control is canceled, and then the normal power control is performed. Will be executed.

  As described above, the induction heating cooker 1 of the present embodiment includes the input power control unit 32 that enables execution of thinning control when the object to be heated 31 is an iron-based material. For this reason, in the configuration using one (one type) induction heating coil 27 having the number of turns according to the aluminum-based material, when heating a high-resistance and high-permeability material such as an iron-based pan. While making it possible to obtain a desired maximum power (for example, 3 kW), it is further possible to reduce the minimum value of the heating power to a value (for example, about 100 W) smaller than the conventional value. Therefore, the induction heating cooker 1 of this embodiment can be heated effectively regardless of the material of the article to be heated 31.

  As another method for lowering the minimum value of the heating power, a method of reducing the heating power as the integrated power by repeatedly turning on / off the power in a long cycle (second unit) is conceivable. However, this method has the following problems. That is, since ON / OFF is repeated with a relatively long cycle, input power at the time of ON becomes higher than in the case of continuous heating, and bumping may occur. Due to the occurrence of bumping, for example, when cooking in stew, there is a risk of scorching on the bottom of the pan. Furthermore, when the pan is removed, there is a problem that the pan detection time becomes long because the state cannot be detected during the period when the power is turned off. On the other hand, according to the induction heating cooker 1 of the present embodiment, when the heating power is reduced to the minimum value, the above-described thinning control is executed, so that the cycle of the drive signal (the high-frequency current flowing through the resonance circuit 4) is increased. Power supply / stop (on / off) is repeated in units of several tens of microseconds. For this reason, the heating state which is hardly different from the state of continuous heating can be maintained. Therefore, according to the present embodiment, the occurrence of bumping is suppressed, and when the pan is removed, the state can be detected immediately.

  During execution of the thinning control, the execution of the thinning control is canceled when the frequency of the thinning control becomes the minimum value fmin of the drive frequency. Thereby, the frequency component of the thinning control included in the high-frequency current flowing in the resonance circuit 4 during the thinning control is not smaller than 20 kHz (minimum frequency). For this reason, the high-frequency current flowing through the resonance circuit 4 with the execution of the thinning-out control does not include an audible frequency component, and the generation of abnormal noise from the pan as the object to be heated 31 is suppressed.

  The power factor correction circuit 71 performs a power factor correction operation including a boosting operation for boosting the DC voltage supplied to the inverter 3 to a predetermined value. However, the power factor improvement circuit 71 performs the power factor improvement operation during a period when the input power control unit 32 does not execute the thinning control, and does not perform the power factor correction operation during the period when the thinning control is executed. The period in which the thinning control is executed is such that power control is performed so that the input power matches the relatively small set power. When the boosting operation is not performed and the DC voltage supplied to the inverter 3 is lowered, the input power is also lowered accordingly. Therefore, according to such a configuration, the minimum value of the input power can be set to a lower value. The power factor improvement circuit 71 may perform a power factor improvement operation regardless of whether or not the input power control unit 32 executes the thinning control.

  Both the normal operation and the signal changing operation performed by the thinning circuit 72 in the thinning control are performed for a period corresponding to two cycles of the drive pulse signal. As a result, after the normal operation in which each of the transistors 18 to 21 is driven ON, until the next normal operation is performed, at least during a period corresponding to two cycles of the drive pulse signal in which the signal changing operation is performed, There is a period during which the transistors 18 to 21 are driven off. By providing such an off period, the next time the transistors 18 to 21 are turned on, the resonance state is not converged, and a situation occurs in which the transistors 18 to 21 are turned on (switched) with an abnormal voltage. Can be prevented. The off period can be changed as appropriate as long as it is a period necessary for the free resonance by the resonance circuit 4 to converge.

  A dead time variable circuit 45 is provided, and the rising timing of the drive pulse signals VG1 to VG4 can be changed in two stages. That is, the length of the dead time of each arm of the inverter 3 can be changed in two stages. The high-frequency current flowing through the resonance circuit 4 changes according to the input power, but the slope of the collector-emitter voltage of the transistors 18 to 21 changes depending on the magnitude of the high-frequency current. Specifically, as shown in FIG. 10, as the input power is smaller, the slope of the fall of the collector-emitter voltage of the transistors 18 to 21 becomes gentler. This is because the time for which the snubber capacitors 22 and 23 are charged increases when the resonance current of the resonance circuit 4 decreases. Therefore, when the resonance current is small, that is, when the input power is small, the dead time is lengthened, and when the resonance current is large, that is, when the input power is large, the dead time is shortened to prevent an arm short circuit. At the same time, the switching loss due to the snubber voltage can be reduced.

  Drive pulse signals VG1 'to VG4' generated based on, for example, a base clock signal output from a general-purpose I / O port of a microcomputer are given to the thinning circuit 72 included in the inverter drive pulse generation unit 25. The thinning circuit 72 performs a thinning operation for changing the ON period to the OFF period for the drive pulse signals VG1 'to VG4'. The thinning circuit 72 is configured as a digital circuit (logic circuit). According to such a configuration, the thinning circuit 72 can easily perform the signal changing operation in units of one cycle of the drive pulse signal.

  The waveform generation circuit 44 is configured as a digital circuit, like the thinning circuit 72. As a result, these circuits can be changed by a circuit including a programmable logic circuit such as an FPGA (Field Programmable Gate Array), a PLD (Programmable Logic Device), and an ASIC (Application Specific Integrated Circuit). Thus, the circuit configuration can be simplified, and the effects of reducing the board area and the number of parts can be obtained. In addition, it is not necessary to newly increase the number of parts in order to enable thinning control. Further, since the circuit configuration excluding a part of the configuration of the multiplier circuit 43 (LPF 47, capacitor constituting the VCO 48, resistor, etc.) is also a digital circuit, if the circuit configuration is incorporated in an ASIC or the like, the above-described effects are further enhanced. It will be.

  The input power control unit 32 performs thinning control when the load determination unit 33 determines that the material of the object to be heated 31 is a high-resistance and high-permeability iron-based material (including stainless steel). Made possible. In the case of a non-magnetic material such as aluminum, the value of Q (a value representing the sharpness of the resonance peak) is high. Therefore, since the change in input power accompanying the change in drive frequency becomes large, it is easy to set the input power to a relatively low value within the changeable range of drive frequency. On the other hand, in the case of a magnetic material such as iron, the value of Q is small. Therefore, since the input power change accompanying the change of the drive frequency is small, it is difficult to set the input power to a relatively low value within the changeable range of the drive frequency. In the present embodiment, the input power can be reduced to a relatively low value by executing the thinning control when heating the object 31 made of an iron-based material whose input power is difficult to set to a low value. ing. Note that, even when the load determining unit 33 determines that the material of the object to be heated 31 is an aluminum material, the thinning control may be performed.

(Other embodiments)
Although one embodiment of the present invention has been described above, this embodiment is presented as an example and is not intended to limit the scope of the invention. The novel embodiment can be implemented in various other forms, and various omissions, replacements, and changes can be made without departing from the scope of the invention.

  The drive pulse signals VG1 to VG4 are not limited to the waveforms shown in the above embodiment. That is, the driving method of the inverter 3 is not limited to the method in which the driving frequency shown in the above embodiment is the same as the frequency of the high-frequency current (resonant current) flowing through the resonance circuit 4. For example, a system that can suffice for the drive frequency to be ½ of the frequency of the resonance current, or a system that can suffice for the drive frequency to be ¼ of the frequency of the resonance current. Furthermore, it is good also as a structure which can switch these systems.

  However, when adopting a method in which the drive frequency as described above is 1 / N times the resonance frequency (N is an integer), the following changes are necessary for the thinning control. That is, during the normal operation period Ta and the signal change operation period Tb, the frequency fa of the thinning control when the drive frequency fb is the maximum value (the value when the frequency of the resonance current becomes the maximum frequency (100 kHz)) What is necessary is just to set to the range used as the frequency which becomes 1 / N times or more the minimum frequency (20kHz) of resonance current. Further, during execution of the thinning control, the thinning control may be canceled when the frequency fa of the thinning control becomes less than 1 / N times the minimum frequency (= 20 kHz) of the high frequency current.

The dead time variable circuit 45 only needs to change the rising timing of the drive pulse signals VG1 ″ to VG4 ″ and output the drive pulse signals VG1 to VG4, and is not limited to the circuit configuration of FIG. Further, the dead time variable circuit 45 may be capable of switching the dead time in three or more stages, or may be one having no dead time switching function. Furthermore, the dead time variable circuit 45 may be provided as necessary.
The waveform generation circuit 44 and the thinning-out circuit 72 may not be configured by circuits including programmable logic circuits such as FPGA, PLD, and ASIC, but may be digital circuits configured individually.

  These embodiments and modifications thereof are included in the scope and gist of the invention, and are included in the invention described in the claims and the equivalents thereof.

  In the drawings, 1 is an induction heating cooker, 2 is a DC power supply circuit, 3 is an inverter, 4 is a resonance circuit, 9 is a control means, 18 to 21 are first to fourth transistors (switching elements), and 27 is induction heating. Coil, 28 and 29 are resonance capacitors, 31 is an object to be heated, 33 is a load determination unit (material determination means), 44 is a waveform generation circuit (drive signal generation circuit), and 71 is a power factor improvement circuit.

Claims (8)

A DC power supply circuit;
A resonance circuit including one induction heating coil and a resonance capacitor for induction heating the object to be heated based on a set heating power;
An inverter configured to include a plurality of switching elements, and converts the DC voltage output from the DC power supply circuit into a high-frequency voltage and supplies the high-frequency voltage to the resonance circuit;
A control means configured to include a drive signal generation circuit that generates a drive signal for driving each of the plurality of switching elements, and controls a switching operation of the inverter;
With
The control means includes
By changing the drive frequency of the drive signal within a range where the frequency of the high-frequency current flowing in the resonant circuit is equal to or lower than the maximum frequency and equal to or higher than the minimum frequency, power control is performed to match the heating power with the set heating power
When performing the power control, even when the driving frequency of the driving signal is changed to the maximum value, the heating power at that time does not match the set heating power,
A signal changing operation for changing the drive signal in units of one cycle of the drive signal so that an ON period for driving the switching element to be an OFF period for driving the switching element OFF, and a normal operation for not changing the drive signal An induction heating cooker that executes thinning control for alternately executing the above. The drive frequency of the drive signal generated by the drive signal generation circuit is 1 / N times the frequency of the high-frequency current flowing in the resonance circuit (where N is an integer),
The control means performs the signal change operation and the normal operation so that the frequency of the thinning control when the drive frequency is the maximum value is equal to or higher than a frequency obtained by multiplying the minimum frequency of the high-frequency current by 1 / N. The induction heating cooker according to claim 1, wherein a period is set. The DC power supply circuit is
It outputs a DC voltage that is obtained by rectifying and smoothing an AC voltage supplied from a commercial AC power source.
Equipped with a power factor correction circuit capable of performing power factor correction operation with boosting operation,
The induction heating cooker according to claim 1 or 2, wherein the power factor correction circuit does not execute the power factor correction operation during a period in which the control unit executes the thinning control. The drive frequency of the drive signal generated by the drive signal generation circuit is 1 / N times the frequency of the high-frequency current flowing in the resonance circuit (where N is an integer),
In the state where the thinning control is executed, the control means changes the driving frequency of the driving signal until the frequency of the thinning control reaches a frequency obtained by multiplying the minimum frequency of the high-frequency current by 1 / N. The induction heating cooker according to any one of claims 1 to 3, wherein when the heating power at that time does not coincide with the set heating power, the execution of the thinning control is canceled.   2. The control unit sets the period of the signal change operation and the period of the normal operation so that the off period necessary for the free resonance by the resonance circuit to converge is provided. The induction heating cooking appliance as described in any one of -4. The inverter includes an arm having two switching elements connected in series,
The said control means changes the said drive signal so that the dead time when two switching elements of the said arm turn off simultaneously becomes long, so that input power is small. The induction heating cooker as described in one.   The induction heating cooker according to any one of claims 1 to 6, wherein the control means is configured by a circuit including a logic circuit whose configuration can be changed by a program. Comprising a material judging means for judging the material of the object to be heated;
The said control means enables execution of the said thinning | decimation control, when it determines with the material of a to-be-heated material having a high resistance and a high magnetic permeability by the said material determination means. The induction heating cooking appliance as described in any one of claim | item 1 -7.

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