JP2004014218A

JP2004014218A - Induction heating device

Info

Publication number: JP2004014218A
Application number: JP2002164136A
Authority: JP
Inventors: Hiroyuki Shoji; 庄司　浩幸; Ryuji Iyotani; 伊予谷　隆二; Tamahiko Kanouda; 叶田　玲彦; Masayuki Isogai; 磯貝　雅之; Junpei Uruno; 宇留野　純平; Yasuhiko Kono; 河野　恭彦; Hideaki Sugawara; 菅原　秀昭
Original assignee: Hitachi Ltd; Hitachi Home Tech Ltd
Current assignee: Hitachi Ltd; Hitachi Appliances Inc
Priority date: 2002-06-05
Filing date: 2002-06-05
Publication date: 2004-01-15
Anticipated expiration: 2022-06-05
Also published as: JP3831298B2

Abstract

<P>PROBLEM TO BE SOLVED: To provide an induction heating device preventing the generation of interference sound when driving a plurality of inverters at one time. <P>SOLUTION: The induction heating device comprises the inverter supplying high frequency electric power from a power source to a circuit including an object to be heated, a heating coil, and a resonant capacitor. The inverter comprises a switching circuit converting direct current voltage supplied from the power source into an alternating current voltage, and a resonance circuit arranged between the output terminal of the switching circuit and at least one electrode of the power source. A bypass circuit connected in parallel with the resonance circuit is formed. <P>COPYRIGHT: (C)2004,JPO

Description

【０００１】
【発明の属する技術分野】
本発明は、複数の加熱部を有する電磁誘導加熱装置に関するものである。
【０００２】
【従来の技術】
近年、火を使わずに鍋などの被加熱物を加熱するインバータ方式の電磁誘導加熱装置が広く用いられるようになってきている。電磁誘導加熱装置は、加熱コイルに高周波電流を流し、コイルに近接して配置された鉄やステンレスなどの材質で作られた被加熱物に渦電流を発生させ、被加熱物自体の電気抵抗により発熱させる。被加熱物の温度制御が可能で安全性が高いことから、新しい熱源として認知されている。
【０００３】
従来、システムキッチン等に組み込まれる電気調理器には、シーズヒータやプレートヒータ、ハロゲンヒータ等の抵抗体を熱源としたものが使われていたが、近年では、一部を誘導加熱調理器に置き換えたもの、あるいは２口以上を誘導加熱調理器にしたものに代わりつつある。電磁誘導加熱装置の入力電力を変化させ被加熱物の温度制御を行う方法としては、インバータの駆動周波数を変化させる方法が一般的である。しかし、加熱コイルを複数個備えてそれぞれ別々の被加熱物を加熱する場合、インバータ間の差分周波数に起因して被加熱物から干渉音が発生するという問題がある。
【０００４】
このような問題を解決する従来例として、特許第２５３２５６５号公報に開示されるような誘導加熱調理器がある。この調理器は、一定の駆動周波数でスイッチング素子の導通期間を変化させ入力電力を制御するものである。
【０００５】
【発明が解決しようとする課題】
上記の従来例では、複数個のインバータを作動しても同一の駆動周波数で入力電力を変えることができるため、干渉音の発生を防ぐことができる。しかしながら、低電力で駆動する場合には、一方のスイッチング素子に発生する損失が大きくなる問題点がある。
【０００６】
本発明の課題は、スイッチング素子の損失発生を抑え、複数のインバータを同時に駆動した場合に干渉音の発生を防止して各々の入力電力を制御する電磁誘導加熱装置を提供することである。
【０００７】
【課題を解決するための手段】
本発明の電磁誘導加熱装置は、直流電圧を高周波の交流電圧に変換するインバータ回路を有し、該インバータ回路はスイッチング回路と共振回路を含み、共振回路は加熱コイルと該加熱コイルに直列接続される共振コンデンサを含み、前記直流電源の両端子（ｐ／ｏ）のいずれか一方とスイッチング回路の出力端子（ｔ）との間に前記共振回路を接続し、前記共振コンデンサに並列に接続されるバイパス回路を有することを特徴とする
【０００８】
【発明の実施の形態】
（実施例１）
図１は本発明の第１の実施形態である電磁誘導加熱装置のブロック図である。図１において、インバータ１００はスイッチング回路２０と加熱コイル５、共振コンデンサ６から成る共振回路６０及びバイパス回路３０によって構成されている。
【０００９】
スイッチング回路２０は電源１０の正電極ｐ点と負電極ｏ点との間に接続されており、電源１０から供給される電圧を高周波電圧に変換して共振回路６０に印加し、加熱コイル５に高周波電力を供給する。バイパス回路３０は共振コンデンサ６に並列に接続されており、共振コンデンサ６に流れる電流をバイパスすることによって、共振回路６０の負荷特性を制御する。一般に共振型のインバータでは、共振負荷の特性が誘導性になるように駆動周波数ｆｓを共振周波数ｆｒよりも高くなるように設定し、インバータの出力電圧に対し共振回路に流れる電流が遅れ位相になるように制御する。図１においては、スイッチング回路２０と共振回路６０の接続点ｔ点の電圧に対して共振回路６０に流れる電流ＩＬ５が遅れ位相になるように制御することになる。
【００１０】
しかしながら、駆動周波数ｆｓを一定にしてスイッチング素子の導通期間を変化させ電力制御を行うと、スイッチング回路２０の導通期間に電流の極性が反転し出力電圧より電流が進み位相になる進相モードへ移行する場合が生じる。これは、スイッチング回路２０の損失増加を招くため、共振型のインバータでは避けなければならないモードである。本発明においては、被加熱物の材質や状態及び設定された入力電力の大きさに応じてバイパス回路の導通状態を変えて、共振コンデンサーの電圧を制御することにより、共振回路６０の負荷特性を誘導性に維持することができる。即ち、共振回路６０の共振周波数をスイッチング回路２０の駆動周波数より常に低くすることができる。
【００１１】
これにより、インバータ１００は駆動周波数ｆｓを一定にしてスイッチング回路２０の導通期間を変化させ、入力電力を制御することが可能となる。このように、バイパス回路３０は、一定の駆動周波数ｆｓで動作を実現するための補助スイッチング回路としての役割を果たす。スイッチング回路２０及びバイパス回路３０はそれぞれドライブ回路６１、６２によって駆動され、ドライブ回路６１、６２は制御回路７０によってコントロールされる。入力電力設定部７５は、使用者が入力電力を設定するためのインターフェースであり、設定された出力に応じて制御回路７０に信号を送る。制御回路７０は入力電力設定部７５からの信号に応じてスイッチング回路２０及びバイパス回路３０を制御する。
【００１２】
（実施例２）
図２は本発明の第２の実施形態である電磁誘導加熱装置の回路構成図である。図２において、電源１０は商用電源１の交流電圧を整流する整流回路２とインダクタ３及びコンデンサ４で構成された平滑回路からなり、交流電圧を直流電圧に変換してインバータ１００に電力を供給する。コンデンサ４の正電極ｐ点と負電極ｏ点との間にはパワー半導体スイッチング素子であるＩＧＢＴ１１とＩＧＢＴ１２が直列に接続されスイッチング回路２０を構成している。ＩＧＢＴ１１，１２にはそれぞれダイオード２１，２２が逆方向に並列接続されている。
【００１３】
また、ＩＧＢＴ１１，１２にはそれぞれ並列にスナバコンデンサ３１，３２が接続されている。スナバコンデンサ３１、３２は、ＩＧＢＴ１１又はＩＧＢＴ１２のターンオフ時の遮断電流によって充電あるいは放電される。スナバコンデンサ３１，３２の容量は、ＩＧＢＴ１１，１２のコレクタとエミッタ間の出力容量より十分に大きいため、ターンオフ時に両ＩＧＢＴに印加される電圧の変化は低減され、ターンオフ損失は抑制される。ＩＧＢＴ１１，１２の接続点、即ち出力端子をｔ点とすると、ｔ点とｏ点間には加熱コイル５と共振コンデンサ６で構成された共振回路６０が接続されている。
【００１４】
共振コンデンサ６には並列にバイパス回路３０が接続されており、バイパス回路３０は直列接続されたＩＧＢＴ１３とＩＧＢＴ１３の逆電流を阻止するダイオード２５、及びＩＧＢＴ１３に逆方向に並列接続されたダイオード２３によって構成されている。ここで、ｔ点から共振コンデンサ６に向かって流れる共振電流ＩＬ５を正とすると、バイパス回路３０に流れるバイパス電流Ｉｃ３は負の一方向である。本実施例では、入力電力を下げる場合、上アームの導通期間を絞り下アームの導通期間を延ばすため、下アームの導通期間内に共振電流ＩＬ５の極性が負から正に反転し易くなる。従って、負の方向に共振電流ＩＬ５が流る場合にバイパス電流Ｉｃ３を流すことによって、共振電流ＩＬ５の極性が負から正に反転することを防止することができる。
【００１５】
スイッチング回路２０及びバイパス回路３０はそれぞれドライブ回路６１、６２によって駆動され、ドライブ回路６１、６２は制御回路７０によってコントロールされる。本実施例では、共振コンデンサ６をインバータ１００のｔ点側に接続することにより、バイパス回路３０のＩＧＢＴ１３はｔ点を基準に駆動することができる。従って、ドライブ回路６２の電源はＩＧＢＴ１１を駆動するドライブ回路６１の電源と共用することが可能となる。
【００１６】
電流検出素子７１は、共振回路に流れる電流を検出し、共振電流検出回路７２は、電流検出素子７１の出力信号レベルを制御回路７０の入力レベルに適した信号に変換する。また、電流検出素子７３（電流検出手段）は、商用電源１から入力する電流を検出し、入力電流検出回路７４は電流検出素子７３の出力信号レベルを制御回路７０の入力レベルに適した信号に変換する。制御回路７０は入力電流と共振電流の関係から被加熱物の材質や状態を判断し、加熱動作の開始又は停止を行う。
【００１７】
また、入力電力設定部７５からの信号に応じてスイッチング回路２０のＩＧＢＴ１１，１２及びバイパス回路３０のＩＧＢＴ１３の導通期間を設定し入力電力を制御する。材質の検知は、過電流や過電圧の発生を防ぐために低電力かつ短時間で実施する必要がある。本実施例において材質検知の初期段階では、バイパス回路３０を導通状態にすることにより、共振回路のインピーダンスを大きくすることができ、過電流や過電圧の発生及び入力電力の急増を防ぐことができる。
【００１８】
次に図３を用いて本実施例の動作について説明する。図２、３において、スイッチング回路２０の上アームに流れる電流をＩｃ１、下アームに流れる電流をＩｃ２、バイパス回路３０に流れる電流をＩｃ３、共振電流をＩＬ５とする。また、上アームのＩＧＢＴ１１のコレクタ、エミッタ間の電圧をＶｃ１、下アームのＩＧＢＴ１２のコレクタ、エミッタ間の電圧をＶｃ２、共振コンデンサ６の共振電圧をＶｃ３、インバータの電源電圧をＶｐとする。
【００１９】
（モード１）図３において、ＩＧＢＴ１１の駆動信号がオンになり加熱コイル５の蓄積エネルギーがゼロになると共振電流ＩＬ５の極性が負から正に変わり、ＩＧＢＴ１１、共振コンデンサ６、加熱コイル５の経路で共振電流ＩＬ５が流れる。共振コンデンサＣ６は共振電流ＩＬ５によって放電され、Ｖｃ３は減少する。
（モード２）次にＩＧＢＴ１１の駆動信号がオフになると、共振電流ＩＬ５は正の極性を有しており、この電流はスナバコンデンサ３１、共振コンデンサ６、加熱コイル５の経路とスナバコンデンサ３２、共振コンデンサ６、加熱コイル５の経路で流れ続け、上アームのスナバコンデンサ３１は充電、下アームのスナバコンデンサ３２は放電される。
【００２０】
従って、ＩＧＢＴ１１のコレクタ、エミッタ間の電圧Ｖｃ１は徐々に増加し、ＩＧＢＴ１２のコレクタ、エミッタ間の電圧Ｖｃ２、即ち出力電圧は徐々に減少する。その後、Ｖｃ１の電圧がインバータの電源電圧Ｖｐに達し、下アームのダイオード２２に順方向の電圧が印加されると共振電流ＩＬ５は環流電流として加熱コイル５、ダイオード２２、共振コンデンサ６の経路で流れ続ける。この期間、ＩＧＢＴ１２、１３の駆動信号はオンになるが、電流の極性が変わらない限り、ダイオード２２を流れ続ける。
【００２１】
（モード３）次に加熱コイル５の蓄積エネルギーがゼロになり共振電流ＩＬ５の極性が正から負に変わると、既にＩＧＢＴ１２はオンしている為、共振電流ＩＬ５は加熱コイル５、共振コンデンサ６、ＩＧＢＴ１２の経路で流れ、共振コンデンサ６はＩＬ５によって充電される。
【００２２】
（モード４）電圧Ｖｃ３が増加しバイパス回路３０のダイオード２５に順方向の電圧が印加されると、既にＩＧＢＴ１３はオンしている為、共振電流ＩＬ５は加熱コイル５、ダイオード２５、ＩＧＢＴ１３、ＩＧＢＴ１２の経路で流れ、共振コンデンサ６の充電は一時停止する。
【００２３】
（モード５）次にＩＧＢＴ１３の駆動信号がオフになると、共振電流ＩＬ５は再び加熱コイル５、共振コンデンサ６、ＩＧＢＴ１２の経路で流れ、共振コンデンサ６はＩＬ５によって充電を開始する。
【００２４】
（モード６）次にＩＧＢＴ１２の駆動信号がオフになると、共振電流ＩＬ５は負の極性を有しており、この電流は加熱コイル５、共振コンデンサ６、スナバコンデンサ３１の経路と加熱コイル５、共振コンデンサ６、スナバコンデンサ３２の経路で流れ続け、上アームのスナバコンデンサ３１は放電、下アームのスナバコンデンサ３２は充電される。従って、ＩＧＢＴ１１のコレクタ、エミッタ間の電圧Ｖｃ１は徐々に減少し、ＩＧＢＴ１２のコレクタ、エミッタ間の電圧Ｖｃ２、即ち出力電圧は徐々に増加する。その後、Ｖｃ２の電圧がインバータの電源電圧Ｖｐに達し、上アームのダイオード２１に順方向の電圧が印加されると共振電流ＩＬ５は環流電流として加熱コイル５、共振コンデンサ６、ダイオード２１の経路で流れ続ける。この期間、ＩＧＢＴ１１の駆動信号はオンになるが、電流の極性が変わらない限り、ダイオード２１を流れ続ける。
【００２５】
以上のように、共振電流ＩＬ５の一周期の間に上記の動作が行われ、以後、この動作を繰り返す。このように本実施例は共振コンデンサ６に並列にバイパス回路３０を設け、負荷の特性を誘導性に維持することによって進相モードを回避することができる。これにより、スイッチング回路２０の導通期間を変化させることにより入力電力を制御することが可能となる。
【００２６】
次に入力電力の制御方法について説明する。図３にＩＧＢＴ１１，１２，１３の導通期間をｔ１、ｔ２、ｔ３で表し、共振電流の一周期、即ち駆動周期をｔｓで表す。図４は、例えば材質が鉄製の被加熱物を高出力で加熱する場合の制御条件を示しており、横軸はＩＧＢＴ１３の導通期間ｔ３、縦軸は入力電力Ｐｉｎを表している。ここでは、ｔ１、ｔ２は変えず、ｔ３を変化させ入力電力Ｐｉｎを制御する場合について説明する。図４より、ｔ３を長くすることにより入力電力Ｐｉｎを減少させることができる。これは、図３より共振電圧Ｖｃ３の電圧振幅値Ｖｃｐがｔ３の増加に伴い減少し、加熱コイル５に印加される電圧が減少するため共振電流ＩＬ５が小さくなるからである。入力電力Ｐｉｎはｔ３が約２５μｓ以上になると約１．１ｋＷで一定になる。これは、ＩＧＢＴ１３のオンしているｔ３の期間に、実際に共振電流がバイパス回路３０を流れる期間は、共振コンデンサ６の電圧Ｖｃ３が負から正に変わり、バイパス回路３０のダイオード２５に順方向の電圧が印加されてから、ＩＧＢＴ１３がオンの状態で共振電流の極性が負から正に反転するまでの期間に限られるからである。
【００２７】
従って、図４においてｔ１、ｔ２を変えずにｔ３のみで調整できる入力電力Ｐｉｎの下限値は約１．１ｋＷとなり、この値以下に設定する場合は、ｔ１、ｔ２を変化させることになる。
【００２８】
次に、ｔ３を約２５μｓ以上とし、ｔ１、ｔ２を変化させ入力電力Ｐｉｎを制御する場合について説明する。図５は、被加熱物を低出力で加熱する場合の制御条件を示しており、横軸は駆動周期ｔｓに対する上アーム導通期間ｔ１の導通比を示すＤｕｔｙ（＝ｔ１／ｔｓ）、縦軸は入力電力Ｐｉｎを表している。図５より、バイパス回路３０を導通状態にしてＤｕｔｙを変えることにより、入力電力Ｐｉｎを変化させることができる。本実施例において、Ｄｕｔｙを絞り入力電力を小さくしていくと、スイッチング回路２０がターンオン損失を発生する場合が生じる。
【００２９】
次に図６を用いてターンオン損失が発生する条件について説明する。図６において、ＩＧＢＴ１２がｔｏｆｆの時刻でターンオフすると、遮断電流Ｉｏｆｆによってスナバコンデンサ３１、３２はそれぞれ放電、充電される。
【００３０】
しかし、Ｉｏｆｆが小さいと、スナバコンデンサ３２の電圧、即ちＩＧＢＴ１２の電圧Ｖｃ２の電圧立ち上がりが遅くなる。ＩＧＢＴ１１，１２のデッドタイムｔｄが経過した後、ＩＧＢＴ１１がｔｏｎの時刻にターンオンすると、Ｖｃ２がインバータの電源電圧Ｖｐよりも低いため、ＩＧＢＴ１１にはスナバコンデンサ３１、３２をそれぞれ放電、充電する大きな電流が流れ、ターンオン損失が発生する。同様に、ＩＧＢＴ１１の遮断電流が小さい場合においても、ＩＧＢＴ１２のターンオン時にスナバコンデンサ３１，３２を充放電する大きな電流が流れ、ターンオン損失が発生することになる。
【００３１】
従って、入力電力を所望の電力以下にする場合、Ｄｕｔｙの最小値を設定し、数秒間インバータを駆動した後、数秒間インバータを停止させる間欠制御に切り替えて入力電力をコントロールすることになる。しかしながら、上記間欠制御では、インバータ駆動時に所望する電力より大きな電力を入力することになり被加熱物を常に一定の温度で加熱することが不可能となる。本実施例では図２に示すように、スナバコンデンサ３１，３２がスイッチング回路２０に常時接続されるため、低出力で加熱する場合にターンオン損失が発生する場合が生じる。
【００３２】
（実施例３）
図７は本発明の第３の実施形態である電磁誘導加熱装置の回路構成図である。図２と同一部分については同一符号を付しており説明は省略する。図７において前記実施例２と異なる点は、インバータ１００のｔ点とｏ点間に、スナバコンデンサ３３とＩＧＢＴ１４を直列に接続したスナバコンデンサ切替回路４０を接続し、ＩＧＢＴ１４をオフすることによってスナバコンデンサ３３をスイッチング回路２０から切り離すことができる点である。ＩＧＢＴ１４にはダイオード２４が逆方向に並列接続されており、スナバコンデンサ３３の電荷を放電する場合に使用される。ＩＧＢＴ１４はドライブ回路６３により駆動され、ドライブ回路６３は制御回路７０によってコントロールされる。
【００３３】
次に図８を用いて本実施例の動作について説明する。図８において、ＩＧＢＴ１２がｔｏｆｆの時刻でターンオフすると、遮断電流ＩｏｆｆによってＩＧＢＴ１１，１２のコレクタとエミッタ間の出力容量（図示せず）はそれぞれ放電、充電される。ここでは、Ｉｏｆｆが小さくても、ＩＧＢＴ１１，１２の出力容量が小さいためＩＧＢＴ１２の電圧Ｖｃ２の電圧立ち上がりは早く、Ｖｃ２がインバータの電源電圧Ｖｐに達するまでに要す時間ｔｒはデッドタイムｔｄよりも短くなる。ＩＧＢＴ１１の並列ダイオード２１に順方向の電圧が印加されるとダイオード２１には還流電流が流れ、還流期間であるｔｏｎの時刻にＩＧＢＴ１１がターンオンすることになる。そのため、ターンオン時はゼロボルトスイッチングを実現することができ、遮断電流が小さくなる場合においてもターンオン損失を無くすことができる。
【００３４】
従って、低出力時においても間欠制御のようにインバータの駆動を停止することなく連続してインバータを駆動することができ、被加熱物を常に一定の温度で加熱することが可能となる。
【００３５】
一方、高出力で加熱を行う場合には、ＩＧＢＴ１４をオンすることによりコンデンサ３３はインバータ１００のｔ点とｏ点間に接続され、前記実施例２と同様にターンオフ時に両ＩＧＢＴに印加される電圧の変化は低減され、ターンオフ損失は抑制される。図７において、スナバコンデンサ３３とＩＧＢＴ１４及びダイオード２４は上アームのＩＧＢＴ１１に接続しても、前述のような効果は得られる。
【００３６】
（実施例４）
図９は本発明の第４の実施形態である電磁誘導加熱装置の回路構成図である。電源１０、スイッチング回路２０は省略しバイパス回路３０及び共振回路６０のみ示す。また、図２と同一部分については同一符号を付しており説明は省略する。
【００３７】
図９において加熱コイル５と共振コンデンサ６で構成された共振回路６０はインバータ１００のｐ点とｔ点との間に接続されている。共振コンデンサ６には並列にバイパス回路３０が接続されており、バイパス回路３０は直列接続されたＩＧＢＴ１３とＩＧＢＴ１３の逆電流を阻止するダイオード２５、及びＩＧＢＴ１３に逆方向に並列接続されたダイオード２３によって構成されている。ここで、ｐ点から共振コンデンサ６に向かって流れる共振電流ＩＬ５を正とすると、バイパス回路３０に流れるバイパス電流Ｉｃ３は負の一方向である。本実施例では、入力電力を下げる場合、下アームの導通期間を絞り上アームの導通期間を延ばすため、上アームの導通期間内に共振電流ＩＬ５の極性が負から正に反転し易くなる。
【００３８】
従って、負の方向に共振電流ＩＬ５が流る場合にバイパス電流Ｉｃ３を流すことによって、共振電流ＩＬ５の極性が負から正に反転することを防止することができる。本実施例では、バイパス回路３０のＩＧＢＴ１３はｐ点を基準に駆動することになり、バイパス回路３０のドライブ回路はスイッチング回路２０のドライブ回路６１と別に新たな駆動電源を設ける必要がある。
【００３９】
本実施例の動作は、前記実施例２の上下アームの制御が逆転することになるが、その他は実施例２と同様であり、説明は省略する。
【００４０】
（実施例５）
図１０は本発明の第５の実施形態である電磁誘導加熱装置の回路の一部である。電源１０、スイッチング回路２０は省略しバイパス回路３０及び共振回路６０のみ示す。また、図２と同一部分については同一符号を付しており説明は省略する。図１０において、バイパス回路３０のスイッチング素子は逆方向の耐圧を有する逆電流阻止形のＩＧＢＴ１６であり、前記図２のＩＧＢＴ１３と逆電流阻止用のダイオード２５が１個のＩＧＢＴ１６に置き換えられた構成となる。前記図２のバイパス回路３０では、ＩＧＢＴ１３とダイオード２５でそれぞれ損失が発生するが、本実施例ではＩＧＢＴ１６の損失分となるため、インバータ１００における回路損失は低減し変換効率は向上する。本実施例の動作について、前記実施例２と同様であり、説明は省略する。
【００４１】
（実施例６）
図１１は本発明の第６の実施形態である電磁誘導加熱装置の回路の一部である。電源１０、スイッチング回路２０は省略しバイパス回路３０及び共振回路６０のみ示す。また、図２と同一部分については同一符号を付しており説明は省略する。図１１において共振コンデンサ６には並列にバイパス回路３０が接続されており、バイパス回路３０はＩＧＢＴ１８とＩＧＢＴ１８に逆方向に並列接続されたダイオード２８によって構成されている。前記実施例２と異なる点は、バイパス回路３０に正負のバイパス電流が流れる点である。本実施例では、入力電力を下げる場合、上アームの導通期間を絞り下アームの導通期間を延ばすため、下アームの導通期間内に共振電流ＩＬ５の極性が負から正に反転し易くなる。
【００４２】
従って、負の方向に共振電流ＩＬ５が流る場合にダイオード２８を介してバイパス電流Ｉｃ４を流すことによって、共振電流ＩＬ５の極性が負から正に反転することを防止することができる。
【００４３】
次に図１２を用いて本実施例の動作について説明する。図１２において、スイッチング回路２０の上アームに流れる電流をＩｃ１、下アームに流れる電流をＩｃ２、バイパス回路３０に流れる電流をＩｃ４、共振電流をＩＬ５とする。また、上アームのＩＧＢＴ１１のコレクタ、エミッタ間の電圧をＶｃ１、下アームのＩＧＢＴ１２のコレクタ、エミッタ間の電圧をＶｃ２、共振コンデンサ６の共振電圧をＶｃ４、インバータの電源電圧をＶｐとする。
【００４４】
（モード１）図１２において、ＩＧＢＴ１１の駆動信号がオンの状態でＩＧＢＴ１８の駆動信号がオフになると、共振電流ＩＬ５はＩＧＢＴ１１、加熱コイル５、共振コンデンサ６の経路で流れ、共振コンデンサＣ６は共振電流ＩＬ５によって充電が開始され、Ｖｃ４は増加する。
【００４５】
（モード２）次にＩＧＢＴ１１の駆動信号がオフになると、共振電流ＩＬ５は正の極性を有しており、この電流はスナバコンデンサ３１、加熱コイル５、共振コンデンサ６の経路とスナバコンデンサ３２、加熱コイル５、共振コンデンサ６の経路で流れ続け、上アームのスナバコンデンサ３１は充電、下アームのスナバコンデンサ３２は放電される。従って、ＩＧＢＴ１１のコレクタ、エミッタ間の電圧Ｖｃ１は徐々に増加し、ＩＧＢＴ１２のコレクタ、エミッタ間の電圧Ｖｃ２、即ち出力電圧は徐々に減少する。その後、Ｖｃ１の電圧がインバータの電源電圧Ｖｐに達し、下アームのダイオード２２に順方向の電圧が印加されると共振電流ＩＬ５は環流電流として加熱コイル５、共振コンデンサ６、ダイオード２２の経路で流れ続ける。この期間、ＩＧＢＴ１２の駆動信号はオンになるが、電流の極性が変わらない限り、ダイオード２２を流れ続ける。
【００４６】
（モード３）次に加熱コイル５の蓄積エネルギーがゼロになり共振電流ＩＬ５の極性が正から負に変わると、既にＩＧＢＴ１２はオンしている為、共振電流ＩＬ５は共振コンデンサ６の電圧Ｖｃ４を電圧源として加熱コイル５、ＩＧＢＴ１２の経路で流れＶｃ４は減少する。
【００４７】
（モード４）Ｖｃ４が減少しバイパス回路３０のダイオード２８に順方向の電圧が印加されると、共振電流ＩＬ５は加熱コイル５、ＩＧＢＴ１２、ダイオード２８の経路で流れる。
【００４８】
（モード５）次にＩＧＢＴ１２の駆動信号がオフになると、共振電流ＩＬ５は負の極性を有しており、この電流はダイオード２８、加熱コイル５、スナバコンデンサ３１の経路とダイオード２８、加熱コイル５、スナバコンデンサ３２の経路で流れ続け、上アームのコンデンサ３１は放電、下アームのコンデンサ３２は充電される。従って、ＩＧＢＴ１１のコレクタ、エミッタ間の電圧Ｖｃ１は徐々に減少し、ＩＧＢＴ１２のコレクタ、エミッタ間の電圧Ｖｃ２、即ち出力電圧は徐々に増加する。その後、Ｖｃ２の電圧がインバータの電源電圧Ｖｐに達し、上アームのダイオード２１に順方向の電圧が印加されると共振電流ＩＬ５は環流電流としてダイオード２８、加熱コイル５、ダイオード２１の経路で流れ続ける。ＩＧＢＴ１２がオフするタイミングをトリガーとしてＩＧＢＴ１８の駆動信号はオンするが、共振電流の極性が変わらない限り、ダイオード２８を流れ続ける。同様に、この環流期間にＩＧＢＴ１１の駆動信号はオンになるが、共振電流の極性が変わらない限り、ダイオード２１を流れ続ける。
【００４９】
（モード６）次に加熱コイル５の蓄積エネルギーがゼロになり共振電流ＩＬ５の極性が負から正に変わると、既にＩＧＢＴ１１、１８はオンしている為、共振電流ＩＬ５はＩＧＢＴ１１、加熱コイル５、ＩＧＢＴ１８の経路で流れる。
【００５０】
以上のように、共振電流ＩＬ５の一周期の間に上記の動作が行われ、以後、この動作を繰り返す。このように本実施例は共振コンデンサ６に並列にバイパス回路３０を設け、負荷の特性を誘導性に維持することによって進相モードを回避することができる。これにより、スイッチング回路２０の導通期間を変化させることにより入力電力を制御することが可能となる。
【００５１】
次に入力電力の制御方法について説明する。図１２にＩＧＢＴ１１，１２，１８の導通期間をｔ１、ｔ２、ｔ４で表し、共振電流の一周期、即ち駆動周期をｔｓで表す。図１２において、ｔ４を制御すると共振電圧Ｖｃ４の電圧振幅値Ｖｃｐは変化する。前述のモード３において、負の方向の共振電流ＩＬ５は共振コンデンサ６の電圧Ｖｃ４を電圧源として加熱コイル５、ＩＧＢＴ１２の経路で流れるため、共振電流ＩＬ５はＶｃ４の大きさに応じて変化する。従って、ｔ４を変化させ入力電力Ｐｉｎを制御することができる。また、実施例２と同様にｔ１、ｔ２の導通期間を変えて入力電力Ｐｉｎを制御することも可能であり、設定された入力電力の大きさに応じて導通期間を制御する。
【００５２】
（実施例７）
図１３は本発明の第７の実施形態である電磁誘導加熱装置の回路の一部である。電源１０、スイッチング回路２０は省略しバイパス回路３０及び共振回路６０のみ示す。また、図２と同一部分については同一符号を付しており説明は省略する。図１３において前記実施例２と異なる点は、バイパス回路３０のダイオード２５に並列にＩＧＢＴ１５が接続されているため、バイパス回路３０に正負のバイパス電流を流すことができる点である。前記実施例６では、バイパス回路３０に正負の電流が流れるが、負方向はダイオード２８を流れるため、ＩＧＢＴ１８で制御できる電流は正方向のみである。本実施例ではＩＧＢＴ１３，１５を駆動することにより正負のバイパス電流を制御することが可能であり、共振回路６０の負荷特性を誘導性に維持しながら共振電圧及びスイッチング回路のＤｕｔｙを制御し入力電力を変える。制御方法については、ＩＧＢＴ１３は前記実施例２と同様の制御となり、ＩＧＢＴ１５は前期実施例６のＩＧＢＴ１８と同様となるため、説明は省略する。
【００５３】
（実施例８）
図１４は本発明の第８の実施形態である電磁誘導加熱装置の回路の一部である。電源１０、スイッチング回路２０は省略しバイパス回路３０及び共振回路６０のみ示す。また、図２と同一部分については同一符号を付しており説明は省略する。図１４において、バイパス回路３０は逆方向の耐圧を有する逆電流阻止形のＩＧＢＴ１６とＩＧＢＴ１７を逆方向に並列接続した構成となる。前記図１３のバイパス回路３０では、正方向の電流がダイオード２３とＩＧＢＴ１５に流れ、負方向の電流がダイオード２５とＩＧＢＴ１３に流れるため、ダイオード及びＩＧＢＴの各々で損失が発生する。本実施例ではＩＧＢＴ１６とＩＧＢＴ１７の損失となるため、インバータ１００における回路損失は低減し変換効率は向上する。本実施例の動作について、前記実施例７と同様であり、説明は省略する。
【００５４】
（実施例９）
図１５は本発明の第９の実施形態である電磁誘導加熱装置の回路の一部である。電源１０、スイッチング回路２０は省略しバイパス回路３０及び共振回路６０のみ示す。また、図２と同一部分については同一符号を付しており説明は省略する。図１５において共振コンデンサ６には直列接続されたＩＧＢＴ１９と補助共振コンデンサ７が並列に接続されており、ＩＧＢＴ１９にはダイオード２９が逆方向に並列接続されている。前記実施例２と異なる点は、バイパス回路３０は補助共振コンデンサ７とＩＧＢＴ１９あるいはダイオード２９を介してバイパス電流を流すことができる点である。
【００５５】
これにより、ＩＧＢＴ１９を導通状態にすると、共振コンデンサ６と補助共振コンデンサ７が並列に接続されるため、共振回路６０の共振周波数が低くなり、誘導性の負荷特性を維持し易くなる。また、非磁性ステンレスやアルミのように低抵抗の材質で作られた被加熱物が負荷となって等価インダクタンスが小さくなった場合においても、補助共振コンデンサ７を導通状態にすることで共振周波数の増加を抑制することができ、誘導性の負荷特性を維持することが可能となる。ＩＧＢＴ１３の制御方法については前記実施例２と同様であり、説明は省略する。
【００５６】
（実施例１０）
図１６は本発明の第１０の実施形態である電磁誘導加熱装置のブロック図である。図１６において、インバータ１００及びインバータ２００はそれぞれドライブ回路６１，６２，６３及びドライブ回路６４，６５，６６により駆動され、各ドライブ回路は制御回路７０によってコントロールされる。制御回路７０はインバータ１００及び２００のスイッチング回路とバイパス回路及びスナバコンデンサ切替回路の導通期間を設定する導通期間設定部８０，８１とインバータの駆動周波数を設定する可変周波数発振器８２を備えている。本発明では図１６に示すようにインバータを複数個備えてそれぞれ別々の被加熱物を加熱する場合、干渉音の発生を防ぐために、各インバータは可変周波数発振器８２から出力される同一周波数で駆動され、導通期間設定部８０，８１でスイッチング回路、バイパス回路及びスナバコンデンサ切替回路の導通期間を制御し入力電力をコントロールする。
【００５７】
図１７は本実施例において、可変周波数発振器８２の発振周波数、即ち駆動周波数ｆｓが変動せず概略一定に制御された場合の共振電流ＩＬ５を表している。図１７において、ＩＬ５の振幅値は、商用交流電圧Ｖａｃを整流平滑して得られたインバータの電源電圧に応じて変化しており、ＩＬ５の包絡線はインバータの電源電圧とほぼ相似となる。この場合、インバータから外部に放射されるノイズレベルは、図１８に示すような周波数特性となり、駆動周波数ｆｓにおけるノイズレベルＮｐを最大値として高次高調波になるに従って減少する。
【００５８】
本実施例において、可変周波数発振器８２の発振周波数を変化させ、駆動周波数ｆｓを可変することにより、ノイズの周波数成分を拡散しノイズレベルを低減することができる。図１９は駆動周波数をインバータの電源電圧に同期して変化させた場合の共振電流ＩＬ５を表している。図１９において、駆動周波数はｆｓを基準に±Δｆｓ分だけ変動させており、インバータの電源電圧が高くなると駆動周波数が高くなり、電圧が低くなると周波数が低くなるように設定している。共振型のインバータでは共振周波数より高い駆動周波数で動作させるため、駆動周波数を高くすると共振電流は減る。
【００５９】
従って、インバータの電源電圧が高いときに駆動周波数を高めにシフトすることにより、共振電流ＩＬ５を小さくすることができ、駆動周波数を一定に制御した場合のＩＬ５の最大値をＩｐとすると図１９に示すようにＩＬ５の最大値をＩｐからＩｄに低減することが可能となる。一方、インバータの電源電圧が低いときは駆動周波数を低めにシフトするため、共振電流ＩＬ５を大きくすることができ、駆動周波数を一定に制御した場合よりＩＬ５を増加させることが可能となる。この場合、インバータから外部に放射されるノイズレベルは、図２０に示すような周波数特性となり、駆動周波数を一定に制御した場合より、駆動周波数ｆｓ付近におけるノイズレベルの最大値をＮｐからＮｄのレベルに低減することができる。
【００６０】
【発明の効果】
本発明によれば、共振コンデンサにバイパス回路を備えて共振電流の一部をバイパスすることにより、共振回路の負荷特性を誘導性に維持することができ、Ｄｕｔｙ制御によって入力電力を制御でき、スイッチング素子の損失発生を抑えることができる。複数のインバータを同時に駆動した場合においても、全てのインバータの駆動周波数を同一にすることができるため、干渉音が発生しない電磁誘導加熱装置を提供することができる。
【図面の簡単な説明】
【図１】本発明の第１の実施形態である電磁誘導加熱装置のブロック図である。
【図２】本発明の第２の実施形態である電磁誘導加熱装置の回路構成図である。
【図３】図２の実施形態における動作説明図である。
【図４】図２の実施形態における制御方法の説明図である。
【図５】図２の実施形態における制御方法の説明図である。
【図６】図２の実施形態における動作説明図である。
【図７】本発明の第３の実施形態である電磁誘導加熱装置の回路構成図である。
【図８】図７の実施形態における動作説明図である。
【図９】本発明の第４の実施形態である電磁誘導加熱装置の回路構成図である。
【図１０】本発明の第５の実施形態である電磁誘導加熱装置の回路の一部である。
【図１１】本発明の第６の実施形態である電磁誘導加熱装置の回路の一部である。
【図１２】図１１の実施形態における動作説明図である。
【図１３】本発明の第７の実施形態である電磁誘導加熱装置の回路の一部である。
【図１４】本発明の第８の実施形態である電磁誘導加熱装置の回路の一部である。
【図１５】本発明の第７の実施形態である電磁誘導加熱装置の回路の一部である。
【図１６】本発明の第１０の実施形態である電磁誘導加熱装置のブロック図である。
【図１７】図１６の実施形態における動作説明図である。
【図１８】図１６の実施形態における説明図である。
【図１９】図１６の実施形態における動作説明図である。
【図２０】図１６の実施形態における説明図である。
【符号の説明】
１…商用交流電源、２…整流回路、３…インダクタ、４，６，７，３１，３２，３３…コンデンサ、５…加熱コイル、１０…電源、１１，１２，１３，１４，１５，１６，１７，１８，１９…半導体スイッチング素子、２０…スイッチング回路、２１，２２，２３，２４，２５，２８，２９…ダイオード、３０…バイパス回路、４０…スナバコンデンサ切換回路、６０…共振回路、６１，６２，６３，６４，６５，６６…ドライブ回路、７０…制御回路、７１，７３…電流検出素子、７２…共振電流検出回路、７４…入力電流検出回路、７５…入力電力設定部、８０，８１…導通期間設定部、８２…可変周波数発振器、１００，２００…インバータ。[0001]
TECHNICAL FIELD OF THE INVENTION
The present invention relates to an electromagnetic induction heating device having a plurality of heating units.
[0002]
[Prior art]
2. Description of the Related Art In recent years, an inverter-type electromagnetic induction heating device that heats an object to be heated such as a pot without using a fire has been widely used. An electromagnetic induction heating device applies a high-frequency current to a heating coil to generate an eddy current in an object to be heated made of a material such as iron or stainless steel arranged close to the coil, and to generate an eddy current by an electric resistance of the object to be heated. Generate heat. Since the temperature of an object to be heated can be controlled and safety is high, it is recognized as a new heat source.
[0003]
Conventionally, electric cookers incorporated in system kitchens, etc., used heaters with heat sources such as sheathed heaters, plate heaters, and halogen heaters. In recent years, some have been replaced with induction heating cookers. Or two or more induction heating cookers. As a method of changing the input power of the electromagnetic induction heating device to control the temperature of the object to be heated, a method of changing the drive frequency of the inverter is generally used. However, when a plurality of heating coils are provided to heat different objects to be heated, there is a problem that an interference sound is generated from the object to be heated due to a difference frequency between the inverters.
[0004]
As a conventional example which solves such a problem, there is an induction heating cooker as disclosed in Japanese Patent No. 2532565. This cooker controls the input power by changing the conduction period of the switching element at a constant drive frequency.
[0005]
[Problems to be solved by the invention]
In the above conventional example, even if a plurality of inverters are operated, the input power can be changed at the same driving frequency, so that the generation of interference noise can be prevented. However, when driving with low power, there is a problem that the loss generated in one switching element increases.
[0006]
SUMMARY OF THE INVENTION An object of the present invention is to provide an electromagnetic induction heating device that suppresses the occurrence of loss of a switching element, prevents the occurrence of interference noise when simultaneously driving a plurality of inverters, and controls each input power.
[0007]
[Means for Solving the Problems]
The electromagnetic induction heating device of the present invention has an inverter circuit that converts a DC voltage to a high-frequency AC voltage, the inverter circuit includes a switching circuit and a resonance circuit, and the resonance circuit is connected in series with the heating coil and the heating coil. And the resonance circuit is connected between one of both terminals (p / o) of the DC power supply and the output terminal (t) of the switching circuit, and is connected in parallel to the resonance capacitor. Having a bypass circuit
[0008]
BEST MODE FOR CARRYING OUT THE INVENTION
(Example 1)
FIG. 1 is a block diagram of an electromagnetic induction heating device according to a first embodiment of the present invention. In FIG. 1, the inverter 100 includes a switching circuit 20, a heating coil 5, a resonance circuit 60 including a resonance capacitor 6, and a bypass circuit 30.
[0009]
The switching circuit 20 is connected between the positive electrode p point and the negative electrode o point of the power supply 10, converts a voltage supplied from the power supply 10 into a high frequency voltage, applies the high frequency voltage to the resonance circuit 60, Supply high frequency power. The bypass circuit 30 is connected in parallel with the resonance capacitor 6, and controls a load characteristic of the resonance circuit 60 by bypassing a current flowing through the resonance capacitor 6. Generally, in a resonance type inverter, the drive frequency fs is set to be higher than the resonance frequency fr so that the characteristic of the resonance load becomes inductive, and the current flowing through the resonance circuit becomes a lag phase with respect to the output voltage of the inverter. Control. In FIG. 1, the current IL5 flowing through the resonance circuit 60 is controlled so as to have a lag phase with respect to the voltage at the connection point t between the switching circuit 20 and the resonance circuit 60.
[0010]
However, if the power control is performed by changing the conduction period of the switching element while keeping the drive frequency fs constant, the polarity of the current is inverted during the conduction period of the switching circuit 20 and the phase shifts to the advanced mode in which the current is advanced in phase with the output voltage. May occur. This is a mode that must be avoided in a resonance-type inverter because loss of the switching circuit 20 increases. In the present invention, the load characteristic of the resonance circuit 60 is controlled by controlling the voltage of the resonance capacitor by changing the conduction state of the bypass circuit according to the material and state of the object to be heated and the magnitude of the set input power. It can be kept inductive. That is, the resonance frequency of the resonance circuit 60 can be always lower than the drive frequency of the switching circuit 20.
[0011]
Accordingly, the inverter 100 can control the input power by changing the conduction period of the switching circuit 20 while keeping the drive frequency fs constant. Thus, the bypass circuit 30 plays a role as an auxiliary switching circuit for realizing operation at a constant drive frequency fs. The switching circuit 20 and the bypass circuit 30 are driven by drive circuits 61 and 62, respectively, and the drive circuits 61 and 62 are controlled by a control circuit 70. The input power setting unit 75 is an interface for the user to set the input power, and sends a signal to the control circuit 70 according to the set output. The control circuit 70 controls the switching circuit 20 and the bypass circuit 30 according to a signal from the input power setting unit 75.
[0012]
(Example 2)
FIG. 2 is a circuit configuration diagram of an electromagnetic induction heating device according to a second embodiment of the present invention. 2, a power supply 10 includes a rectifying circuit 2 for rectifying an AC voltage of a commercial power supply 1, a smoothing circuit including an inductor 3 and a capacitor 4, and converts an AC voltage into a DC voltage to supply power to an inverter 100. . IGBT11 and IGBT12, which are power semiconductor switching elements, are connected in series between the positive electrode p point and the negative electrode o point of the capacitor 4 to form a switching circuit 20. Diodes 21 and 22 are connected in parallel to the IGBTs 11 and 12, respectively, in opposite directions.
[0013]
In addition, snubber capacitors 31 and 32 are connected in parallel to the IGBTs 11 and 12, respectively. The snubber capacitors 31 and 32 are charged or discharged by a cutoff current when the IGBT 11 or IGBT 12 is turned off. Since the capacitance of the snubber capacitors 31 and 32 is sufficiently larger than the output capacitance between the collector and the emitter of the IGBTs 11 and 12, the change in the voltage applied to both IGBTs at the time of turn-off is reduced, and the turn-off loss is suppressed. Assuming that the connection point of the IGBTs 11 and 12, that is, the output terminal is a point t, a resonance circuit 60 including the heating coil 5 and the resonance capacitor 6 is connected between the points t and o.
[0014]
A bypass circuit 30 is connected to the resonance capacitor 6 in parallel. The bypass circuit 30 includes an IGBT 13 connected in series, a diode 25 for blocking a reverse current of the IGBT 13, and a diode 23 connected in parallel to the IGBT 13 in the reverse direction. Have been. Here, assuming that the resonance current IL5 flowing from the point t toward the resonance capacitor 6 is positive, the bypass current Ic3 flowing through the bypass circuit 30 is in one negative direction. In this embodiment, when lowering the input power, the conduction period of the upper arm is narrowed and the conduction period of the lower arm is extended, so that the polarity of the resonance current IL5 is easily inverted from negative to positive during the conduction period of the lower arm. Therefore, by flowing the bypass current Ic3 when the resonance current IL5 flows in the negative direction, it is possible to prevent the polarity of the resonance current IL5 from being inverted from negative to positive.
[0015]
The switching circuit 20 and the bypass circuit 30 are driven by drive circuits 61 and 62, respectively, and the drive circuits 61 and 62 are controlled by a control circuit 70. In this embodiment, the IGBT 13 of the bypass circuit 30 can be driven based on the point t by connecting the resonance capacitor 6 to the point t of the inverter 100. Therefore, the power supply of the drive circuit 62 can be shared with the power supply of the drive circuit 61 that drives the IGBT 11.
[0016]
The current detection element 71 detects a current flowing through the resonance circuit, and the resonance current detection circuit 72 converts an output signal level of the current detection element 71 into a signal suitable for an input level of the control circuit 70. The current detection element 73 (current detection means) detects a current input from the commercial power supply 1, and the input current detection circuit 74 converts the output signal level of the current detection element 73 into a signal suitable for the input level of the control circuit 70. Convert. The control circuit 70 determines the material and state of the object to be heated from the relationship between the input current and the resonance current, and starts or stops the heating operation.
[0017]
In addition, according to a signal from the input power setting unit 75, the conduction period of the IGBTs 11 and 12 of the switching circuit 20 and the IGBT 13 of the bypass circuit 30 is set to control the input power. The detection of the material needs to be performed with low power and in a short time in order to prevent occurrence of overcurrent or overvoltage. In the present embodiment, in the initial stage of material detection, the impedance of the resonance circuit can be increased by making the bypass circuit 30 conductive, thereby preventing the occurrence of overcurrent or overvoltage and the rapid increase in input power.
[0018]
Next, the operation of this embodiment will be described with reference to FIG. 2 and 3, the current flowing to the upper arm of the switching circuit 20 is Ic1, the current flowing to the lower arm is Ic2, the current flowing to the bypass circuit 30 is Ic3, and the resonance current is IL5. The voltage between the collector and the emitter of the upper arm IGBT 11 is Vc1, the voltage between the collector and the emitter of the lower arm IGBT 12 is Vc2, the resonance voltage of the resonance capacitor 6 is Vc3, and the power supply voltage of the inverter is Vp.
[0019]
(Mode 1) In FIG. 3, when the drive signal of the IGBT 11 is turned on and the energy stored in the heating coil 5 becomes zero, the polarity of the resonance current IL5 changes from negative to positive, and the path of the IGBT 11, the resonance capacitor 6, and the heating coil 5 The resonance current IL5 flows. The resonance capacitor C6 is discharged by the resonance current IL5, and Vc3 decreases.
(Mode 2) Next, when the drive signal of the IGBT 11 is turned off, the resonance current IL5 has a positive polarity, and this current flows through the snubber capacitor 31, the resonance capacitor 6, the path of the heating coil 5 and the snubber capacitor 32, the resonance The flow continues through the path of the condenser 6 and the heating coil 5, so that the upper arm snubber condenser 31 is charged and the lower arm snubber condenser 32 is discharged.
[0020]
Therefore, the voltage Vc1 between the collector and the emitter of the IGBT 11 gradually increases, and the voltage Vc2 between the collector and the emitter of the IGBT 12, that is, the output voltage gradually decreases. After that, when the voltage of Vc1 reaches the power supply voltage Vp of the inverter and a forward voltage is applied to the diode 22 of the lower arm, the resonance current IL5 flows through the path of the heating coil 5, the diode 22, and the resonance capacitor 6 as a circulating current. to continue. During this period, the drive signals of the IGBTs 12 and 13 are turned on, but continue to flow through the diode 22 unless the polarity of the current changes.
[0021]
(Mode 3) Next, when the energy stored in the heating coil 5 becomes zero and the polarity of the resonance current IL5 changes from positive to negative, the IGBT 12 has already been turned on. Flowing on the path of the IGBT 12, the resonance capacitor 6 is charged by the IL5.
[0022]
(Mode 4) When the voltage Vc3 increases and a forward voltage is applied to the diode 25 of the bypass circuit 30, the IGBT 13 has already been turned on. Therefore, the resonance current IL5 is applied to the heating coil 5, the diode 25, the IGBT 13, and the IGBT 12. Flowing through the path, charging of the resonance capacitor 6 is temporarily stopped.
[0023]
(Mode 5) Next, when the drive signal of the IGBT 13 is turned off, the resonance current IL5 flows again through the path of the heating coil 5, the resonance capacitor 6, and the IGBT 12, and the resonance capacitor 6 starts charging by the IL5.
[0024]
(Mode 6) Next, when the drive signal of the IGBT 12 is turned off, the resonance current IL5 has a negative polarity, and this current flows through the path of the heating coil 5, the resonance capacitor 6, the snubber capacitor 31, the heating coil 5, the resonance The capacitor 6 and the snubber capacitor 32 continue to flow, and the upper arm snubber capacitor 31 is discharged and the lower arm snubber capacitor 32 is charged. Therefore, the voltage Vc1 between the collector and the emitter of the IGBT 11 gradually decreases, and the voltage Vc2 between the collector and the emitter of the IGBT 12, that is, the output voltage gradually increases. Thereafter, when the voltage of Vc2 reaches the power supply voltage Vp of the inverter and a forward voltage is applied to the diode 21 of the upper arm, the resonance current IL5 flows as a circulating current through the path of the heating coil 5, the resonance capacitor 6, and the diode 21. to continue. During this period, the drive signal of the IGBT 11 is turned on, but continues to flow through the diode 21 unless the polarity of the current changes.
[0025]
As described above, the above operation is performed during one cycle of the resonance current IL5, and thereafter, this operation is repeated. As described above, in the present embodiment, the bypass circuit 30 is provided in parallel with the resonance capacitor 6, and the phase characteristic can be avoided by maintaining the load characteristics inductive. This makes it possible to control the input power by changing the conduction period of the switching circuit 20.
[0026]
Next, a control method of the input power will be described. In FIG. 3, the conduction periods of the IGBTs 11, 12, and 13 are represented by t1, t2, and t3, and one cycle of the resonance current, that is, the drive cycle is represented by ts. FIG. 4 shows a control condition in a case where a material to be heated made of iron, for example, is heated at a high output. The horizontal axis represents the conduction period t3 of the IGBT 13, and the vertical axis represents the input power Pin. Here, the case where the input power Pin is controlled by changing t3 without changing t1 and t2 will be described. From FIG. 4, it is possible to reduce the input power Pin by increasing t3. This is because, as shown in FIG. 3, the voltage amplitude value Vcp of the resonance voltage Vc3 decreases as t3 increases, and the voltage applied to the heating coil 5 decreases, so that the resonance current IL5 decreases. The input power Pin becomes constant at about 1.1 kW when t3 becomes about 25 μs or more. This is because the voltage Vc3 of the resonance capacitor 6 changes from negative to positive during the period when the resonance current actually flows through the bypass circuit 30 during the period t3 when the IGBT 13 is on, and the diode 25 of the bypass circuit 30 This is because it is limited to a period from when the voltage is applied to when the polarity of the resonance current is inverted from negative to positive while the IGBT 13 is on.
[0027]
Therefore, in FIG. 4, the lower limit value of the input power Pin that can be adjusted only at t3 without changing t1 and t2 is about 1.1 kW, and when set below this value, t1 and t2 are changed.
[0028]
Next, a case will be described in which t3 is set to about 25 μs or more, and t1 and t2 are changed to control the input power Pin. FIG. 5 shows control conditions in the case where the object to be heated is heated at a low output. The horizontal axis represents Duty (= t1 / ts) representing the conduction ratio of the upper arm conduction period t1 to the drive cycle ts, and the vertical axis represents the control cycle. Represents the input power Pin. As shown in FIG. 5, the input power Pin can be changed by changing the Duty by setting the bypass circuit 30 to the conductive state. In this embodiment, when the duty is reduced and the input power is reduced, the switching circuit 20 may generate a turn-on loss.
[0029]
Next, the conditions under which turn-on loss occurs will be described with reference to FIG. In FIG. 6, when the IGBT 12 is turned off at the time of toff, the snubber capacitors 31 and 32 are discharged and charged by the cutoff current Ioff.
[0030]
However, when Ioff is small, the voltage of the snubber capacitor 32, that is, the voltage Vc2 of the IGBT 12 rises slowly. When the IGBT 11 is turned on at the time of ton after the dead time td of the IGBTs 11 and 12 has elapsed, a large current for discharging and charging the snubber capacitors 31 and 32 respectively is supplied to the IGBT 11 because Vc2 is lower than the power supply voltage Vp of the inverter. Flow and turn-on losses occur. Similarly, even when the cutoff current of the IGBT 11 is small, a large current for charging and discharging the snubber capacitors 31 and 32 flows when the IGBT 12 is turned on, and a turn-on loss occurs.
[0031]
Therefore, when the input power is set to be equal to or lower than the desired power, the input power is controlled by setting the minimum value of the duty, driving the inverter for several seconds, and then switching to intermittent control for stopping the inverter for several seconds. However, in the intermittent control described above, electric power larger than desired electric power is input at the time of driving the inverter, and it becomes impossible to always heat the object to be heated at a constant temperature. In the present embodiment, as shown in FIG. 2, since the snubber capacitors 31 and 32 are always connected to the switching circuit 20, a turn-on loss may occur when heating at a low output.
[0032]
(Example 3)
FIG. 7 is a circuit configuration diagram of an electromagnetic induction heating device according to a third embodiment of the present invention. The same parts as those in FIG. 2 are denoted by the same reference numerals, and description thereof will be omitted. 7 is different from the second embodiment in that a snubber capacitor switching circuit 40 in which a snubber capacitor 33 and an IGBT 14 are connected in series is connected between points t and o of the inverter 100, and the snubber capacitor is turned off by turning off the IGBT 14. 33 can be separated from the switching circuit 20. The diode 24 is connected to the IGBT 14 in parallel in the reverse direction, and is used when discharging the charge of the snubber capacitor 33. The IGBT 14 is driven by a drive circuit 63, and the drive circuit 63 is controlled by a control circuit 70.
[0033]
Next, the operation of this embodiment will be described with reference to FIG. In FIG. 8, when the IGBT 12 is turned off at the time of toff, the output capacitance (not shown) between the collector and the emitter of the IGBTs 11 and 12 is discharged and charged by the cutoff current Ioff. Here, even if Ioff is small, the voltage rise of the voltage Vc2 of the IGBT 12 is fast because the output capacitance of the IGBTs 11 and 12 is small, and the time tr required for Vc2 to reach the power supply voltage Vp of the inverter is shorter than the dead time td. Become. When a forward voltage is applied to the parallel diode 21 of the IGBT 11, a return current flows through the diode 21, and the IGBT 11 is turned on at the time ton, which is a return period. Therefore, at the time of turn-on, zero volt switching can be realized, and even when the cutoff current becomes small, the turn-on loss can be eliminated.
[0034]
Therefore, even at the time of low output, the inverter can be driven continuously without stopping the drive of the inverter as in the case of the intermittent control, and the object to be heated can always be heated at a constant temperature.
[0035]
On the other hand, when heating at high output, the capacitor 33 is connected between the point t and the point o of the inverter 100 by turning on the IGBT 14, and the voltage applied to both IGBTs at the time of turn-off as in the second embodiment. Is reduced, and the turn-off loss is suppressed. In FIG. 7, even if the snubber capacitor 33, the IGBT 14, and the diode 24 are connected to the upper arm IGBT 11, the above-described effects can be obtained.
[0036]
(Example 4)
FIG. 9 is a circuit configuration diagram of an electromagnetic induction heating device according to a fourth embodiment of the present invention. The power supply 10 and the switching circuit 20 are omitted, and only the bypass circuit 30 and the resonance circuit 60 are shown. In addition, the same parts as those in FIG. 2 are denoted by the same reference numerals, and description thereof will be omitted.
[0037]
In FIG. 9, a resonance circuit 60 composed of a heating coil 5 and a resonance capacitor 6 is connected between a point p and a point t of the inverter 100. A bypass circuit 30 is connected to the resonance capacitor 6 in parallel. The bypass circuit 30 includes an IGBT 13 connected in series, a diode 25 for blocking a reverse current of the IGBT 13, and a diode 23 connected in parallel to the IGBT 13 in the reverse direction. Have been. Here, assuming that the resonance current IL5 flowing from the point p toward the resonance capacitor 6 is positive, the bypass current Ic3 flowing through the bypass circuit 30 is in one negative direction. In this embodiment, when the input power is reduced, the conduction period of the lower arm is narrowed to extend the conduction period of the upper arm, so that the polarity of the resonance current IL5 is easily inverted from negative to positive during the conduction period of the upper arm.
[0038]
Therefore, by flowing the bypass current Ic3 when the resonance current IL5 flows in the negative direction, it is possible to prevent the polarity of the resonance current IL5 from being inverted from negative to positive. In the present embodiment, the IGBT 13 of the bypass circuit 30 is driven based on the point p, and the drive circuit of the bypass circuit 30 needs to provide a new drive power supply separately from the drive circuit 61 of the switching circuit 20.
[0039]
The operation of this embodiment is the same as that of the second embodiment except that the control of the upper and lower arms of the second embodiment is reversed.
[0040]
(Example 5)
FIG. 10 shows a part of a circuit of an electromagnetic induction heating apparatus according to a fifth embodiment of the present invention. The power supply 10 and the switching circuit 20 are omitted, and only the bypass circuit 30 and the resonance circuit 60 are shown. In addition, the same parts as those in FIG. 2 are denoted by the same reference numerals, and description thereof will be omitted. In FIG. 10, the switching element of the bypass circuit 30 is a reverse current blocking type IGBT 16 having a reverse withstand voltage. The configuration in which the IGBT 13 and the reverse current blocking diode 25 in FIG. Become. In the bypass circuit 30 shown in FIG. 2, losses occur in the IGBT 13 and the diode 25, respectively. However, in the present embodiment, the loss is the loss of the IGBT 16, so that the circuit loss in the inverter 100 is reduced and the conversion efficiency is improved. The operation of this embodiment is the same as that of the second embodiment, and the description is omitted.
[0041]
(Example 6)
FIG. 11 shows a part of a circuit of an electromagnetic induction heating device according to a sixth embodiment of the present invention. The power supply 10 and the switching circuit 20 are omitted, and only the bypass circuit 30 and the resonance circuit 60 are shown. In addition, the same parts as those in FIG. 2 are denoted by the same reference numerals, and description thereof will be omitted. In FIG. 11, a bypass circuit 30 is connected to the resonance capacitor 6 in parallel, and the bypass circuit 30 includes an IGBT 18 and a diode 28 connected in parallel in the opposite direction to the IGBT 18. The difference from the second embodiment is that positive and negative bypass currents flow through the bypass circuit 30. In this embodiment, when lowering the input power, the conduction period of the upper arm is narrowed and the conduction period of the lower arm is extended, so that the polarity of the resonance current IL5 is easily inverted from negative to positive during the conduction period of the lower arm.
[0042]
Therefore, when the resonance current IL5 flows in the negative direction, by flowing the bypass current Ic4 through the diode 28, the polarity of the resonance current IL5 can be prevented from being inverted from negative to positive.
[0043]
Next, the operation of this embodiment will be described with reference to FIG. In FIG. 12, the current flowing in the upper arm of the switching circuit 20 is Ic1, the current flowing in the lower arm is Ic2, the current flowing in the bypass circuit 30 is Ic4, and the resonance current is IL5. The voltage between the collector and the emitter of the upper arm IGBT 11 is Vc1, the voltage between the collector and the emitter of the lower arm IGBT 12 is Vc2, the resonance voltage of the resonance capacitor 6 is Vc4, and the power supply voltage of the inverter is Vp.
[0044]
(Mode 1) In FIG. 12, when the drive signal of the IGBT 18 is turned off while the drive signal of the IGBT 11 is on, the resonance current IL5 flows through the path of the IGBT 11, the heating coil 5, and the resonance capacitor 6, and the resonance capacitor C6 is the resonance current. Charging is started by IL5, and Vc4 increases.
[0045]
(Mode 2) Next, when the drive signal of the IGBT 11 is turned off, the resonance current IL5 has a positive polarity, and this current flows through the path of the snubber capacitor 31, the heating coil 5, the resonance capacitor 6 and the snubber capacitor 32, The coil 5 and the resonance capacitor 6 continue to flow, and the snubber capacitor 31 of the upper arm is charged and the snubber capacitor 32 of the lower arm is discharged. Therefore, the voltage Vc1 between the collector and the emitter of the IGBT 11 gradually increases, and the voltage Vc2 between the collector and the emitter of the IGBT 12, that is, the output voltage gradually decreases. Thereafter, when the voltage of Vc1 reaches the power supply voltage Vp of the inverter and a forward voltage is applied to the diode 22 of the lower arm, the resonance current IL5 flows as a circulating current through the path of the heating coil 5, the resonance capacitor 6, and the diode 22. to continue. During this period, the drive signal of the IGBT 12 is turned on, but continues to flow through the diode 22 as long as the polarity of the current does not change.
[0046]
(Mode 3) Next, when the energy stored in the heating coil 5 becomes zero and the polarity of the resonance current IL5 changes from positive to negative, the IGBT 12 has already been turned on, and the resonance current IL5 changes the voltage Vc4 of the resonance capacitor 6 to the voltage. The flow Vc4 in the path of the heating coil 5 and the IGBT 12 as a source decreases.
[0047]
(Mode 4) When Vc4 decreases and a forward voltage is applied to the diode 28 of the bypass circuit 30, the resonance current IL5 flows through the path of the heating coil 5, the IGBT 12, and the diode 28.
[0048]
(Mode 5) Next, when the drive signal of the IGBT 12 is turned off, the resonance current IL5 has a negative polarity, and this current flows through the diode 28, the heating coil 5, the path of the snubber capacitor 31 and the diode 28, the heating coil 5 , Continue to flow through the path of the snubber capacitor 32, the capacitor 31 of the upper arm is discharged, and the capacitor 32 of the lower arm is charged. Therefore, the voltage Vc1 between the collector and the emitter of the IGBT 11 gradually decreases, and the voltage Vc2 between the collector and the emitter of the IGBT 12, that is, the output voltage gradually increases. Thereafter, when the voltage of Vc2 reaches the power supply voltage Vp of the inverter and a forward voltage is applied to the diode 21 of the upper arm, the resonance current IL5 continues to flow as a circulating current through the path of the diode 28, the heating coil 5, and the diode 21. . The drive signal of the IGBT 18 turns on with the timing of turning off the IGBT 12 as a trigger, but continues to flow through the diode 28 unless the polarity of the resonance current changes. Similarly, the drive signal of the IGBT 11 is turned on during this reflux period, but continues to flow through the diode 21 as long as the polarity of the resonance current does not change.
[0049]
(Mode 6) Next, when the energy stored in the heating coil 5 becomes zero and the polarity of the resonance current IL5 changes from negative to positive, the IGBTs 11 and 18 have already been turned on. It flows on the path of the IGBT 18.
[0050]
As described above, the above operation is performed during one cycle of the resonance current IL5, and thereafter, this operation is repeated. As described above, in the present embodiment, the bypass circuit 30 is provided in parallel with the resonance capacitor 6, and the phase characteristic can be avoided by maintaining the load characteristics inductive. This makes it possible to control the input power by changing the conduction period of the switching circuit 20.
[0051]
Next, a control method of the input power will be described. In FIG. 12, conduction periods of the IGBTs 11, 12, and 18 are represented by t1, t2, and t4, and one cycle of the resonance current, that is, a drive cycle is represented by ts. In FIG. 12, when t4 is controlled, the voltage amplitude value Vcp of the resonance voltage Vc4 changes. In the mode 3 described above, the resonance current IL5 in the negative direction flows through the path of the heating coil 5 and the IGBT 12 using the voltage Vc4 of the resonance capacitor 6 as a voltage source, so that the resonance current IL5 changes according to the magnitude of Vc4. Therefore, input power Pin can be controlled by changing t4. Further, similarly to the second embodiment, it is also possible to control the input power Pin by changing the conduction period of t1 and t2, and to control the conduction period according to the set magnitude of the input power.
[0052]
(Example 7)
FIG. 13 shows a part of a circuit of an electromagnetic induction heating apparatus according to a seventh embodiment of the present invention. The power supply 10 and the switching circuit 20 are omitted, and only the bypass circuit 30 and the resonance circuit 60 are shown. In addition, the same parts as those in FIG. 2 are denoted by the same reference numerals, and description thereof will be omitted. 13 differs from the second embodiment in that the IGBT 15 is connected in parallel with the diode 25 of the bypass circuit 30, so that positive and negative bypass currents can flow through the bypass circuit 30. In the sixth embodiment, positive and negative currents flow through the bypass circuit 30. However, since the current flows through the diode 28 in the negative direction, the current that can be controlled by the IGBT 18 is only the positive direction. In this embodiment, it is possible to control the positive and negative bypass currents by driving the IGBTs 13 and 15, and to control the resonance voltage and the duty of the switching circuit while maintaining the load characteristics of the resonance circuit 60 inductive, thereby controlling the input power. change. The control method of the IGBT 13 is the same as that of the second embodiment, and the IGBT 15 is the same as the IGBT 18 of the sixth embodiment.
[0053]
(Example 8)
FIG. 14 shows a part of a circuit of an electromagnetic induction heating apparatus according to an eighth embodiment of the present invention. The power supply 10 and the switching circuit 20 are omitted, and only the bypass circuit 30 and the resonance circuit 60 are shown. In addition, the same parts as those in FIG. 2 are denoted by the same reference numerals, and description thereof will be omitted. In FIG. 14, the bypass circuit 30 has a configuration in which a reverse current blocking IGBT 16 and an IGBT 17 having a reverse breakdown voltage are connected in parallel in the reverse direction. In the bypass circuit 30 shown in FIG. 13, since a positive current flows through the diode 23 and the IGBT 15, and a negative current flows through the diode 25 and the IGBT 13, a loss occurs in each of the diode and the IGBT. In the present embodiment, since the loss is caused by the IGBT 16 and the IGBT 17, the circuit loss in the inverter 100 is reduced and the conversion efficiency is improved. The operation of this embodiment is the same as that of the seventh embodiment, and the description is omitted.
[0054]
(Example 9)
FIG. 15 shows a part of a circuit of an electromagnetic induction heating apparatus according to a ninth embodiment of the present invention. The power supply 10 and the switching circuit 20 are omitted, and only the bypass circuit 30 and the resonance circuit 60 are shown. In addition, the same parts as those in FIG. 2 are denoted by the same reference numerals, and description thereof will be omitted. In FIG. 15, an IGBT 19 and an auxiliary resonance capacitor 7 connected in series are connected to the resonance capacitor 6 in parallel, and a diode 29 is connected in parallel to the IGBT 19 in the reverse direction. The difference from the second embodiment is that the bypass circuit 30 allows a bypass current to flow through the auxiliary resonance capacitor 7 and the IGBT 19 or the diode 29.
[0055]
Accordingly, when the IGBT 19 is turned on, the resonance capacitor 6 and the auxiliary resonance capacitor 7 are connected in parallel, so that the resonance frequency of the resonance circuit 60 is reduced, and it is easy to maintain inductive load characteristics. Further, even when the object to be heated made of a low-resistance material such as non-magnetic stainless steel or aluminum acts as a load and the equivalent inductance becomes small, the auxiliary resonance capacitor 7 is made conductive to reduce the resonance frequency. The increase can be suppressed, and the inductive load characteristics can be maintained. The control method of the IGBT 13 is the same as that of the second embodiment, and the description is omitted.
[0056]
(Example 10)
FIG. 16 is a block diagram of an electromagnetic induction heating device according to a tenth embodiment of the present invention. 16, inverters 100 and 200 are driven by drive circuits 61, 62, 63 and drive circuits 64, 65, 66, respectively, and each drive circuit is controlled by a control circuit 70. The control circuit 70 includes conduction period setting units 80 and 81 for setting the conduction periods of the switching circuits, bypass circuits and snubber capacitor switching circuits of the inverters 100 and 200, and a variable frequency oscillator 82 for setting the drive frequency of the inverter. In the present invention, when a plurality of inverters are provided to heat different objects to be heated as shown in FIG. 16, each inverter is driven at the same frequency output from the variable frequency oscillator 82 in order to prevent the generation of interference sound. The conduction period setting sections 80 and 81 control the conduction period of the switching circuit, bypass circuit and snubber capacitor switching circuit to control the input power.
[0057]
FIG. 17 shows the resonance current IL5 when the oscillation frequency of the variable frequency oscillator 82, that is, the drive frequency fs is controlled to be substantially constant without fluctuating in this embodiment. In FIG. 17, the amplitude value of IL5 changes in accordance with the power supply voltage of the inverter obtained by rectifying and smoothing the commercial AC voltage Vac, and the envelope of IL5 is substantially similar to the power supply voltage of the inverter. In this case, the noise level radiated from the inverter to the outside has frequency characteristics as shown in FIG. 18, and decreases as the noise level Np at the drive frequency fs becomes the maximum value and becomes higher harmonics.
[0058]
In this embodiment, by changing the oscillation frequency of the variable frequency oscillator 82 and changing the drive frequency fs, the frequency component of noise can be diffused and the noise level can be reduced. FIG. 19 shows the resonance current IL5 when the drive frequency is changed in synchronization with the power supply voltage of the inverter. In FIG. 19, the drive frequency is varied by ± Δfs with respect to fs. The drive frequency is set to be higher when the power supply voltage of the inverter is higher, and to be lower when the voltage is lower. Since the resonance type inverter operates at a drive frequency higher than the resonance frequency, the resonance current decreases when the drive frequency is increased.
[0059]
Therefore, by shifting the drive frequency higher when the power supply voltage of the inverter is high, the resonance current IL5 can be reduced. Assuming that the maximum value of IL5 when the drive frequency is controlled to be constant is Ip, FIG. As shown, the maximum value of IL5 can be reduced from Ip to Id. On the other hand, when the power supply voltage of the inverter is low, the driving frequency is shifted lower, so that the resonance current IL5 can be increased, and it is possible to increase IL5 as compared with the case where the driving frequency is controlled to be constant. In this case, the noise level radiated from the inverter to the outside has a frequency characteristic as shown in FIG. Can be reduced.
[0060]
【The invention's effect】
According to the present invention, by providing a bypass circuit in a resonance capacitor and bypassing a part of the resonance current, the load characteristics of the resonance circuit can be maintained inductive, the input power can be controlled by the Duty control, and the switching can be performed. The occurrence of element loss can be suppressed. Even when a plurality of inverters are driven at the same time, the drive frequency of all the inverters can be made the same, so that an electromagnetic induction heating device that does not generate interference noise can be provided.
[Brief description of the drawings]
FIG. 1 is a block diagram of an electromagnetic induction heating device according to a first embodiment of the present invention.
FIG. 2 is a circuit configuration diagram of an electromagnetic induction heating device according to a second embodiment of the present invention.
FIG. 3 is an operation explanatory diagram in the embodiment of FIG. 2;
FIG. 4 is an explanatory diagram of a control method in the embodiment of FIG.
FIG. 5 is an explanatory diagram of a control method in the embodiment of FIG.
FIG. 6 is an operation explanatory diagram in the embodiment of FIG. 2;
FIG. 7 is a circuit configuration diagram of an electromagnetic induction heating device according to a third embodiment of the present invention.
FIG. 8 is an operation explanatory diagram in the embodiment of FIG. 7;
FIG. 9 is a circuit configuration diagram of an electromagnetic induction heating device according to a fourth embodiment of the present invention.
FIG. 10 is a part of a circuit of an electromagnetic induction heating device according to a fifth embodiment of the present invention.
FIG. 11 is a part of a circuit of an electromagnetic induction heating device according to a sixth embodiment of the present invention.
FIG. 12 is an operation explanatory diagram in the embodiment of FIG. 11;
FIG. 13 shows a part of a circuit of an electromagnetic induction heating apparatus according to a seventh embodiment of the present invention.
FIG. 14 is a part of a circuit of an electromagnetic induction heating device according to an eighth embodiment of the present invention.
FIG. 15 shows a part of a circuit of an electromagnetic induction heating device according to a seventh embodiment of the present invention.
FIG. 16 is a block diagram of an electromagnetic induction heating device according to a tenth embodiment of the present invention.
FIG. 17 is an operation explanatory diagram in the embodiment of FIG. 16;
FIG. 18 is an explanatory diagram in the embodiment of FIG.
FIG. 19 is an operation explanatory diagram in the embodiment of FIG. 16;
FIG. 20 is an explanatory diagram in the embodiment of FIG. 16;
[Explanation of symbols]
DESCRIPTION OF SYMBOLS 1 ... Commercial AC power supply, 2 ... Rectifier circuit, 3 ... Inductor, 4, 6, 7, 31, 32, 33 ... Capacitor, 5 ... Heating coil, 10 ... Power supply, 11, 12, 13, 14, 15, 16, 17, 18, 19: semiconductor switching element, 20: switching circuit, 21, 22, 23, 24, 25, 28, 29: diode, 30: bypass circuit, 40: snubber capacitor switching circuit, 60: resonance circuit, 61, 62, 63, 64, 65, 66 drive circuit, 70 control circuit, 71, 73 current detection element, 72 resonance current detection circuit, 74 input current detection circuit, 75 input power setting section, 80, 81 ... Conduction period setting unit, 82, variable frequency oscillator, 100, 200, inverter.

Claims

A DC power supply, and an inverter circuit that converts a DC voltage supplied from the DC power supply into a high-frequency AC voltage, the inverter circuit includes a switching circuit and a resonance circuit, and the resonance circuit includes a heating coil and a heating coil. Including a resonance capacitor connected in series, connecting the resonance circuit between one of both terminals (p / o) of the DC power supply and an output terminal (t) of a switching circuit, and connecting the resonance circuit in parallel with the resonance capacitor. An electromagnetic induction heating device having a bypass circuit to be connected.

A DC power supply, and an inverter circuit that converts a DC voltage supplied from the DC power supply into a high-frequency AC voltage, the inverter circuit includes a switching circuit and a resonance circuit, and the resonance circuit includes a heating coil and a heating coil. Including a resonance capacitor connected in series, connecting the resonance circuit between one of both terminals (p / o) of the DC power supply and an output terminal (t) of a switching circuit, and connecting the resonance circuit in parallel with the resonance capacitor. Having a bypass circuit to be connected,
A first control method for varying the conduction period of the bypass circuit to control power supplied to the resonance circuit, and a second control method for varying the conduction period of the switching circuit to control power supplied to the resonance circuit. An operation control method for an electromagnetic induction heating device, comprising a control method, wherein at least one of the first and second control methods is selected to control electric power.

In the operation control method of the electromagnetic induction heating device according to claim 2,
First current detecting means for detecting a current flowing through the resonance circuit, second current detecting means for detecting an input current flowing to the DC power supply, and a signal from the first and second current detecting means. A control circuit for determining the material and state of the object to be heated,
An operation control method for an electromagnetic induction heating apparatus, wherein the control circuit controls at least one of the first and second control methods to control electric power.

In any one of claims 1 to 3,
The bypass circuit includes a first switching element and a second diode connected in series, and the first switching element includes a first diode connected in anti-parallel. apparatus.

In any one of claims 1 to 3,
The electromagnetic induction heating device, wherein the bypass circuit includes a first switching element having a reverse withstand voltage function.

In any one of claims 1 to 3,
The electromagnetic induction heating device, wherein the bypass circuit includes a third switching element and a third diode connected in anti-parallel to the third switching element.

In any one of claims 1 to 3,
The bypass circuit includes a first switching element and a second switching element connected in anti-series, and the first and second switching elements include first and second diodes connected in anti-parallel, respectively. An electromagnetic induction heating device, comprising:

In any one of claims 1 to 3,
The electromagnetic induction heating device, wherein the bypass circuit includes first and second switching elements having a reverse withstand voltage function connected in anti-parallel.

In any one of claims 1 to 8,
The electromagnetic induction heating apparatus, wherein the bypass circuit includes an auxiliary resonance capacitor connected in series and a fourth switching element, and has a fourth diode connected in anti-parallel to the fourth switching element. .

In any one of claims 1 to 9,
The switching circuit includes fifth and sixth switching elements connected in series, and the fifth and sixth switching elements include fifth and sixth diodes connected in anti-parallel, respectively. And an electromagnetic induction heating device.

The electromagnetic induction heating device according to claim 10,
The electromagnetic induction heating device, wherein the switching circuit includes a first snubber capacitor in parallel with at least one of the fifth and sixth switching elements.

The electromagnetic induction heating device according to claim 10,
The switching circuit includes a snubber capacitor switching circuit in parallel with at least one of the fifth and sixth switching elements, and the snubber capacitor switching circuit includes a second snubber capacitor and a seventh switching element connected in series. An electromagnetic induction heating device, comprising: a seventh diode, wherein the seventh switching element includes a seventh diode connected in anti-parallel.

The electromagnetic induction heating device according to claim 12,
A first drive circuit that drives a switching element of the switching circuit, a second drive circuit that drives a switching element of the bypass circuit, and a third drive circuit that drives a switching element of the snubber capacitor switching circuit, An input power setting unit for setting power to be supplied to the object to be heated, wherein the control circuit controls the first, second, and third drive circuits according to an output signal from the input power setting unit. An electromagnetic induction heating device characterized by the following.

In any one of claims 3 to 13,
The control circuit includes a variable frequency oscillator capable of changing an oscillating frequency, and at least one conduction period setting unit that sets a conduction period of each switching element, and at least one inverter is output from the variable frequency oscillator. An electromagnetic induction heating device driven at an oscillation frequency.

The electromagnetic induction heating device according to claim 14,
The variable frequency oscillator changes an oscillating frequency in accordance with a voltage of the DC power supply.

A DC power supply, and an inverter circuit that converts a DC voltage supplied from the DC power supply into a high-frequency AC voltage, the inverter circuit includes a switching circuit and a resonance circuit, and the resonance circuit includes a heating coil and a heating coil. Including a resonance capacitor connected in series, the resonance circuit is connected between one of both terminals (p / o) of the DC power supply and an output terminal (t) of a switching circuit, and supplied to the resonance circuit. A control circuit that controls power, the control circuit includes a variable frequency oscillator capable of changing an oscillating frequency, and at least one conduction period setting unit that sets a conduction period of each switching element of the switching circuit. An electromagnetic induction heating device, wherein one inverter is driven at an oscillation frequency output from the variable frequency oscillator.

The electromagnetic induction heating device according to claim 16,
The variable frequency oscillator changes an oscillating frequency in accordance with a voltage of the DC power supply.

A DC power supply, and an inverter circuit that converts a DC voltage supplied from the DC power supply into a high-frequency AC voltage, the inverter circuit includes a switching circuit and a resonance circuit, and the resonance circuit includes a heating coil and a heating coil. A resonance capacitor connected in series, the resonance circuit being connected between one of both terminals (p / o) of the DC power supply and an output terminal (t) of a switching circuit, and a resonance frequency of the resonance circuit being Wherein the voltage of the resonance capacitor is controlled so as to make the driving frequency lower than the driving frequency of the inverter.