JP2010063324A - Induced power transmission circuit - Google Patents

Induced power transmission circuit Download PDF

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JP2010063324A
JP2010063324A JP2008229171A JP2008229171A JP2010063324A JP 2010063324 A JP2010063324 A JP 2010063324A JP 2008229171 A JP2008229171 A JP 2008229171A JP 2008229171 A JP2008229171 A JP 2008229171A JP 2010063324 A JP2010063324 A JP 2010063324A
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antenna
circuit
inductive
power
power transmission
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Hideo Kikuchi
秀雄 菊地
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Kikuchi Hideo
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Abstract

<P>PROBLEM TO BE SOLVED: To obtain a parameter of a circuit element for transmitting an AC power having an angular frequency ω from a transmission antenna connected to a power supply circuit to a reception antenna at a distance with excellent efficiency and transmitting it to a load circuit, without trial and error. <P>SOLUTION: The induced power transmission circuit includes: a circuit whose ends are connected by a capacitor C1, and in which a power supply circuit is connected in series to the transmission antenna of an effective self-inductance L1; and a circuit whose ends are connected by a capacitor C2, and in which a load circuit is connected in series to the reception antenna of an effective self-inductance L2. The distance between the transmission antenna and the reception antenna is set to be equal to or less than 1/2π of the wavelength of an electromagnetic field, in which power is transmitted. Regarding a phase angle β having a value from 0 to π radians, the angular frequency ω is set to the square root of the reciprocal of a value of L2×C2×(1+k×cos(β)), the output impedance of the power supply circuit is set to approximately kωL1×sin(β)≡r1, and the input impedance of the load circuit is set to approximately kωL2×sin(β)≡r2. <P>COPYRIGHT: (C)2010,JPO&INPIT

Description

本発明は、電力を無線誘導手段を介して空間を越えて給電する誘導電力伝送回路に関する。   The present invention relates to an inductive power transmission circuit that feeds electric power across a space via wireless induction means.

従来、特許文献１で、電動車などへの応用を見込んで、電源装置の一次巻線から被給電装置の二次巻線に非接触で電力を供給する誘導電力伝送回路が提案されていた。この誘導電力伝送回路では、電気端子を接触させないので、電気端子の接点の接触不良が発生しない利点がある。ここで、誘導電力伝送回路の電源装置の一次巻線に電流を流して電磁界を発生させ、その電磁界が被給電装置の二次巻線に電磁誘導させて電力を伝達するが、特に、二次巻線の両端にコンデンサを接続して二次巻線のインダクタンスとコンデンサの容量とによる共振回路を形成し、その共振回路の共振を利用して二次巻線が電力を受け取る電力を大きくしていた。そして、その共振回路に並列に被給電装置を接続し、被給電装置に電力を供給する効率を向上させていた。また、この種の誘導電力伝送回路の応用製品として、歯ブラシや携帯電話などに非接触で電力を伝送するシステムが実用化されている。特許文献１では、一次巻線と被給電装置の二次巻線との距離は変えずに一定に保って電力を伝送し、歯ブラシや携帯電話では被給電装置をホルダーに設置して所定の一定位置に保持して電力を伝送していた。しかし、特許文献１の構成では、一次巻線と二次巻線の間隔が大きくなり一次巻線と二次巻線の相互インダクタンスＭが密結合な相互誘導回路の相互インダクタンスより小さくなると電力の伝送効率が悪くなり、伝送できる電力の大きさが小さくなってしまう問題があった。   Conventionally, Patent Document 1 has proposed an induction power transmission circuit that supplies power in a non-contact manner from a primary winding of a power supply device to a secondary winding of a power-supplied device in anticipation of application to an electric vehicle or the like. In this inductive power transmission circuit, since the electrical terminals are not brought into contact with each other, there is an advantage that contact failure of the contacts of the electrical terminals does not occur. Here, a current flows through the primary winding of the power supply device of the induction power transmission circuit to generate an electromagnetic field, and the electromagnetic field electromagnetically induces the secondary winding of the power-supplied device to transmit power. A capacitor is connected to both ends of the secondary winding to form a resonance circuit with the inductance of the secondary winding and the capacitance of the capacitor. The resonance of the resonance circuit is used to increase the power received by the secondary winding. Was. And the power supply apparatus was connected in parallel with the resonance circuit, and the efficiency which supplies electric power to the power supply apparatus was improved. In addition, as an application product of this type of inductive power transmission circuit, a system for transmitting power in a contactless manner to a toothbrush or a mobile phone has been put into practical use. In Patent Document 1, power is transmitted while keeping the distance between the primary winding and the secondary winding of the power-supplied device constant, and in a toothbrush or a mobile phone, the power-supplied device is installed in a holder and a predetermined constant The power was transmitted in the position. However, in the configuration of Patent Document 1, when the distance between the primary winding and the secondary winding becomes large and the mutual inductance M between the primary winding and the secondary winding becomes smaller than the mutual inductance of the tightly coupled mutual induction circuit, electric power is transmitted. There is a problem in that the efficiency is lowered and the amount of power that can be transmitted is reduced.

特許文献２では、被給電装置であるＩＣカードの二次巻線の位置を、リーダーライタの電源装置の一次巻線から空間を隔てた遠隔位置に置き、その一次巻線と二次巻線の間隔を一定位置に保持せずに、リーダーライタの電源装置からＩＣカードの被給電装置（遠隔装置）に電力を供給する技術が開示されていた。その被給電装置の二次巻線の両端にコンデンサを接続し、二次巻線のインダクタンスとコンデンサとで共振回路を構成して受信する電力を大きくしていた。そして、一次巻線と二次巻線の間の距離が変わると電源装置から遠隔装置に供給される電力が変動する問題を解決するために、電源装置を電源回路と整合回路と一次巻線で構成して電力を伝送した。また、ＩＣカードの遠隔装置には、電力を受信する二次巻線と容量による共振回路を構成し、その共振回路に並列に可変インピーダンス回路を接続し、その先に整流回路から成る誘起電圧発生部を接続し、その先にＩＣチップの負荷回路を接続した。そして、負荷回路に加わる電圧を検出してその電圧を安定させるべく可変インピーダンス回路を調整した。この構成により、負荷回路へ供給する電力を安定させた。しかし、電源回路から供給される電力のうち、負荷回路に供給される電力以外の電力は無駄に消費され、電源回路から負荷回路までの電力の伝送効率が良くない問題があった。   In Patent Document 2, the position of the secondary winding of the IC card, which is a power-supplied device, is placed at a remote position separated from the primary winding of the power supply device of the reader / writer, and the primary winding and the secondary winding There has been disclosed a technique for supplying power from a power supply device of a reader / writer to a power-supplied device (remote device) of an IC card without holding the interval at a fixed position. A capacitor is connected to both ends of the secondary winding of the power-supplied device, and a resonance circuit is constituted by the inductance of the secondary winding and the capacitor to increase the received power. In order to solve the problem that the power supplied from the power supply device to the remote device fluctuates when the distance between the primary winding and the secondary winding changes, the power supply device is composed of a power supply circuit, a matching circuit, and a primary winding. Configured and transmitted power. In addition, the remote device of the IC card is configured with a secondary circuit that receives power and a resonant circuit with a capacity, and a variable impedance circuit is connected in parallel with the resonant circuit, and an induced voltage is generated ahead of the rectifier circuit. The IC chip load circuit was connected to the end. Then, the variable impedance circuit was adjusted in order to detect the voltage applied to the load circuit and stabilize the voltage. With this configuration, the power supplied to the load circuit is stabilized. However, of the power supplied from the power supply circuit, power other than the power supplied to the load circuit is consumed wastefully, and there is a problem that the power transmission efficiency from the power supply circuit to the load circuit is not good.

特許文献３では、特許文献２と同様にＩＣカードにおいて、検出手段で電力伝送効率を検出して、２つのコンデンサの容量を変化させるか、コンデンサとインダクタンスのパラメータを変化させるインピーダンス可変手段によるか、あるいは、同様な２つの回路素子のパラメータを変化させるインピーダンス可変手段により調整してリーダーライタからＩＣカードまでの電力の伝送効率を良くした。   In Patent Document 3, as in Patent Document 2, in the IC card, whether the power transmission efficiency is detected by the detection means and the capacitance of the two capacitors is changed, or the impedance variable means for changing the parameters of the capacitor and the inductance is used. Alternatively, the power transmission efficiency from the reader / writer to the IC card is improved by adjusting the impedance variable means for changing the parameters of the two similar circuit elements.

特許文献４では、特許文献２及び３と同様に、電源装置から物理的に大きな空間を隔てて配置された遠隔装置へ電力を伝送し、遠隔装置から、そのエネルギー受信状況を、電源装置側に通知し、その情報により電源装置が電力の供給を調整していた。すなわち、電源装置のコンデンサとインダクタンスとの２つの回路素子のパラメータを、パラメータ可変手段により変えることで、電源装置から遠隔装置に高い電力伝送効率で電力を供給していた。   In Patent Document 4, similarly to Patent Documents 2 and 3, power is transmitted from a power supply device to a remote device that is physically separated from a large space, and the energy reception status is transmitted from the remote device to the power supply device side. The power supply device adjusted the supply of power according to the information. That is, power is supplied from the power supply device to the remote device with high power transmission efficiency by changing the parameters of the two circuit elements of the capacitor and inductance of the power supply device by the parameter variable means.

以下に公知文献を記す。
特表平６−５０６０９９号公報 特開平１０−１４５９８７号公報 特開２００１−２３８３７２号公報 特表２００６−５１７７７８号公報 The known literature is described below.
JP-T 6-506099 Japanese Patent Laid-Open No. 10-145987 JP 2001-238372 A JP-T-2006-517778

特許文献２から４では、一次巻線と二次巻線の間隔が変化する場合に対応して電力を伝送し、遠隔装置の二次巻線（あるいはスパイラルアンテナ）の両端にコンデンサを接続して二次巻線（スパイラルアンテナ）のインダクタンスとコンデンサから成る共振回路を共振させることで、一次巻線から二次巻線への電力の伝送効率を高くしていた。それらは、二次巻線の両端にコンデンサを接続した共振回路に並列に負荷回路を接続して電力を受信していた。そのうち、特許文献３では、リーダーライタからＩＣカードまで、一定の電力を効率良く伝送するために、２つの回路素子のパラメータを変えて共振回路のインピーダンスを調整することで効率良く電力を伝送できる技術が開示されていた。しかし、特許文献３では、リーダーライタからＩＣカードまでの距離を変えた場合に電力の伝送の良い効率を維持するには、どのように２つの回路素子パラメータを調整したらよいかが開示されておらず、試行錯誤して調整する必要があった。特許文献４も同様に、電源装置のコンデンサとインダクタンスとの２つの回路素子のパラメータを試行錯誤して調整する必要があった。   In Patent Documents 2 to 4, electric power is transmitted in response to a change in the interval between the primary winding and the secondary winding, and a capacitor is connected to both ends of the secondary winding (or spiral antenna) of the remote device. By resonating a resonance circuit composed of an inductance and a capacitor of the secondary winding (spiral antenna), the power transmission efficiency from the primary winding to the secondary winding is increased. They received power by connecting a load circuit in parallel with a resonant circuit having capacitors connected to both ends of the secondary winding. Among them, in Patent Document 3, in order to efficiently transmit constant power from a reader / writer to an IC card, a technology that can efficiently transmit power by adjusting the impedance of the resonance circuit by changing the parameters of two circuit elements. Has been disclosed. However, Patent Document 3 does not disclose how to adjust two circuit element parameters in order to maintain good power transmission efficiency when the distance from the reader / writer to the IC card is changed. It was necessary to adjust by trial and error. Similarly, in Patent Document 4, it is necessary to adjust parameters of two circuit elements, that is, a capacitor and an inductance of the power supply device, by trial and error.

そのため、本発明の第１の目的は、電源装置の電源回路からの電力を、前記電源回路に接続した送信アンテナから受信アンテナまでの空間を伝送し、前記受信アンテナに接続した負荷回路で電力を消費する誘導電力伝送回路において、回路素子のパラメータを試行錯誤せずに調整でき、電力を高い伝送効率で伝送できる誘導電力伝送回路を得ることにある。   Therefore, a first object of the present invention is to transmit power from a power supply circuit of a power supply device through a space from a transmission antenna to a reception antenna connected to the power supply circuit, and to supply power by a load circuit connected to the reception antenna. In the inductive power transmission circuit to be consumed, an object is to obtain an inductive power transmission circuit capable of adjusting the parameters of circuit elements without trial and error and transmitting power with high transmission efficiency.

また、特許文献３と特許文献４では、一次巻線と二次巻線の間隔の変動に対応して電力の伝送効率を高い効率に維持しようとすると、同時に２つの回路素子のパラメータを適切な値に調整することでインピーダンスを整合しなければならず、前記２つの回路素子のパラメータが共に適切な値に設定しないとインピーダンスが整合せず、調整が難しい問題があった。そのため、本発明の第２の目的は、１つの回路素子のパラメータの調整のみで電源回路から負荷回路まで空間を隔てて電力を効率良く伝送できる誘導電力伝送回路を得ることにある。   Further, in Patent Document 3 and Patent Document 4, if it is attempted to maintain high power transmission efficiency in response to fluctuations in the distance between the primary winding and the secondary winding, the parameters of the two circuit elements are simultaneously set appropriately. The impedance must be matched by adjusting the value, and unless both the parameters of the two circuit elements are set to appropriate values, the impedance is not matched and there is a problem that adjustment is difficult. Therefore, a second object of the present invention is to obtain an inductive power transmission circuit capable of efficiently transmitting power across a space from a power supply circuit to a load circuit only by adjusting parameters of one circuit element.

この課題を解決するために鋭意研究の結果、電源回路及び負荷回路のインピーダンスをアンテナに合わせたある特定の抵抗値（誘導抵抗）まで下げることで効率良く電力を伝送できることを見出した。その誘導抵抗は、アンテナに誘導される電圧をそのアンテナ電流で割り算した値であり、それぞれのアンテナの誘導抵抗に、それぞれのアンテナに接続する電源回路及び負荷回路のインピーダンスを整合させれば良い効率で電力を伝送できることを見出し、本発明に至った。   As a result of earnest research to solve this problem, it was found that power can be efficiently transmitted by lowering the impedance of the power supply circuit and the load circuit to a specific resistance value (inductive resistance) matched to the antenna. The inductive resistance is a value obtained by dividing the voltage induced in the antenna by the antenna current, and it is sufficient that the impedance of each power supply circuit and load circuit connected to each antenna is matched to the inductive resistance of each antenna. It was found that power can be transmitted with this method, and the present invention has been achieved.

すなわち、本発明は、電力を電源回路に接続した送信アンテナから角周波数ωの交流電力を空間を隔てた受信アンテナに伝送し負荷回路に伝送する誘導電力伝送回路であって、両端を容量Ｃ１でつないだ、実効的自己インダクタンスがＬ１の送信アンテナに電源回路を直列に接続した回路と、両端を容量Ｃ２でつないだ、実効的自己インダクタンスがＬ２の受信アンテナに負荷回路を直列に接続した回路を有し、前記送信アンテナと前記受信アンテナの間の距離を、電力を伝送する電磁界の波長の２π分の１以下の距離にして前記送信アンテナと前記受信アンテナの電磁誘導の結合係数をｋにし、０ラジアン以上πラジアン以下の値の位相角βに関して、前記角周波数ωを、Ｌ２×Ｃ２×（１＋ｋ・ｃｏｓ（β））の値の逆数の平方根にして、前記電源回路の出力インピーダンスを約ｋωＬ１・ｓｉｎ（β）≡ｒ１にし、前記負荷回路の入力インピーダンスを約ｋωＬ２・ｓｉｎ（β）≡ｒ２にして前記電源回路から前記負荷回路に効率良く電力を伝送することを特徴とする誘導電力伝送回路である。   That is, the present invention is an inductive power transmission circuit that transmits AC power of angular frequency ω from a transmitting antenna connected to a power supply circuit to a receiving antenna separated by a space and transmits it to a load circuit, with both ends having a capacitance C1. A circuit in which a power supply circuit is connected in series to a transmitting antenna having an effective self-inductance L1, and a circuit in which a load circuit is connected in series to a receiving antenna having an effective self-inductance L2 in which both ends are connected by a capacitor C2. And the distance between the transmitting antenna and the receiving antenna is set to a distance equal to or less than 1 / 2π of the wavelength of the electromagnetic field transmitting power, and the coupling coefficient of electromagnetic induction between the transmitting antenna and the receiving antenna is set to k. , With respect to the phase angle β having a value not less than 0 radians and not more than π radians, the angular frequency ω is defined as the square root of the reciprocal of the value of L2 × C2 × (1 + k · cos (β)). The output impedance of the power supply circuit is set to about kωL1 · sin (β) ≡r1 and the input impedance of the load circuit is set to about kωL2 · sin (β) ≡r2 to efficiently transmit power from the power supply circuit to the load circuit. An inductive power transmission circuit characterized by the above.

また、本発明は、上記の誘導電力伝送回路において、前記電源回路と前記送信アンテナの回路を、前記送信アンテナに第１の誘導結合配線が相互インダクタンスＭ１で誘導結合し、前記第１の誘導結合配線の両端に元の電源回路を接続した回路に代え、前記元の電源回路の出力インピーダンスを約（２πｆ×Ｍ１）２／ｒ１にしたことを特徴とする誘導電力伝送回路である。 Further, according to the present invention, in the inductive power transmission circuit, the power supply circuit and the transmission antenna circuit are inductively coupled to the transmission antenna by a first inductive coupling wiring with a mutual inductance M1. Instead of a circuit in which the original power supply circuit is connected to both ends of the wiring, the output power impedance of the original power supply circuit is about (2πf × M1) 2 / r1.

また、本発明は、上記の誘導電力伝送回路において、前記負荷回路と前記受信アンテナの回路を、前記受信アンテナに第２の誘導結合配線が相互インダクタンスＭ２で誘導結合し、前記第２の誘導結合配線の両端に元の負荷回路を接続した回路に代え、前記元の負荷回路の入力インピーダンスを約（２πｆ×Ｍ２）２／ｒ２にしたことを特徴とする誘導電力伝送回路である。 Further, according to the present invention, in the inductive power transmission circuit, the load circuit and the receiving antenna circuit are inductively coupled to the receiving antenna by a second inductive coupling wiring with a mutual inductance M2, and the second inductive coupling is provided. Instead of a circuit in which the original load circuit is connected to both ends of the wiring, the inductive power transmission circuit is characterized in that the input impedance of the original load circuit is about (2πf × M2) 2 / r2.

また、本発明は、上記の誘導電力伝送回路において、前記第１の誘導結合配線を前記送信アンテナが共用し、前記元の電源回路の出力インピーダンスを約（２πｆ×Ｌ１）２／ｒ１にしたことを特徴とする誘導電力伝送回路である。 In the inductive power transmission circuit according to the present invention, the first inductive coupling wiring is shared by the transmitting antenna, and the output impedance of the original power supply circuit is about (2πf × L1) 2 / r1. An inductive power transmission circuit characterized by the above.

また、本発明は、上記の誘導電力伝送回路において、前記第２の誘導結合配線を前記受信アンテナが共用し、前記元の負荷回路の入力インピーダンスを約（２πｆ×Ｌ２）２／ｒ２にしたことを特徴とする誘導電力伝送回路である。 Further, according to the present invention, in the inductive power transmission circuit, the second inductive coupling wiring is shared by the receiving antenna, and the input impedance of the original load circuit is about (2πf × L2) 2 / r2. An inductive power transmission circuit characterized by the above.

また、本発明は、上記の誘導電力伝送回路において、前記電源回路と前記送信アンテナと前記容量Ｃ１の回路を、空間から電磁波を受け取るアンテナに代えたことを特徴とする誘導電力伝送回路である。   The present invention is the inductive power transmission circuit according to the above-described inductive power transmission circuit, wherein the power supply circuit, the transmission antenna, and the capacitor C1 are replaced with antennas that receive electromagnetic waves from space.

また、本発明は、上記の誘導電力伝送回路において、前記負荷回路と前記受信アンテナと前記容量Ｃ２の回路を、空間に電磁波を放射するアンテナに代えたことを特徴とする誘導電力伝送回路である。   The present invention is the inductive power transmission circuit according to the above-described inductive power transmission circuit, wherein the load circuit, the receiving antenna, and the capacitor C2 are replaced with an antenna that radiates electromagnetic waves in space. .

また、本発明は、電源回路に接続した送信アンテナから角周波数ωの交流電力を空間を隔てた受信アンテナに伝送し負荷回路に伝送する誘導電力伝送回路であって、両端を容量Ｃ１でつないだ、実効的自己インダクタンスがＬ１の送信アンテナに電源回路を直列に接続した回路と、両端を容量Ｃ２でつないだ、実効的自己インダクタンスがＬ２の受信アンテナに負荷回路を直列に接続した回路を有し、前記送信アンテナと前記受信アンテナの間の距離を、電力を伝送する電磁界の波長の２π分の１以下にして相互インダクタンスをＭにし、前記角周波数ωの値をＬ２×Ｃ２の逆数の平方根の値にし、前記電源回路の出力インピーダンスＺ１を、前記負荷回路側での誘導インピーダンス値（ωＭ）２／Ｚ１に変換し、前記誘導インピーダンス値に前記負荷回路の入力入力インピーダンスを等しくして電力を伝送することを特徴とする誘導電力伝送回路である。 The present invention is also an inductive power transmission circuit that transmits AC power of angular frequency ω from a transmitting antenna connected to a power supply circuit to a receiving antenna spaced apart and transmits it to a load circuit, with both ends connected by a capacitor C1. A circuit in which a power supply circuit is connected in series to a transmission antenna having an effective self-inductance of L1, and a circuit in which a load circuit is connected in series to a receiving antenna having an effective self-inductance of L2 connected by a capacitor C2. The distance between the transmitting antenna and the receiving antenna is set to be equal to or less than 1 / 2π of the wavelength of the electromagnetic field transmitting power, the mutual inductance is set to M, and the value of the angular frequency ω is the square root of the reciprocal of L2 × C2. the value, the output impedance Z1 of the power supply circuit, and converts the inductive impedance values (ωM) 2 / Z1 in the load circuit side, the inductive impedance value And equal input input impedance of the serial load circuit is an inductive electric power transfer circuit, characterized in that transmitting power.

また、本発明は、上記の誘導電力伝送回路において、前記電源回路と前記送信アンテナの回路を、前記送信アンテナに第１の誘導結合配線が相互インダクタンスＭ１で誘導結合し、前記第１の誘導結合配線の両端に元の電源回路を接続した回路に代え、前記元の電源回路の出力インピーダンスＺ３を、前記負荷回路側での誘導抵抗値（Ｍ／Ｍ１）２×Ｚ３に変換することを特徴とする誘導電力伝送回路である。 Further, according to the present invention, in the inductive power transmission circuit, the power supply circuit and the transmission antenna circuit are inductively coupled to the transmission antenna by a first inductive coupling wiring with a mutual inductance M1. Instead of a circuit in which the original power supply circuit is connected to both ends of the wiring, the output impedance Z3 of the original power supply circuit is converted into an inductive resistance value (M / M1) 2 × Z3 on the load circuit side. Inductive power transmission circuit.

また、本発明は、上記の誘導電力伝送回路において、前記負荷回路と前記受信アンテナの回路を、前記受信アンテナに第２の誘導結合配線が相互インダクタンスＭ２で誘導結合し、前記第２の誘導結合配線の両端に元の負荷回路を接続した回路に代え、前記電源回路の出力インピーダンスＺ１を、前記元の負荷回路側での誘導抵抗値（Ｍ２／Ｍ）２×Ｚ１に変換することを特徴とする誘導電力伝送回路である。 Further, according to the present invention, in the inductive power transmission circuit, the load circuit and the receiving antenna circuit are inductively coupled to the receiving antenna by a second inductive coupling wiring with a mutual inductance M2, and the second inductive coupling is provided. Instead of a circuit in which the original load circuit is connected to both ends of the wiring, the output impedance Z1 of the power supply circuit is converted into an inductive resistance value (M2 / M) 2 × Z1 on the original load circuit side. Inductive power transmission circuit.

また、本発明は、上記の誘導電力伝送回路において、前記第１の誘導結合配線を前記送信アンテナが共用し、前記元の電源回路の出力インピーダンスＺ３を、前記負荷回路側での誘導抵抗値（Ｍ／Ｌ１）２×Ｚ３に変換することを特徴とする誘導電力伝送回路である。 Further, according to the present invention, in the above-described inductive power transmission circuit, the first inductive coupling wiring is shared by the transmission antenna, and the output impedance Z3 of the original power supply circuit is converted to an inductive resistance value ( M / L1) An inductive power transmission circuit characterized by converting to 2 × Z3.

また、本発明は、上記の誘導電力伝送回路において、前記第２の誘導結合配線を前記受信アンテナが共用し、前記電源回路の出力インピーダンスＺ１を、前記元の負荷回路側での誘導抵抗値（Ｌ２／Ｍ）２×Ｚ１に変換することを特徴とする誘導電力伝送回路である。 Further, according to the present invention, in the above-described inductive power transmission circuit, the second inductive coupling wiring is shared by the receiving antenna, and the output impedance Z1 of the power circuit is changed to an inductive resistance value ( L2 / M) 2 × Z1 is an inductive power transmission circuit characterized by conversion.

本発明は、電源回路に接続した送信アンテナから角周波数ωの交流電力を空間を隔てた受信アンテナに伝送し負荷回路に伝送する誘導電力伝送回路であり、電源回路及び負荷回路のインピーダンスをアンテナに合わせたある特定の抵抗値（誘導抵抗）まで下げることで効率良く電力を伝送できる効果がある。その誘導抵抗は本発明により容易に計算でき、電源回路から負荷回路まで電力を完全な効率で伝送できる誘導電力伝送回路が得られる効果がある。また、本発明は、空芯コイルの構成で誘導電力伝送回路のインピーダンスを容易に変換できるインピーダンス変換回路が得られる効果がある。   The present invention is an inductive power transmission circuit that transmits AC power of an angular frequency ω from a transmitting antenna connected to a power circuit to a receiving antenna that is separated from the space, and transmits the power to the load circuit. The impedance of the power circuit and the load circuit is used as the antenna. There is an effect that electric power can be efficiently transmitted by lowering to a certain specific resistance value (inductive resistance). The inductive resistance can be easily calculated by the present invention, and there is an effect that an inductive power transmission circuit capable of transmitting power from the power supply circuit to the load circuit with complete efficiency can be obtained. In addition, the present invention has an effect of obtaining an impedance conversion circuit that can easily convert the impedance of the induction power transmission circuit with an air-core coil configuration.

＜本発明の原理＞
図１（ａ）に、本発明の誘導電力伝送回路の送信アンテナ１と受信アンテナ２の平面図（ＸＹ図）を示し、図１（ｂ）に側面図を示す。すなわち、本発明の誘導電力伝送回路は、実効的自己インダクタンスＬ１のコイル状の送信アンテナ１のリボン状の銅の配線の両端を容量Ｃ１でつないだ共振回路を作り、その共振回路のアンテナの配線の中間に直列に電源回路３を接続する。同様に、実効的自己インダクタンスＬ２のコイル状の受信アンテナ２のリボン状の銅の配線の両端を容量Ｃ２でつないだ共振回路を作り、その共振回路のアンテナの配線の中間に直列に負荷回路４を接続する。図１のように、互いに電磁誘導する両アンテナを（アンテナが共振する電磁界の波長）／（２π）以下の距離の近傍に近づける。この距離以下の近傍に近づけることで、両アンテナは、電波を放射しない電磁誘導の相互作用が優勢になる。また、その送信アンテナ１の配線と受信アンテナ２の配線を少し離すと、両アンテナの配線の相互インダクタンスＭは、密結合な相互誘導回路の相互インダクタンス√（Ｌ１×Ｌ２）の６割以下になる。 <Principle of the present invention>
FIG. 1 (a) shows a plan view (XY diagram) of the transmitting antenna 1 and the receiving antenna 2 of the inductive power transmission circuit of the present invention, and FIG. 1 (b) shows a side view. That is, the inductive power transmission circuit of the present invention creates a resonance circuit in which both ends of the ribbon-shaped copper wiring of the coiled transmission antenna 1 having an effective self-inductance L1 are connected by the capacitor C1, and the antenna wiring of the resonance circuit is formed. Is connected in series with the power supply circuit 3. Similarly, a resonance circuit is formed by connecting both ends of the ribbon-shaped copper wiring of the coil-shaped receiving antenna 2 of the effective self-inductance L2 with the capacitor C2, and the load circuit 4 is connected in series between the antenna wiring of the resonance circuit in series. Connect. As shown in FIG. 1, both antennas that electromagnetically induce each other are brought close to a distance of (wavelength of electromagnetic field at which the antenna resonates) / (2π) or less. By approaching the vicinity of this distance or less, the interaction of electromagnetic induction that does not radiate radio waves becomes dominant in both antennas. Further, when the wiring of the transmitting antenna 1 and the wiring of the receiving antenna 2 are slightly separated, the mutual inductance M of the wirings of both antennas becomes 60% or less of the mutual inductance √ (L1 × L2) of the tightly coupled mutual induction circuit. .

図１では、送信アンテナ１のコイル状の配線が乗るアンテナの面（ＸＹ面）と受信アンテナ２のアンテナの面を平行にし、送信アンテナ１と受信アンテナ２のコイル状のアンテナの配線の中心軸を一致させて、アンテナ同士をその中心軸の方向に近づけて設置したが、両アンテナの配置は、（アンテナが共振する電磁界の波長）／（２π）以下の距離の近傍に設置するだけで良く、後の図２０（ａ）に示すように両アンテナを同一平面上に並べて配置しても良い。また、アンテナの形状は、アンテナの配線の両端を容量でつなぐだけで良く、アンテナの配線の巻数は１巻でも多数巻きのコイル状又は螺旋状でも良い。アンテナ配線の両端を結ぶ容量Ｃ１あるいはＣ２は、コンデンサを接続しないでも、アンテナ配線の両端を開放して両端間に発生する寄生容量でアンテナ配線の両端をつなぐだけの構成でも良い。アンテナの形状及び寸法は送信アンテナ１と受信アンテナ２で異なっていても良い。更に、送信アンテナ１と受信アンテナ２は、図１のようなコイル状に形成しないで、例えば、ダイポールアンテナでも、共鳴させることで電力を完全な効率（後に説明する式２９で与えられる電力伝送効率Ｐｅで電力を伝送できることを完全な効率と呼ぶ）で伝送できると考える。   In FIG. 1, the antenna surface (XY plane) on which the coiled wiring of the transmitting antenna 1 rides and the antenna surface of the receiving antenna 2 are parallel, and the central axis of the coiled antenna wiring of the transmitting antenna 1 and the receiving antenna 2 The antennas are placed close to each other in the direction of the central axis, but the arrangement of both antennas can be done only by placing them near a distance of (wavelength of the electromagnetic field at which the antenna resonates) / (2π) or less. Alternatively, both antennas may be arranged side by side on the same plane as shown in FIG. The antenna may be formed by simply connecting both ends of the antenna wiring with a capacitance, and the number of turns of the antenna wiring may be one or many coils or a spiral. The capacitor C1 or C2 connecting both ends of the antenna wiring may have a configuration in which both ends of the antenna wiring are connected by a parasitic capacitance generated between both ends by opening both ends of the antenna wiring without connecting a capacitor. The shape and dimensions of the antenna may be different between the transmitting antenna 1 and the receiving antenna 2. Further, the transmitting antenna 1 and the receiving antenna 2 are not formed in a coil shape as shown in FIG. 1, and for example, even a dipole antenna resonates to achieve complete efficiency (power transmission efficiency given by Equation 29 described later). It is considered that transmission of power with Pe is called perfect efficiency).

本発明の誘導電力伝送回路は、図２（ａ）の回路図でモデル化でき、その回路は、実効的自己インダクタンスＬ１の送信アンテナ１と実効的自己インダクタンスＬ２の受信アンテナ２が、相互インダクタンスＭを介して、送信アンテナ１に接続した電源回路３から受信アンテナ２に接続した負荷回路４まで非接触で電力を供給する誘導電力伝送回路である。ここで、送信アンテナ１の両端をつなぐ容量Ｃ１は、電源回路３から角周波数ωの電流を送信アンテナ１に供給するタンク回路における容量で構成しても良い。また、送信アンテナ１に直列に接続する電源回路３の端子（ポート１）は、送信アンテナ１の配線の中間に設置しても良いし、寄生容量に比べて大きな容量Ｃ１をアンテナ配線の両端に接続した送信アンテナ１のその容量Ｃ１の一方の接続点と送信アンテナ１のアンテナ配線との接続部分に設置しても良い。受信アンテナ２に直列に接続する負荷回路４の端子（ポート２）の位置も、受信アンテナ２の配線の中間に設置しても良いし、寄生容量に比べて大きな容量Ｃ２をアンテナ配線の両端に接続した受信アンテナ２のその容量Ｃ２の一方の接続点と受信アンテナ２のアンテナ配線との接続部分に設置しても良い。   The inductive power transmission circuit of the present invention can be modeled by the circuit diagram of FIG. 2A, and the circuit includes a transmission antenna 1 having an effective self-inductance L1 and a receiving antenna 2 having an effective self-inductance L2, and a mutual inductance M. , An inductive power transmission circuit that supplies power in a non-contact manner from the power supply circuit 3 connected to the transmission antenna 1 to the load circuit 4 connected to the reception antenna 2. Here, the capacitor C <b> 1 that connects both ends of the transmission antenna 1 may be configured by a capacitor in a tank circuit that supplies a current having an angular frequency ω from the power supply circuit 3 to the transmission antenna 1. Further, the terminal (port 1) of the power supply circuit 3 connected in series to the transmission antenna 1 may be installed in the middle of the wiring of the transmission antenna 1, or a capacitance C1 larger than the parasitic capacitance is provided at both ends of the antenna wiring. You may install in the connection part of one connection point of the capacity | capacitance C1 of the connected transmitting antenna 1, and the antenna wiring of the transmitting antenna 1. FIG. The position of the terminal (port 2) of the load circuit 4 connected in series to the receiving antenna 2 may also be installed in the middle of the wiring of the receiving antenna 2, or a capacitance C2 that is larger than the parasitic capacitance is provided at both ends of the antenna wiring. You may install in the connection part of one connection point of the capacity | capacitance C2 of the connected receiving antenna 2, and the antenna wiring of the receiving antenna 2. FIG.

鋭意研究の結果、以上で説明した回路構成で誘導電力伝送回路を構成すると、送信アンテナ１と容量Ｃ１により共振回路を構成した送信回路と、受信アンテナ２と容量Ｃ２により共振回路を構成した受信回路を共鳴させることができ、以下の条件を満たせば、電源回路３から負荷回路４まで電力を完全な効率で伝送することができることを見出し、本発明に至った。電力を効率良く伝送する条件を以下で説明する。図２（ａ）の電源回路３を送信アンテナ回路１に接続するポート１の電圧Ｅｉｎは以下の式１であらわせる。
Ｅｉｎ＝ｊ｛ωＬ１−（１／（ωＣ１））｝×Ｉ１＋ｊωＭ×Ｉ２ （式１）
ここで、Ｉ１は送信アンテナ１に流れるアンテナ電流、Ｉ２は受信アンテナ２に流れるアンテナ電流であり、ω＝２πｆはその角周波数であり、ｆは高周波電流の周波数である。また、図２（ａ）の電源回路３側から見た、送信アンテナ回路１の入力インピーダンスＺｉｎに関しては、以下の式２が成り立つ。
Ｅｉｎ＝Ｚｉｎ×Ｉ１ （式２）
式１の右辺の最後の項のｊωＭ×Ｉ２は、受信アンテナ２に流れる高周波のアンテナ電流Ｉ２が送信アンテナ１の配線の近傍の電磁界を時間変化させ、それが送信アンテナ１に誘導する誘導電圧Ｅ１である。それを式３であらわす。
Ｅ１＝ｊωＭ×Ｉ２ （式３）
式１と式２と式３から、アンテナ回路の入力インピーダンスＺｉｎが以下の式４であらわされる。
Ｚｉｎ＝ｊ｛ωＬ１−（１／（ωＣ１））｝＋Ｅ１／Ｉ１ （式４） As a result of earnest research, when an inductive power transmission circuit is configured with the circuit configuration described above, a transmission circuit in which a resonance circuit is configured by the transmission antenna 1 and the capacitor C1, and a reception circuit in which a resonance circuit is configured by the reception antenna 2 and the capacitance C2 It has been found that if the following conditions are satisfied, power can be transmitted from the power supply circuit 3 to the load circuit 4 with complete efficiency, and the present invention has been achieved. The conditions for efficiently transmitting power will be described below. The voltage Ein of the port 1 that connects the power supply circuit 3 of FIG. 2A to the transmitting antenna circuit 1 is expressed by the following equation 1.
Ein = j {ωL1- (1 / (ωC1))} × I1 + jωM × I2 (Formula 1)
Here, I1 is the antenna current flowing through the transmitting antenna 1, I2 is the antenna current flowing through the receiving antenna 2, ω = 2πf is the angular frequency, and f is the frequency of the high-frequency current. Further, with respect to the input impedance Zin of the transmission antenna circuit 1 as viewed from the power supply circuit 3 side in FIG.
Ein = Zin × I1 (Formula 2)
JωM × I2 in the last term on the right side of Equation 1 is the induced voltage that the high-frequency antenna current I2 flowing in the receiving antenna 2 changes the electromagnetic field in the vicinity of the wiring of the transmitting antenna 1 over time, and that induces the transmitting antenna 1 E1. This is expressed by Equation 3.
E1 = jωM × I2 (Formula 3)
From Equation 1, Equation 2, and Equation 3, the input impedance Zin of the antenna circuit is expressed by Equation 4 below.
Zin = j {ωL1- (1 / (ωC1))} + E1 / I1 (Formula 4)

この入力インピーダンスＺｉｎは、誘導電圧Ｅ１に起因する見かけのインピーダンス（Ｅ１／Ｉ１）と回路のインピーダンスの和になる。そして、その加わったインピーダンス（Ｅ１／Ｉ１）の実数成分により、Ｚｉｎは実数成分を持つ。そのＺｉｎの実数成分を誘導抵抗ｒ１として、以下の式５であらわす。
ｒ１≡Ｒｅａｌ（Ｅ１／Ｉ１）＝Ｒｅａｌ（ｊωＭ×Ｉ２／Ｉ１） （式５）
ここで、受信アンテナ２の電流Ｉ２と送信アンテナ１の電流Ｉ１の比を、実数のパラメータαと位相角βを用いて以下の式６であらわすと、誘導抵抗ｒ１は式７であらわせる。
Ｉ２／Ｉ１≡α・ｅｘｐ（−ｊβ） （式６）
ｒ１＝α・ωＭ・ｓｉｎ（β） （式７）
ここで、電源回路３から最も効率良く送信アンテナ１に電力を供給する条件は、電源回路３のインピーダンスＺ１が図２（ａ）の電源回路３から送信アンテナ１側を見た回路のインピーダンスＺｉｎと整合する（等しくなる）ことである。電源回路３の出力インピーダンスＺ１が純抵抗の場合は、そのＺ１が誘導抵抗ｒ１に等しくなることが整合の条件であると考える。この誘導抵抗ｒ１は、受信アンテナ２の電流Ｉ２が送信アンテナ１に誘導する電圧Ｅ１のうち送信アンテナ１の電流Ｉ１と同位相の成分を送信アンテナの電流Ｉ１で割り算した値である。 This input impedance Zin is the sum of the apparent impedance (E1 / I1) caused by the induced voltage E1 and the impedance of the circuit. And Zin has a real component by the real component of the added impedance (E1 / I1). The real number component of the Zin is expressed by the following formula 5 as an induction resistance r1.
r1≡Real (E1 / I1) = Real (jωM × I2 / I1) (Formula 5)
Here, when the ratio of the current I2 of the receiving antenna 2 and the current I1 of the transmitting antenna 1 is expressed by the following expression 6 using the real parameter α and the phase angle β, the induction resistance r1 is expressed by expression 7.
I2 / I1≡α · exp (−jβ) (Formula 6)
r1 = α · ωM · sin (β) (Formula 7)
Here, the condition for supplying power from the power supply circuit 3 to the transmission antenna 1 most efficiently is that the impedance Z1 of the power supply circuit 3 is the impedance Zin of the circuit viewed from the power supply circuit 3 in FIG. Match (become equal). When the output impedance Z1 of the power supply circuit 3 is a pure resistance, it is considered that the matching condition is that Z1 becomes equal to the induction resistance r1. The induction resistance r1 is a value obtained by dividing a component having the same phase as the current I1 of the transmission antenna 1 out of the voltage E1 induced by the current I2 of the reception antenna 2 by the transmission antenna 1 by the current I1 of the transmission antenna.

一方、受信アンテナ２にも送信アンテナ１のアンテナ電流Ｉ１で誘導される誘導電圧Ｅ２が発生する。その誘導電圧Ｅ２を式８であらわす。
Ｅ２＝ｊωＭ×Ｉ１ （式８）
すると、受信アンテナ２を負荷回路４に接続するポート２の電圧Ｅｏｕｔは以下の式９と式１０であらわされる。
Ｅｏｕｔ＝Ｅ２＋ｊ｛ωＬ２−（１／（ωＣ２））｝×Ｉ２ （式９）
Ｅｏｕｔ＝−Ｚ２×Ｉ２ （式１０）
この誘導電圧Ｅ２の出力インピーダンスの実数成分を誘導抵抗ｒ２とすると、誘導抵抗ｒ２は以下の式１１であらわされる。
ｒ２≡Ｒｅａｌ（Ｅ２／（−Ｉ２））＝Ｒｅａｌ（−ｊωＭ×Ｉ１／Ｉ２）
＝（１／α）・ωＭ・ｓｉｎ（β） （式１１）
ここで、受信アンテナ２から最も効率良く負荷回路４に電力を供給する条件は、受信アンテナ２に加わった誘導電圧Ｅ２を電源と見なして、その電源の出力インピーダンスが負荷回路４の入力インピーダンスＺ２と整合する（等しくなる）ことであると考える。そのため、負荷回路４の入力インピーダンスＺ２が純抵抗の場合は、そのＺ２が誘導抵抗ｒ２に等しくなることが整合の条件であると考える。この誘導抵抗ｒ２は、送信アンテナ１の電流Ｉ１が受信アンテナ２に誘導する電圧Ｅ２のうち受信アンテナ２の電流Ｉ２と同位相の成分を受信アンテナの電流Ｉ２で割り算した値である。 On the other hand, an induction voltage E2 induced by the antenna current I1 of the transmission antenna 1 is also generated in the reception antenna 2. The induced voltage E2 is expressed by Equation 8.
E2 = jωM × I1 (Formula 8)
Then, the voltage Eout of the port 2 that connects the receiving antenna 2 to the load circuit 4 is expressed by the following equations 9 and 10.
Eout = E2 + j {ωL2- (1 / (ωC2))} × I2 (Equation 9)
Eout = −Z2 × I2 (Formula 10)
When the real component of the output impedance of the induction voltage E2 is an induction resistance r2, the induction resistance r2 is expressed by the following equation 11.
r2≡Real (E2 / (− I2)) = Real (−jωM × I1 / I2)
= (1 / α) · ωM · sin (β) (Formula 11)
Here, the most efficient condition for supplying power from the receiving antenna 2 to the load circuit 4 is that the induced voltage E2 applied to the receiving antenna 2 is regarded as a power source, and the output impedance of the power source is the input impedance Z2 of the load circuit 4. Think of it as being consistent (equal). Therefore, when the input impedance Z2 of the load circuit 4 is a pure resistance, it is considered that the matching condition is that the Z2 is equal to the induction resistance r2. This induction resistance r2 is a value obtained by dividing a component having the same phase as the current I2 of the receiving antenna 2 out of the voltage E2 that the current I1 of the transmitting antenna 1 induces to the receiving antenna 2 by the current I2 of the receiving antenna.

また、図２（ｂ）のように、受信アンテナ２に直列に変成器の一次巻線を接続し、その変成器の二次巻線に負荷回路４を接続する受信回路を構成することもできる。この受信回路も、図２（ａ）の右側の回路であらわす、受信アンテナ２と容量Ｃ２に直列に直列負荷回路を接続した受信回路に等価な回路である。図２（ｂ）の受信回路の場合は、その図２（ａ）の右側の等価受信回路と見なして調整すれば良い。この変成器を用いる一実施形態として、図２１（ａ）のように、受信アンテナ２のコイル状（螺旋状）の配線自体をその変成器の一次巻線とし、その受信配線が乗るＸＹ面に平行に近接して設置したコイル状又は螺旋状の誘導結合配線６を、受信アンテナ２の配線に誘導結合する変成器の二次巻線とする受信回路を構成することもできる。この場合、受信アンテナ２に誘導結合配線６が相互インダクタンスＭ２で誘導結合し、誘導結合配線６の両端をポート３として負荷回路４に接続する回路構成にする。図２１（ａ）の場合においては、誘導結合配線６の有する自己インダクタンスが誘導抵抗ｒ２に比べて小さいことが影響し、ポート３の負荷回路４に接続するアンテナのインピーダンスが（２πｆ×Ｍ１）２／ｒ２になる。 Further, as shown in FIG. 2B, a receiving circuit in which the primary winding of the transformer is connected in series to the receiving antenna 2 and the load circuit 4 is connected to the secondary winding of the transformer can be configured. . This receiving circuit is also a circuit equivalent to a receiving circuit represented by a circuit on the right side of FIG. 2A in which a series load circuit is connected in series to the receiving antenna 2 and the capacitor C2. In the case of the receiving circuit of FIG. 2 (b), the adjustment may be performed considering that it is the equivalent receiving circuit on the right side of FIG. 2 (a). As an embodiment using this transformer, as shown in FIG. 21A, the coiled (spiral) wiring itself of the receiving antenna 2 is used as the primary winding of the transformer, and the XY plane on which the receiving wiring rides is placed. It is also possible to configure a receiving circuit in which a coiled or spiral inductive coupling wiring 6 installed close to the parallel is used as a secondary winding of a transformer inductively coupled to the wiring of the receiving antenna 2. In this case, the inductive coupling wiring 6 is inductively coupled to the receiving antenna 2 with the mutual inductance M 2, and both ends of the inductive coupling wiring 6 are connected to the load circuit 4 as ports 3. In the case of FIG. 21A, it is influenced by the fact that the self-inductance of the inductive coupling wiring 6 is smaller than the induction resistance r2, and the impedance of the antenna connected to the load circuit 4 of the port 3 is (2πf × M1) 2. / R2.

また、この受信回路は、受信アンテナ２自身を誘導結合配線６にすることもできる。その場合は、誘導結合配線６の両端のポート３は受信アンテナ２のアンテナ配線の両端になり、そのポート３に、容量Ｃ２に並列に、負荷回路４を接続して図２（ｃ）のように受信回路を構成することもできる。その場合は、その負荷回路４に接続するアンテナのインピーダンスが（ω・Ｌ２）２／ｒ２になる。すなわち、図２（ｃ）の回路は、図２（ｂ）の回路の誘導結合配線６の相互インダクタンスＭ２を、受信アンテナ２の実効的自己インダクタンスＬ２に置き換えた回路になる。 In the receiving circuit, the receiving antenna 2 itself can be an inductive coupling wiring 6. In that case, the ports 3 at both ends of the inductive coupling wiring 6 become both ends of the antenna wiring of the receiving antenna 2, and a load circuit 4 is connected to the port 3 in parallel with the capacitor C2 as shown in FIG. It is also possible to configure a receiving circuit. In that case, the impedance of the antenna connected to the load circuit 4 becomes (ω · L2) 2 / r2. That is, the circuit of FIG. 2C is a circuit in which the mutual inductance M2 of the inductive coupling wiring 6 in the circuit of FIG. 2B is replaced with the effective self-inductance L2 of the receiving antenna 2.

また、図２（ａ）の回路の受信アンテナ２は、コイル状（螺旋状）のアンテナ配線にし、一方、送信アンテナ１とそれに接続する電源回路３は、図２５（ａ）のように、空間から電磁波を受け取るダイポールアンテナにすることもできる。その場合は、そのダイポールアンテナが空間の電磁波から電力を受け取って回路に供給する電源になり、その電力を、ダイポールアンテナが兼用する送信アンテナ１が受信アンテナ２に伝達する。その電力が受信アンテナ２に誘導抵抗ｒ２を発生させ、その誘導抵抗ｒ２に負荷回路４の入力インピーダンスＺ２を等しくして空間の電磁波から効率良く電力を受け取る誘導電力伝送回路を構成することができる。また、その逆に、送信アンテナ１は、コイル状（螺旋状）のアンテナ配線にし、一方、受信アンテナ２とそれに接続する負荷回路４は、空間に電磁波を放射するダイポールアンテナにし、空間へ電磁波を放射して電力を消費する負荷にすることもできる。その場合は、送信アンテナ１からダイポールアンテナで構成される受信アンテナ２に接続する負荷となる放射抵抗に受信アンテナ２の誘導抵抗ｒ２を整合（等しくする）させることで、空間に効率良く電磁波を放射する誘導電力伝送回路を構成することもできる。   In addition, the receiving antenna 2 of the circuit of FIG. 2A is a coiled (spiral) antenna wiring, while the transmitting antenna 1 and the power supply circuit 3 connected to the transmitting antenna 1 are a space as shown in FIG. It can also be a dipole antenna that receives electromagnetic waves from. In that case, the dipole antenna serves as a power source that receives power from electromagnetic waves in the space and supplies it to the circuit, and the transmitting antenna 1 that is also used by the dipole antenna transmits the power to the receiving antenna 2. An induction power transmission circuit that efficiently receives power from electromagnetic waves in the space can be configured by generating the induction resistance r2 in the receiving antenna 2 and making the input resistance Z2 of the load circuit 4 equal to the induction resistance r2. On the contrary, the transmitting antenna 1 is a coiled (spiral) antenna wiring, while the receiving antenna 2 and the load circuit 4 connected to the receiving antenna 2 are dipole antennas that radiate electromagnetic waves into the space. It can be a load that radiates and consumes power. In that case, electromagnetic waves can be efficiently radiated into the space by matching (equalizing) the induction resistance r2 of the receiving antenna 2 with the radiation resistance serving as a load connected from the transmitting antenna 1 to the receiving antenna 2 composed of a dipole antenna. An inductive power transmission circuit can be configured.

以下では、このようにインピーダンスを整合させる場合に、共鳴をおこす角周波数ωがどうなるかを詳しく解析する。すなわち、電源回路３の出力インピーダンスＺ１が誘導抵抗ｒ１に等しい値に整合されて、負荷回路４の入力インピーダンスＺ２が誘導抵抗ｒ２に整合（等しく）される場合には、式１と式９等から、以下の式１２と式１３が成り立つ。
ωＬ１−（１／（ωＣ１））＝−α・ωＭ・ｃｏｓ（β） （式１２）
ωＬ２−（１／（ωＣ２））＝−（１／α）・ωＭ・ｃｏｓ（β） （式１３）
これらの式１２と式１３が成り立ちＺ１＝ｒ１、Ｚ２＝ｒ２の場合には、アンテナ１の電磁界とアンテナ２の電磁界が共鳴し、それにより、電源回路３から負荷回路４まで電力が完全な効率で伝送されると考える。 Hereinafter, in the case of matching the impedance in this way, a detailed analysis will be made of what happens to the angular frequency ω causing resonance. That is, when the output impedance Z1 of the power supply circuit 3 is matched to a value equal to the induction resistance r1, and the input impedance Z2 of the load circuit 4 is matched (equal) to the induction resistance r2, The following expressions 12 and 13 hold.
ωL1- (1 / (ωC1)) = − α · ωM · cos (β) (Formula 12)
ωL2- (1 / (ωC2)) = − (1 / α) · ωM · cos (β) (Formula 13)
When these equations 12 and 13 hold and Z1 = r1 and Z2 = r2, the electromagnetic field of the antenna 1 and the electromagnetic field of the antenna 2 resonate, so that the power is completely transmitted from the power supply circuit 3 to the load circuit 4. Is considered to be transmitted with high efficiency.

ここで、ｃｏｓ（β）が０で無い場合には、式７、式１１、式１２、式１３から以下の式が得られる。
ｒ１・ｒ２＝（ωＭ）２−ｇ１×ｇ２ （式１４）
ｇ１≡ωＬ１−（１／（ωＣ１）） （式１５）
ｇ２≡ωＬ２−（１／（ωＣ２）） （式１６）
α２＝ｇ１／ｇ２ （式１７）
ｓｉｎ（β）２＝Ｚ１・Ｚ２／（ωＭ）２ （式１８）
ＭとＬ１とＣ１とＬ２とＣ２が定まっている場合は、式１４により、ωに応じてｒ１・ｒ２が求まり、そして式１７からαが求まる。次に、式１８から、βが求まる。次に、式７と式１１によって、ｒ１とｒ２が求まる。特に、式１７から、ｇ１×ｇ２は正である。そして、ωＭが小さい値の場合にも式１４が成り立つには、ωがあるωｏの場合にｇ１＝ｇ２＝０になる必要がある。そのための条件は以下の式１９である。
Ｌ１・Ｃ１＝Ｌ２・Ｃ２≡（１／ωｏ）２ （式１９） Here, when cos (β) is not 0, the following expressions are obtained from Expression 7, Expression 11, Expression 12, and Expression 13.
r1 · r2 = (ωM) 2 −g1 × g2 (Formula 14)
g1≡ωL1- (1 / (ωC1)) (Formula 15)
g2≡ωL2- (1 / (ωC2)) (Formula 16)
α 2 = g1 / g2 (Formula 17)
sin (β) 2 = Z1 · Z2 / (ωM) 2 (Formula 18)
When M, L1, C1, L2, and C2 are determined, r1 · r2 is obtained according to ω by Equation 14, and α is obtained from Equation 17. Next, β is obtained from Equation 18. Next, r1 and r2 are obtained by Expression 7 and Expression 11. In particular, from Equation 17, g1 × g2 is positive. In order for Equation 14 to hold even when ωM is a small value, it is necessary that g1 = g2 = 0 when ω is some ωo. The condition for this is Equation 19 below.
L1 · C1 = L2 · C2≡ (1 / ωo) 2 (Formula 19)

（第１の場合）
式１９が成り立つ場合に、ω＝ωｏのとき、すなわち、式１２及び式１３でｃｏｓ（β）が０でｓｉｎ（β）が１になる場合に、ｇ１＝ｇ２＝０が成り立つ。この第１の場合については後で説明する。
（第２の場合）
式１９が成り立つ場合に、ωがωｏ以外のときには、式６と式７と式１１と式１５から式１７を使うと、以下の式２０が成り立つ。これを第２の場合と呼ぶ。
｜Ｉ２／Ｉ１｜２＝α２＝Ｌ１／Ｌ２＝Ｃ２／Ｃ１＝ｒ１／ｒ２ （式２０）
この式２０から以下の式２１が成り立つ。
Ｌ１×｜Ｉ１｜２＝Ｌ２×｜Ｉ２｜２ （式２１）
この式２１は、送信アンテナ１に蓄積される電磁界のエネルギーと受信アンテナ２に蓄積される電磁界のエネルギーが等しく、両アンテナが互いにその電磁界エネルギーを交換して共鳴している状態をあらわしていると考える。
この式２１を変形して、以下の式２２を得る。
｜Ｉ２｜＝｜Ｉ１｜×√（Ｌ１／Ｌ２） （式２２）
すなわち、共鳴した送信アンテナ１の電流Ｉ１と受信アンテナ２の電流Ｉ２の比が、受信アンテナ２の配線の実効的自己インダクタンスＬ２と送信アンテナ１の配線の実効的自己インダクタンスＬ１の比の平方根である。電磁界シミュレーションの結果でも、共振して効率良く（１００％近い効率で）エネルギーを伝送するアンテナ回路では式２２の関係が成り立っていた。式２２のように受信アンテナ２に多くの電流が流れるので、受信アンテナ２に流れる高周波のアンテナ電流Ｉ２が電磁界を時間変化させ、それにより送信アンテナ１に誘導電圧Ｅ１を発生させると考える。 (First case)
When Equation 19 holds, when ω = ωo, that is, when cos (β) is 0 and sin (β) is 1 in Equations 12 and 13, g1 = g2 = 0 holds. This first case will be described later.
(Second case)
When Equation 19 is satisfied and ω is other than ωo, using Equation 6, Equation 7, Equation 11, and Equation 15 to Equation 17, the following Equation 20 is established. This is called the second case.
| I2 / I1 | 2 = α 2 = L1 / L2 = C2 / C1 = r1 / r2 (Formula 20)
From this equation 20, the following equation 21 holds.
L1 × | I1 | 2 = L2 × | I2 | 2 (Formula 21)
This equation 21 represents a state in which the electromagnetic field energy stored in the transmitting antenna 1 and the electromagnetic field energy stored in the receiving antenna 2 are equal, and both antennas resonate with each other by exchanging their electromagnetic energy. I think.
This equation 21 is modified to obtain the following equation 22.
| I2 | = | I1 | × √ (L1 / L2) (Formula 22)
That is, the ratio of the resonant current I1 of the transmitting antenna 1 to the current I2 of the receiving antenna 2 is the square root of the ratio of the effective self inductance L2 of the wiring of the receiving antenna 2 to the effective self inductance L1 of the wiring of the transmitting antenna 1. . As a result of the electromagnetic field simulation, the relationship of Formula 22 is established in the antenna circuit that resonates and efficiently transmits energy (with an efficiency close to 100%). Since a large amount of current flows through the receiving antenna 2 as shown in Equation 22, it is considered that the high-frequency antenna current I2 flowing through the receiving antenna 2 changes the electromagnetic field over time, thereby generating an induced voltage E1 at the transmitting antenna 1.

ここで、式１９が成り立つ場合は、送信アンテナ１と受信アンテナ２の電磁誘導の結合係数ｋ（式２３であらわす）を用いて、式７と式１１と式１２と式１３が、式２４から式２７に書き換えられる。
ｋ≡Ｍ／√（Ｌ１×Ｌ２） （式２３）
ｒ１＝ｋωＬ１・ｓｉｎ（β） （式２４）
ｒ２＝ｋωＬ２・ｓｉｎ（β） （式２５）
ωＬ１−１／（ωＣ１）＝−ｋωＬ１・ｃｏｓ（β） （式２６）
ωＬ２−１／（ωＣ２）＝−ｋωＬ２・ｃｏｓ（β） （式２７）
式２６と式２７から、以下の式２８が得られる。
ω＝ωｏ／√（１＋ｋ・ｃｏｓ（β）） （式２８） Here, when Expression 19 holds, Expression 7, Expression 11, Expression 12, and Expression 13 are obtained from Expression 24 using the electromagnetic induction coupling coefficient k (expressed by Expression 23) of the transmission antenna 1 and the reception antenna 2. Equation 27 is rewritten.
k≡M / √ (L1 × L2) (Equation 23)
r1 = kωL1 · sin (β) (Equation 24)
r2 = kωL2 · sin (β) (Equation 25)
ωL1-1 / (ωC1) = − kωL1 · cos (β) (Formula 26)
ωL2-1 / (ωC2) = − kωL2 · cos (β) (Expression 27)
From Expression 26 and Expression 27, the following Expression 28 is obtained.
ω = ωo / √ (1 + k · cos (β)) (Formula 28)

以上の関係は、以下のように言い換えることができる。すなわち、Ｌ１×Ｃ１＝Ｌ２×Ｃ２＝１／ωｏ２であるアンテナ系において、０からπラジアンまでの値の任意の位相角βに関して、電力を伝送する交流の角周波数ωを、Ｌ１×Ｃ１×（１＋ｋ・ｃｏｓ（β））の値の逆数の平方根にして、送信アンテナ１に直列に接続する電源回路３の出力インピーダンスＺ１をｒ１＝ｋωＬ１・ｓｉｎ（β）にし、受信アンテナに直列に接続する負荷回路４の入力インピーダンスＺ２をｒ２＝ｋωＬ２・ｓｉｎ（β）にすると、電力を完全な効率で伝送できる。そして、電源回路３から負荷回路４まで、電力を完全な効率で伝送できる誘導抵抗ｒの値に上限がある。空芯コイルの送信アンテナ１と受信アンテナ２を対向させ近づけると結合係数ｋが大きくなり電力を伝送できる誘導抵抗ｒの上限が大きくなる。誘導抵抗ｒ１の上限がｋωＬ１で、誘導抵抗ｒ２の上限がｋωＬ２であり、その上限以下の誘導抵抗ｒ１に電源回路３の出力インピーダンスＺ１を等しくし、誘導抵抗ｒ２に負荷回路４の入力インピーダンスＺ２を等しくすることで電力を完全な効率で伝送できる。電源回路３の出力インピーダンスＺ１と負荷回路４の入力インピーダンスＺ２を誘導抵抗ｒの上限より小さく設定する場合は、式２４と式２５でｓｉｎ（β）が１より小さい値でそれらのインピーダンスが誘導抵抗に等しくなりインピーダンスが整合して電力が伝送できる。そして、その場合には、ｃｏｓ（β）が０では無く、式２８により、ωｏからずれた共振角周波数ωで送信アンテナ１と受信アンテナ２が共鳴する。 The above relationship can be paraphrased as follows. That is, in an antenna system in which L1 × C1 = L2 × C2 = 1 / ωo 2 , the angular frequency ω of the alternating current for transmitting power is expressed as L1 × C1 × for an arbitrary phase angle β having a value from 0 to π radians. Using the square root of the reciprocal of the value of (1 + k · cos (β)), the output impedance Z1 of the power supply circuit 3 connected in series to the transmitting antenna 1 is set to r1 = kωL1 · sin (β) and connected in series to the receiving antenna. When the input impedance Z2 of the load circuit 4 is r2 = kωL2 · sin (β), power can be transmitted with complete efficiency. There is an upper limit to the value of the induction resistor r that can transmit power from the power supply circuit 3 to the load circuit 4 with complete efficiency. When the transmitting antenna 1 and the receiving antenna 2 of the air-core coil face each other and approach each other, the coupling coefficient k increases and the upper limit of the inductive resistance r that can transmit power increases. The upper limit of the induction resistance r1 is kωL1, the upper limit of the induction resistance r2 is kωL2, the output impedance Z1 of the power supply circuit 3 is made equal to the induction resistance r1 below the upper limit, and the input impedance Z2 of the load circuit 4 is set to the induction resistance r2. By making them equal, power can be transmitted with complete efficiency. When the output impedance Z1 of the power supply circuit 3 and the input impedance Z2 of the load circuit 4 are set to be smaller than the upper limit of the induction resistance r, the sin (β) is a value smaller than 1 in Expression 24 and Expression 25, and those impedances are induction resistance. The impedance is matched and power can be transmitted. In this case, cos (β) is not 0, and the transmission antenna 1 and the reception antenna 2 resonate at a resonance angular frequency ω deviated from ωo according to Equation 28.

この第２の場合の現象を利用して、送信アンテナ１と受信アンテナ２の位置が安定せず電磁誘導の結合係数ｋが変動する場合にも誘導抵抗ｒ１とｒ２に整合（等しい）する電源回路３と負荷回路４のインピーダンスを一定に保つことができる誘導電力伝送回路を構成できる。それは、上限の値より小さな値の誘導抵抗ｒ１に等しい値の固定した値の出力インピーダンスＺ１を有する電源回路３と、その際の誘導抵抗ｒ２に等しい値の固定した値の入力インピーダンスＺ２を有する負荷回路４を用い、結合係数ｋの値の変化に応じて位相角βを変え共振角周波数ωを変えるように電源回路３を適応させて共鳴させる。そのように適応する電源回路３の構成は、送信アンテナ１の共振電流を正帰還回路により電源回路３に正帰還させて、その電流を増幅して出力する電源回路３を構成することで実現できる。これにより、電源回路３から共振角周波数ωに適応した周波数の電流Ｉ１を取り出すことができ、電磁誘導の結合係数ｋの変化に適応してアンテナ回路を共鳴させ、結合係数ｋの変化があっても完全な効率の電力伝送を維持させる誘導電力伝送回路が構成できる効果がある。   By utilizing the phenomenon of the second case, a power supply circuit that matches (equals) the induction resistances r1 and r2 even when the positions of the transmission antenna 1 and the reception antenna 2 are not stable and the coupling coefficient k of electromagnetic induction varies. 3 and an inductive power transmission circuit that can keep the impedance of the load circuit 4 constant. The power supply circuit 3 has a fixed output impedance Z1 having a value equal to the induction resistance r1 having a value smaller than the upper limit value, and a load having the fixed input impedance Z2 having a value equal to the induction resistance r2 at that time. Using the circuit 4, the power supply circuit 3 is adapted to resonate so as to change the phase angle β and change the resonance angular frequency ω in accordance with the change in the value of the coupling coefficient k. The configuration of the power supply circuit 3 that is adapted in this way can be realized by configuring the power supply circuit 3 that amplifies and outputs the current by positively feeding back the resonance current of the transmission antenna 1 to the power supply circuit 3 using a positive feedback circuit. . As a result, the current I1 having a frequency adapted to the resonance angular frequency ω can be extracted from the power supply circuit 3, and the antenna circuit is resonated in accordance with the change of the coupling coefficient k of electromagnetic induction, and the coupling coefficient k is changed. In addition, there is an effect that an inductive power transmission circuit capable of maintaining perfect power transmission can be configured.

ここで、アンテナ系の電力伝送効率Ｐｅは、ｒ１＝ｒ２の場合は、以下の式２９で計算できると考える。
Ｐｅ＝ｒ１／（ｒ１＋（ｒｅｆ１＋ｒｅｆ２）） （式２９）
ここで、ｒｅｆ１は送信アンテナ１の実効的抵抗、ｒｅｆ２は受信アンテナ２の実効的抵抗である。送信アンテナ１の実効的抵抗ｒｅｆ１と受信アンテナ２の実効的抵抗ｒｅｆ２が誘導抵抗ｒ１より小さいほど電力の伝送効率が良く、アンテナ配線の実効的抵抗が誘導抵抗ｒに比べて無視できるほど小さい場合はほぼ１００％の電力が伝送できると考える。また、送信アンテナ１と受信アンテナ２は、ダイポールアンテナ状にして共鳴させて電力を伝送することもできるが、図１のようなコイル状（渦巻き状）にアンテナ配線を巻くと、アンテナの実効的自己インダクタンスＬ１およびＬ２がダイポールアンテナ形の場合より大きくなるので、式２４と式２４により誘導抵抗ｒ１およびｒ２が大きくなり、式２９で示すアンテナ系の電力伝送効率Ｐｅが大きくなる効果がある。 Here, it is considered that the power transmission efficiency Pe of the antenna system can be calculated by the following equation 29 when r1 = r2.
Pe = r1 / (r1 + (ref1 + ref2)) (Formula 29)
Here, ref1 is an effective resistance of the transmitting antenna 1, and ref2 is an effective resistance of the receiving antenna 2. When the effective resistance ref1 of the transmitting antenna 1 and the effective resistance ref2 of the receiving antenna 2 are smaller than the induction resistance r1, the power transmission efficiency is better, and the effective resistance of the antenna wiring is small enough to be ignored compared to the induction resistance r. It is considered that almost 100% of power can be transmitted. Further, the transmitting antenna 1 and the receiving antenna 2 can resonate in the form of a dipole antenna to transmit electric power. However, if the antenna wiring is wound in a coil shape (spiral shape) as shown in FIG. Since the self-inductances L1 and L2 are larger than those in the case of the dipole antenna type, the induction resistances r1 and r2 are increased by Expression 24 and Expression 24, and the power transmission efficiency Pe of the antenna system indicated by Expression 29 is increased.

特に、ω≒ωｏの場合は、式２６と式２７の左辺が０に近くなるので、その右辺も０に近くなるため、βがπ／２ラジアンに近くなり、ｃｏｓ（β）が０に近くなり、ｓｉｎ（β）が１に近くなり、以上の式２４と式２５は以下の式３０と式３１になる。
ｒ１≒ｋωＬ１ （式３０）
ｒ２≒ｋωＬ２ （式３１）
このように、ω≒ωｏの場合に、図２（ａ）に示す電源回路３の出力インピーダンスＺ１を式３０の誘導抵抗ｒ１に整合させ、負荷回路４の負荷インピーダンスＺ２を式３１の誘導抵抗ｒ２に整合させると、電源回路３から負荷回路４まで、電力を完全な効率で伝送できる。 In particular, in the case of ω≈ωo, the left side of Equation 26 and Equation 27 is close to 0, and the right side thereof is also close to 0. Therefore, β is close to π / 2 radians, and cos (β) is close to 0. Thus, sin (β) becomes close to 1, and the above Expression 24 and Expression 25 become Expression 30 and Expression 31 below.
r1≈kωL1 (Formula 30)
r2≈kωL2 (Formula 31)
As described above, when ω≈ωo, the output impedance Z1 of the power supply circuit 3 shown in FIG. 2A is matched with the induction resistance r1 of Expression 30, and the load impedance Z2 of the load circuit 4 is changed to the induction resistance r2 of Expression 31. Therefore, power can be transmitted from the power supply circuit 3 to the load circuit 4 with complete efficiency.

（第１の場合）
以下で、先に示した第１の場合について詳しく説明する。第１の場合は、式１９が成り立つ場合に、ｓｉｎ（β）が１になる場合であって、ｇ１＝ｇ２＝０になる。この場合は、アンテナ電流Ｉ１とアンテナ電流Ｉ２の位相差をあらわす位相角βが９０度（π／４ラジアン）の場合である。この場合は、角周波数ω＝ωｏで共振し、以下の式３２から式３５の状態でアンテナ系が共鳴する。
ｃｏｓ（β）＝０ （式３２）
ｒ１＝ωＭ・α （式３３）
ｒ２＝ωＭ／α （式３４）
Ｉ２／Ｉ１＝−ｊα （式３５）
以上の関係は、以下のように言い換えることができる。すなわち、Ｌ１×Ｃ１＝Ｌ２×Ｃ２＝１／ωｏ２であるアンテナ系において、電力を伝送する交流の角周波数ωをωｏにし、任意の正の数αに関して、送信アンテナ１に直列に接続する電源回路３の出力インピーダンスＺ１をｒ１＝ωＭ・αにし、受信アンテナに直列に接続する負荷回路４の入力インピーダンスＺ２をｒ２＝ωＭ／αにすると、電力を完全な効率で伝送できる。つまり、この共鳴の場合は、式９で示すようにアンテナの共鳴の角周波数ω＝２πｆがωｏに一致して共鳴するが、任意のアンテナ電流の比αで電力を伝送できる特徴がある。アンテナ電流の比αが任意であるという意味は、送信アンテナ１の電流Ｉ１を大きくして大きな電磁界を発生させれば、受信アンテナ２に流れる電流Ｉ２が小さくても良い効率で電力を伝送できることを意味する。逆に、受信アンテナ２に流れる電流Ｉ２が大きければ、送信アンテナ１の電流Ｉ１が小さくても良い効率で電力を伝送できることを意味する。なお、誘導抵抗ｒ１とｒ２の積が（ωＭ）の二乗の一定値である。また、誘導抵抗ｒ２に対するｒ１の比は、アンテナ電流の比αの二乗であって任意に変えることができる。 (First case)
Hereinafter, the first case described above will be described in detail. The first case is a case where sin (β) is 1 when Equation 19 holds, and g1 = g2 = 0. In this case, the phase angle β representing the phase difference between the antenna current I1 and the antenna current I2 is 90 degrees (π / 4 radians). In this case, the antenna system resonates at an angular frequency ω = ωo, and the antenna system resonates in the following equations 32 to 35.
cos (β) = 0 (Formula 32)
r1 = ωM · α (Formula 33)
r2 = ωM / α (Formula 34)
I2 / I1 = −jα (Formula 35)
The above relationship can be paraphrased as follows. That is, in an antenna system in which L1 × C1 = L2 × C2 = 1 / ωo 2 , an AC angular frequency ω for transmitting power is set to ωo, and an arbitrary positive number α is connected to the transmission antenna 1 in series. When the output impedance Z1 of the circuit 3 is set to r1 = ωM · α, and the input impedance Z2 of the load circuit 4 connected in series to the receiving antenna is set to r2 = ωM / α, power can be transmitted with complete efficiency. That is, in the case of this resonance, the resonance frequency ω = 2πf of the antenna resonates in accordance with ωo as shown in Equation 9, but there is a feature that power can be transmitted with an arbitrary antenna current ratio α. The meaning that the ratio α of the antenna current is arbitrary means that if the current I1 of the transmitting antenna 1 is increased to generate a large electromagnetic field, power can be transmitted with an efficiency that allows the current I2 flowing through the receiving antenna 2 to be small. Means. Conversely, if the current I2 flowing through the receiving antenna 2 is large, it means that power can be transmitted with sufficient efficiency even if the current I1 of the transmitting antenna 1 is small. The product of the inductive resistances r1 and r2 is a constant value of the square of (ωM). The ratio of r1 to the induction resistance r2 is a square of the ratio α of the antenna current and can be arbitrarily changed.

この第１の場合の現象を利用して、空芯コイルによる送信アンテナ１の誘導抵抗ｒ１と受信アンテナ２の誘導抵抗ｒ２を変換するインピーダンス変換回路を構成する誘導電力伝送回路が得られる。すなわち、図２（ａ）の回路図で、電源回路３のインピーダンスｒ１を負荷回路４のインピーダンスｒ２＝（ωＭ）２／ｒ１に変換するインピーダンス変換回路を構成できる。また、このインピーダンス変換回路用の送信アンテナ１と受信アンテナ２の間隔を変えて結合係数ｋすなわち相互インダクタンスＭを変えれば、送信アンテナのインピーダンスｒ１を元のまま変えないで受信アンテナ２のインピーダンスｒ２の値だけを変えることができるインピーダンス変換回路を構成できる。このインピーダンス変換回路は、１つのパラメータｋを変えるだけで変換するインピーダンスｒ２の値を変えることができ、回路パラメータの調整が簡単であり容易にインピーダンスを調整できる効果がある。 By utilizing the phenomenon in the first case, an inductive power transmission circuit constituting an impedance conversion circuit for converting the induction resistance r1 of the transmission antenna 1 and the induction resistance r2 of the reception antenna 2 by the air-core coil is obtained. That is, in the circuit diagram of FIG. 2A, an impedance conversion circuit that converts the impedance r1 of the power supply circuit 3 into the impedance r2 = (ωM) 2 / r1 of the load circuit 4 can be configured. Further, if the coupling coefficient k, that is, the mutual inductance M is changed by changing the distance between the transmission antenna 1 and the reception antenna 2 for the impedance conversion circuit, the impedance r2 of the reception antenna 2 is not changed without changing the impedance r1 of the transmission antenna. An impedance conversion circuit that can change only the value can be configured. This impedance conversion circuit can change the value of the impedance r2 to be converted only by changing one parameter k, and the adjustment of the circuit parameter is simple and the impedance can be easily adjusted.

以下では、各実施形態毎に、電磁界シミュレーションで、電源回路３から負荷回路４まで電力を完全な効率で伝送する電源回路３の出力インピーダンスＺ１と負荷回路４の入力インピーダンスＺ２を求め、その値を誘導抵抗ｒ１とｒ２として誘導抵抗を求める。その誘導抵抗ｒ１はωＭα・ｓｉｎ（β）であって、誘導抵抗ｒ２はωＭ・ｓｉｎ（β）／αである。以下に示す電磁界シミュレーションでは、送信アンテナ１と受信アンテナ２を空芯コイル状（螺旋状）に形成し、両アンテナの電磁誘導の結合係数ｋを０．０１ぐらいに小さくするまで両アンテナを離して空間をあけた場合でも、両アンテナ回路を共鳴させることができた。そして、その共鳴させたアンテナ回路では、本発明の原理の第２の場合には、インピーダンスＺを式２４と式２５（式３０と式３１）であらわす誘導抵抗ｒに整合（等しく）させれば、電源回路３から負荷回路４へ電力を完全な伝送効率で伝送できる効果があることを見出した。また、本発明の原理の第１の場合には、インピーダンスＺを式３３と式３４であらわす誘導抵抗ｒに等しくすれば、電源回路３から負荷回路４へ電力を完全な伝送効率で伝送できる効果があることを見出した。なお、本発明は、共鳴させるアンテナ間の空間に真空や空気以外の、例えば誘電体媒質を充填した回路にも適用でき、また、常磁性体を充填した回路にも適用できる。また、本発明の誘導電力伝送回路は、電力をエネルギー供給のために伝送する用途だけに限定されず、信号伝達のために電力を送信アンテナ１から受信アンテナ２に伝送する用途の誘導電力伝送回路に用いることもできる。   In the following, for each embodiment, an electromagnetic field simulation determines the output impedance Z1 of the power supply circuit 3 and the input impedance Z2 of the load circuit 4 that transmit power from the power supply circuit 3 to the load circuit 4 with complete efficiency, and the values Is determined as induction resistances r1 and r2. The induction resistance r1 is ωMα · sin (β), and the induction resistance r2 is ωM · sin (β) / α. In the electromagnetic field simulation shown below, the transmitting antenna 1 and the receiving antenna 2 are formed in an air-core coil shape (spiral shape), and the antennas are separated until the electromagnetic induction coupling coefficient k of both antennas is reduced to about 0.01. Both antenna circuits were able to resonate even when the space was opened. In the second case of the principle of the present invention, in the resonant antenna circuit, if the impedance Z is matched (equal) to the inductive resistance r expressed by Equation 24 and Equation 25 (Equation 30 and Equation 31), The present inventors have found that power can be transmitted from the power supply circuit 3 to the load circuit 4 with complete transmission efficiency. Further, in the first case of the principle of the present invention, if the impedance Z is made equal to the induction resistance r expressed by the equations 33 and 34, the power can be transmitted from the power supply circuit 3 to the load circuit 4 with perfect transmission efficiency. Found that there is. The present invention can be applied to a circuit in which a space between antennas to be resonated is filled with, for example, a dielectric medium other than vacuum or air, and can also be applied to a circuit in which a paramagnetic material is filled. In addition, the inductive power transmission circuit of the present invention is not limited to the purpose of transmitting power for energy supply, but the inductive power transmission circuit for the purpose of transmitting power from the transmitting antenna 1 to the receiving antenna 2 for signal transmission. It can also be used.

＜第１の実施形態＞
第１の実施形態として、生体内に受信アンテナを埋め込み、生体外に送信アンテナから皮膚を隔てて生体内の受信アンテナまで電力を伝送する誘導電力伝送回路を説明する。図１から図１２により、第１の実施形態を説明する。図１では、送信アンテナ１は、平面上に形成した幅が１ｍｍで厚さが５０μｍの銅の配線でコイル径Ｄが４６ｍｍの１巻のコイルを形成した。その送信アンテナ１は例えば厚さ０．０２５ｍｍのポリイミドフィルム上に形成し、また、受信アンテナは、厚さが０．２５ｍｍのポリイミド層で覆って作り、生体内に手術によって埋め込むことができる。その送信アンテナ１の配線の中間に電源回路３の端子（ポート１）を直列に接続して給電する。電源回路３から送信アンテナ１まで接続する給電線は電源回路３の出力インピーダンスＺ１に整合する特性インピーダンスの給電線を用いる。例えば、電源回路３の出力インピーダンスＺ１が４Ωの場合は、比誘電率が３．５のポリイミドの５０μｍの厚さのフィルムの両面に、幅が２．４ｍｍの銅の配線を対向させて形成した、特性インピーダンスが４Ωの給電線を用いる。送信アンテナ１には、その両端をつなぐ１００ｐＦの容量Ｃ１を設置した。配線パターンで形成する１００ｐＦの容量Ｃ１は、一辺の長さが３４ｍｍの正方形の電極を２つ平行に配置し０．１ｍｍの空気の間隔をあけることで形成できる。また、厚さ０．０２５ｍｍで誘電率が３．５のポリイミドの両面に４６ｍｍ×１．８ｍｍの矩形の電極を形成することでも１００ｐＦの容量を形成できる。受信アンテナ２は、幅が１ｍｍで厚さが５０μｍの銅の配線でコイル径Ｇが５０ｍｍの１巻のコイルを厚さ０．０２５ｍｍのポリイミドフィルムで覆って形成する。この受信アンテナ２の径は送信アンテナ１の径と異ならせても良い。その受信アンテナ２の配線の中間に負荷回路４の端子（ポート２）を直列に接続する。また、受信アンテナ２の両端をつなぐ９０ｐＦの容量Ｃ２を設置する。図１（ｂ）の側面図のように、送信アンテナ１と受信アンテナ２は、アンテナのコイルの軸方向（ＸＹ面に垂直方向）にアンテナ間隔ｈの距離を隔てて配置する。そして、図２（ａ）の回路図の誘導電力伝送回路を構成し、電源回路３は、アンテナ回路が共鳴する角周波数ωの電流Ｉ１を送信アンテナ１に出力するように、出力電流Ｉ１を正帰還して増幅する電源回路に構成し、アンテナの共鳴角周波数ωで発振させる。 <First Embodiment>
As a first embodiment, an inductive power transmission circuit that embeds a receiving antenna in a living body and transmits power to the receiving antenna in the living body across the skin from the transmitting antenna outside the living body will be described. The first embodiment will be described with reference to FIGS. In FIG. 1, the transmitting antenna 1 is formed of a single coil having a width of 1 mm and a thickness of 50 μm formed on a plane and a coil diameter D of 46 mm. The transmitting antenna 1 is formed on, for example, a polyimide film having a thickness of 0.025 mm, and the receiving antenna can be covered with a polyimide layer having a thickness of 0.25 mm and can be embedded in a living body by surgery. The terminal (port 1) of the power supply circuit 3 is connected in series between the wirings of the transmission antenna 1 to supply power. A power supply line connected from the power supply circuit 3 to the transmission antenna 1 uses a power supply line having a characteristic impedance that matches the output impedance Z1 of the power supply circuit 3. For example, when the output impedance Z1 of the power supply circuit 3 is 4Ω, a copper wiring having a width of 2.4 mm is formed on both sides of a 50 μm-thick polyimide film having a relative dielectric constant of 3.5. A feed line having a characteristic impedance of 4Ω is used. The transmitting antenna 1 was provided with a capacitance C1 of 100 pF connecting both ends thereof. The capacitance C1 of 100 pF formed by the wiring pattern can be formed by arranging two square electrodes each having a length of 34 mm in parallel and leaving an air gap of 0.1 mm. A capacitance of 100 pF can also be formed by forming a rectangular electrode of 46 mm × 1.8 mm on both surfaces of polyimide having a thickness of 0.025 mm and a dielectric constant of 3.5. The receiving antenna 2 is formed by covering a single coil having a width of 1 mm and a thickness of 50 μm and a coil diameter G of 50 mm with a polyimide film having a thickness of 0.025 mm. The diameter of the receiving antenna 2 may be different from the diameter of the transmitting antenna 1. The terminal (port 2) of the load circuit 4 is connected in series between the wires of the receiving antenna 2. In addition, a capacitance C2 of 90 pF that connects both ends of the receiving antenna 2 is installed. As shown in the side view of FIG. 1B, the transmitting antenna 1 and the receiving antenna 2 are arranged with an antenna interval h in the axial direction of the antenna coil (perpendicular to the XY plane). Then, the inductive power transmission circuit of the circuit diagram of FIG. 2A is configured, and the power supply circuit 3 positively outputs the output current I1 so as to output the current I1 of the angular frequency ω at which the antenna circuit resonates to the transmitting antenna 1. A power supply circuit that is fed back and amplified is configured to oscillate at the resonance angular frequency ω of the antenna.

（電源回路３と負荷回路４のインピーダンスを整合させるアンテナの誘導抵抗値）
この誘導電力伝送回路の誘導抵抗ｒ１とｒ２を、電磁界シミュレーションで、以下のように求めた。すなわち、電源回路３から負荷回路４へ最も効率良く電力を伝送する電源回路３の出力インピーダンスＺ１の値を求め、その値が送信アンテナ１の誘導抵抗ｒ１であるとし、同じく、最も効率良く電力を伝送する場合の負荷回路４の負荷インピーダンスＺ２の値が受信アンテナ２の誘導抵抗ｒ２であるとする。図３に、アンテナ間隔ｈを種々に変えてシミュレーションした結果の、電源回路３から負荷回路４までの電力の伝送のＳパラメータ（Ｓ２１）をｄＢ（デシベル）であらわして縦軸に示す。その横軸は、電源回路が送信アンテナ１に流すアンテナ電流Ｉ１の周波数ｆをあらわすグラフを示す。図３（ａ）は、図１のアンテナ間隔ｈが１ｍｍの場合を示し、図３（ｂ）はｈ＝１０ｍｍの場合を示し、図３（ｃ）は、ｈ＝２０ｍｍの場合を示す。図３（ａ）で、アンテナ間隔ｈが１ｍｍの場合は、送信アンテナ１の誘導抵抗ｒ１が２０Ωであり、受信アンテナ２の誘導抵抗ｒ２が２３Ωである。この誘導抵抗ｒに電源回路３と負荷回路４のインピーダンスＺを一致させた場合にアンテナが共鳴して電力の伝送効率が最も良くなり、アンテナ電流Ｉ１の周波数ｆが４０ＭＨｚの場合の電力の伝送効率は１００％に近かった。アンテナ間の距離は（アンテナが共振する電磁界の波長）／（２π）以下の近傍にする。本実施形態では、周波数ｆ＝４０ＭＨｚの電磁界の波長は約７．５ｍの波長であり、２０ｍｍのアンテナ間隔ｈで離しても、そのアンテナ間隔ｈは（アンテナが共振する電磁界の波長）／（２π）の６０分の１であり十分近い。図３（ｂ）で、アンテナ間隔ｈが１０ｍｍの場合は、ｒ１＝８Ωでｒ２＝９Ωであり、図３（ｃ）で、アンテナ間隔ｈが２０ｍｍの場合は、ｒ１＝４Ωでｒ２＝４Ωである。図３（ｃ）の、アンテナ間隔ｈが２０ｍｍの場合でも、Ｓ２１は−０．３ｄＢであり９２％の電力を伝送できた。 (Inductive resistance value of antenna for matching impedance of power supply circuit 3 and load circuit 4)
Inductive resistances r1 and r2 of this inductive power transmission circuit were obtained by electromagnetic field simulation as follows. That is, the value of the output impedance Z1 of the power supply circuit 3 that transmits the power most efficiently from the power supply circuit 3 to the load circuit 4 is obtained, and the value is the induction resistor r1 of the transmission antenna 1, and the power is also transmitted most efficiently. It is assumed that the value of the load impedance Z2 of the load circuit 4 when transmitting is the induction resistance r2 of the receiving antenna 2. FIG. 3 shows the S parameter (S21) of power transmission from the power supply circuit 3 to the load circuit 4 as a result of simulation by changing the antenna interval h in various ways in dB (decibel). The horizontal axis shows a graph representing the frequency f of the antenna current I1 that the power supply circuit passes through the transmitting antenna 1. 3A shows the case where the antenna interval h in FIG. 1 is 1 mm, FIG. 3B shows the case where h = 10 mm, and FIG. 3C shows the case where h = 20 mm. In FIG. 3A, when the antenna interval h is 1 mm, the induction resistance r1 of the transmission antenna 1 is 20Ω, and the induction resistance r2 of the reception antenna 2 is 23Ω. When the impedance Z of the power supply circuit 3 and the load circuit 4 is made to coincide with the induction resistor r, the antenna resonates and the power transmission efficiency becomes the best, and the power transmission efficiency when the frequency f of the antenna current I1 is 40 MHz. Was close to 100%. The distance between the antennas is set in the vicinity of (wavelength of electromagnetic field at which the antenna resonates) / (2π) or less. In the present embodiment, the wavelength of the electromagnetic field having a frequency f = 40 MHz is a wavelength of about 7.5 m, and even if the antenna distance h is separated by 20 mm, the antenna distance h is (the wavelength of the electromagnetic field at which the antenna resonates) / It is 1/60 of (2π) and is close enough. 3B, when the antenna interval h is 10 mm, r1 = 8Ω and r2 = 9Ω. In FIG. 3C, when the antenna interval h is 20 mm, r1 = 4Ω and r2 = 4Ω. is there. Even in the case where the antenna interval h in FIG. 3C is 20 mm, S21 was −0.3 dB, and 92% of power could be transmitted.

以上の場合は、アンテナ間隔ｈが１ｍｍ、１０ｍｍ、２０ｍｍの場合とも、ほとんど１００％の電力を伝送した。電力伝送効率が１００％となる周波数ｆには周波数帯域（共鳴周波数帯域）の帯域幅があり、それは、図３（ａ）の、アンテナ間隔ｈが１ｍｍの場合は、３５ＭＨｚから５５ＭＨｚまでの周波数帯域であり、約２０ＭＨｚの周波数帯域幅がある。電力伝送効率がほぼ１００％となる共鳴周波数帯域は、アンテナ間隔ｈが大きくなるにつれて狭まり、図３（ｂ）の、アンテナ間隔ｈが１０ｍｍの場合は、共鳴周波数帯域は３６ＭＨｚから４６ＭＨｚまでの約８ＭＨｚの周波数帯域幅になる。図３（ｃ）の、アンテナ間隔ｈが２０ｍｍの場合は、共鳴周波数帯域は３８ＭＨｚから４１ＭＨｚまでの約３ＭＨｚの周波数帯域幅になる。   In the above case, almost 100% of power was transmitted even when the antenna interval h was 1 mm, 10 mm, and 20 mm. The frequency f at which the power transmission efficiency is 100% has a frequency band (resonance frequency band), which is a frequency band from 35 MHz to 55 MHz when the antenna interval h is 1 mm in FIG. And has a frequency bandwidth of about 20 MHz. The resonance frequency band in which the power transmission efficiency is almost 100% narrows as the antenna interval h increases. When the antenna interval h is 10 mm in FIG. 3B, the resonance frequency band is about 8 MHz from 36 MHz to 46 MHz. Frequency bandwidth. When the antenna interval h in FIG. 3C is 20 mm, the resonance frequency band is a frequency bandwidth of about 3 MHz from 38 MHz to 41 MHz.

図３のシミュレーションの結果の共振周波数ｆが４０ＭＨｚであり、送信アンテナ１の両端をつないだ容量Ｃ１が１００ｐＦであることから、送信アンテナ１の実効的自己インダクタンスＬ１が計算でき、Ｌ１は１６０ｎＨである。また、受信アンテナ２の両端をつないだ容量Ｃ２が９０ｐＦであることから、受信アンテナ２の実効的自己インダクタンスＬ２は１８０ｎＨである。   Since the resonance frequency f of the simulation result of FIG. 3 is 40 MHz and the capacitance C1 connecting both ends of the transmission antenna 1 is 100 pF, the effective self-inductance L1 of the transmission antenna 1 can be calculated, and L1 is 160 nH. . Further, since the capacitance C2 connecting both ends of the receiving antenna 2 is 90 pF, the effective self-inductance L2 of the receiving antenna 2 is 180 nH.

図４のグラフは、縦軸にシミュレーションで得た誘導抵抗ｒ１を（２πｆＬ１）で割り算して無次元量にした値ｒ１／（２πｆＬ１）を黒丸印で示し、ｒ２を（２πｆＬ２）で割り算して無次元量にした値ｒ２／（２πｆＬ２）を白丸印で示す。図４のグラフの横軸は、アンテナ間隔ｈを、コイル径Ｄとコイル径Ｇの積の平方根で割り算して無次元量にした（ｈ／√（Ｄ×Ｇ））をあらわす。図４のグラフで、実線は、以下の近似式３６で計算した結果を示す。
ｒ／（２πｆＬ）＝1.8ＥＸＰ（−4.3√（0.04＋√（１／ｔ２−１））） （式３６）
ｔ＝√（Ｄ×Ｇ／（（（Ｄ＋Ｇ）／２）２＋ｈ２）） （式３７）
近似式３６のｔは式３７で計算される値である。図４で、実線で示す近似式３６は、シミュレーション結果と良く一致する。 In the graph of FIG. 4, the value r1 / (2πfL1) obtained by dividing the induction resistance r1 obtained by the simulation by (2πfL1) to a dimensionless amount is indicated by a black circle on the vertical axis, and r2 is divided by (2πfL2). The dimensionless value r2 / (2πfL2) is indicated by a white circle. The horizontal axis of the graph in FIG. 4 represents the antenna interval h divided by the square root of the product of the coil diameter D and the coil diameter G to obtain a dimensionless amount (h / √ (D × G)). In the graph of FIG. 4, the solid line indicates the result calculated by the following approximate expression 36.
r / (2πfL) = 1.8EXP (−4.3√ (0.04 + √ (1 / t 2 −1))) (Formula 36)
t = √ (D × G / (((D + G) / 2) 2 + h 2 )) (Formula 37)
T in the approximate expression 36 is a value calculated by the expression 37. In FIG. 4, the approximate expression 36 indicated by the solid line is in good agreement with the simulation result.

図４の縦軸の値は、式３０と式３１と同じであるから、送信アンテナ１と受信アンテナ２の電磁誘導の結合係数ｋをあらわすことになる。それを確認するため、送信アンテナ１が直径Ｄの円形コイルで受信アンテナ２が直径Ｇの円形コイルの場合の、両コイル間の電磁誘導の結合係数ｋを理論的に厳密に計算し、以下の式３９の係数Ａであらわす式３８を得た。
ｋ＝Ａ×（（−ｔ＋２／ｔ）×Ｋ（ｔ）−（２／ｔ）×Ｅ（ｔ）） （式３８）
Ａ＝μ√（Ｄ×Ｇ／（Ｌ１×Ｌ２））／２ （式３９）
ここで、μは透磁率であり、ｔは式３７で定義し、Ｋ（ｔ）は第１種完全楕円積分関数、Ｅ（ｔ）は第２種完全楕円積分関数である。 Since the values on the vertical axis in FIG. 4 are the same as those in Expression 30 and Expression 31, the electromagnetic induction coupling coefficient k between the transmission antenna 1 and the reception antenna 2 is represented. In order to confirm this, when the transmitting antenna 1 is a circular coil with a diameter D and the receiving antenna 2 is a circular coil with a diameter G, the coupling coefficient k of electromagnetic induction between the two coils is theoretically strictly calculated. Equation 38 expressed by the coefficient A in Equation 39 was obtained.
k = A × ((− t + 2 / t) × K (t) − (2 / t) × E (t)) (Formula 38)
A = μ√ (D × G / (L1 × L2)) / 2 (Formula 39)
Here, μ is the magnetic permeability, t is defined by Equation 37, K (t) is the first type complete elliptic integral function, and E (t) is the second type complete elliptic integral function.

この式から以下の近似式４０が得られる。
ｋ＝4.86Ａ｛1.8ＥＸＰ（−4.3√（0.04＋√（１／ｔ２−１）））｝ （式４０）
式４０の最初の係数の４．８６ＡにＬ１＝１６０ｎＨとＬ２＝１８０ｎＨと真空の透磁率μ＝１．２６μΩ・ｓ／ｍを代入すると、それは約０．８８で１割強程度の誤差で１になる。すなわち、式４０で計算される両コイル間の電磁誘導の結合係数ｋは、１割強の誤差で近似式３６の値に一致する。そのため、図４の縦軸は送信アンテナ１と受信アンテナ２の電磁誘導の結合係数ｋに一致すると言える。そして、シミュレーションから得た誘導抵抗ｒ１とｒ２は式３０と式３１の誘導抵抗ｒに一致すると言える。また、図４のグラフで、実線は、近似式３６で計算した結果であり、それは、ほぼ両アンテナの配線の電磁誘導の結合係数ｋの計算結果をあらわしている。図４において、その結合係数ｋとシミュレーション結果のｒ／（２πｆＬ）は良く一致し、式３０と式３１を満足している。 From this equation, the following approximate equation 40 is obtained.
k = 4.86A {1.8 EXP (−4.3√ (0.04 + √ (1 / t 2 −1)))} (Formula 40)
Substituting L1 = 160 nH, L2 = 180 nH, and vacuum permeability μ = 1.26 μΩ · s / m into the initial coefficient of 4.86 A in Equation 40, it is about 0.88, which is an error of about 10%. become. That is, the coupling coefficient k of the electromagnetic induction between both coils calculated by Expression 40 matches the value of the approximate expression 36 with an error of more than 10%. Therefore, it can be said that the vertical axis in FIG. 4 matches the electromagnetic induction coupling coefficient k of the transmission antenna 1 and the reception antenna 2. Then, it can be said that the induction resistances r1 and r2 obtained from the simulation coincide with the induction resistance r of Expression 30 and Expression 31. Further, in the graph of FIG. 4, the solid line is the result calculated by the approximate expression 36, which almost represents the calculation result of the electromagnetic induction coupling coefficient k of the wirings of both antennas. In FIG. 4, the coupling coefficient k and r / (2πfL) of the simulation result are in good agreement, and Expressions 30 and 31 are satisfied.

（電力伝送効率を飽和させる周波数の帯域幅）
図４から、ｈ＝１ｍｍで（ｈ／√（Ｄ×Ｇ））が０．０２の場合は、結合係数ｋが０．５であり誘導抵抗ｒが約２０Ωであり、ｈ＝１０ｍｍで（ｈ／√（Ｄ×Ｇ））が０．２の場合は結合係数ｋが０．２であり誘導抵抗ｒが８Ωであり、ｈ＝２０ｍｍで（ｈ／√（Ｄ×Ｇ））が０．４の場合は結合係数ｋが０．１であり誘導抵抗ｒが４Ωである。図３（ａ）のグラフは、ｈ＝１ｍｍで結合係数ｋ＝０．５であり誘導抵抗ｒが約２０Ωの場合の電力の伝送効率Ｐｅの周波数特性をＳパラメータＳ２１で示す。図３（ｂ）のグラフは、ｈ＝１０ｍｍで結合係数ｋ＝０．２で誘導抵抗ｒが８Ωの場合を示し、図３（ｃ）のグラフは、ｈ＝２０ｍｍで結合係数ｋ＝０．１で誘導抵抗ｒが４Ωの場合を示す。図３のこれらのグラフでは、電力の伝送効率Ｐｅが飽和する周波数ｆの帯域の上限は、ほぼｆ／ｆｏ＝１／√（１−ｋ）であり、下限は、ほぼｆ／ｆｏ＝１／√（１＋ｋ）になっている。このため、結合係数ｋを大きくすると、電力の伝送効率Ｐｅが飽和する周波数ｆの帯域幅の割合ｆ／ｆｏが結合係数ｋ程度の幅を確保できる効果がある。それゆえ、結合係数ｋを大きくすると、送信アンテナ１から受信アンテナ２への電力伝送効率Ｐｅを飽和させる周波数帯域幅を大きくでき、両アンテナの共振周波数同士を緩い精度で一致させれば十分であり、両アンテナ回路の製造と調整が容易になる効果がある。 (Frequency bandwidth that saturates power transmission efficiency)
From FIG. 4, when h = 1 mm and (h / √ (D × G)) is 0.02, the coupling coefficient k is 0.5, the induction resistance r is about 20Ω, and h = 10 mm (h / √ (D × G)) is 0.2, the coupling coefficient k is 0.2, the induction resistance r is 8Ω, h = 20 mm, and (h / √ (D × G)) is 0.4. In this case, the coupling coefficient k is 0.1 and the induction resistance r is 4Ω. In the graph of FIG. 3A, the frequency characteristic of the power transmission efficiency Pe when h = 1 mm, the coupling coefficient k = 0.5, and the inductive resistance r is about 20Ω is indicated by the S parameter S21. The graph of FIG. 3B shows a case where h = 10 mm, the coupling coefficient k = 0.2, and the induction resistance r is 8Ω, and the graph of FIG. 3C is a graph where h = 20 mm and the coupling coefficient k = 0. 1 and the induction resistance r is 4Ω. In these graphs of FIG. 3, the upper limit of the band of the frequency f at which the power transmission efficiency Pe is saturated is approximately f / fo = 1 / √ (1-k), and the lower limit is approximately f / fo = 1 / k. √ (1 + k). For this reason, when the coupling coefficient k is increased, there is an effect that the bandwidth ratio f / fo of the frequency f at which the power transmission efficiency Pe is saturated can secure a width of about the coupling coefficient k. Therefore, if the coupling coefficient k is increased, the frequency bandwidth for saturating the power transmission efficiency Pe from the transmitting antenna 1 to the receiving antenna 2 can be increased, and it is sufficient to match the resonance frequencies of both antennas with a low degree of accuracy. This has the effect of facilitating the manufacture and adjustment of both antenna circuits.

本実施形態で、電力の伝送効率Ｐｅが飽和する周波数ｆの帯域幅の割合ｆ／ｆｏが結合係数ｋ程度の幅を確保できるので、誘導電力伝送回路の送信アンテナ１と受信アンテナ２の結合係数ｋを０．００４以上に設定して電力を伝送すると、周波数ｆの帯域幅が共振周波数の０．４％以上の幅を確保することができる。そのため、結合係数ｋを０．００４以上にすることが望ましい。そうすれば、誘導電力伝送回路に用いる部品の特性のバラツキを０．４％以内にして送信アンテナ１と受信アンテナ２の共振周波数のバラツキを０．４％以内にすることが比較的容易にできるので、この両アンテナの共振周波数のずれは電力を伝送するのに支障が無い程度の範囲内に収めることができる効果が得られる。   In this embodiment, since the bandwidth ratio f / fo at which the power transmission efficiency Pe saturates can secure a width of about the coupling coefficient k, the coupling coefficient between the transmission antenna 1 and the reception antenna 2 of the inductive power transmission circuit. When power is transmitted with k set to 0.004 or more, the bandwidth of the frequency f can be ensured to be 0.4% or more of the resonance frequency. For this reason, it is desirable that the coupling coefficient k is 0.004 or more. By doing so, it is relatively easy to make the variation in the characteristics of the components used in the inductive power transmission circuit within 0.4% and the variation in the resonance frequency between the transmitting antenna 1 and the receiving antenna 2 within 0.4%. Therefore, it is possible to obtain an effect that the difference between the resonance frequencies of the two antennas can be kept within a range that does not hinder the transmission of power.

このように、本実施形態では、生体外の送信アンテナ１から生体内の受信アンテナ２に、周波数ｆ＝４０ＭＨｚの高周波で電力を伝送し、電源回路３と負荷回路４のインピーダンスＺ１とＺ２を２０Ωから４Ωの誘導抵抗ｒ１とｒ２に整合することで電力を効率良く伝送できる効果がある。この効果は、アンテナ間の距離を（アンテナが共振する電磁界の波長）／（２π）以下の近傍にすることで得られる。本実施形態Ｎ周波数ｆ＝４０ＭＨｚの電磁界の波長は約７．５ｍの波長であり、その周波数ｆで共振する送信アンテナ１と受信アンテナの間隔が２０ｍｍの場合は、その間隔は波長／（２π）の近傍距離の６０分の１であり、十分短い。本実施形態の誘導電力伝送回路は、電力を伝送するために誘導抵抗ｒに整合する電源回路３の出力インピーダンスＺ１及び負荷回路４の入力インピーダンスＺ２は、２０Ωから４Ωで小さいため、所定の電力を伝送するための回路の電圧を低くでき、電力伝送回路の安全性が高い効果がある。本実施形態では、生体の外から、非接触で２０ｍｍの厚さの生体組織を隔てた生体内にも９２％の効率で電力を伝送できる効果がある。また、生体内に埋め込む受信アンテナ２には、縦横５０ｍｍで幅が１ｍｍで厚さが、銅の厚さ５０μｍを２５μｍの厚さの絶縁体で覆った薄いアンテナを用いることができ、生体内でアンテナが占有する体積が小さいので、生体内へ埋め込み易い効果がある。   As described above, in this embodiment, power is transmitted from the transmitting antenna 1 outside the living body to the receiving antenna 2 inside the living body at a high frequency of frequency f = 40 MHz, and the impedances Z1 and Z2 of the power supply circuit 3 and the load circuit 4 are 20Ω. Therefore, there is an effect that power can be efficiently transmitted by matching with the induction resistances r1 and r2 of 4Ω. This effect can be obtained by setting the distance between the antennas to be in the vicinity of (the wavelength of the electromagnetic field at which the antenna resonates) / (2π) or less. In this embodiment, the wavelength of the electromagnetic field having the frequency f = 40 MHz is a wavelength of about 7.5 m, and when the distance between the transmitting antenna 1 and the receiving antenna that resonates at the frequency f is 20 mm, the distance is the wavelength / (2π ) And is sufficiently short. In the inductive power transmission circuit of the present embodiment, the output impedance Z1 of the power supply circuit 3 and the input impedance Z2 of the load circuit 4 matched to the inductive resistor r to transmit power are as small as 20Ω to 4Ω. The voltage of the circuit for transmission can be lowered, and the power transmission circuit is highly safe. In the present embodiment, there is an effect that power can be transmitted with an efficiency of 92% from outside the living body to a living body that is separated from a living tissue having a thickness of 20 mm without contact. The receiving antenna 2 embedded in the living body can be a thin antenna in which the length and width are 50 mm, the width is 1 mm, and the thickness is 50 μm of copper covered with an insulator having a thickness of 25 μm. Since the volume occupied by the antenna is small, there is an effect that the antenna can be easily embedded in the living body.

（変形例１）
変形例１として、生体内に埋め込んだ受信アンテナ２に電力を供給する交流の角周波数ωを低下させる実施形態を説明する。変形例１では、第１の実施形態の送信アンテナ１と受信アンテナ２の端部間容量Ｃ１とＣ２を、ほぼ４倍の、Ｃ１＝４００ｐＦと、Ｃ２＝３６０ｐＦにする。図５のグラフに、縦軸にシミュレーション結果の電力の伝送のＳパラメータ（Ｓ２１）を、横軸を周波数ｆで表す。図５のグラフでは、電力を伝送する共振周波数ｆが第１の実施形態の半分の２０ＭＨｚになった。図６に、縦軸に、変形例１における誘導抵抗ｒを無次元量のｒ／（２πｆＬ）、すなわち結合係数ｋであらわし、横軸をアンテナ間隔ｈを無次元量の（ｈ／√（Ｄ×Ｇ））であらわすグラフを示す。図６も、図４と同様に、黒丸印と白丸印はシミュレーション結果を示し、実線は、近似式３６の計算値を示す。図６でも、シミュレーション結果は近似式３６の計算結果と良く一致した。図６で、ｈ＝１ｍｍで（ｈ／√（Ｄ×Ｇ））が０．０２の場合は、結合係数ｋが０．５であり誘導抵抗ｒが約１０Ωであり、ｈ＝１０ｍｍで（ｈ／√（Ｄ×Ｇ））が０．２の場合は結合係数ｋが０．２であり誘導抵抗ｒが４Ωであり、ｈ＝２０ｍｍで（ｈ／√（Ｄ×Ｇ））が０．４の場合は結合係数ｋが０．１であり誘導抵抗ｒが２Ωであり、共振周波数ｆが半分になることで誘導抵抗ｒが半分になった。 (Modification 1)
As a first modification, an embodiment in which the AC angular frequency ω for supplying power to the receiving antenna 2 embedded in the living body is reduced will be described. In the first modification, the end-to-end capacitances C1 and C2 of the transmission antenna 1 and the reception antenna 2 of the first embodiment are approximately four times C1 = 400 pF and C2 = 360 pF. In the graph of FIG. 5, the vertical axis represents the S parameter (S21) of power transmission as a simulation result, and the horizontal axis represents the frequency f. In the graph of FIG. 5, the resonance frequency f for transmitting electric power is 20 MHz, which is half of the first embodiment. In FIG. 6, the vertical axis represents the inductive resistance r in Modification 1 as a dimensionless amount r / (2πfL), that is, the coupling coefficient k, and the horizontal axis represents the antenna interval h as a dimensionless amount (h / √ (D XG)) represents a graph. In FIG. 6, as in FIG. 4, black circles and white circles indicate simulation results, and a solid line indicates the calculated value of the approximate expression 36. Also in FIG. 6, the simulation result agrees well with the calculation result of the approximate expression 36. In FIG. 6, when h = 1 mm and (h / √ (D × G)) is 0.02, the coupling coefficient k is 0.5, the induction resistance r is about 10Ω, and h = 10 mm (h / √ (D × G)) is 0.2, the coupling coefficient k is 0.2, the induction resistance r is 4Ω, and h = 20 mm and (h / √ (D × G)) is 0.4. In this case, the coupling coefficient k is 0.1, the induction resistance r is 2Ω, and the induction resistance r is halved when the resonance frequency f is halved.

（変形例２）
変形例２として、家屋の壁を隔てて電力を伝送する誘導電力伝送回路の実施形態を説明する。図７のように送信アンテナ１と受信アンテナ２の縦横の径Ｄをともに５０ｍｍの同じ寸法のアンテナにし、アンテナの配線は、厚さが５０μｍのポリイミド膜の上に形成した、幅が１ｍｍで厚さが５０μｍの銅の配線にし、アンテナの両端間の容量Ｃ１とＣ２をともに１００ｐＦにする。アンテナの配線の上は厚さが３０μｍ程度のソルダーレジストを印刷するかポリイミド膜を被せる等で形成した絶縁膜で覆う。この場合において、送信アンテナ１に対して受信アンテナ２を水平面（ＸＹ面）の方向で縦（Ｙ方向）と横（Ｘ方向）にずらすずれ距離ｄを０にした場合をシミュレーションした。その結果、アンテナ電流が３７．４ＭＨｚの周波数ｆで共振することを確認し、アンテナの実効的自己インダクタンスはＬ１＝Ｌ２＝Ｌ＝１７６ｎＨであることがわかった。図８に、縦軸に誘導抵抗ｒを無次元量のｒ／（２πｆＬ）＝結合係数ｋであらわし、横軸にアンテナ間隔ｈを無次元量の（ｈ／Ｄ）であらわすグラフを示す。図８のグラフの横軸は、アンテナの軸方向のアンテナ間隔ｈを２ｍｍから５０ｍｍまで種々に変えた場合をあらわす。図８のグラフの黒丸印はシミュレーションから得たｒをあらわし、実線は、近似式３６の計算結果を示す。シミュレーション結果は近似式３６と良く一致した。 (Modification 2)
As a second modification, an embodiment of an inductive power transmission circuit that transmits power across a wall of a house will be described. As shown in FIG. 7, the transmitting antenna 1 and the receiving antenna 2 have the same vertical and horizontal diameters D of 50 mm, and the antenna wiring is formed on a polyimide film having a thickness of 50 μm. The copper wiring with a thickness of 50 μm is used, and the capacitances C1 and C2 between both ends of the antenna are both set to 100 pF. The antenna wiring is covered with an insulating film formed by printing a solder resist having a thickness of about 30 μm or covering with a polyimide film. In this case, a case was simulated in which the shift distance d for shifting the reception antenna 2 from the transmission antenna 1 in the direction of the horizontal plane (XY plane) in the vertical (Y direction) and horizontal (X direction) was zero. As a result, it was confirmed that the antenna current resonated at a frequency f of 37.4 MHz, and the effective self-inductance of the antenna was found to be L1 = L2 = L = 176 nH. FIG. 8 is a graph in which the vertical axis represents the inductive resistance r as a dimensionless amount r / (2πfL) = coupling coefficient k, and the horizontal axis represents the antenna interval h as a dimensionless amount (h / D). The horizontal axis of the graph of FIG. 8 represents the case where the antenna interval h in the axial direction of the antenna is variously changed from 2 mm to 50 mm. The black circles in the graph of FIG. 8 indicate r obtained from the simulation, and the solid line indicates the calculation result of the approximate expression 36. The simulation result agreed well with the approximate expression 36.

また、図８で、（ｈ／Ｄ）が１の場合に誘導抵抗ｒが０．８Ωになり結合係数ｋが約０．０２になった。また、（ｈ／Ｄ）が２になる場合は、シミュレーションの結果の誘導抵抗ｒは０．２４Ωになり、結合係数ｋは０．００６になった。そして電力伝送効率Ｐｅは２２％あった。結合係数ｋが０．００６程度あれば、先に説明したように電力の伝送効率Ｐｅが飽和する周波数ｆの帯域幅の割合ｆ／ｆｏは結合係数ｋ程度の幅を確保できるため、電力伝送効率Ｐｅの飽和する周波数ｆの帯域幅は共振周波数ｆｏの０．６％程度ある。そのため、０．６％程度の特性のバラツキのある部品を使っても、送信アンテナ１と受信アンテナ２の共振周波数のバラツキを許容範囲内に留めることができる効果がある。このように、直径Ｄのコイルの送信アンテナ１と受信アンテナ２の間隔ｈをアンテナのコイルの直径Ｄの２倍以下にすることで、実用的な誘導電力伝送回路が構成できる効果がある。   In FIG. 8, when (h / D) is 1, the induction resistance r is 0.8Ω, and the coupling coefficient k is about 0.02. When (h / D) is 2, the inductive resistance r as a result of the simulation is 0.24Ω, and the coupling coefficient k is 0.006. The power transmission efficiency Pe was 22%. If the coupling coefficient k is about 0.006, the bandwidth ratio f / fo of the frequency f at which the power transmission efficiency Pe is saturated as described above can secure a width of about the coupling coefficient k. The bandwidth of the frequency f at which Pe is saturated is about 0.6% of the resonance frequency fo. For this reason, even if a component having a characteristic variation of about 0.6% is used, there is an effect that the variation in the resonance frequency of the transmitting antenna 1 and the receiving antenna 2 can be kept within an allowable range. Thus, by setting the distance h between the transmitting antenna 1 and the receiving antenna 2 of the coil having a diameter D to be not more than twice the diameter D of the coil of the antenna, there is an effect that a practical inductive power transmission circuit can be configured.

図９のグラフは、縦軸が、変形例２の幅１ｍｍの銅の配線のアンテナでの、共振周波数ｆの３７ＭＨｚにおける電力伝送効率Ｐｅをあらわす。横軸は（ｈ／Ｄ）をあらわす。図９では、電力伝送効率Ｐｅは（ｈ／Ｄ）が大きくなるとともに低下する。その理由は、（ｈ／Ｄ）が大きくなると結合係数ｋが小さくなり、式２４と式２５であらわされる誘導抵抗ｒ１とｒ２が小さくなり、式２９の電力伝送効率Ｐｅが小さくなるからである。図９では、アンテナ間隔ｈをコイル径Ｄと同じ距離の５０ｍｍ離した場合（ｈ／Ｄ＝１の場合）でも、電力伝送効率が約７５％あり、十分効率良く電力を伝送できた。このように、家屋内の電源回路３から５０ｍｍ程度の厚さの壁を隔ててコイル径Ｄが５０ｍｍの送信アンテナ１と受信アンテナ２を非接触で対向させて、家屋の壁に配線のための孔をあけずに、家屋外の照明装置や表示装置などの負荷回路４に約７５％の効率で屋外に電力を供給する装置が製造できる。   In the graph of FIG. 9, the vertical axis represents the power transmission efficiency Pe at a resonance frequency f of 37 MHz in the antenna of copper wiring having a width of 1 mm according to the second modification. The horizontal axis represents (h / D). In FIG. 9, the power transmission efficiency Pe decreases as (h / D) increases. The reason is that as (h / D) increases, the coupling coefficient k decreases, the induction resistances r1 and r2 expressed by Expression 24 and Expression 25 decrease, and the power transmission efficiency Pe of Expression 29 decreases. In FIG. 9, even when the antenna interval h is 50 mm apart from the coil diameter D (when h / D = 1), the power transmission efficiency is about 75%, and the power can be transmitted sufficiently efficiently. In this way, the transmitting antenna 1 and the receiving antenna 2 having a coil diameter D of 50 mm are opposed to each other in a non-contact manner with a wall having a thickness of about 50 mm from the power supply circuit 3 in the house, so that the wiring is placed on the wall of the house. A device that supplies power to the load circuit 4 such as a lighting device or a display device outside the house at an efficiency of about 75% without making a hole can be manufactured.

なお、アンテナの配線のコイル（螺旋）の巻き数を増して実効的インダクタンスＬを大きくすることにより、アンテナの配線の導通抵抗に比して誘導抵抗ｒを大きくできるので、電力伝送効率Ｐｅの式２９に従って、電力伝送効率Ｐｅを大きくできる効果がある。一方、アンテナの両端を大きな容量Ｃ１とＣ２のコンデンサ（キャパシタンス素子）で接続すると、共振角周波数ωが小さくなり、それにより誘導抵抗ｒが小さくなるので、式２９に従って電力伝送効率Ｐｅが小さくなる。   Note that by increasing the number of turns of the coil (spiral) of the antenna wiring and increasing the effective inductance L, the induction resistance r can be increased as compared to the conduction resistance of the antenna wiring. 29 is effective in increasing the power transmission efficiency Pe. On the other hand, when both ends of the antenna are connected by capacitors (capacitance elements) having large capacitances C1 and C2, the resonance angular frequency ω is decreased, and thereby the induction resistance r is decreased. Therefore, the power transmission efficiency Pe is decreased according to Equation 29.

（変形例３）
変形例３として、家屋の壁を隔てて電力を伝送する誘導電力伝送回路において、電力を伝送する交流の角周波数ωを低下させる実施例を説明する。変形例２の送信アンテナ１と受信アンテナ２に、変形例２の４倍の４００ｐＦの端部間容量Ｃ１とＣ２を設置した場合をシミュレーションし、共振周波数ｆが変形例２の半分の約２０ＭＨｚに低下させることができた。図１０に、縦軸に誘導抵抗ｒを無次元量のｒ／（２πｆＬ）であらわし、横軸にアンテナ間隔ｈを無次元量の（ｈ／Ｄ）であらわすグラフを示す。図１０で、黒丸印はシミュレーション結果の誘導抵抗ｒ＝ｒ１＝ｒ２を示し、実線は近似式３６の結果を示す。図１０でも、シミュレーション結果は近似式３６に良く一致した。 (Modification 3)
As a third modification, a description will be given of an embodiment in which the angular frequency ω of alternating current that transmits power is reduced in an inductive power transmission circuit that transmits power across a wall of a house. The case where 400 pF end-to-end capacitances C1 and C2 that are four times that of the second modification are installed in the transmission antenna 1 and the reception antenna 2 of the second modification is simulated, and the resonance frequency f is about 20 MHz, which is half that of the second modification. It was possible to reduce. FIG. 10 is a graph in which the vertical axis represents the inductive resistance r as a dimensionless amount r / (2πfL), and the horizontal axis represents the antenna interval h as a dimensionless amount (h / D). In FIG. 10, black circles indicate the inductive resistance r = r1 = r2 of the simulation result, and the solid line indicates the result of the approximate expression 36. Also in FIG. 10, the simulation result agreed well with the approximate expression 36.

（変形例４）
変形例４として、家屋の壁等の絶縁体を隔てて電力を伝送する誘導電力伝送回路において、図７の送信アンテナ１と受信アンテナ２の相対位置を縦横にずれ距離ｄずらして用いる誘導電力伝送回路を示す。この誘導電力伝送回路において、送信アンテナ１と受信アンテナ２の端部間容量Ｃ１とＣ２を１００ｐＦに固定し、アンテナ間隔ｈを２ｍｍに固定した場合について、種々のｄの場合についてシミュレーションして以下に説明する回路の整合条件を求めた。この場合のアンテナ回路の共振周波数ｆは、変形例２と同じ３７．４ＭＨｚのｆｏに固定した。図１１に、この共振周波数ｆ＝３７．４ＭＨｚにおけるシミュレーション結果を黒丸印であらわし、縦軸に誘導抵抗ｒを無次元量のｒ／（２πｆＬ）であらわし、横軸にコイルのずれ距離ｄを無次元量（ｄ／Ｄ）であらわすグラフを示す。コイルの位置を水平方向にずらすと、アンテナに現われるインピーダンスｒが低下した。この原因は、コイルの位置をずらすと、コイル同士の電磁誘導の結合係数ｋが小さくなる為であると考える。 (Modification 4)
As a fourth modification, in an inductive power transmission circuit that transmits power across an insulator such as a wall of a house, inductive power transmission is used in which the relative positions of the transmitting antenna 1 and the receiving antenna 2 in FIG. The circuit is shown. In this inductive power transmission circuit, the case where the end-to-end capacitances C1 and C2 of the transmitting antenna 1 and the receiving antenna 2 are fixed to 100 pF and the antenna interval h is fixed to 2 mm is simulated for various d cases. The matching condition of the circuit to be explained was obtained. In this case, the resonance frequency f of the antenna circuit was fixed to fo of 37.4 MHz as in the second modification. In FIG. 11, the simulation results at this resonance frequency f = 37.4 MHz are represented by black circles, the vertical axis represents the inductive resistance r by a dimensionless amount r / (2πfL), and the horizontal axis represents the coil displacement distance d. The graph represented by a dimensional quantity (d / D) is shown. When the position of the coil was shifted in the horizontal direction, the impedance r appearing on the antenna was lowered. This is considered to be because the coupling coefficient k of electromagnetic induction between the coils decreases when the position of the coils is shifted.

図１２に、電源回路３から負荷回路４までの電力伝送効率（縦軸）を、横軸をコイルのずれ距離ｄを無次元量（ｄ／Ｄ）であらわすグラフを示す。図１２から、（ｄ／Ｄ）が０．４以下、すなわち、コイルのずれ距離ｄが２０ｍｍ以下ならば９０％以上の電力伝送効率があり、十分効率良く電力を伝送できる。コイルのずれ距離ｄが（ｄ／Ｄ）＝０．６６になる位置では電力伝送効率が略０になる。この位置では、一方のアンテナのコイルが発生して他方のアンテナのコイルを横切る磁界の方向がアンテナのコイル内の場所により反転して磁界の正負が逆になるため、磁界の総和が０になり、誘導電圧及び結合係数ｋが０になる為と考える。図１２のように、（ｄ／Ｄ）が０．４を超えると電力伝送効率が回復して来る。   FIG. 12 is a graph showing the power transmission efficiency (vertical axis) from the power supply circuit 3 to the load circuit 4, and the horizontal axis representing the coil shift distance d as a dimensionless amount (d / D). From FIG. 12, if (d / D) is 0.4 or less, that is, if the coil shift distance d is 20 mm or less, there is a power transmission efficiency of 90% or more, and power can be transmitted sufficiently efficiently. At a position where the coil shift distance d is (d / D) = 0.66, the power transmission efficiency is substantially zero. At this position, the coil of one antenna is generated and the direction of the magnetic field crossing the coil of the other antenna is reversed depending on the location in the coil of the antenna so that the polarity of the magnetic field is reversed. This is because the induced voltage and the coupling coefficient k become zero. As shown in FIG. 12, when (d / D) exceeds 0.4, the power transmission efficiency is restored.

この送信アンテナ１と受信アンテナ２のコイルの軸を横方向にのみ、アンテナの径Ｄ＝５０ｍｍの２倍の距離の１００ｍｍずらすと、誘導抵抗ｒが０．２３Ωになり、アンテナ系のコイルの結合係数ｋと等しいｒ／（２πｆＬ）が０．００５になり、電力伝送効率Ｐｅが２０％の効率で電力を伝送できる。このように、アンテナのコイルの軸をずらす距離をアンテナの径Ｄの２倍以下にすることで、アンテナの結合係数ｋが０．００５以下になるので、電力伝送効率Ｐｅの飽和する周波数ｆの帯域幅は共振周波数ｆｏの０．５％程度あるので、０．５％程度の特性のバラツキのある部品を使っても、送信アンテナ１と受信アンテナ２の共振周波数のバラツキを許容範囲内に留めることができる。そのため、直径Ｄのコイルの送信アンテナ１と受信アンテナ２の間隔ｄを、コイルの直径Ｄの２倍以下にすることで実用的な誘導電力伝送回路を構成できる。   If the axes of the coils of the transmission antenna 1 and the reception antenna 2 are shifted 100 mm, which is twice the distance of the antenna diameter D = 50 mm, only in the horizontal direction, the induction resistance r becomes 0.23Ω, and the coupling of the antenna system coils R / (2πfL) equal to the coefficient k is 0.005, and the power transmission efficiency Pe can be transmitted with an efficiency of 20%. Since the antenna coupling coefficient k becomes 0.005 or less by making the distance for shifting the axis of the antenna coil less than twice the antenna diameter D in this way, the frequency f at which the power transmission efficiency Pe saturates is reduced. Since the bandwidth is about 0.5% of the resonance frequency fo, the variation in the resonance frequency between the transmission antenna 1 and the reception antenna 2 is kept within the allowable range even if a component having a characteristic variation of about 0.5% is used. be able to. Therefore, a practical inductive power transmission circuit can be configured by setting the distance d between the transmitting antenna 1 and the receiving antenna 2 of the coil having a diameter D to be not more than twice the diameter D of the coil.

なお、送信アンテナ１と受信アンテナ２の結合係数ｋは０．００４以上に限定されず、結合係数ｋがそれより小さい場合も、送信アンテナ１側で送信アンテナ１の共振周波数ｆを受信アンテナ２の共振周波数に同調させる回路を加えれば、効率良く電力を伝送する誘導電力伝送回路を構成できる。   Note that the coupling coefficient k between the transmission antenna 1 and the reception antenna 2 is not limited to 0.004 or more. Even when the coupling coefficient k is smaller than that, the resonance frequency f of the transmission antenna 1 is set to the value of the reception antenna 2 on the transmission antenna 1 side. If a circuit that is tuned to the resonance frequency is added, an inductive power transmission circuit that efficiently transmits power can be configured.

＜第２の実施形態＞
第２の実施形態として、生体外の送信アンテナ１から皮膚を隔てて生体内の受信アンテナ２まで電力を伝送する誘導電力伝送回路において、送信アンテナ１と受信アンテナ２の位置が安定せず電磁誘導の結合係数ｋが変動することに適応して安定して電力を供給する誘導電力伝送回路を示す。この誘導電力伝送回路のアンテナは、図７の送信アンテナ１と受信アンテナ２を用いる。 <Second Embodiment>
As a second embodiment, in the induction power transmission circuit that transmits power from the in vitro transmitting antenna 1 to the receiving antenna 2 in the living body across the skin, the positions of the transmitting antenna 1 and the receiving antenna 2 are not stable and electromagnetic induction is performed. 1 shows an inductive power transmission circuit that stably supplies power in response to fluctuations in the coupling coefficient k. As the antenna of this induction power transmission circuit, the transmission antenna 1 and the reception antenna 2 of FIG. 7 are used.

図１３（ａ）に、図７の送信アンテナ１と受信アンテナ２の間隔ｈ＝１０ｍｍ（ｈ／Ｄ＝０．２）の場合について、電源回路３と負荷回路４のインピーダンスＺを変えた場合の電力伝送効率（％）をあらわす。ただし、図１３（ａ）で、インピーダンスＺをあらわす横軸は、無次元量のＺ／（２πｆＬ）であらわす。図１３（ａ）において、実線は、第２の実施形態以外の場合で、周波数ｆを３７．４ＭＨｚに固定した場合の電力の伝送効率を示し、破線は、第２の実施形態の場合であり、アンテナ電流の周波数ｆを、最大の電力を伝送する周波数ｆに適合させて変えた場合の電力の伝送効率を示す。   FIG. 13A shows a case where the impedance Z between the power supply circuit 3 and the load circuit 4 is changed in the case where the distance h = 10 mm (h / D = 0.2) between the transmission antenna 1 and the reception antenna 2 in FIG. Represents the power transmission efficiency (%). However, in FIG. 13A, the horizontal axis representing the impedance Z is a dimensionless amount of Z / (2πfL). In FIG. 13A, the solid line indicates the power transmission efficiency when the frequency f is fixed to 37.4 MHz in the case other than the second embodiment, and the broken line indicates the case of the second embodiment. The power transmission efficiency when the frequency f of the antenna current is changed in conformity with the frequency f at which the maximum power is transmitted is shown.

図１３（ｂ）には、図１３（ａ）の横軸の特定のインピーダンスＺに関して、そのインピーダンスＺでの誘導電力伝送回路の電力の伝送のＳパラメータ（Ｓ２１）をｄＢ表示であらわす。図１３（ｂ）の横軸は、アンテナ電流の周波数ｆをあらわす。図１３（ａ）で、Ｚ／（２πｆＬ）が０．２２、すなわちＺが９Ω、より大きい場合は、電力の伝送効率は低下する。０．２２は電磁誘導の結合係数ｋである。一方、Ｚ／（２πｆＬ）が結合係数ｋの０．２２、すなわちＺが９Ω、より小さい場合は、以下の２つの場合に分かれる。伝送する電力の周波数ｆを３７．４ＭＨｚに固定する場合は、電力の伝送効率は、電源回路３および負荷回路４のインピーダンスＺが誘導抵抗ｒより低下するにつれて低下する。一方、アンテナ電流の周波数ｆを、最大の電力を伝送する周波数ｆに適合させて変える場合は、図１３（ａ）の破線で示すように、電力の伝送効率はほとんど低下しない。第２の実施形態では、そのように、周波数ｆを、最大の電力を伝送する周波数ｆに適合させて変える。   In FIG. 13B, regarding the specific impedance Z on the horizontal axis of FIG. 13A, the S parameter (S21) of power transmission of the inductive power transmission circuit at the impedance Z is expressed in dB. The horizontal axis in FIG. 13B represents the frequency f of the antenna current. In FIG. 13A, when Z / (2πfL) is 0.22, that is, Z is larger than 9Ω, the power transmission efficiency decreases. 0.22 is a coupling coefficient k of electromagnetic induction. On the other hand, when Z / (2πfL) is 0.22 of the coupling coefficient k, that is, Z is smaller than 9Ω, it is divided into the following two cases. When the frequency f of the power to be transmitted is fixed to 37.4 MHz, the power transmission efficiency decreases as the impedance Z of the power supply circuit 3 and the load circuit 4 decreases below the induction resistance r. On the other hand, when the frequency f of the antenna current is changed in conformity with the frequency f at which the maximum power is transmitted, the power transmission efficiency is hardly lowered as shown by the broken line in FIG. In the second embodiment, as described above, the frequency f is changed in conformity with the frequency f at which the maximum power is transmitted.

第２の実施形態の誘導電力伝送回路は、本発明の原理の説明におけるωがωｏ以外の場合である第２の場合を利用する。すなわち、送信アンテナ１と受信アンテナ２の位置が安定せず電磁誘導の結合係数ｋが変動する場合に適応する誘導電力伝送回路として、誘導抵抗ｒ１の上限の値より小さな値に出力インピーダンスＺ１を固定した電源回路３と、Ｚ１×（Ｌ２／Ｌ１）の値に入力インピーダンスＺ２を固定した負荷回路４を用いる。そして、送信アンテナ１と受信アンテナ２の位置の変化に伴う結合係数ｋの値の変化に応じて電源回路３の電流の共振角周波数ωを適応させて変えて両アンテナを共鳴させる共鳴周波数調整回路を電源回路３に設置する。その共鳴周波数調整回路は、送信アンテナ１の共振電流を正帰還回路により電源回路３に正帰還させて、その電流を増幅して電源回路３から出力させる回路で構成する。これにより、電源回路３から共振角周波数ωに適応した周波数の電流Ｉ１を取り出す。すなわち、電力を伝送する交流の角周波数ωを、Ｌ１×Ｃ１×（１＋ｋ・ｃｏｓ（β））の値の逆数の平方根に適応させて、電力を最大の効率で伝送する誘導電力伝送回路である。   The inductive power transmission circuit according to the second embodiment uses the second case where ω is other than ωo in the description of the principle of the present invention. In other words, the output impedance Z1 is fixed to a value smaller than the upper limit value of the induction resistance r1 as an induction power transmission circuit adapted to the case where the positions of the transmission antenna 1 and the reception antenna 2 are not stable and the coupling coefficient k of electromagnetic induction varies. And the load circuit 4 in which the input impedance Z2 is fixed to the value of Z1 × (L2 / L1). Then, a resonance frequency adjustment circuit that resonates both antennas by adaptively changing the resonance angular frequency ω of the current of the power supply circuit 3 in accordance with a change in the value of the coupling coefficient k accompanying a change in the position of the transmitting antenna 1 and the receiving antenna 2. Is installed in the power supply circuit 3. The resonance frequency adjustment circuit is configured by a circuit that positively feeds back the resonance current of the transmission antenna 1 to the power supply circuit 3 by a positive feedback circuit, and amplifies the current to output from the power supply circuit 3. As a result, a current I1 having a frequency adapted to the resonance angular frequency ω is extracted from the power supply circuit 3. That is, the induction power transmission circuit transmits the power with the maximum efficiency by adapting the angular frequency ω of the alternating current for transmitting the power to the square root of the reciprocal of the value of L1 × C1 × (1 + k · cos (β)). .

これにより、第２の実施形態は、ωをωｏ以外にし、式２０から式３１の関係を成り立たせる。式２８に示すωが式１９の値ωｏからずれるに従い、すなわち、βがπ／２からずれるに従い、ｃｏｓ（β）を０から±１にまで変え、式２８の共鳴角周波数ωをωｏから式４１の値にまで変える。
ω≒ωｏ／√（１±ｋ） （式４１）
そして、ｓｉｎ（β）を１より小さくし０に近づけ、式２４と式２５に示す誘導抵抗ｒ１とｒ２を上限値のｋωＬ１およびｋωＬ２より小さくする。このように、第２の実施形態は、共鳴角周波数ωがωｏからずれた値で、この式２４と式２５に示す小さな誘導抵抗ｒ１とｒ２に電源回路３の出力インピーダンスＺ１と、負荷回路４の負荷インピーダンスＺ２を等しくすることにより、電力を完全な効率で伝送できる。この場合に、誘導抵抗ｒ１とｒ２の比は実効的自己インダクタンスＬ１とＬ２の比になる。このように、第２の実施形態の誘導電力伝送回路は、アンテナ間の距離が変化し安定しない場合でも、その変化に適応してアンテナ回路を共鳴させ、完全な効率の電力伝送を維持させることができる効果がある。 Thus, in the second embodiment, ω is set to other than ωo, and the relationship of Expressions 20 to 31 is established. As ω shown in Equation 28 deviates from the value ωo in Equation 19, that is, as β deviates from π / 2, cos (β) is changed from 0 to ± 1, and the resonance angular frequency ω in Equation 28 is changed from ωo to Equation (28). Change to a value of 41.
ω≈ωo / √ (1 ± k) (Formula 41)
Then, sin (β) is made smaller than 1 and brought close to 0, and the induction resistances r1 and r2 shown in Expression 24 and Expression 25 are made smaller than the upper limit values kωL1 and kωL2. As described above, in the second embodiment, the resonance angular frequency ω is deviated from ωo, and the output impedance Z1 of the power supply circuit 3 and the load circuit 4 are added to the small inductive resistances r1 and r2 shown in Expression 24 and Expression 25. By making the load impedances Z2 equal, power can be transmitted with complete efficiency. In this case, the ratio of the induction resistances r1 and r2 is the ratio of the effective self-inductances L1 and L2. Thus, even when the distance between the antennas changes and is not stable, the inductive power transmission circuit of the second embodiment adapts to the change and resonates the antenna circuit to maintain perfect efficiency power transmission. There is an effect that can.

＜第３の実施形態＞
第３の実施形態として、家屋の壁を隔てて電力を伝送する誘導電力伝送回路で、アンテナの渦巻き配線の巻数を増すことでアンテナの誘導抵抗を大きくし、その誘導抵抗を電源回路３から送信アンテナ１までの電力の給電線の特性インピーダンスに近づける誘導電力伝送回路の実施形態を示す。図１４（ａ）に第３の実施形態の誘導電力伝送回路の送信アンテナ１と受信アンテナ２の平面図を示し、図１４（ｂ）に側面図を示す。第１の実施形態と同様に、送信アンテナ１に電源回路３を接続し、受信アンテナ２に負荷回路４を接続する。図１４では、送信アンテナ１として、厚さ５０μｍのポリイミドフィルム上に幅が１ｍｍで厚さが５０μｍの銅の配線で形成したコイル径Ｄが５４ｍｍの３巻のコイルのアンテナの両端を２８０ｐＦの端部間容量Ｃ１でつなぎ、アンテナの中間に電源回路３から給電する端子（ポート１）を設置した。受信アンテナ２は、送信アンテナ１と同じ形で同じ寸法のアンテナとし、縦横５４ｍｍの３巻のコイル状アンテナの両端を２８０ｐＦの端部間容量Ｃ２でつなぎ、そのアンテナの中間に負荷回路４の端子（ポート２）を設置した。送信アンテナ１のコイルと受信アンテナ２のコイルは、図１４（ｂ）の側面図のように、コイル面を平行にし、コイル面（ＸＹ面）に垂直方向のアンテナ間隔ｈだけ離して配置し、更に、両者のコイルの軸を横（Ｘ）方向にのみ７ｍｍのずれ距離ｄでずらして、縦（Ｙ）方向にはずらさず配置した。 <Third Embodiment>
In an inductive power transmission circuit that transmits power across a wall of a house as a third embodiment, the inductive resistance of the antenna is increased by increasing the number of turns of the spiral wiring of the antenna, and the inductive resistance is transmitted from the power supply circuit 3 3 shows an embodiment of an inductive power transmission circuit that approximates the characteristic impedance of the power feed line to the antenna 1. FIG. 14A shows a plan view of the transmitting antenna 1 and the receiving antenna 2 of the inductive power transmission circuit of the third embodiment, and FIG. 14B shows a side view. As in the first embodiment, the power supply circuit 3 is connected to the transmission antenna 1 and the load circuit 4 is connected to the reception antenna 2. In FIG. 14, as the transmitting antenna 1, both ends of the antenna of a three-coil coil having a coil diameter D of 54 mm and formed of copper wiring having a width of 1 mm and a thickness of 50 μm on a polyimide film having a thickness of 50 μm are connected to ends of 280 pF. A terminal (port 1) for supplying power from the power supply circuit 3 was installed in the middle of the antenna by connecting with the inter-unit capacitance C1. The receiving antenna 2 is an antenna having the same shape and the same dimensions as the transmitting antenna 1, and both ends of a coiled antenna having a length and width of 54 mm are connected by an end-to-end capacitance C2 of 280 pF, and a terminal of the load circuit 4 is interposed between the antennas. (Port 2) was installed. The coil of the transmitting antenna 1 and the coil of the receiving antenna 2 are arranged so that the coil surfaces are parallel and separated from the coil surface (XY plane) by the antenna interval h in the vertical direction, as shown in the side view of FIG. Further, the axes of both coils were shifted only in the horizontal (X) direction by a shift distance d of 7 mm, and were not shifted in the vertical (Y) direction.

シミュレーションの結果、共振周波数ｆは９ＭＨｚになり、送信アンテナ１の実効的自己インダクタンスＬ１と受信アンテナ２の実効的自己インダクタンスＬ２は、ともに１．２μＨであった。本実施形態では、送信アンテナ１と受信アンテナ２は、コイルが３巻であり、第１の実施形態のコイルの巻き数の３倍あるので、このコイルのアンテナの実効的自己インダクタンスＬ＝Ｌ１＝Ｌ２はコイルの巻き数の二乗に近い約７倍に大きくなった。そして、共振周波数ｆ＝９ＭＨｚの前後に、図３と同様な形のグラフで電力の伝送効率の周波数特性グラフを得た。この実施形態の実効的自己インダクタンスＬは第１の実施形態の７倍になったので、共振の周波数ｆが第１の実施形態の４０ＭＨｚから９ＭＨｚに下がっても、誘導抵抗ｒの値ｋωＬは第１の実施形態より大きくなり、それに整合させる電源回路３及び負荷回路４のインピーダンスＺを大きくできる。そして、誘導抵抗ｒが大きくなると、式２９であらわせる電力伝送効率Ｐｅを向上させる効果がある。   As a result of the simulation, the resonance frequency f was 9 MHz, and the effective self-inductance L1 of the transmitting antenna 1 and the effective self-inductance L2 of the receiving antenna 2 were both 1.2 μH. In the present embodiment, the transmitting antenna 1 and the receiving antenna 2 have three coils, which is three times the number of turns of the coil of the first embodiment, so that the effective self-inductance L = L1 = L2 became about 7 times as large as the square of the number of turns of the coil. Then, before and after the resonance frequency f = 9 MHz, a frequency characteristic graph of power transmission efficiency was obtained with a graph having the same shape as in FIG. Since the effective self-inductance L of this embodiment is seven times that of the first embodiment, the value kωL of the inductive resistance r remains the same even when the resonance frequency f decreases from 40 MHz to 9 MHz of the first embodiment. The impedance Z of the power supply circuit 3 and the load circuit 4 matched with that of the first embodiment can be increased. When the induction resistance r increases, there is an effect of improving the power transmission efficiency Pe expressed by Equation 29.

図１５（ａ）に、第３の実施形態の共振周波数ｆ＝９ＭＨｚにおける誘導抵抗ｒを無次元量のｒ／（２πｆＬ）にして縦軸であらわし、横軸にアンテナ間隔ｈを無次元量の（ｈ／Ｄ）であらわすグラフを示す。図１５（ａ）で、黒丸印はシミュレーション結果を示し、実線は、近似式３６の値を示す。図１５（ａ）は、シミュレーション結果は近似式３６に概ね一致した。図１５（ｂ）に、縦軸に電源回路３から負荷回路４までの電力伝送効率をあらわし、横軸にアンテナ間隔ｈを無次元量（ｈ／Ｄ）であらわすグラフを示す。図１５（ｂ）から、アンテナ間隔ｈをコイル径Ｄの６割程度の約３０ｍｍ離した場合（ｈ／Ｄ＝０．６の場合）でも、電力伝送効率が約９０％あり十分効率良く電力を伝送できる。   In FIG. 15A, the inductive resistance r at the resonance frequency f = 9 MHz of the third embodiment is expressed as a dimensionless amount r / (2πfL), and the abscissa represents the antenna interval h as the dimensionless amount. The graph represented by (h / D) is shown. In FIG. 15A, the black circles indicate the simulation results, and the solid line indicates the value of the approximate expression 36. In FIG. 15 (a), the simulation result almost coincides with the approximate expression 36. FIG. 15B shows a graph in which the vertical axis represents the power transmission efficiency from the power supply circuit 3 to the load circuit 4, and the horizontal axis represents the antenna interval h as a dimensionless amount (h / D). From FIG. 15 (b), even when the antenna interval h is about 30 mm, which is about 60% of the coil diameter D (when h / D = 0.6), the power transmission efficiency is about 90% and the power can be sufficiently efficiently supplied. Can be transmitted.

（変形例５）
変形例５として、第３の実施形態の送信アンテナ１と受信アンテナ２それぞれの両端をつなぐ端部間容量Ｃ１とＣ２の容量を第３の実施形態の容量の１６分の１の１７ｐＦにして共振周波数ｆを高くした場合を示す。変形例５の場合は共振周波数ｆ＝３５ＭＨｚで共振した。図１６（ａ）に、その場合の誘導抵抗ｒを無次元量のｒ／（２πｆＬ）で縦軸であらわし、横軸にアンテナ間隔ｈを無次元量の（ｈ／Ｄ）であらわすグラフを示す。このように、端部間容量Ｃ１とＣ２の容量を約１６分の１に小さくすると、共振周波数ｆは約４倍の３５ＭＨｚに大きくなった。図１６（ａ）は、黒丸印で示すシミュレーション結果は実線で示す近似式３６に概ね一致した。図１６（ｂ）に、縦軸に電源回路３から負荷回路４までの電力伝送効率をあらわし、横軸にアンテナ間隔ｈを無次元量（ｈ／Ｄ）であらわすグラフを示す。図１６（ｂ）から、アンテナ間隔ｈをコイル径Ｄの８割程度の約４０ｍｍ離した場合（ｈ／Ｄ＝０．８の場合）でも、電力伝送効率が約９０％あり十分効率良く電力を伝送できる。図１６（ａ）から、アンテナ間隔ｈを１ｍｍ離して（ｈ／Ｄ）を約０．６にする場合に、ｒ／（２πｆＬ）＝結合係数ｋが０．６になり誘導抵抗ｒが１５８Ωに高くなる。この実施形態の実効的インダクタンスＬは第１の実施形態の７倍になったので、第１の実施形態の４０ＭＨｚとほぼ同じ周波数ｆの３５ＭＨｚの場合では、誘導抵抗ｒの値ｋωＬは第１の実施形態の７倍に大きくなり、それに整合させる電源回路３及び負荷回路４のインピーダンスＺが７倍に大きくなる。そして、誘導抵抗ｒが大きくなるので式２９であらわせる電力伝送効率Ｐｅを向上させる効果がある。 (Modification 5)
As a fifth modification, the capacitance of the end-to-end capacitances C1 and C2 connecting both ends of the transmission antenna 1 and the reception antenna 2 of the third embodiment is set to 17 pF, which is 1/16 of the capacitance of the third embodiment. A case where the frequency f is increased is shown. In the case of the modified example 5, resonance occurred at a resonance frequency f = 35 MHz. FIG. 16A shows a graph in which the inductive resistance r in that case is represented by a dimensionless amount r / (2πfL) on the vertical axis, and the antenna interval h is represented by a dimensionless amount (h / D) on the horizontal axis. . As described above, when the end-to-end capacities C1 and C2 were reduced to about 1/16, the resonance frequency f was increased to about 4 times 35 MHz. In FIG. 16 (a), the simulation result indicated by the black circle generally matches the approximate expression 36 indicated by the solid line. FIG. 16B shows a graph in which the vertical axis represents the power transmission efficiency from the power supply circuit 3 to the load circuit 4 and the horizontal axis represents the antenna interval h as a dimensionless amount (h / D). From FIG. 16 (b), even when the antenna interval h is about 40 mm, which is about 80% of the coil diameter D (when h / D = 0.8), the power transmission efficiency is about 90%, and the power can be sufficiently efficiently supplied. Can be transmitted. From FIG. 16A, when the antenna interval h is 1 mm apart and (h / D) is set to about 0.6, r / (2πfL) = coupling coefficient k becomes 0.6 and the induction resistance r becomes 158Ω. Get higher. Since the effective inductance L of this embodiment is seven times that of the first embodiment, the value kωL of the inductive resistance r is the first value in the case of 35 MHz, which is substantially the same frequency f as 40 MHz of the first embodiment. The impedance Z of the power supply circuit 3 and the load circuit 4 to be matched is increased seven times as much as that in the embodiment. And since the induction resistance r becomes large, there is an effect of improving the power transmission efficiency Pe expressed by the equation 29.

＜第４の実施形態＞
第４の実施形態として、第３の実施形態のアンテナの両端に外部コンデンサを加えずに共振周波数ｆを高くすることでアンテナの誘導抵抗を高くし、電源回路３から送信アンテナ１までの電力の給電線に特性インピーダンスの高い給電線を用いる場合を示す。すなわち、第４の実施形態では、図１４の送信アンテナ１と受信アンテナ２の両端を開放し、アンテナの端部間を寄生容量のみでつなぎ、外部コンデンサは加えない。第４の実施形態の誘導電力伝送回路のアンテナ系は、周波数ｆ＝１５４ＭＨｚで共振する。共振周波数ｆ＝１５４ＭＨｚと、外部コンデンサを加えた場合に得たアンテナのコイルの実効的自己インダクタンスＬ＝１．２μＨとから計算すると、このアンテナのコイルの両端は実効的に約１ｐＦの寄生容量でつながれていることがわかる。このため、アンテナのコイル端を開放した状態でも、図２の回路図で送信アンテナ１と受信アンテナ２それぞれのアンテナの両端に約１ｐＦのコンデンサをつないだ回路とみなせる。 <Fourth Embodiment>
As a fourth embodiment, the induction resistance of the antenna is increased by increasing the resonance frequency f without adding an external capacitor to both ends of the antenna of the third embodiment, and the power from the power supply circuit 3 to the transmission antenna 1 is increased. A case where a power supply line having high characteristic impedance is used as the power supply line is shown. That is, in the fourth embodiment, both ends of the transmission antenna 1 and the reception antenna 2 in FIG. 14 are opened, the end portions of the antenna are connected only by the parasitic capacitance, and no external capacitor is added. The antenna system of the inductive power transmission circuit according to the fourth embodiment resonates at a frequency f = 154 MHz. When calculated from the resonance frequency f = 154 MHz and the effective self-inductance L = 1.2 μH of the antenna coil obtained when an external capacitor is added, both ends of this antenna coil are effectively about 1 pF of parasitic capacitance. You can see that they are connected. For this reason, even when the coil end of the antenna is opened, it can be regarded as a circuit in which a capacitor of about 1 pF is connected to both ends of the antennas of the transmitting antenna 1 and the receiving antenna 2 in the circuit diagram of FIG.

図１７（ａ）に、第４の実施形態の共振周波数ｆ＝１５４ＭＨｚにおける誘導抵抗ｒを無次元量のｒ／（２πｆＬ）で縦軸であらわし、横軸にアンテナ間隔ｈを無次元量の（ｈ／Ｄ）であらわすグラフを示す。ここでｒ／（２πｆＬ）を計算する基礎にするＬの値としては、外部コンデンサを加えた場合に得たアンテナのコイルの実効的自己インダクタンスＬ＝１．２μＨを用いた。黒丸印はシミュレーション結果を示し、点線は近似式３６の値×（２／π）を示す。アンテナ端部間に寄生容量以外には外部コンデンサを加えずアンテナ端を開放した場合は、アンテナ電流分布がアンテナの端部間を個別のコンデンサで接続した場合と異なり、アンテナの端部に近づくにつれアンテナ電流が小さくなる。そして、アンテナ電流の平均値がアンテナの中間のポートでの電流値の（２／π）倍になる。そのため、点線のグラフは、アンテナ電流の平均値のポート電流に対する減少分を補正するために近似式３６の右辺の式に（２／π）を掛け算した値を示す。その補正結果の値がシミュレーション結果に一致した。この補正が必要になった理由は、アンテナ端を開放した場合のアンテナのコイルの実効的自己インダクタンスＬは、外部コンデンサを加えた場合に得たアンテナのコイルの実効的自己インダクタンスＬとは変わってしまうことに依ると考える。近似式３６に用いるＬは、アンテナ電流の分布に応じて変化する実行的自己インダクタンスＬを用いるべきである。このアンテナでアンテナ端を開放した場合の実効的インダクタンスＬの値は、アンテナ端に大きな容量を接続した場合の実効的インダクタンスの値の概ね（２／π）倍になるので、近似式３６を用いる場合にＬの値として、外部コンデンサを加えた場合に得た実効的自己インダクタンスＬを用いる場合は、計算結果を補正する必要がある考える。   In FIG. 17A, the inductive resistance r at the resonance frequency f = 154 MHz of the fourth embodiment is represented by a dimensionless amount r / (2πfL) on the vertical axis, and the antenna interval h is represented by a dimensionless amount ( A graph represented by h / D) is shown. Here, as the value of L used as a basis for calculating r / (2πfL), the effective self-inductance L = 1.2 μH of the coil of the antenna obtained when an external capacitor is added was used. Black circles indicate simulation results, and dotted lines indicate the value of the approximate expression 36 × (2 / π). When the antenna end is opened without adding an external capacitor other than the parasitic capacitance between the antenna ends, the antenna current distribution differs from the case where the antenna ends are connected by individual capacitors, as the antenna ends are approached. The antenna current is reduced. The average value of the antenna current is (2 / π) times the current value at the intermediate port of the antenna. Therefore, the dotted line graph represents a value obtained by multiplying the expression on the right side of the approximate expression 36 by (2 / π) in order to correct the decrease of the average value of the antenna current with respect to the port current. The value of the correction result coincided with the simulation result. The reason why this correction is necessary is that the effective self-inductance L of the antenna coil when the antenna end is opened is different from the effective self-inductance L of the antenna coil obtained when an external capacitor is added. I think that it depends on. The effective self-inductance L that changes according to the distribution of the antenna current should be used as L used in the approximate expression 36. Since the effective inductance L when the antenna end is opened with this antenna is approximately (2 / π) times the effective inductance when a large capacitance is connected to the antenna end, the approximate expression 36 is used. In this case, when the effective self-inductance L obtained when an external capacitor is added is used as the value of L, the calculation result needs to be corrected.

図１７（ａ）から、アンテナ間隔ｈを４ｍｍ離して（ｈ／Ｄ）を０．０７４にする場合に、ｒ／（２πｆＬ）が０．２２になり誘導抵抗ｒが２６０Ωになる。第４の実施形態では、アンテナ間隔ｈを約４ｍｍ離すことで、誘導抵抗ｒを約２６０Ωにする。この誘導抵抗ｒには、平行２線の給電線の特性インピーダンスが良く整合する。平行２線の給電線の特性インピーダンスは、給電線の２線間の距離が給電線の半径の１０倍の場合は給電線の特性インピーダンスが２７７Ωになり２６０Ωの誘導抵抗ｒに近いからである。第４の実施形態では、誘導抵抗ｒを高くしそれに整合する電源回路３と給電線の特性インピーダンスを高くしたので、電力を伝送するための電流が少なくなり給電線の導体抵抗に電流が流れることによる損失を少なくできる効果がある。   From FIG. 17A, when the antenna interval h is 4 mm apart and (h / D) is 0.074, r / (2πfL) is 0.22, and the induction resistance r is 260Ω. In the fourth embodiment, the induction resistance r is set to about 260Ω by separating the antenna interval h by about 4 mm. The characteristic impedance of the parallel two-wire feed line is well matched with the induction resistance r. This is because the characteristic impedance of the two parallel feed lines is close to the induction resistance r of 260Ω when the distance between the two feed lines is 10 times the radius of the feed line and the characteristic impedance of the feed line is 277Ω. In the fourth embodiment, the inductive resistance r is increased and the characteristic impedance of the power supply circuit 3 and the feeder line matched therewith is increased, so that the current for transmitting power is reduced and the current flows through the conductor resistance of the feeder line. There is an effect that the loss due to can be reduced.

図１７（ｂ）に、縦軸に電源回路３から負荷回路４までの電力伝送効率をあらわし、横軸にアンテナ間隔ｈを無次元量（ｈ／Ｄ）であらわすグラフを示す。図１７（ｂ）から、アンテナ間隔ｈをコイル径Ｄの８割（ｈ／Ｄ＝０．８）の４３ｍｍ程度離した場合でも、電力伝送効率が約９０％あり十分効率良く電力を伝送できる。   FIG. 17B is a graph in which the vertical axis represents the power transmission efficiency from the power supply circuit 3 to the load circuit 4 and the horizontal axis represents the antenna interval h as a dimensionless amount (h / D). From FIG. 17B, even when the antenna interval h is about 43 mm, which is 80% of the coil diameter D (h / D = 0.8), the power transmission efficiency is about 90% and power can be transmitted sufficiently efficiently.

＜第５の実施形態＞
第５の実施形態は、本発明の原理の第２の場合を利用して、変成器を構成する誘導電力伝送回路を示す。すなわち、図２（ａ）の回路図で、空芯コイルの送信アンテナ１と受信アンテナ２を共鳴電磁界の波長の２π分の１より近い距離に設置し、コイルの巻き数を変えて実効的自己インダクタンスＬ１とＬ２を変えることで、電源回路３の出力インピーダンスＺ１を、（Ｌ２／Ｌ１）倍の、負荷回路４の負荷インピーダンスＺ２に変換する変成器を構成する誘導電力伝送回路を示す。第５の実施形態は、共振角周波数ωがωｏと異なる場合の本発明の原理の第２の場合には、アンテナのコイルの誘導抵抗ｒが、式２４と式２５に従って、アンテナの配線のコイル（巻線）の実効的自己インダクタンスＬに比例して変わり、その比例係数は結合係数ｋと２πｆとｓｉｎ（β）の積であることを利用した。この変成器は、送信アンテナ１側に接続した電源回路３のｋωＬ１以下のインピーダンスを、受信アンテナ側で見るとｋωＬ２以下の出力インピーダンスに変換できる効果がある。本実施形態で、ポリイミドフィルム上に銅の配線をエッチングして螺旋状のパターン形成した空芯コイルの送信アンテナ１と受信アンテナ２の間隔に磁性体を設けず、アンテナ同士を空気中（あるいは絶縁体中）で対向させ近づけると、結合係数ｋが大きくなり、インピーダンス変換できるインピーダンスの値の上限が大きくなるので、両アンテナを空気中（あるいは絶縁樹脂などの絶縁体中）で近づけることが望ましい。この変成器では、送信アンテナ１の実効的自己インダクタンスＬ１と受信アンテナ２の実効的自己インダクタンスＬ２を、アンテナのコイルの巻数を変えて調整することでインピーダンスの変換率（Ｌ２／Ｌ１）を調整する。アンテナの配線のコイルの実効的自己インダクタンスＬは、概ね巻数の二乗に比例して変わる。このように、第５の実施形態の変成器は、異なるインピーダンスＺ１を持つ電源回路３と負荷回路４のインピーダンスＺ２を変換して、電源回路３から負荷回路４へ略１００％の電力を伝送する誘導電力伝送回路である。 <Fifth Embodiment>
The fifth embodiment shows an inductive power transmission circuit constituting a transformer by using the second case of the principle of the present invention. In other words, in the circuit diagram of FIG. 2 (a), the transmitting antenna 1 and the receiving antenna 2 of the air-core coil are installed at a distance closer than 1 / 2π of the wavelength of the resonance electromagnetic field, and the number of turns of the coil is changed. An inductive power transmission circuit constituting a transformer that converts the output impedance Z1 of the power supply circuit 3 into the load impedance Z2 of the load circuit 4 by changing the self-inductances L1 and L2 is shown. According to the fifth embodiment, in the second case of the principle of the present invention when the resonance angular frequency ω is different from ωo, the induction resistance r of the antenna coil is determined by the coil of the antenna wiring according to Equations 24 and 25. It changed in proportion to the effective self-inductance L of the (winding), and the proportionality coefficient used was the product of the coupling coefficient k, 2πf and sin (β). This transformer has an effect that the impedance of kωL1 or less of the power supply circuit 3 connected to the transmitting antenna 1 side can be converted to an output impedance of kωL2 or less when viewed on the receiving antenna side. In this embodiment, a magnetic material is not provided in the space between the transmitting antenna 1 and the receiving antenna 2 of the air-core coil formed by etching a copper wiring on the polyimide film to form a spiral pattern. Since the coupling coefficient k increases and the upper limit of the impedance value that can be converted into impedance increases, it is desirable to bring both antennas close to each other in the air (or in an insulator such as an insulating resin). In this transformer, the impedance conversion rate (L2 / L1) is adjusted by adjusting the effective self-inductance L1 of the transmitting antenna 1 and the effective self-inductance L2 of the receiving antenna 2 by changing the number of turns of the coil of the antenna. . The effective self-inductance L of the antenna wiring coil changes approximately in proportion to the square of the number of turns. As described above, the transformer according to the fifth embodiment converts the impedance Z2 of the power circuit 3 and the load circuit 4 having different impedances Z1, and transmits approximately 100% of the power from the power circuit 3 to the load circuit 4. Inductive power transmission circuit.

第５の実施形態の変成器は電源回路３の出力インピーダンスＺ１を送信アンテナ１の誘導抵抗ｒ１に一致させ、負荷回路４の入力インピーダンスＺ２を受信アンテナ２の誘導抵抗ｒ２に一致させる。誘導抵抗ｒ１とｒ２は式２４と式２５のｓｉｎ（β）を小さくすることで小さくしてインピーダンスＺ１とＺ２に整合させる。その整合条件を満たす共鳴の角周波数ωはそのβの値に従って、式２８であらわされるように、ωｏから式４１の値（ωｏ／√（１±ｋ））にまで変わる。式２８の共振角周波数ωは、ｃｏｓ（β）が負の共鳴条件の場合は、ωｏより高い角周波数にシフトし、図１３（ｂ）の２山のグラフの右の山を形成する。受信アンテナ２の電流Ｉ２は送信アンテナ１の電流Ｉ１を打ち消すように逆向きに流れ、送信アンテナ１と受信アンテナ２の総体のアンテナ系が外部に発生する電磁界が小さくなり、この変成器の発生する不要電磁波ノイズ（ＥＭＩ）が小さくなる効果がある。   The transformer according to the fifth embodiment makes the output impedance Z1 of the power supply circuit 3 coincide with the induction resistance r1 of the transmission antenna 1, and makes the input impedance Z2 of the load circuit 4 coincide with the induction resistance r2 of the reception antenna 2. The inductive resistances r1 and r2 are made smaller by reducing sin (β) in the expressions 24 and 25 to match the impedances Z1 and Z2. The resonance angular frequency ω that satisfies the matching condition changes from ωo to the value of Equation 41 (ωo / √ (1 ± k)) as represented by Equation 28 according to the value of β. The resonance angular frequency ω of Equation 28 is shifted to an angular frequency higher than ωo when cos (β) is a negative resonance condition, and forms a right peak in the two-peak graph of FIG. The current I2 of the receiving antenna 2 flows in the opposite direction so as to cancel the current I1 of the transmitting antenna 1, and the electromagnetic field generated by the overall antenna system of the transmitting antenna 1 and the receiving antenna 2 is reduced, and this transformer is generated. This is effective in reducing unnecessary electromagnetic noise (EMI).

また、ｃｏｓ（β）が正の共鳴条件の場合は、共振角周波数ωはωｏより低い角周波数にシフトし、図１３（ｂ）の２山のグラフの左の山を形成する。受信アンテナ２の電流Ｉ２は送信アンテナ１の電流Ｉ１と同じ方向に流れ、送信アンテナ１と受信アンテナ２の総体のアンテナ系が外部に発生する電磁界が大きくなるが、その一方、この場合には、１つの送信アンテナ１のアンテナ配線のコイルの軸に、アンテナ配線のコイルの軸を共有する複数の受信アンテナ２を平行に設置して、同時に複数の受信アンテナ２の回路に無線電力を給電できる効果がある。   When cos (β) is a positive resonance condition, the resonance angular frequency ω is shifted to an angular frequency lower than ωo, forming the left peak of the two peak graph in FIG. The current I2 of the receiving antenna 2 flows in the same direction as the current I1 of the transmitting antenna 1, and the total antenna system of the transmitting antenna 1 and the receiving antenna 2 generates a large electromagnetic field. On the other hand, in this case, A plurality of receiving antennas 2 sharing the axis of the coil of the antenna wiring can be installed in parallel to the axis of the coil of the antenna wiring of one transmitting antenna 1, and wireless power can be supplied to the circuits of the plurality of receiving antennas 2 at the same time. effective.

なお、空芯コイルの送信アンテナ１と受信アンテナ２を（共鳴電磁界の波長）／（２π）よりも近い距離に設置して構成する第５の実施形態の変成器は、空芯コイルを空気中あるいは絶縁樹脂中で対向させて用いると、従来の変成器に比べ、重量と寸法を軽減できる効果がある。また、第５の実施形態の変成器は、コアに強磁性体の材料を用いる従来の変成器がその強磁性体の周波数特性により制約されて使えなかった高周波においても、制約無く変成器を構成できる効果がある。   Note that the transformer of the fifth embodiment in which the transmitting antenna 1 and the receiving antenna 2 of the air-core coil are installed at a distance closer than (wavelength of the resonance electromagnetic field) / (2π) is the air transformer in which the air-core coil is the air. When used in the middle or in an insulating resin, the weight and dimensions can be reduced as compared with a conventional transformer. Further, the transformer according to the fifth embodiment is configured without restriction even at a high frequency where a conventional transformer using a ferromagnetic material for the core is restricted due to the frequency characteristics of the ferromagnetic material. There is an effect that can be done.

＜第６の実施形態＞
本発明の第６の実施形態として、集積回路の配線層間に電力を伝送する誘導電力伝送回路を示す。図１８（ａ）に、第６の実施形態の送信アンテナ１と受信アンテナ２の平面図を示し、図１８（ｂ）に側面図を示す。送信アンテナ１の配線の中間に電源回路３の端子（ポート１）を設置し、受信アンテナ２の配線の中間に負荷回路４の端子（ポート２）を設置する。図１８では、送信アンテナ１と受信アンテナ２として、集積回路チップの配線層に形成した、厚さが１μｍの銅で、第３の実施形態の１０００分の１の寸法のコイルを用いた。すなわち、送信アンテナ１と受信アンテナ２は、配線幅が１μｍでコイル径Ｄが５４μｍの３巻のコイルを用いた。このアンテナのコイルは集積回路チップ内の絶縁樹脂等の上層に形成したグローバル配線層に形成することが望ましい。また、このアンテナのコイル状の配線の両端は開放してその両端間は寄生容量で結合させる。集積回路チップの配線層に受信アンテナ２の配線を形成し、その集積回路を設置する基板の配線層に送信アンテナ１の銅の配線を形成し、基板側から集積回路チップに無線で電力を供給する誘導電力伝送回路が構成できる。図１８（ｂ）の側面図のように、送信アンテナ１のコイルと受信アンテナ２のコイルは、両者のコイルの軸を横（Ｘ）方向にのみ７μｍのずれ距離ｄでずらして、縦（Ｙ）方向にはずらさず、コイルの面に平行にアンテナ間隔ｈを保って配置した。 <Sixth Embodiment>
As an sixth embodiment of the present invention, an inductive power transmission circuit for transmitting power between wiring layers of an integrated circuit is shown. FIG. 18A shows a plan view of the transmitting antenna 1 and the receiving antenna 2 of the sixth embodiment, and FIG. 18B shows a side view. A terminal (port 1) of the power supply circuit 3 is installed in the middle of the wiring of the transmitting antenna 1, and a terminal (port 2) of the load circuit 4 is installed in the middle of the wiring of the receiving antenna 2. In FIG. 18, the transmission antenna 1 and the reception antenna 2 are made of copper having a thickness of 1 μm and having a thickness of 1/1000 of that of the third embodiment, which is formed in the wiring layer of the integrated circuit chip. That is, for the transmission antenna 1 and the reception antenna 2, three winding coils having a wiring width of 1 μm and a coil diameter D of 54 μm were used. The antenna coil is preferably formed on a global wiring layer formed on an insulating resin or the like in the integrated circuit chip. Also, both ends of the coiled wiring of the antenna are opened and the ends are coupled by a parasitic capacitance. The wiring of the receiving antenna 2 is formed on the wiring layer of the integrated circuit chip, the copper wiring of the transmitting antenna 1 is formed on the wiring layer of the substrate on which the integrated circuit is installed, and power is supplied to the integrated circuit chip wirelessly from the substrate side. An inductive power transmission circuit can be configured. As shown in the side view of FIG. 18B, the coil of the transmitting antenna 1 and the coil of the receiving antenna 2 are shifted vertically (Y) by shifting the axes of the coils only in the lateral (X) direction by a deviation distance d of 7 μm. The antenna spacing h is arranged in parallel to the coil surface without shifting in the direction.

図１９（ａ）に、この場合の共振周波数ｆ＝１４０ＧＨｚにおける誘導抵抗ｒを無次元量のｒ／（２πｆＬ）で縦軸であらわし、横軸にアンテナ間隔ｈを無次元量の（ｈ／Ｄ）であらわすグラフを示す。このアンテナのコイルの実効的自己インダクタンスＬは１．３ｎＨであり、このアンテナのコイルの両端は実効的に約０．００１ｐＦの寄生容量でつながれた。図１９（ａ）で、黒丸印はシミュレーション結果を示し、点線は近似式１３の値×（２／π）を示す。このグラフは、第４の実施形態の結果と同様に黒丸印のシミュレーション結果が点線のグラフに一致した。アンテナ間隔ｈをコイル径Ｄの２割（ｈ／Ｄ＝０．２）程度の約１０μｍ離す場合はｒ／（２πｆＬ）が０．１２で誘導抵抗ｒが約１４０Ωあり、アンテナ間隔ｈをコイル径Ｄの４割（ｈ／Ｄ＝０．４）程度の約２０μｍ離す場合はｒ／（２πｆＬ）が０．０８で誘導抵抗ｒが約９０Ωある。電源回路３及び負荷回路４のインピーダンスはこれらの誘導抵抗に整合させる。図１９（ｂ）に、縦軸に電源回路３から負荷回路４までの電力伝送効率をあらわし、横軸にアンテナ間隔ｈを無次元量（ｈ／Ｄ）であらわす。図１９（ｂ）から、アンテナ間隔ｈをコイル径Ｄの２割（ｈ／Ｄ＝０．２）程度の約１０μｍ離す場合、電力伝送効率が約８０％あり十分効率良く電力を伝送できる。コイル径Ｄの４割（ｈ／Ｄ＝０．４）程度の約２０μｍ離す場合も電力伝送効率が約６０％ある。この現象を利用して、半導体集積回路内に５４μｍ程度の直径のコイル状の送信アンテナ１と受信アンテナ２間のアンテナ間隔ｈを１０μｍから２０μｍ離してアンテナを対向させて非接触で効率良く電力を伝送する回路を構成できる。また、電力を伝送する信号層間の距離、すなわちアンテナ間隔ｈを数倍大きくする場合は、アンテナの直径Ｄを数倍に大きくすることで、効率良く電力を伝送する誘導電力伝送回路を構成できる。また、集積回路チップの配線層に受信アンテナ２の配線を形成し、その集積回路を設置する基板の配線層に送信アンテナ１の銅の配線を形成する場合に、図２３の例のように、基板側の送信アンテナ１の寸法を集積回路チップ側の受信アンテナ２の寸法より大きく形成することもできる。。   In FIG. 19A, the inductive resistance r at the resonance frequency f = 140 GHz in this case is represented by a dimensionless amount r / (2πfL) on the vertical axis, and the antenna interval h is represented by a dimensionless amount (h / D) on the horizontal axis. ). The effective self-inductance L of this antenna coil was 1.3 nH, and both ends of the coil of this antenna were effectively connected with a parasitic capacitance of about 0.001 pF. In FIG. 19A, the black circles indicate the simulation results, and the dotted lines indicate the value of the approximate expression 13 × (2 / π). In this graph, similar to the result of the fourth embodiment, the simulation result indicated by the black circle coincides with the dotted line graph. When the antenna interval h is about 10 μm apart, which is about 20% of the coil diameter D (h / D = 0.2), r / (2πfL) is 0.12, the induction resistance r is about 140Ω, and the antenna interval h is the coil diameter. When about 20 μm, which is about 40% of D (h / D = 0.4), r / (2πfL) is 0.08 and the induction resistance r is about 90Ω. The impedances of the power supply circuit 3 and the load circuit 4 are matched to these inductive resistors. In FIG. 19B, the vertical axis represents the power transmission efficiency from the power supply circuit 3 to the load circuit 4, and the horizontal axis represents the antenna interval h as a dimensionless amount (h / D). From FIG. 19B, when the antenna interval h is about 10 μm, which is about 20% of the coil diameter D (h / D = 0.2), the power transmission efficiency is about 80%, and power can be transmitted sufficiently efficiently. The power transmission efficiency is about 60% even when the coil diameter D is about 20 μm, which is about 40% of the coil diameter D (h / D = 0.4). By utilizing this phenomenon, the antenna interval h between the coiled transmitting antenna 1 and the receiving antenna 2 having a diameter of about 54 μm is separated from 10 μm to 20 μm in the semiconductor integrated circuit, and the antennas are opposed to each other so as to efficiently supply power without contact. A circuit for transmission can be configured. Further, when the distance between signal layers for transmitting power, that is, the antenna interval h is increased several times, an induction power transmission circuit that efficiently transmits power can be configured by increasing the antenna diameter D several times. Further, when the wiring of the receiving antenna 2 is formed in the wiring layer of the integrated circuit chip and the copper wiring of the transmitting antenna 1 is formed in the wiring layer of the substrate on which the integrated circuit is installed, as in the example of FIG. The size of the transmitting antenna 1 on the substrate side can be made larger than the size of the receiving antenna 2 on the integrated circuit chip side. .

＜第７の実施形態＞
第７の実施形態として、電源回路３に直径が約３００ｍｍの送信アンテナ１を接続し、電子ディスプレイ装置に埋め込んだ約３００ｍｍの直径の受信アンテナ２までの空間を電力を伝送し、その受信アンテナ２に電子ディスプレイ装置の負荷回路４を接続して電子ディスプレイ装置を動作させる誘導電力伝送回路を示す。第７の実施形態では、第３の実施形態の図１４（ａ）の３巻きのコイル状のアンテナの寸法を約６倍の３００ｍｍに拡大し、厚さ５０μｍで幅が１０ｍｍの銅の配線を３巻きした送信アンテナ１と受信アンテナ２を対向させてアンテナ間に電力を伝送する。このアンテナ配線は、ポリイミドフィルムに厚さ５０μｍで積層した銅箔をエッチングすることで形成することができる。この各アンテナのコイルの実効的自己インダクタンスＬ１とＬ２は４．９μＨである。各アンテナコイルの配線の両端に１００ｐＦの容量Ｃ１とＣ２を接続する。この場合は、共振周波数ｆが７．３ＭＨｚになる。送信アンテナ１と受信アンテナ２のアンテナ間隔ｈをアンテナの寸法Ｄ程度の３００ｍｍ離した場合に、両アンテナの結合係数ｋが約０．０２になり、誘導抵抗ｒ１とｒ２が約４Ωになる。この整合した回路での電力伝送効率Ｐｅは約９４％になり電力が効率良く伝送できる。 <Seventh Embodiment>
As a seventh embodiment, a transmitting antenna 1 having a diameter of about 300 mm is connected to a power supply circuit 3, and power is transmitted through a space up to a receiving antenna 2 having a diameter of about 300 mm embedded in an electronic display device. An inductive power transmission circuit for operating the electronic display device by connecting the load circuit 4 of the electronic display device is shown. In the seventh embodiment, the size of the three-turn coiled antenna of FIG. 14A of the third embodiment is increased to about 6 times by 300 mm, and copper wiring having a thickness of 50 μm and a width of 10 mm is formed. The transmitting antenna 1 and the receiving antenna 2 that are wound three times are opposed to each other, and power is transmitted between the antennas. This antenna wiring can be formed by etching a copper foil laminated on a polyimide film with a thickness of 50 μm. The effective self-inductances L1 and L2 of the coils of each antenna are 4.9 μH. Capacitors C1 and C2 of 100 pF are connected to both ends of the wiring of each antenna coil. In this case, the resonance frequency f is 7.3 MHz. When the antenna interval h between the transmitting antenna 1 and the receiving antenna 2 is about 300 mm apart from the antenna dimension D, the coupling coefficient k of both antennas is about 0.02, and the induction resistances r1 and r2 are about 4Ω. The power transmission efficiency Pe in this matched circuit is about 94%, and power can be transmitted efficiently.

この電力伝送効率Ｐｅが良い原因は、アンテナの寸法が大きくなって共振周波数ｆが低くなったため、アンテナ配線の表皮効果による損失が小さくなったからである。このように大きな寸法のアンテナを用いて電子ディスプレイ装置に電力を供給することで、送信アンテナ１と受信アンテナ２のアンテナ間隔ｈを大きく取っても電力を効率良く電子ディスプレイ装置に伝送できる効果がある。   The reason why the power transmission efficiency Pe is good is that the loss due to the skin effect of the antenna wiring is reduced because the dimensions of the antenna are increased and the resonance frequency f is decreased. By supplying power to the electronic display device using the antenna having such a large size, there is an effect that the power can be efficiently transmitted to the electronic display device even when the antenna interval h between the transmitting antenna 1 and the receiving antenna 2 is large. .

（変形例６）
変形例６として、直径が３００ｍｍの大きなアンテナを用いて、電力を動力として利用する車両などに電力を供給する誘導電力伝送システムを示す。すなわち、変形例６では、電力供給設備の電源回路３から電力を、周波数が約７．３ＭＨｚの高周波電流にして、幅１０ｍｍの銅の配線の３巻きの矩形のコイルで直径Ｄが３００ｍｍの送信アンテナ１に電流を供給し、その送信アンテナ１から３００ｍｍ程度の距離を隔てて対向する車両の受信アンテナ２に約９４％の効率で電力を送信し、その受信アンテナ２から、その電力をその車両の充電池などの負荷回路４に伝送する誘導電力伝送システムを提案する。 (Modification 6)
As a sixth modification, an inductive power transmission system that supplies electric power to a vehicle that uses electric power as power by using a large antenna having a diameter of 300 mm is shown. That is, in the modified example 6, the power from the power supply circuit 3 of the power supply facility is changed to a high-frequency current having a frequency of about 7.3 MHz, and a three-turn rectangular coil of copper wiring having a width of 10 mm is used to transmit a diameter D of 300 mm. A current is supplied to the antenna 1, and power is transmitted to the receiving antenna 2 of the vehicle opposed to the transmitting antenna 1 at a distance of about 300 mm with an efficiency of about 94%, and the power is transmitted from the receiving antenna 2 to the vehicle. An inductive power transmission system for transmitting to a load circuit 4 such as a rechargeable battery is proposed.

＜第８の実施形態＞
第８の実施形態として、例えば携帯電話などの電子機器に非接触で電力を伝送する回路として、図２０（ａ）の平面図のように、送信アンテナ１と受信アンテナ２を同一平面上に並べて配置して電力を伝送する誘導電力伝送回路を示す。すなわち、縦横４７ｍｍの矩形の送信アンテナ１と受信アンテナ２を同一平面上に２０ｍｍ隔てて並べた構造の誘導電力伝送回路である。このアンテナ間の結合係数ｋは０．０１３である。送信アンテナ１と受信アンテナ２を配線するアンテナ面（ＸＹ面）は紙面に平行にし、紙面の表側の配線が７巻あり紙面の裏側の配線が７巻あるコイル状にアンテナを形成し、紙面の表側の配線と裏側の配線は、紙面に垂直方向に１ｍｍの間隔をあけて配置する。アンテナの紙面の表側の配線と紙面の裏側の配線とはＸＹ面に投影した配線パターンがＹ軸に関して左右対称になるように配線する。そして、アンテナの両端は開放し、アンテナの両端間を接続する容量は寄生容量にする。送信アンテナ１の配線の中間に電源回路３の端子のポート１を設置し、受信アンテナ２の配線の中間に負荷回路４の端子のポート２を設置した。このアンテナは、アンテナ端を大きな容量で結んだ場合に実効的自己インダクタンスＬを求めると８．９μＨであった。ただし、アンテナの実効的インダクタンスＬはアンテナ電流の分布に依存して変わり、アンテナ端を開放した場合の実効的インダクタンスＬは、アンテナ端に大きな容量を接続した場合の実効的インダクタンスより小さいと考える。このアンテナは周波数ｆ＝２３．８ＭＨｚで共振した。アンテナの自己インダクタンスを８．９μＨとして計算すると、アンテナの両端は概ね５ｐＦの寄生容量でつながれている。この寄生容量は、紙面の表側の配線と裏側の配線の、紙面に垂直方向の間隔を大きくすると容量値が小さくなる。 <Eighth Embodiment>
As an eighth embodiment, for example, as a circuit for transmitting power to an electronic device such as a mobile phone in a non-contact manner, the transmitting antenna 1 and the receiving antenna 2 are arranged on the same plane as shown in the plan view of FIG. An inductive power transmission circuit that arranges and transmits electric power is shown. That is, the induction power transmission circuit has a structure in which a rectangular transmission antenna 1 and a reception antenna 2 of 47 mm in length and width are arranged on the same plane with a distance of 20 mm. The coupling coefficient k between the antennas is 0.013. The antenna surface (XY plane) for wiring the transmitting antenna 1 and the receiving antenna 2 is parallel to the paper surface, and the antenna is formed in a coil shape with seven windings on the front side of the paper and seven windings on the back side of the paper. The front-side wiring and the back-side wiring are arranged with an interval of 1 mm perpendicular to the paper surface. The wiring on the front side of the paper of the antenna and the wiring on the back side of the paper are wired so that the wiring pattern projected on the XY plane is symmetrical with respect to the Y axis. The both ends of the antenna are opened, and the capacitance connecting the both ends of the antenna is a parasitic capacitance. The port 1 of the terminal of the power supply circuit 3 is installed in the middle of the wiring of the transmitting antenna 1, and the port 2 of the terminal of the load circuit 4 is installed in the middle of the wiring of the receiving antenna 2. This antenna had an effective self-inductance L of 8.9 μH when the antenna ends were connected with a large capacity. However, the effective inductance L of the antenna changes depending on the distribution of the antenna current, and the effective inductance L when the antenna end is opened is considered to be smaller than the effective inductance when a large capacitance is connected to the antenna end. This antenna resonated at a frequency f = 23.8 MHz. When the antenna self-inductance is calculated as 8.9 μH, both ends of the antenna are connected with a parasitic capacitance of approximately 5 pF. The parasitic capacitance decreases as the distance between the front side wiring and the back side wiring in the direction perpendicular to the paper is increased.

図２０（ａ）のアンテナの誘導抵抗ｒは、ｒ＝ｒ１＝ｒ２＝７Ωであった。このアンテナでは、近似式３６で計算する元にする実効的自己インダクタンスＬの値は３．６μＨであり、アンテナの両端を大きな容量で結んだ場合の８．９μＨの約４０％であると考える。このように実効的自己インダクタンスが大きく変わる原因は、このアンテナにおいてアンテナ面（ＸＹ）に垂直方向で対向する配線間に発生する寄生容量が実効的自己インダクタンスを下げる効果が大きいことに依ると考える。図２０（ｂ）に、縦軸に、送信アンテナ１から受信アンテナ２への電力伝送のＳ２１をあらわし、横軸を周波数ｆにしたグラフを示す。この誘導抵抗ｒにインピーダンスＺを整合させて電力を伝送することでＳ２１は−０．７３ｄＢが得られ、約８５％の効率で効率良く電力を伝送できた。このように同一平面上に離して置いたアンテナで、アンテナ間の電磁誘導の結合係数ｋが０．０１３と小さい場合でも、十分な電力伝送効率が得られた。なお、この電力の伝送効率Ｐｅは、式２９に、誘導抵抗ｒ１＝７Ωを代入して、送信アンテナ１の実効的抵抗ｒｅｆ１および受信アンテナ２の実効的抵抗ｒｅｆ２を、表皮効果を加味して約０．６２Ωと見積もって、式２９に代入すると、電力伝送効率Ｐｅ＝０．８５が得られた。このように、式２９を用いて計算しても、電磁界シミュレーションの結果とほぼ一致する結果が得られた。   The induction resistance r of the antenna of FIG. 20A was r = r1 = r2 = 7Ω. In this antenna, the effective self-inductance L calculated based on the approximate expression 36 is 3.6 μH, which is considered to be about 40% of 8.9 μH when both ends of the antenna are connected with a large capacity. The reason why the effective self-inductance changes greatly in this way is considered to be that the parasitic capacitance generated between the wirings facing the antenna surface (XY) in the vertical direction in this antenna has a large effect of reducing the effective self-inductance. FIG. 20B is a graph in which the vertical axis represents S21 of power transmission from the transmitting antenna 1 to the receiving antenna 2, and the horizontal axis represents the frequency f. By transmitting the power by matching the impedance Z to the induction resistance r, S21 was −0.73 dB, and the power could be transmitted efficiently with an efficiency of about 85%. Thus, even when the antennas placed on the same plane are separated from each other and the electromagnetic induction coupling coefficient k between the antennas is as small as 0.013, sufficient power transmission efficiency is obtained. The power transmission efficiency Pe is calculated by substituting the induction resistance r1 = 7Ω into the equation 29, and the effective resistance ref1 of the transmission antenna 1 and the effective resistance ref2 of the reception antenna 2 are approximately equal to the skin effect. When it was estimated to be 0.62Ω and substituted into Equation 29, power transmission efficiency Pe = 0.85 was obtained. As described above, even when the calculation was performed using Expression 29, a result almost identical to the result of the electromagnetic field simulation was obtained.

第８の実施形態では、コイル状の送信アンテナ１のアンテナ面（ＸＹ面）と同一平面上の周囲に複数の電子機器のコイル状の受信アンテナ２を並べて設置し、一度に多数の電子機器の受信アンテナ２に電力を伝送する電力伝送システムを構成できる。そのように１つの送信アンテナ１に対して複数の受信アンテナ２を設置する場合は、送信アンテナ１に直列に接続する電源回路３の出力インピーダンスＺ１に関するＺ１／（２πｆＬ１）は、各受信アンテナ２毎の結合係数ｋの総和に調整して、電源回路３の出力インピーダンスＺ１を複数の受信アンテナ２が発生する誘導抵抗ｒ１の総和に等しくさせる。   In the eighth embodiment, coil-shaped receiving antennas 2 of a plurality of electronic devices are arranged side by side around the same plane as the antenna surface (XY surface) of the coil-shaped transmitting antenna 1, and a large number of electronic devices can be A power transmission system for transmitting power to the receiving antenna 2 can be configured. When a plurality of receiving antennas 2 are installed for one transmitting antenna 1 as described above, Z1 / (2πfL1) related to the output impedance Z1 of the power supply circuit 3 connected in series to the transmitting antenna 1 is set for each receiving antenna 2. The output impedance Z1 of the power supply circuit 3 is made equal to the sum of the inductive resistors r1 generated by the plurality of receiving antennas 2.

（変形例７）
変形例７として、第８の実施形態のアンテナの紙面の表側の７巻の配線とそれと平行な紙面の裏側の７巻の配線の、紙面に垂直方向の間隔を４倍の４ｍｍにする場合を示す。このモデルを電磁界シミュレーションした結果、４０．１ＭＨｚで共振した。このアンテナの両端を大きな容量で結んだ場合の実効的自己インダクタンスは８μＨになったので、それを用いて計算すると、アンテナの両端を結ぶ寄生容量は２ｐＦになり４割に小さくなった。このアンテナの誘導抵抗は、ｒ＝ｒ１＝ｒ２＝１８Ωになり、この誘導抵抗ｒにインピーダンスＺを整合させて電力を伝送するとＳ２１は−０．４５５ｄＢが得られ、約９０％の効率で、より良い効率で電力を伝送できる。このアンテナでは、近似式３６で計算する元にする実効的自己インダクタンスＬの値は５．５μＨであり、アンテナの両端を大きな容量で結んだ場合の８μＨの約７０％であると考える。更に、アンテナの紙面の表側の７巻の配線とそれと平行な紙面の裏側の７巻の配線の、紙面に垂直方向の間隔を更に大きく８ｍｍにすると、アンテナは５１．４ＭＨｚで共振し、送信アンテナ１と受信アンテナ２のアンテナ系の誘導抵抗ｒ＝ｒ１＝ｒ２＝３４Ωになり、Ｓ２１は−０．３２ｄＢになり約９３％の効率で電力を伝送する。 (Modification 7)
As a modified example 7, the case where the distance between the front surface of the antenna of the eighth embodiment and the seven wires on the back side of the paper parallel to it is set to 4 mm, which is four times the vertical direction of the paper. Show. As a result of electromagnetic simulation of this model, it resonated at 40.1 MHz. The effective self-inductance when connecting both ends of the antenna with a large capacitance was 8 μH, and when calculated using this, the parasitic capacitance connecting both ends of the antenna was 2 pF, which was 40% smaller. The induction resistance of this antenna is r = r1 = r2 = 18Ω. When power is transmitted by matching the impedance Z to this induction resistance r, S21 is −0.455 dB, with an efficiency of about 90%. Power can be transmitted with good efficiency. In this antenna, the effective self-inductance L based on the approximate expression 36 is 5.5 μH, which is considered to be about 70% of 8 μH when both ends of the antenna are connected with a large capacity. Further, when the distance between the front surface of the antenna and the 7 turns of the wiring on the back side of the paper and the 7 turns of the wiring on the back side of the paper parallel to it is further increased to 8 mm in the direction perpendicular to the paper, the antenna resonates at 51.4 MHz. Inductive resistance r = r1 = r2 = 34Ω of the antenna system of 1 and the receiving antenna 2, S21 becomes −0.32 dB, and power is transmitted with an efficiency of about 93%.

（変形例８）
変形例８として、第８の実施形態の図２０（ａ）のアンテナを、紙面の片側だけの７巻のアンテナにし、図２０（ａ）と同様に同一平面上に２０ｍｍ隔てて配置する場合を示す。その送信アンテナ１の配線の中間に電源回路３の端子のポート１を設置し、受信アンテナ２の配線の中間に負荷回路４の端子のポート２を設置して電磁界シミュレーションした。この変形例８のアンテナは、自己インダクタンスＬが２．３μＨであり、アンテナの両端間の寄生容量Ｃは０．８ｐＦあり、１１５ＭＨｚで共振した。図２０（ａ）の配置でのアンテナの誘導抵抗は、ｒ＝ｒ１＝ｒ２＝１４Ωになり、この誘導抵抗ｒにインピーダンスＺを整合させて電力を伝送するとＳ２１は−０．４９ｄＢが得られ、約８９％の効率で電力を伝送した。 (Modification 8)
As a modification 8, the antenna of FIG. 20A of the eighth embodiment is changed to a seven-turn antenna on only one side of the paper, and is arranged 20 mm apart on the same plane as in FIG. Show. The port 1 of the terminal of the power supply circuit 3 is installed in the middle of the wiring of the transmission antenna 1, and the port 2 of the terminal of the load circuit 4 is installed in the middle of the wiring of the receiving antenna 2. The antenna of this modification 8 had a self-inductance L of 2.3 μH, a parasitic capacitance C between both ends of the antenna of 0.8 pF, and resonated at 115 MHz. The induction resistance of the antenna in the arrangement shown in FIG. 20A is r = r1 = r2 = 14Ω. When power is transmitted by matching the impedance Z to this induction resistance r, S21 is −0.49 dB, Electric power was transmitted with an efficiency of about 89%.

＜第９の実施形態＞
第９の実施形態の誘導電力伝送回路は、以上で説明した誘導電力伝送回路において、送信アンテナ１あるいは受信アンテナ２に電源回路３あるいは負荷回路４を直接には接続せず、そのアンテナはコイル状の配線の両端に容量Ｃ２を結んだのみの構成にし、そのアンテナに誘導結合する誘導結合配線６の両端を負荷回路４に接続した誘導電力伝送回路である。第９の実施形態を、図２１により説明する。図２１（ａ）は、第９の実施形態の送信アンテナ１と受信アンテナ２と誘導結合配線６の平面図を示す。図２１（ａ）の平面図は、第９の実施形態を説明するために、第８の実施形態の図２０（ａ）に示した構成の送信アンテナ１と受信アンテナ２をＸＹ平面上に並べて配置し、その受信配線２の螺旋状の配線の中にＸＹ平面に設置した螺旋状の誘導結合配線６を設置する。すなわち、縦横４７ｍｍの矩形の送信アンテナ１と受信アンテナ２をＸＹ平面上に２０ｍｍ隔てて並べ、その受信アンテナ２で囲まれる中に、厚さが５μｍで幅が１ｍｍの銅の配線で、径が１０ｍｍから１５ｍｍのループ状の誘導結合配線６を設置した構造の誘導電力伝送回路である。その誘導結合配線６の両端のポート３を負荷回路４に接続する。誘導結合配線６のループの大きさが縦横１０ｍｍの場合、電源回路３から負荷回路４まで最も効率良く電力が伝送するための負荷回路のインピーダンスＺ２は４Ωである。一方、誘導結合配線６のループの大きさを縦横１５ｍｍに大きくして受信回路２との相互インダクタンスＭ２を大きくした場合、誘導結合配線６に誘導される誘導インピーダンスに整合する負荷回路のインピーダンスＺ２は２０Ωに大きくなる。図２１（ｂ）に、誘導結合配線６の径が１０ｍｍから１５ｍｍの場合の、送信アンテナ１から誘導結合配線６への電力伝送効率ＰｅをＳ２１であらわした周波数特性のグラフを示すが、効率良く電力が伝送できた。本実施形態の図２１（ａ）の場合においては、誘導結合配線６の有する自己インダクタンスが受信アンテナ２の誘導抵抗ｒ２＝７Ωに比べて小さい。ただし、誘導抵抗ｒ２は第８の実施形態における受信アンテナ２の誘導抵抗ｒ２を指す。そのため、ポート３に接続する負荷回路４の入力インピーダンスＺ２は、誘導結合配線６が受信アンテナ２と誘導結合する相互インダクタンスをＭ２とあらわすと、Ｚ２を（２πｆ×Ｍ２）２／ｒ２にすると電源回路３から負荷回路４まで最も効率良く電力が伝送できる。 <Ninth Embodiment>
In the inductive power transmission circuit of the ninth embodiment, the power circuit 3 or the load circuit 4 is not directly connected to the transmitting antenna 1 or the receiving antenna 2 in the inductive power transmission circuit described above, and the antenna is coiled. This is an inductive power transmission circuit having a configuration in which the capacitor C2 is simply connected to both ends of the wiring, and both ends of the inductively coupled wiring 6 inductively coupled to the antenna are connected to the load circuit 4. A ninth embodiment will be described with reference to FIG. FIG. 21A is a plan view of the transmitting antenna 1, the receiving antenna 2, and the inductive coupling wiring 6 according to the ninth embodiment. In the plan view of FIG. 21A, in order to explain the ninth embodiment, the transmitting antenna 1 and the receiving antenna 2 having the configuration shown in FIG. 20A of the eighth embodiment are arranged on the XY plane. The spiral inductive coupling wiring 6 placed on the XY plane is installed in the spiral wiring of the reception wiring 2. That is, a rectangular transmission antenna 1 and a reception antenna 2 of 47 mm in length and width are arranged 20 mm apart on the XY plane, and surrounded by the reception antenna 2, a copper wiring having a thickness of 5 μm and a width of 1 mm and a diameter of This is an inductive power transmission circuit having a structure in which a loop-like inductive coupling wiring 6 of 10 mm to 15 mm is installed. The ports 3 at both ends of the inductive coupling wiring 6 are connected to the load circuit 4. When the size of the loop of the inductive coupling wiring 6 is 10 mm in length and width, the impedance Z2 of the load circuit for transmitting power most efficiently from the power supply circuit 3 to the load circuit 4 is 4Ω. On the other hand, when the size of the loop of the inductive coupling wiring 6 is increased to 15 mm in length and width and the mutual inductance M2 with the receiving circuit 2 is increased, the impedance Z2 of the load circuit that matches the inductive impedance induced in the inductive coupling wiring 6 is Increase to 20Ω. FIG. 21B shows a frequency characteristic graph in which the power transmission efficiency Pe from the transmitting antenna 1 to the inductive coupling wiring 6 is expressed by S21 when the diameter of the inductive coupling wiring 6 is 10 mm to 15 mm. Electric power was transmitted. In the case of FIG. 21A of this embodiment, the self-inductance of the inductive coupling wiring 6 is smaller than the induction resistance r2 = 7Ω of the receiving antenna 2. However, the induction resistance r2 indicates the induction resistance r2 of the receiving antenna 2 in the eighth embodiment. Therefore, the input impedance Z2 of the load circuit 4 connected to the port 3 is expressed as M2 when the inductive coupling wiring 6 is inductively coupled with the receiving antenna 2 as M2. If Z2 is (2πf × M2) 2 / r2, the power supply circuit 3 to the load circuit 4 can transmit power most efficiently.

なお、受信回路４自体をこの誘導結合配線６として用いた誘導電力伝送回路も構成できる。また、この実施形態の受信アンテナ２と誘導結合配線６と受信回路４の組み合わせを、同様に送信アンテナ１と電源回路３の組み合わせに用いて構成した誘導電力伝送回路も同様に構成できる。   An inductive power transmission circuit using the receiving circuit 4 itself as the inductive coupling wiring 6 can also be configured. In addition, an induction power transmission circuit configured by using the combination of the reception antenna 2, the inductive coupling wiring 6, and the reception circuit 4 of this embodiment as the combination of the transmission antenna 1 and the power supply circuit 3 can be similarly configured.

＜第１０の実施形態＞
第１０の実施形態は、本発明の原理の第１の場合を利用して、空芯コイルを用いる誘導電力伝送回路によるインピーダンス変換回路５を構成する誘導電力伝送回路を示す。すなわち、図２の回路図で、Ｌ１×Ｃ１＝Ｌ２×Ｃ２＝１／ωｏ２であるアンテナ系において、電力を伝送する交流の角周波数ωをωｏにし、任意の正の数αに関して、送信アンテナ１にポート１で直列に接続する電源回路３の出力インピーダンスＺ１をｒ１＝ωＭ・αにすると、受信アンテナに直列なポート２の出力インピーダンスがｒ２＝ωＭ／αになる。こうして、インピーダンスを変換して電源回路３から負荷回路４に電力を伝送する誘導電力伝送回路を構成できる。すなわち、電源回路３と負荷回路４のインピーダンスは、その積のみを一定にし、両者のインピーダンスの比を任意に変換できるインピーダンス変換回路５を構成できる。 <Tenth Embodiment>
The tenth embodiment shows an inductive power transmission circuit constituting the impedance conversion circuit 5 by an inductive power transmission circuit using an air-core coil, using the first case of the principle of the present invention. That is, in the circuit diagram of FIG. 2, in the antenna system where L1 × C1 = L2 × C2 = 1 / ωo 2 , the angular frequency ω of the alternating current for transmitting power is set to ωo, and the transmission antenna is transmitted with respect to an arbitrary positive number α. When the output impedance Z1 of the power supply circuit 3 connected in series with the port 1 to 1 is r1 = ωM · α, the output impedance of the port 2 in series with the receiving antenna is r2 = ωM / α. In this way, an inductive power transmission circuit that converts impedance and transmits power from the power supply circuit 3 to the load circuit 4 can be configured. That is, the impedance of the power supply circuit 3 and the load circuit 4 is constant only in the product, and the impedance conversion circuit 5 that can arbitrarily convert the ratio of the impedances of the two can be configured.

第１０の実施形態のインピーダンス変換回路５は、電磁界の４分の１波長の伝送線路によるインピーダンス変換回路と同じ機能を有する。４分の１波長の伝送線路によるインピーダンス変換回路は、その伝送線路の配線の長さを４分の１波長にまで長く形成する必要があるが、この第１０の実施形態のインピーダンス変換回路５は、送信アンテナ１及び受信アンテナ２それぞれの両端間を結容量Ｃで結ぶことにより、アンテナの配線長を電磁界の４分の１波長より短くしたインピーダンス変換回路５が得られる効果がある。また、４分の１波長の伝送線路によるインピーダンス変換回路は、伝送線路の特性インピーダンスをＺｏとすると、任意の正の数αに関して、Ｚｏ・αのインピーダンスを、Ｚｏ／αのインピーダンスに変換するので、変換するインピーダンスの基礎にする特性インピーダンスＺｏが固定して調整が難しい問題があるが、第１０の実施形態のインピーダンス変換回路５は、送信アンテナ１と受信アンテナ２の間の距離を変えることで相互インダクタンスＭを変えることで、変換するインピーダンスの基礎にするインピーダンスωＭの値を容易に変えることができる効果がある。   The impedance conversion circuit 5 of the tenth embodiment has the same function as the impedance conversion circuit using a transmission line having a quarter wavelength of the electromagnetic field. The impedance conversion circuit using a quarter-wave transmission line needs to be formed so that the length of the transmission line wiring is increased to a quarter wavelength. The impedance conversion circuit 5 according to the tenth embodiment By connecting both ends of each of the transmitting antenna 1 and the receiving antenna 2 with the coupling capacitor C, there is an effect that the impedance conversion circuit 5 in which the wiring length of the antenna is shorter than a quarter wavelength of the electromagnetic field can be obtained. In addition, the impedance conversion circuit using a quarter-wavelength transmission line converts the impedance of Zo · α into an impedance of Zo / α for any positive number α, where Zo is the characteristic impedance of the transmission line. The characteristic impedance Zo based on the impedance to be converted is fixed and difficult to adjust, but the impedance conversion circuit 5 of the tenth embodiment can change the distance between the transmission antenna 1 and the reception antenna 2. By changing the mutual inductance M, there is an effect that the value of the impedance ωM as the basis of the impedance to be converted can be easily changed.

第１０の実施形態では、送信アンテナ１が所定の強さの電磁界を発生させることで、コイル状の受信アンテナ２に所定の大きさの誘導抵抗ｒ２が発生する。その誘導抵抗ｒ２に等しい入力インピーダンスの負荷回路４を接続することで、受信アンテナ２に効率良く電力を受信させることができる。その受信アンテナ２に誘起される電圧は、送信アンテナ１が受信アンテナ２の位置に発生する磁界の強度に比例し、その磁界の強度が弱くなると、受信アンテナ２に誘起される誘導起電力（電圧）が小さくなる。その際に、受信アンテナ２の負荷回路４の入力インピーダンスを小さくすると受信アンテナ２のアンテナ電流Ｉ２が大きくなり、受信アンテナ２の電流Ｉ２が大きい程大きな電力を受信し、受信アンテナ２の導体損失が小さければ、送信アンテナ１が供給する電力を１００％受信するに至る所定の電流値まで受信アンテナ２の電流Ｉ２が大きくなる。受信アンテナ２の電流Ｉ２がその所定電流値を越えると、その受信アンテナ２の電流Ｉ２が送信アンテナ１に誘起する誘導抵抗ｒ１が大きくなり、送信アンテナ１により大きな電力を供給させようとする。こうして、送信アンテナ１に接続した電源回路３の供給する電力が１００％近い効率で受信回路２に接続した負荷回路に伝送できる。   In the tenth embodiment, when the transmitting antenna 1 generates an electromagnetic field having a predetermined strength, an induction resistor r2 having a predetermined magnitude is generated in the coiled receiving antenna 2. By connecting the load circuit 4 having an input impedance equal to the induction resistor r2, the receiving antenna 2 can receive power efficiently. The voltage induced in the reception antenna 2 is proportional to the strength of the magnetic field generated at the position of the reception antenna 2 by the transmission antenna 1. When the strength of the magnetic field becomes weak, the induced electromotive force (voltage) induced in the reception antenna 2. ) Becomes smaller. At that time, if the input impedance of the load circuit 4 of the receiving antenna 2 is reduced, the antenna current I2 of the receiving antenna 2 is increased, and the larger the current I2 of the receiving antenna 2 is, the larger power is received, and the conductor loss of the receiving antenna 2 is reduced. If it is smaller, the current I2 of the receiving antenna 2 becomes larger up to a predetermined current value that reaches 100% of the power supplied by the transmitting antenna 1. When the current I2 of the receiving antenna 2 exceeds the predetermined current value, the induction resistance r1 induced in the transmitting antenna 1 by the current I2 of the receiving antenna 2 increases, and the transmitting antenna 1 tries to supply a large amount of power. Thus, the power supplied from the power supply circuit 3 connected to the transmitting antenna 1 can be transmitted to the load circuit connected to the receiving circuit 2 with an efficiency of nearly 100%.

なお、第１０の実施形態でも、送信アンテナ１と受信アンテナ２の結合係数ｋは０．００４以上に限定されず、結合係数ｋがそれより小さい場合も、送信アンテナ１側で送信アンテナ１の共振周波数ｆを受信アンテナ２の共振周波数に同調させることで、効率良く電力を伝送する誘導電力伝送回路を構成できる。特に、電力伝送効率を良くするために、式２９の電力伝送効率Ｐｅにおいて、受信アンテナ２の誘導抵抗ｒ２を受信アンテナ２の配線の実効的抵抗ｒ４より大きくし、一方、送信アンテナ１の誘導抵抗ｒ１は例えば送信アンテナ１の配線を超伝導体で形成するなどしてその配線の実効的抵抗ｒ３を小さくし、その送信アンテナ１に流す電流Ｉ１を受信アンテナの流す電流Ｉ２より大きくすることで、式２９で計算する電力伝送効率Ｐｅを高くした誘導電力伝送回路も構成できる。   Also in the tenth embodiment, the coupling coefficient k between the transmission antenna 1 and the reception antenna 2 is not limited to 0.004 or more, and even when the coupling coefficient k is smaller than that, the resonance of the transmission antenna 1 on the transmission antenna 1 side. By tuning the frequency f to the resonance frequency of the receiving antenna 2, an inductive power transmission circuit that efficiently transmits power can be configured. In particular, in order to improve the power transmission efficiency, in the power transmission efficiency Pe of Expression 29, the induction resistance r2 of the reception antenna 2 is made larger than the effective resistance r4 of the wiring of the reception antenna 2, while the induction resistance of the transmission antenna 1 is increased. For example, the effective resistance r3 of the wiring is reduced by, for example, forming the wiring of the transmission antenna 1 with a superconductor, and the current I1 flowing through the transmission antenna 1 is made larger than the current I2 flowing through the reception antenna. An inductive power transmission circuit in which the power transmission efficiency Pe calculated by Expression 29 is increased can also be configured.

（変形例９）
変形例９として、第１０の実施形態の誘導電力伝送回路を、第９の実施形態の図２１（ａ）に示すように、負荷回路４と受信アンテナ２の回路を、受信アンテナ２に誘導結合配線６が相互インダクタンスＭ２で誘導結合し、誘導結合配線６の両端に負荷回路４を接続した回路に代える。変形例９では、この構成で、電源回路３の出力インピーダンスＺ１を、その負荷回路４側での誘導抵抗値（Ｍ２／Ｍ）２×Ｚ１に変換し、その誘導抵抗値に負荷回路４の入力インピーダンスＺ２を等しくすると最も効率良く電力を伝送できる。また、更に、実効的インダクタンスＬ２を持つ受信アンテナ２自身をその誘導結合配線６にし、その受信アンテナ２のアンテナ配線の両端の容量Ｃ２に並列に負荷回路４を接続する構成にもできる。その場合は、電源回路３の出力インピーダンスＺ１を、その負荷回路４側での誘導抵抗値（Ｌ２／Ｍ）２×Ｚ１に変換し、その誘導抵抗値に負荷回路４の入力インピーダンスＺ２を等しくすると最も効率良く電力を伝送できる。 (Modification 9)
As a modification 9, the inductive power transmission circuit of the tenth embodiment is inductively coupled to the reception antenna 2 by connecting the load circuit 4 and the reception antenna 2 circuit as shown in FIG. 21A of the ninth embodiment. Instead of the circuit in which the wiring 6 is inductively coupled with the mutual inductance M 2 and the load circuit 4 is connected to both ends of the inductive coupling wiring 6. In the modification 9, with this configuration, the output impedance Z1 of the power supply circuit 3 is converted into an induction resistance value (M2 / M) 2 × Z1 on the load circuit 4 side, and the input resistance value is input to the load circuit 4 When impedance Z2 is made equal, power can be transmitted most efficiently. Further, the receiving antenna 2 itself having an effective inductance L2 can be used as the inductive coupling wiring 6 and the load circuit 4 can be connected in parallel to the capacitance C2 at both ends of the antenna wiring of the receiving antenna 2. In that case, if the output impedance Z1 of the power supply circuit 3 is converted into an induction resistance value (L2 / M) 2 × Z1 on the load circuit 4 side, and the input impedance Z2 of the load circuit 4 is made equal to the induction resistance value It can transmit power most efficiently.

（変形例１０）
変形例１０として、変形例９の誘導電力伝送回路における、図２１（ａ）の送信アンテナ１と電源回路３の組み合わせを、受信アンテナ２と負荷回路４の組み合わせに入れ替えた回路構成の誘導電力伝送回路を示す。変形例１０では、電源回路３と送信アンテナ１の回路を、送信アンテナ１に誘導結合配線６が相互インダクタンスＭ１で誘導結合し、誘導結合配線６の両端に電源回路３を接続した回路に代えることができる。その場合は、その電源回路３の出力インピーダンスＺ３を、負荷回路４側での誘導抵抗値（Ｍ／Ｍ１）２×Ｚ３に変換し、その誘導抵抗値に負荷回路４の入力インピーダンスＺ２を等しくすると最も効率良く電力を伝送できる。また、更に、実効的インダクタンスＬ１を持つ送信アンテナ１自身をその誘導結合配線６にし、その送信アンテナ１のアンテナ配線の両端の容量Ｃ１に並列に電源回路３を接続する構成にもできる。その場合は、その電源回路３の出力インピーダンスＺ３を、負荷回路４側での誘導抵抗値（Ｍ／Ｌ１）２×Ｚ３に変換し、その誘導抵抗値に負荷回路４の入力インピーダンスＺ２を等しくすると最も効率良く電力を伝送できる。 (Modification 10)
As a tenth modification, inductive power transmission having a circuit configuration in which the combination of the transmission antenna 1 and the power supply circuit 3 in FIG. 21A in the inductive power transmission circuit of the ninth modification is replaced with a combination of the reception antenna 2 and the load circuit 4. The circuit is shown. In the modification 10, the circuit of the power supply circuit 3 and the transmission antenna 1 is replaced with a circuit in which the inductive coupling wiring 6 is inductively coupled to the transmission antenna 1 with the mutual inductance M1 and the power supply circuit 3 is connected to both ends of the inductive coupling wiring 6. Can do. In that case, the output impedance Z3 of the power supply circuit 3 is converted into an induction resistance value (M / M1) 2 × Z3 on the load circuit 4 side, and the input impedance Z2 of the load circuit 4 is made equal to the induction resistance value. It can transmit power most efficiently. Further, the transmission antenna 1 itself having an effective inductance L1 can be used as the inductive coupling wiring 6, and the power supply circuit 3 can be connected in parallel to the capacitance C1 at both ends of the antenna wiring of the transmission antenna 1. In that case, the output impedance Z3 of the power supply circuit 3 is converted into an induction resistance value (M / L1) 2 × Z3 on the load circuit 4 side, and the input impedance Z2 of the load circuit 4 is made equal to the induction resistance value. It can transmit power most efficiently.

＜第１１の実施形態＞
第１１の実施形態は、送信アンテナ１と受信アンテナ２の間の距離が変動してインピーダンスが変動する場合に、第１０の実施形態のインピーダンス変換回路５を用いてインピーダンスの変動を補正する誘導電力伝送回路を示す。図２２（ｂ）に第１１の実施形態の誘導電力伝送回路の平面図を示す。図２２（ｂ）では、送信アンテナ１のコイルのインダクタンスＬ１と受信アンテナ２のコイルのインダクタンスＬ２が同じインダクタンスＬ＝Ｌ１＝Ｌ２の場合を示すが、両者のインピーダンスが異なる場合にも本実施形態に思想を適用できる。先ず、図２２（ａ）には、インピーダンス変換回路５を用いない場合の誘導電力伝送回路の送信アンテナ１と受信アンテナ２の平面図を示す。図２２（ａ）のように送信アンテナ１と受信アンテナ２が結合係数ｋの値ｋｏで電磁結合する場合に相互インダクタンスＭが値Ｍｏを持つものとする。その場合に、両アンテナを共振角周波数ω＝ωｏ＝１／√（Ｌ１・Ｃ１）で共鳴させると、式３０と式３１により、電源回路３の出力インピーダンスＺ１と負荷回路４の入力インピーダンスＺ２をωＭｏにして整合させて完全な効率で電力を伝送させることができる。 <Eleventh embodiment>
In the eleventh embodiment, when the distance between the transmitting antenna 1 and the receiving antenna 2 fluctuates and the impedance fluctuates, the inductive power that corrects the fluctuation of the impedance using the impedance conversion circuit 5 of the tenth embodiment A transmission circuit is shown. FIG. 22B is a plan view of the inductive power transmission circuit according to the eleventh embodiment. FIG. 22B shows a case where the inductance L1 of the coil of the transmission antenna 1 and the inductance L2 of the coil of the reception antenna 2 are the same inductance L = L1 = L2. The idea can be applied. First, FIG. 22A shows a plan view of the transmission antenna 1 and the reception antenna 2 of the induction power transmission circuit when the impedance conversion circuit 5 is not used. Assume that the mutual inductance M has a value Mo when the transmitting antenna 1 and the receiving antenna 2 are electromagnetically coupled with the value ko of the coupling coefficient k as shown in FIG. In this case, when both antennas are made to resonate at the resonance angular frequency ω = ωo = 1 / √ (L1 · C1), the output impedance Z1 of the power supply circuit 3 and the input impedance Z2 of the load circuit 4 are obtained by Expression 30 and Expression 31. The power can be transmitted with perfect efficiency by matching with ωMo.

次に、図２２（ｂ）の回路を説明する。図２２（ｂ）では、送信アンテナ１と受信アンテナ２の間の距離が変動し両アンテナの結合係数ｋが変動する誘導電力伝送回路の場合であり、その回路の電源回路３と送信アンテナ１の間に第１０の実施形態のインピーダンス変換回路５を挿入することでその変動を補正する。すなわち、出力インピーダンスがＺ１＝ωｏＭｏに固定されている電源回路３に直列に、インピーダンス変換回路５の送信アンテナの中間の入力端子を接続し、インピーダンス変換回路５の受信アンテナの中間の出力端子を送信アンテナ１の配線の中間の入力端子（ポート１）に接続する。このインピーダンス変換回路５は、その送信アンテナと受信アンテナが対向する距離を自由に変えるかあるいは両アンテナのコイルの軸を自由にずらすことで、両アンテナ間の相互インダクタンスを自由に調整できるように構成する。   Next, the circuit of FIG. 22B will be described. FIG. 22B shows a case of an inductive power transmission circuit in which the distance between the transmission antenna 1 and the reception antenna 2 varies and the coupling coefficient k of both antennas varies, and the power supply circuit 3 of the circuit and the transmission antenna 1 The fluctuation is corrected by inserting the impedance conversion circuit 5 of the tenth embodiment in between. That is, the intermediate input terminal of the transmission antenna of the impedance conversion circuit 5 is connected in series with the power supply circuit 3 whose output impedance is fixed to Z1 = ωoMo, and the intermediate output terminal of the reception antenna of the impedance conversion circuit 5 is transmitted. Connect to the input terminal (port 1) in the middle of the antenna 1 wiring. The impedance conversion circuit 5 is configured such that the mutual inductance between the two antennas can be freely adjusted by freely changing the distance between the transmitting antenna and the receiving antenna or by freely shifting the axes of the coils of the two antennas. To do.

この回路で、送信アンテナ１と受信アンテナ２配置が変化し、アンテナ間の距離が遠くなると、送信アンテナ１と受信アンテナ２の結合係数ｋが小さくなり相互インダクタンスＭが小さくなる。その相互インダクタンスＭがＭｏ／αに小さくなったものとする。ここで、αは１より大きい実数とする。この場合に受信アンテナ２に接続する負荷回路４の入力インピーダンスＺ２がωＭｏに固定されているので、その受信アンテナ２側のインピーダンスが送信アンテナ１側では、Ｚ３＝ωＭｏ／α２に変換されてしまう。そのため、インピーダンス変換回路５は、ωＭｏに固定されている電源回路３の出力インピーダンスＺ１を送信アンテナ側のインピーダンスＺ３＝ωＭｏ／α２に変換する必要がある。これを実現するため、インピーダンス変換回路５は、その送信アンテナと受信アンテナの対向する距離を変えて結合係数ｋを変えて相互インダクタンスをωｏＭｏ／αに変える。これにより目的のインピーダンス変換が行なえ、電源回路３から負荷回路４までの回路のインピーダンスを整合させることができる。 In this circuit, when the arrangement of the transmission antenna 1 and the reception antenna 2 is changed and the distance between the antennas is increased, the coupling coefficient k between the transmission antenna 1 and the reception antenna 2 is reduced and the mutual inductance M is reduced. It is assumed that the mutual inductance M is reduced to Mo / α. Here, α is a real number larger than 1. Since the input impedance Z2 of the load circuit 4 connected in this case the receiving antenna 2 is fixed to OmegaMo, the impedance of the reception antenna 2 side in the transmission antenna 1 side, thereby it is converted into Z3 = ωMo / α 2 . Therefore, the impedance conversion circuit 5, it is necessary to convert the output impedance Z1 of the power supply circuit 3 which is fixed to OmegaMo the impedance Z3 = ωMo / α 2 of the transmitting antenna side. In order to realize this, the impedance conversion circuit 5 changes the mutual inductance to ωoMo / α by changing the coupling coefficient k by changing the distance between the transmitting antenna and the receiving antenna. Thereby, the target impedance conversion can be performed, and the impedance of the circuit from the power supply circuit 3 to the load circuit 4 can be matched.

このように、送信アンテナ１のコイルと受信アンテナ２のコイルの位置がずれることによる相互インダクタンスの変動を、インピーダンス変換回路５が補正することで、固定した出力インピーダンスＺ１の電源回路３から固定した入力インピーダンスＺ２の負荷回路４までのインピーダンスを整合させて完全な効率で電力を伝送できる効果がある。また、このように、送信アンテナ１と受信アンテナ２の相互インダクタンスＭが変化しても、その変化によるインピーダンスの変動を、インピーダンス変換回路５が、共鳴の角周波数ωを一定の角周波数ωｏに保ったままで、アンテナ間の距離という１つのパラメータを調整するだけで相互インダクタンスを可変にして、適正な値にインピーダンスを変換して回路のインピーダンスを整合させることができる。本実施形態では、共鳴の周波数が変わらないので誘導電力伝送回路の構成が簡単になる効果があり、また、１つのパラメータを調整するだけで変換すべきインピーダンスを変えられるので調整が容易である効果がある。   As described above, the impedance conversion circuit 5 corrects the mutual inductance variation caused by the displacement of the position of the coil of the transmission antenna 1 and that of the reception antenna 2, thereby fixing the input from the power supply circuit 3 having the fixed output impedance Z <b> 1. There is an effect that power can be transmitted with perfect efficiency by matching the impedance of the impedance Z2 up to the load circuit 4. As described above, even if the mutual inductance M between the transmitting antenna 1 and the receiving antenna 2 changes, the impedance conversion circuit 5 keeps the angular frequency ω of resonance at a constant angular frequency ωo, due to the change in impedance due to the change. The impedance of the circuit can be matched by converting the impedance to an appropriate value by making the mutual inductance variable only by adjusting one parameter of the distance between the antennas. In this embodiment, since the resonance frequency does not change, there is an effect that the configuration of the inductive power transmission circuit is simplified, and the impedance to be converted can be changed only by adjusting one parameter, and the adjustment is easy. There is.

＜第１２の実施形態＞
第１２の実施形態の誘導電力伝送回路は、図２３（ａ）のように、ＸＹ平面上に置いた縦横の直径Ｇ＝３００ｍｍの１巻きの幅１ｍｍの銅の螺旋状のアンテナ配線の送信アンテナ１と、送信アンテナ１の真中の同じＸＹ平面上に、送信アンテナ１の直径Ｇの約６分の１の直径Ｄの受信アンテナ２を置いた誘導電力伝送回路である。この受信アンテナ２は、第８の実施形態の図２０（ａ）の２つの７巻きコイルをポート２の両端に接続してＸＹ面に垂直方向に１ｍｍ離して対向させた直径Ｄ＝４７ｍｍの受信アンテナ２である。この送信アンテナ１の両端は容量Ｃ１が１１６ｐＦのコンデンサ（キャパシタンス素子）で接続し、受信アンテナ２の両端の容量Ｃ２は、浮遊容量５．２ｐＦに加えて１５．６ｐＦのコンデンサ（キャパシタンス素子）を接続した。この送信アンテナ１の自己インダクタンスＬ１は１．５μＨであり、受信アンテナ２の実効的自己インダクタンスＬ２は８．９μＨである。 <Twelfth Embodiment>
As shown in FIG. 23A, the inductive power transmission circuit according to the twelfth embodiment is a transmission antenna having a copper spiral antenna wiring of 1 mm in width and 1 mm in width and having a diameter G of 300 mm placed on the XY plane. 1 and an inductive power transmission circuit in which a receiving antenna 2 having a diameter D that is about one-sixth of the diameter G of the transmitting antenna 1 is placed on the same XY plane in the middle of the transmitting antenna 1. This receiving antenna 2 has a diameter D = 47 mm received by connecting the two 7-turn coils of FIG. 20A of the eighth embodiment to both ends of the port 2 and facing the XY plane at a distance of 1 mm in the vertical direction. Antenna 2. Both ends of the transmission antenna 1 are connected by a capacitor (capacitance element) having a capacitance C1 of 116 pF, and a capacitance C2 of both ends of the reception antenna 2 is connected by a capacitor (capacitance element) of 15.6 pF in addition to the stray capacitance 5.2 pF. did. The self-inductance L1 of the transmitting antenna 1 is 1.5 μH, and the effective self-inductance L2 of the receiving antenna 2 is 8.9 μH.

この誘導電力伝送回路の送信アンテナ１と受信アンテナ２は、本発明の原理の第１の場合を利用して、共振角周波数ωをωｏにしてアンテナ系を共鳴させ、１２ＭＨｚで共振させた。この送信アンテナ１には３．３Ωの誘導抵抗ｒ１が発生し電源回路３の出力インピーダンスに整合し、受信アンテナ２には４．３Ωの誘導抵抗ｒ２が発生し負荷回路４の入力インピーダンスに整合して電力を伝送した。ただし、この誘導抵抗ｒ１とｒ２は、式３３と式３４に従って、送信アンテナ１の共振電流Ｉ１と受信アンテナ２の共振電流Ｉ２の比によって変わり、ｒ１とｒ２の積の値が同じ他の値のｒ１とｒ２の組み合わせになり得る。この送信アンテナ１と受信アンテナ２の結合係数ｋは、シミュレーションで得られた誘導抵抗から計算すると、結合係数ｋ＝√（ｒ１・ｒ２／（Ｌ１・Ｌ２））／（２πｆ）＝０．０１４である。   Using the first case of the principle of the present invention, the transmitting antenna 1 and the receiving antenna 2 of this inductive power transmission circuit resonate the antenna system with the resonance angular frequency ωo and resonate at 12 MHz. A 3.3Ω induction resistor r1 is generated in the transmission antenna 1 to match the output impedance of the power supply circuit 3, and a 4.3Ω induction resistor r2 is generated in the reception antenna 2 to match the input impedance of the load circuit 4. Power was transmitted. However, the induction resistances r1 and r2 change according to the ratio of the resonance current I1 of the transmission antenna 1 and the resonance current I2 of the reception antenna 2 according to Equations 33 and 34, and the product of r1 and r2 has the same value. It can be a combination of r1 and r2. The coupling coefficient k between the transmitting antenna 1 and the receiving antenna 2 is calculated from the inductive resistance obtained by the simulation. The coupling coefficient k = √ (r1 · r2 / (L1 · L2)) / (2πf) = 0.014 is there.

図２３（ｂ）に、送信アンテナ１から受信アンテナ２への電力伝送効率ＰｅをＳ２１であらわした周波数特性のグラフを示す。１２ＭＨｚの共振周波数ｆでＳ２１は−１．３３ｄＢ、すなわち、電力伝送効率Ｐｅが約７４％の効率良い電力伝送ができる。このように送信アンテナ１と、その直径Ｇの６分の１の直径の受信アンテナ２の組み合わせでも良い効率で電力を伝送できる誘導電力伝送回路が得られた。この送信アンテナ１と受信アンテナ２の結合係数ｋを０．００４以上にすれば、第１の実施形態で説明したように、この誘導電力伝送回路の回路の部品に特性のバラツキが少々あっても両アンテナの共振周波数のずれは電力を伝送するのに支障が無い程度の範囲内に収めることができる効果が得られる。受信アンテナ２の直径Ｄが送信アンテナ１の直径Ｇの１２分の１程度になると、両アンテナの結合係数ｋが０．００４程度になると考えられるため、両アンテナの直径の違いを１２倍以内にすることで支障無く電力を伝送できる効果がある。なお、受信アンテナ２のコイルの位置が送信アンテナ１のコイルの位置からずれる距離が、大きい方のアンテナの直径の２倍以内のずれ距離ならば、第１の実施形態の変形例２及び変形例４の場合のように、電力を伝送するのに支障が無い実用的効果が得られる。   FIG. 23B shows a graph of frequency characteristics in which the power transmission efficiency Pe from the transmitting antenna 1 to the receiving antenna 2 is represented by S21. S21 is −1.33 dB at a resonance frequency f of 12 MHz, that is, efficient power transmission with a power transmission efficiency Pe of about 74% can be performed. In this way, an inductive power transmission circuit capable of transmitting electric power with sufficient efficiency even with a combination of the transmitting antenna 1 and the receiving antenna 2 having a diameter of 1/6 of the diameter G was obtained. If the coupling coefficient k between the transmitting antenna 1 and the receiving antenna 2 is set to 0.004 or more, as described in the first embodiment, even if there is a slight variation in characteristics of the circuit components of the inductive power transmission circuit. It is possible to obtain an effect that the difference between the resonance frequencies of the two antennas can be kept within a range that does not hinder the transmission of power. When the diameter D of the receiving antenna 2 is about 1/12 of the diameter G of the transmitting antenna 1, the coupling coefficient k of both antennas is considered to be about 0.004. By doing so, there is an effect that power can be transmitted without any trouble. If the distance that the position of the coil of the receiving antenna 2 deviates from the position of the coil of the transmitting antenna 1 is a deviation distance that is within twice the diameter of the larger antenna, Modification 2 and Modification of the first embodiment As in the case of No. 4, a practical effect that does not hinder the transmission of power can be obtained.

＜第１３の実施形態＞
第１３の実施形態の誘導電力伝送回路は、送信アンテナ１と電源回路３の組み合わせ、あるいは、受信アンテナ２と電源回路４の組み合わせを空間の電磁波を受け取りあるいは放射するアンテナに置き換えた誘導電力伝送システムである。すなわち、図２４（ａ）のように、送信アンテナ１を、ＸＹ平面上に横（Ｘ方向）に置いた長さ９４０ｍｍで幅１ｍｍの銅のアンテナ配線で形成したダイポールアンテナにし、全体系では、空間の電磁波を受け取って電力に変換するアンテナにすることで、ダイポールアンテナが空間電磁波を受け取り回路に電力に供給する作用を電源回路３とした誘導電力伝送回路である。図２４（ａ）の受信アンテナ２は、ポリイミドフィルム上に形成した直径Ｄ＝５４ｍｍのコイル状の銅のアンテナ配線にし、その受信アンテナ２の端を送信アンテナ１からＹ方向に１ｍｍ隔て、受信アンテナ２と送信アンテナ１をＸＹ面に垂直方向にアンテナ間隔ｈを１０ｍｍ離して設置する。この受信アンテナ２は、第３の実施形態の図１４の、両端を開放した３巻きコイルであり、その中間にポート２を設置し、コイルの両端には外付けコンデンサは接続しない。この誘導電力伝送回路の送信アンテナ１と受信アンテナ２は、アンテナ電流Ｉ１とアンテナ電流Ｉ２の位相差βを９０度にし、ｃｏｓ（β）を０にし、共振角周波数ωをωｏにして、先の式３３から式３５の状態でアンテナ系を共鳴させ、１５４ＭＨｚで共振させる。この送信アンテナ１と受信アンテナ２には３０Ωの誘導抵抗ｒ１とｒ２が発生し電源回路３の出力インピーダンスと負荷回路４の入力インピーダンスに整合して電力を伝送する。ただし、この誘導抵抗ｒ１とｒ２は、式３３と式３４に従って、送信アンテナ１の共振電流Ｉ１と受信アンテナ２の共振電流Ｉ２の比によって変わり、ｒ１とｒ２の積の値が同じ他の値のｒ１とｒ２の組み合わせになり得る。 <13th Embodiment>
The induction power transmission circuit according to the thirteenth embodiment is an induction power transmission system in which the combination of the transmission antenna 1 and the power supply circuit 3 or the combination of the reception antenna 2 and the power supply circuit 4 is replaced with an antenna that receives or radiates electromagnetic waves in the space. It is. That is, as shown in FIG. 24A, the transmission antenna 1 is a dipole antenna formed of copper antenna wiring having a length of 940 mm and a width of 1 mm placed horizontally (X direction) on the XY plane. This is an inductive power transmission circuit in which the power supply circuit 3 has an action in which a dipole antenna receives spatial electromagnetic waves and supplies the electric power to the circuit by using an antenna that receives electromagnetic waves in the space and converts it into electric power. The receiving antenna 2 in FIG. 24A is a coiled copper antenna wiring having a diameter D = 54 mm formed on a polyimide film, and the receiving antenna 2 is separated from the transmitting antenna 1 by 1 mm in the Y direction. 2 and the transmitting antenna 1 are installed in a direction perpendicular to the XY plane with an antenna interval h of 10 mm. The receiving antenna 2 is a three-turn coil having both ends opened as shown in FIG. 14 of the third embodiment. A port 2 is installed in the middle of the coil, and no external capacitor is connected to both ends of the coil. The transmitting antenna 1 and the receiving antenna 2 of this inductive power transmission circuit have a phase difference β between the antenna current I1 and the antenna current I2 of 90 degrees, cos (β) is set to 0, the resonance angular frequency ω is set to ωo, The antenna system is resonated in the state of Expression 33 to Expression 35, and is resonated at 154 MHz. Inductive resistances r1 and r2 of 30Ω are generated in the transmitting antenna 1 and the receiving antenna 2, and power is transmitted in conformity with the output impedance of the power supply circuit 3 and the input impedance of the load circuit 4. However, the induction resistances r1 and r2 change according to the ratio of the resonance current I1 of the transmission antenna 1 and the resonance current I2 of the reception antenna 2 according to Equations 33 and 34, and the product of r1 and r2 has the same value. It can be a combination of r1 and r2.

図２４（ｂ）に、送信アンテナ１から受信アンテナ２への電力伝送効率ＰｅをＳ２１であらわした周波数特性のグラフを示す。１５４ＭＨｚの共振周波数ｆでＳ２１が−１．２１ｄＢで電力伝送効率Ｐｅが約７６％の効率良い電力伝送ができる。このようにダイポール型の送信アンテナ１と、コイル状の受信アンテナ２の組み合わせでも良い効率で電力を伝送できる誘導電力伝送回路が得られる。ここで、受信アンテナ２のアンテナ配線の抵抗ｒ４は変えずに、ダイポールアンテナの送信アンテナ１のアンテナ配線の幅を９倍の１０ｍｍ程度に広くすることでアンテナ配線の抵抗ｒ３を９分の１にする。そして、第１のアンテナの電流Ｉ１を第２のアンテナの電流Ｉ２の３倍にすると、式３３と式３４のαが３分の１になるのでｒ１が３分の１になり、ｒ２が３倍になる。そのとき、式２９に従って、電力伝送効率Ｐｅが高くなり、電力の損失率（１−Ｐｅ）がほぼ３分の１に低減できる。また、本実施形態において、受信アンテナ２と送信アンテナ１をＸＹ面に垂直方向に離すアンテナ間隔ｈを５分の１の２ｍｍに近づけると、誘導抵抗ｒ１＝ｒ２＝４０Ωにおいて、Ｓ２１がー０．６３ｄＢで電力伝送効率Ｐｅが約８６％の効率良い電力伝送ができる。本実施形態の誘導電力伝送回路では、電源回路３と送信アンテナ１の組み合わせは空間電磁波を受け取って回路に電力を供給するダイポールアンテナであるが、先の実施例と同様に効率良く電力が伝送できた。   FIG. 24B shows a graph of frequency characteristics in which the power transmission efficiency Pe from the transmission antenna 1 to the reception antenna 2 is represented by S21. Efficient power transmission is possible with a resonance frequency f of 154 MHz, S21 of -1.21 dB, and power transmission efficiency Pe of about 76%. In this way, an inductive power transmission circuit capable of transmitting power with an efficiency that may be a combination of the dipole transmission antenna 1 and the coiled reception antenna 2 is obtained. Here, without changing the resistance r4 of the antenna wiring of the receiving antenna 2, the antenna wiring resistance r3 is reduced to 1/9 by increasing the width of the antenna wiring of the transmitting antenna 1 of the dipole antenna to about 10 mm, which is nine times. To do. Then, if the current I1 of the first antenna is made three times the current I2 of the second antenna, α in Expression 33 and Expression 34 becomes one third, so r1 becomes one third, and r2 becomes 3 Double. At that time, according to Equation 29, the power transmission efficiency Pe is increased, and the power loss rate (1-Pe) can be reduced to approximately one third. In the present embodiment, when the antenna interval h separating the receiving antenna 2 and the transmitting antenna 1 in the direction perpendicular to the XY plane is close to 1/5 of 2 mm, the inductive resistance r1 = r2 = 40Ω and S21 is −0. Efficient power transmission with a power transmission efficiency Pe of about 86% is possible at 63 dB. In the inductive power transmission circuit of this embodiment, the combination of the power supply circuit 3 and the transmission antenna 1 is a dipole antenna that receives spatial electromagnetic waves and supplies power to the circuit, but can transmit power efficiently as in the previous embodiment. It was.

また、本実施形態の送信アンテナ１と電源回路３の組み合わせと受信アンテナ２と負荷回路４の組み合わせを入れ替えた構成の誘導電力伝送回路も構成できる。すなわち、その誘導電力伝送回路において、受信アンテナ２と負荷回路４の組み合わせをダイポールアンテナにし、そのダイポールアンテナを全体系では空間に電磁波を放射するアンテナとして用い、そのアンテナからの電磁波の放射による電力の消費を負荷回路４とする誘導電力伝送回路を構成することもできる。   In addition, an induction power transmission circuit having a configuration in which the combination of the transmission antenna 1 and the power supply circuit 3 and the combination of the reception antenna 2 and the load circuit 4 of the present embodiment are replaced can be configured. That is, in the inductive power transmission circuit, a combination of the receiving antenna 2 and the load circuit 4 is a dipole antenna, and the dipole antenna is used as an antenna that radiates electromagnetic waves in space in the entire system. An inductive power transmission circuit whose consumption is the load circuit 4 can also be configured.

また、本実施形態は、その寸法を一桁小さくしたアンテナ系を、ポリイミド等のフレキシブル基材上に金属パターンで形成することで実施することもできる。そのダイポールアンテナあるいは、そのダイポールアンテナを螺旋状に巻いたＲＦ受信アンテナを送信アンテナ１及び電源回路３の系として設置し、その送信アンテナ１の近くに設置した縦横６ｍｍの有機樹脂基板あるいはセラミックス基板に金属の配線パターンで、両端を開放した直径約５ｍｍの３巻きの螺旋状（コイル状）の受信アンテナ１を形成する。その受信アンテナ１のアンテナ配線の中間のポート２に直列にＩＣチップの受信回路４を接続することで誘導電力伝送回路を構成できる。アンテナの寸法が一桁小さくなることで共振周波数は一桁大きくなり１．５ＧＨｚ近くになるが誘導抵抗ｒの大きさはほぼ同じ値になる。ここで、アンテナの巻き数を増すか、または、アンテナの両端間に大きな容量の外付けコンデンサを接続することで、その共振周波数を小さくすることもできる。   Moreover, this embodiment can also be implemented by forming an antenna system whose dimensions are reduced by one digit on a flexible substrate such as polyimide with a metal pattern. The dipole antenna or an RF receiving antenna in which the dipole antenna is spirally wound is installed as a system of the transmitting antenna 1 and the power supply circuit 3, and the organic resin substrate or ceramic substrate of 6 mm in length and width installed near the transmitting antenna 1. A three-turn spiral (coiled) receiving antenna 1 having a diameter of about 5 mm with both ends open is formed with a metal wiring pattern. An inductive power transmission circuit can be configured by connecting a receiving circuit 4 of an IC chip in series to a port 2 in the middle of the antenna wiring of the receiving antenna 1. By reducing the size of the antenna by an order of magnitude, the resonance frequency is increased by an order of magnitude and is close to 1.5 GHz, but the size of the induction resistor r is almost the same value. Here, the resonance frequency can be reduced by increasing the number of turns of the antenna or by connecting an external capacitor having a large capacity between both ends of the antenna.

本発明の誘導電力伝送回路は、車両などに電力供給設備から非接触で電力を供給する用途に適用できる。また、ディスプレイ装置等に、家屋の壁を隔てて誘導エネルギーを供給する用途に適用できる。また、生体への非侵襲なシステム構成で、生体内に埋め込んだ電子装置にエネルギーを供給する用途に適用できる。また、半導体集積回路内で集積回路の配線層間で電力を非接触で伝送する用途に適用できる。また、空間の電磁波を受け取るアンテナを電源と送信アンテナの組み合わせとし、あるいは空間に電磁波を放射するアンテナを受信アンテナと負荷回路の組み合わせとし、アンテナと電子機器の間に空間を介して電力を伝送する用途に適用できる。   The inductive power transmission circuit of the present invention can be applied to an application for supplying power to a vehicle or the like from a power supply facility in a contactless manner. Further, the present invention can be applied to an application in which induction energy is supplied to a display device or the like across a wall of a house. Further, it can be applied to an application for supplying energy to an electronic device embedded in a living body with a non-invasive system configuration for the living body. Further, the present invention can be applied to a use in which power is transmitted in a non-contact manner between wiring layers of an integrated circuit within a semiconductor integrated circuit. In addition, the antenna that receives electromagnetic waves in the space is a combination of a power supply and a transmitting antenna, or the antenna that radiates electromagnetic waves in the space is a combination of a receiving antenna and a load circuit, and power is transmitted between the antenna and the electronic device via the space. Applicable to usage.

本発明の第１の実施形態の送信アンテナと受信アンテナの平面図および側面図である。It is the top view and side view of a transmitting antenna and a receiving antenna of a 1st embodiment of the present invention. 本発明の誘導電力伝送回路の回路図である。It is a circuit diagram of the induction power transmission circuit of the present invention. 本発明の第１の実施形態の電力伝送のＳパラメータ（Ｓ２１）のグラフである。It is a graph of S parameter (S21) of electric power transmission of a 1st embodiment of the present invention. 本発明の第１の実施形態のアンテナ間隔ｈによる誘導抵抗ｒのグラフである。It is a graph of the induction resistance r by the antenna space | interval h of the 1st Embodiment of this invention. 本発明の第１の実施形態の変形例１の電力伝送のＳパラメータ（Ｓ２１）のグラフである。It is a graph of S parameter (S21) of electric power transmission of modification 1 of the 1st embodiment of the present invention. 本発明の第１の実施形態の変形例１のアンテナ間隔ｈによる誘導抵抗ｒのグラフである。It is a graph of the induction resistance r by the antenna space | interval h of the modification 1 of the 1st Embodiment of this invention. 本発明の第１の実施形態の変形例２の送信アンテナと受信アンテナの平面図および側面図である。It is the top view and side view of a transmitting antenna and a receiving antenna of the modification 2 of the 1st Embodiment of this invention. 本発明の第１の実施形態の変形例２のアンテナ間隔ｈによる誘導抵抗ｒのグラフである。It is a graph of the induction resistance r by the antenna space | interval h of the modification 2 of the 1st Embodiment of this invention. 本発明の第１の実施形態の変形例２のアンテナ間隔ｈによる電力伝送効率のグラフである。It is a graph of the power transmission efficiency by the antenna space | interval h of the modification 2 of the 1st Embodiment of this invention. 本発明の第１の実施形態の変形例３のアンテナ間隔ｈによる誘導抵抗ｒのグラフである。It is a graph of the induction resistance r by the antenna space | interval h of the modification 3 of the 1st Embodiment of this invention. 本発明の第１の実施形態の変形例４のアンテナのずれ距離ｄによる誘導抵抗ｒのグラフである。It is a graph of the induction resistance r by the deviation | shift distance d of the antenna of the modification 4 of the 1st Embodiment of this invention. 本発明の第１の実施形態の変形例４のアンテナのずれ距離ｄによる電力伝送効率のグラフである。It is a graph of the power transmission efficiency by the deviation | shift distance d of the antenna of the modification 4 of the 1st Embodiment of this invention. 本発明の第２の実施形態の電源回路と負荷回路のインピーダンスＺによる電力伝送効率のグラフである。It is a graph of the power transmission efficiency by the impedance Z of the power supply circuit and load circuit of the 2nd Embodiment of this invention. 本発明の第３の実施形態の送信アンテナと受信アンテナの平面図および側面図である。It is the top view and side view of a transmitting antenna and a receiving antenna of the 3rd Embodiment of this invention. （ａ）本発明の第３の実施形態のアンテナ間隔ｈによる誘導抵抗ｒのグラフである。（ｂ）本発明の第３の実施形態のアンテナ間隔ｈによる電力伝送効率のグラフである。(A) It is a graph of the induction resistance r by the antenna space | interval h of the 3rd Embodiment of this invention. (B) It is a graph of the power transmission efficiency by the antenna space | interval h of the 3rd Embodiment of this invention. （ａ）本発明の第３の実施形態の変形例５のアンテナ間隔ｈによる誘導抵抗ｒのグラフである。（ｂ）本発明の第３の実施形態の変形例５のアンテナ間隔ｈによる電力伝送効率のグラフである。(A) It is a graph of the induction resistance r by the antenna space | interval h of the modification 5 of the 3rd Embodiment of this invention. (B) It is a graph of the power transmission efficiency by the antenna space | interval h of the modification 5 of the 3rd Embodiment of this invention. （ａ）本発明の第４の実施形態のアンテナ間隔ｈによる誘導抵抗ｒのグラフである。（ｂ）本発明の第４の実施形態のアンテナ間隔ｈによる電力伝送効率のグラフである。(A) It is a graph of the induction resistance r by the antenna space | interval h of the 4th Embodiment of this invention. (B) It is a graph of the power transmission efficiency by the antenna space | interval h of the 4th Embodiment of this invention. 本発明の第６の実施形態の送信アンテナと受信アンテナの平面図および側面図である。It is the top view and side view of a transmitting antenna and a receiving antenna of a 6th embodiment of the present invention. （ａ）本発明の第６の実施形態のアンテナ間隔ｈによる誘導抵抗ｒのグラフである。（ｂ）本発明の第６の実施形態のアンテナ間隔ｈによる電力伝送効率のグラフである。(A) It is a graph of the induction resistance r by the antenna space | interval h of the 6th Embodiment of this invention. (B) It is a graph of the power transmission efficiency by the antenna space | interval h of the 6th Embodiment of this invention. （ａ）本発明の第８の実施形態の送信アンテナと受信アンテナの平面図および側面図である。（ｂ）本発明の第８の実施形態の電力伝送のＳパラメータ（Ｓ２１）のグラフである。(A) It is the top view and side view of a transmitting antenna and a receiving antenna of the 8th Embodiment of this invention. (B) It is a graph of S parameter (S21) of the power transmission of the 8th Embodiment of this invention. （ａ）本発明の第９の実施形態の送信アンテナと受信アンテナの平面図および側面図である。（ｂ）本発明の第９の実施形態の電力伝送のＳパラメータ（Ｓ２１）のグラフである。(A) It is the top view and side view of a transmitting antenna and a receiving antenna of the 9th Embodiment of this invention. (B) It is a graph of S parameter (S21) of electric power transmission of the 9th embodiment of the present invention. （ａ）本発明の送信アンテナと受信アンテナを示す平面図である。（ｂ）本発明の第１１の実施形態の変成器を示す平面図である。(A) It is a top view which shows the transmitting antenna and receiving antenna of this invention. (B) It is a top view which shows the transformer of the 11th Embodiment of this invention. （ａ）本発明の第１２の実施形態の送信アンテナと受信アンテナの平面図である。（ｂ）本発明の第１２の実施形態の電力伝送のＳパラメータ（Ｓ２１）のグラフである。(A) It is a top view of the transmitting antenna and receiving antenna of 12th Embodiment of this invention. (B) It is a graph of S parameter (S21) of electric power transmission of the 12th embodiment of the present invention. （ａ）本発明の第１３の実施形態の送信アンテナと受信アンテナの平面図である。（ｂ）本発明の第１３の実施形態の電力伝送のＳパラメータ（Ｓ２１）のグラフである。(A) It is a top view of the transmitting antenna and receiving antenna of 13th Embodiment of this invention. (B) It is a graph of S parameter (S21) of electric power transmission of the 13th embodiment of the present invention.

符号の説明Explanation of symbols

１・・・送信アンテナ
２・・・受信アンテナ
３・・・電源回路
４・・・負荷回路
５・・・インピーダンス変換回路
６・・・誘導結合配線
Ｃ１、Ｃ２・・・容量
ｄ・・・ずれ距離
Ｄ・・・送信アンテナ径
Ｇ・・・受信アンテナ径
ｈ・・・アンテナ間隔
Ｉ１、Ｉ２・・・アンテナ電流
Ｌ１、Ｌ２・・・実効的自己インダクタンス
ｒ、ｒ１、ｒ２・・・誘導抵抗
Ｚ１、Ｚ２・・・インピーダンス DESCRIPTION OF SYMBOLS 1 ... Transmission antenna 2 ... Reception antenna 3 ... Power supply circuit 4 ... Load circuit 5 ... Impedance conversion circuit 6 ... Inductive coupling wiring C1, C2 ... Capacitance d ... Deviation Distance D ... Transmitting antenna diameter G ... Receiving antenna diameter h ... Antenna spacing I1, I2 ... Antenna current L1, L2 ... Effective self-inductance r, r1, r2 ... Inductive resistance Z1 , Z2 ... impedance

Claims (12)

電源回路に接続した送信アンテナから角周波数ωの交流電力を空間を隔てた受信アンテナに伝送し負荷回路に伝送する誘導電力伝送回路であって、両端を容量Ｃ１でつないだ、実効的自己インダクタンスがＬ１の送信アンテナに電源回路を直列に接続した回路と、両端を容量Ｃ２でつないだ、実効的自己インダクタンスがＬ２の受信アンテナに負荷回路を直列に接続した回路を有し、前記送信アンテナと前記受信アンテナの間の距離を、電力を伝送する電磁界の波長の２π分の１以下の距離にして前記送信アンテナと前記受信アンテナの電磁誘導の結合係数をｋにし、０ラジアン以上πラジアン以下の値の位相角βに関して、前記角周波数ωを、Ｌ２×Ｃ２×（１＋ｋ・ｃｏｓ（β））の値の逆数の平方根にして、前記電源回路の出力インピーダンスを約ｋωＬ１・ｓｉｎ（β）≡ｒ１にし、前記負荷回路の入力インピーダンスを約ｋωＬ２・ｓｉｎ（β）≡ｒ２にして前記電源回路から前記負荷回路に効率良く電力を伝送することを特徴とする誘導電力伝送回路。   An inductive power transmission circuit that transmits AC power of an angular frequency ω from a transmitting antenna connected to a power supply circuit to a receiving antenna spaced apart and transmits it to a load circuit, and has an effective self-inductance with both ends connected by a capacitor C1. A circuit in which a power supply circuit is connected in series to a transmission antenna of L1, and a circuit in which a load circuit is connected in series to a reception antenna having an effective self-inductance of L2, in which both ends are connected by a capacitor C2. The distance between the receiving antennas is set to a distance equal to or less than 1 / 2π of the wavelength of the electromagnetic field transmitting power, and the coupling coefficient of electromagnetic induction between the transmitting antenna and the receiving antenna is set to k, and is 0 radians or more and π radians or less. With respect to the phase angle β of the value, the angular frequency ω is set to the square root of the reciprocal of the value of L2 × C2 × (1 + k · cos (β)), and the output impedance of the power supply circuit The power is efficiently transmitted from the power supply circuit to the load circuit with the input impedance of the load circuit set to about kωL2 · sin (β) ≡r2 and the input impedance of the load circuit is set to about kωL1 · sin (β) ≡r1. Inductive power transmission circuit. 請求項１に記載の誘導電力伝送回路において、前記電源回路と前記送信アンテナの回路を、前記送信アンテナに第１の誘導結合配線が相互インダクタンスＭ１で誘導結合し、前記第１の誘導結合配線の両端に元の電源回路を接続した回路に代え、前記元の電源回路の出力インピーダンスを約（２πｆ×Ｍ１）２／ｒ１にしたことを特徴とする誘導電力伝送回路。 2. The inductive power transmission circuit according to claim 1, wherein the power supply circuit and the transmission antenna circuit are inductively coupled to the transmission antenna by a first inductance coupling wire with a mutual inductance M <b> 1. An inductive power transmission circuit characterized in that the output impedance of the original power supply circuit is set to about (2πf × M1) 2 / r1 instead of a circuit in which the original power supply circuit is connected to both ends. 請求項１に記載の誘導電力伝送回路において、前記負荷回路と前記受信アンテナの回路を、前記受信アンテナに第２の誘導結合配線が相互インダクタンスＭ２で誘導結合し、前記第２の誘導結合配線の両端に元の負荷回路を接続した回路に代え、前記元の負荷回路の入力インピーダンスを約（２πｆ×Ｍ２）２／ｒ２にしたことを特徴とする誘導電力伝送回路。 2. The inductive power transmission circuit according to claim 1, wherein the load circuit and the receiving antenna circuit are inductively coupled to the receiving antenna by a second inductance coupling wire with a mutual inductance M <b> 2. An inductive power transmission circuit characterized in that the input impedance of the original load circuit is about (2πf × M2) 2 / r2 instead of a circuit in which the original load circuit is connected to both ends. 請求項２に記載の誘導電力伝送回路において、前記第１の誘導結合配線を前記送信アンテナが共用し、前記元の電源回路の出力インピーダンスを約（２πｆ×Ｌ１）２／ｒ１にしたことを特徴とする誘導電力伝送回路。 3. The inductive power transmission circuit according to claim 2, wherein the first inductive coupling wiring is shared by the transmitting antenna, and the output impedance of the original power supply circuit is about (2πf × L1) 2 / r1. Inductive power transmission circuit. 請求項３に記載の誘導電力伝送回路において、前記第２の誘導結合配線を前記受信アンテナが共用し、前記元の負荷回路の入力インピーダンスを約（２πｆ×Ｌ２）２／ｒ２にしたことを特徴とする誘導電力伝送回路。 4. The inductive power transmission circuit according to claim 3, wherein the second inductive coupling wiring is shared by the receiving antenna, and an input impedance of the original load circuit is set to about (2πf × L2) 2 / r2. Inductive power transmission circuit. 請求項１に記載の誘導電力伝送回路において、前記電源回路と前記送信アンテナと前記容量Ｃ１の回路を、空間から電磁波を受け取るアンテナに代えたことを特徴とする誘導電力伝送回路。   The inductive power transmission circuit according to claim 1, wherein the power supply circuit, the transmission antenna, and the circuit of the capacitor C1 are replaced with an antenna that receives electromagnetic waves from a space. 請求項１に記載の誘導電力伝送回路において、前記負荷回路と前記受信アンテナと前記容量Ｃ２の回路を、空間に電磁波を放射するアンテナに代えたことを特徴とする誘導電力伝送回路。   The inductive power transmission circuit according to claim 1, wherein the load circuit, the reception antenna, and the circuit of the capacitor C <b> 2 are replaced with an antenna that radiates electromagnetic waves in space. 電源回路に接続した送信アンテナから角周波数ωの交流電力を空間を隔てた受信アンテナに伝送し負荷回路に伝送する誘導電力伝送回路であって、両端を容量Ｃ１でつないだ、実効的自己インダクタンスがＬ１の送信アンテナに電源回路を直列に接続した回路と、両端を容量Ｃ２でつないだ、実効的自己インダクタンスがＬ２の受信アンテナに負荷回路を直列に接続した回路を有し、前記送信アンテナと前記受信アンテナの間の距離を、電力を伝送する電磁界の波長の２π分の１以下にして相互インダクタンスをＭにし、前記角周波数ωの値をＬ２×Ｃ２の逆数の平方根の値にし、前記電源回路の出力インピーダンスＺ１を、前記負荷回路側での誘導インピーダンス値（ωＭ）２／Ｚ１に変換し、前記誘導インピーダンス値に前記負荷回路の入力入力インピーダンスを等しくして電力を伝送することを特徴とする誘導電力伝送回路。 An inductive power transmission circuit that transmits AC power of an angular frequency ω from a transmitting antenna connected to a power supply circuit to a receiving antenna spaced apart and transmits it to a load circuit, and has an effective self-inductance with both ends connected by a capacitor C1. A circuit in which a power supply circuit is connected in series to a transmission antenna of L1, and a circuit in which a load circuit is connected in series to a reception antenna having an effective self-inductance of L2, in which both ends are connected by a capacitor C2. The distance between the receiving antennas is set to be equal to or less than 1 / 2π of the wavelength of the electromagnetic field transmitting power, the mutual inductance is set to M, the value of the angular frequency ω is set to the square root of the reciprocal of L2 × C2, and the power source the output impedance Z1 of the circuit, and converts the inductive impedance values (ωM) 2 / Z1 in the load circuit side, the input of the load circuit to the inductive impedance value Inductive power transfer circuit is made equal to the input impedance, characterized in that transmitting power. 請求項８に記載の誘導電力伝送回路において、前記電源回路と前記送信アンテナの回路を、前記送信アンテナに第１の誘導結合配線が相互インダクタンスＭ１で誘導結合し、前記第１の誘導結合配線の両端に元の電源回路を接続した回路に代え、前記元の電源回路の出力インピーダンスＺ３を、前記負荷回路側での誘導抵抗値（Ｍ／Ｍ１）２×Ｚ３に変換することを特徴とする誘導電力伝送回路。 9. The inductive power transmission circuit according to claim 8, wherein the power supply circuit and the transmission antenna circuit are inductively coupled to the transmission antenna by a first inductance coupling wire with a mutual inductance M1. Instead of a circuit in which the original power supply circuit is connected to both ends, the output impedance Z3 of the original power supply circuit is converted into an induction resistance value (M / M1) 2 × Z3 on the load circuit side. Power transmission circuit. 請求項８に記載の誘導電力伝送回路において、前記負荷回路と前記受信アンテナの回路を、前記受信アンテナに第２の誘導結合配線が相互インダクタンスＭ２で誘導結合し、前記第２の誘導結合配線の両端に元の負荷回路を接続した回路に代え、前記電源回路の出力インピーダンスＺ１を、前記元の負荷回路側での誘導抵抗値（Ｍ２／Ｍ）２×Ｚ１に変換することを特徴とする誘導電力伝送回路。 9. The inductive power transmission circuit according to claim 8, wherein the load circuit and the reception antenna circuit are inductively coupled to the reception antenna by a second inductive coupling wiring with a mutual inductance M2. Instead of a circuit in which an original load circuit is connected to both ends, an output impedance Z1 of the power supply circuit is converted into an induction resistance value (M2 / M) 2 × Z1 on the original load circuit side. Power transmission circuit. 請求項９に記載の誘導電力伝送回路において、前記第１の誘導結合配線を前記送信アンテナが共用し、前記元の電源回路の出力インピーダンスＺ３を、前記負荷回路側での誘導抵抗値（Ｍ／Ｌ１）２×Ｚ３に変換することを特徴とする誘導電力伝送回路。 The inductive power transmission circuit according to claim 9, wherein the transmission antenna shares the first inductive coupling wiring, and the output impedance Z3 of the original power supply circuit is an inductive resistance value (M / L1) An inductive power transmission circuit characterized by converting to 2 × Z3. 請求項１０に記載の誘導電力伝送回路において、前記第２の誘導結合配線を前記受信アンテナが共用し、前記電源回路の出力インピーダンスＺ１を、前記元の負荷回路側での誘導抵抗値（Ｌ２／Ｍ）２×Ｚ１に変換することを特徴とする誘導電力伝送回路。 11. The inductive power transmission circuit according to claim 10, wherein the receiving antenna shares the second inductive coupling wiring, and the output impedance Z1 of the power circuit is changed to an inductive resistance value (L2 / M) Inductive power transmission circuit characterized by converting to 2 × Z1.
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