JP2008505596A - Method and system for starting permanent magnet synchronous motor drive - Google Patents

Abstract

  A method for determining a rotor angle of an electric motor, comprising: (a) determining a rotor magnetic flux in the electric motor; (b) estimating a rotor angle based on the rotor magnetic flux; and (c) an electric motor. Correcting the estimated rotor angle based on the reactive power input to. Step (a) non-ideally integrates the voltage value and current value of the stator, and step (b) corrects a phase error caused by the non-ideal integration by a PLL circuit (F) having phase compensation, In step (c), (1) the first reactive power input value is set to 1.5 * We * (C_Lq * I * I), and the second reactive power input value is set to 1.5 * (Vq * id− Vd * iq), (2) determining the difference between the first and second reactive power input values, and (3) applying the difference to the rotor angle estimated in step (b) to correct Obtaining a determined rotor angle. The estimated rotor angle is based on the rotor flux at high frequencies and on the motor load model used with the start-up sequence logic at low frequencies.

Description

  The present invention relates to motor drive control, and more particularly to techniques for estimating rotor angle in permanent magnet synchronous motor (PMSM) drive, particularly when PMSM is started from a stationary state.

  Rotor position information is generally required for stable operation of permanent magnet AC motors with sinusoidal current excitation. The continuous rotor position is conventionally obtained from an encoder attached to the shaft of the motor or indirectly from an estimation algorithm based on voltage and current feedback. The latter is preferred because it results in reduced system and operating costs.

  However, most passive rotor angle estimation schemes based on measured voltage and current are complex and require accurate information on motor parameters such as resistance and inductance. These parameters, especially the stator resistance, vary widely with temperature. This makes the rotor angle estimation inaccurate and leads to control stability issues, reduced torque performance per ampere, and reduced motor operating efficiency.

  Patent Document 1 describes a method in which the rotor angle is estimated by a phase-locked loop (PLL) that continuously tracks the magnetic flux of the motor during the normal operation mode.

  However, the inventor of the present invention has realized that additional problems arise at start-up. Under zero speed or low speed (<10%) conditions, it is difficult to accurately measure or estimate the motor voltage because the back electromotive force (BEMF) of the motor has a low amplitude.

  For most sensorless (no axis encoder) control drives, BEMF based rotor angle tracking usually fails at low speeds (<5%). Therefore, some means for starting the motor is necessary for sensorless control of the permanent magnet motor drive.

  In most cases, the motor is started in an open loop (no feedback) manner. As soon as the motor speed increases (generally over 10%), the drive is switched to a closed-loop (using current or voltage feedback) control mode. However, while switching from open loop to closed loop, torque and current pulsations may occur due to mode transitions.

  Therefore, it would be desirable to provide a rotor angle estimation method that maximizes torque performance per ampere without requiring accurate information on stator resistance or other motor parameters.

Furthermore, it would be desirable to provide a method for estimating the rotor angle at start-up so as to provide a powerful start-up and a reduction in the occurrence of torque pulsations during motor start-up.
US Application No. 10 / 294,201

  The present invention provides a novel method for estimating rotor angle information for controlling a permanent magnet AC motor having a sinusoidal counter electromotive force.

  The rotor angle is estimated by a phase locked loop (with phase error compensation) that receives an estimate of the rotor flux. The rotor flux is derived from the stator voltage (actual or command voltage), current, resistance, and inductance.

  The rotor angle estimation error (change in stator resistance with temperature) is then removed by using a novel angle error corrector. This corrector is based on reactive power compensation and is insensitive to resistance changes. Furthermore, only one inductance parameter is required for the reference model of the angle error corrector.

  In order to enable powerful start-up and reduce torque pulsation during motor start-up, the present invention utilizes the above-described PLL together with newly developed start-up logic. This activation logic includes an activation sequencer and a machine model. These components can be very simple because they use the same PLL integrator used in US Pat.

  The estimated flux information is used in a closed loop mode by the PLL to track the rotor angle. A simple machine model is used in open loop mode by the PLL to predict the rotor angle. Startup sequence logic is used to allow a strong and smooth transition from open loop to closed loop control mode during motor startup.

  Accordingly, a rotor angle estimation algorithm for sensorless control of a PM motor will include one or more of the following exemplary functions.

  (1) A magnetic flux estimator having feedforward inductance compensation. The vector PLL is synchronized with the output of the flux estimator. A non-ideal flux estimator is used with a magnetic flux PLL for rotor angle estimation.

  (2) A phase compensation circuit F included in the PLL for removing the phase error derived by the non-ideal magnetic flux estimator.

  (3) A compensation scheme based on reactive power eliminates the sensitivity of the rotor angle estimation method to stator resistance fluctuations.

  (4) The startup sequencer and load model operate with a vector PLL to achieve strong startup and smooth speed increase.

  Other features and advantages of the present invention will become apparent from the following description of embodiments of the invention with reference to the accompanying drawings.

  The present invention will be described herein with reference to a motor control algorithm executed in firmware. However, the scope of the present invention includes embodiments that combine hardware, firmware, and software that can be foreseen by those skilled in the art.

  FIG. 1 shows a block diagram of a first embodiment of the control method. The d-axis is in the same direction as the rotor magnet axis (specifications used in the literature).

  The following is a definition of the quantities described in FIG.

  The rotor angle estimation block of FIG. 1 is shown in detail in FIG. Flx_A and Flx_B are the rotor fluxes obtained by non-ideal integration of the motor back electromotive force formed by the stator current, voltage, resistance and inductance, as shown in FIG. It is. In these figures, Tf represents the time constant of the non-ideal integrator.

  The inputs (V_A, V_B, I_A, and I_B) to the magnetic flux estimator in FIG. 3 are signals obtained by converting three phases (ia, ib, Vab, Vbc) into two phases for simplicity.

  The rotor angle estimator (FIG. 2) utilizes a novel flux phase locked loop system. The frequency feedforward circuit F compensates for phase errors due to non-ideal integration of the stator voltage used in FIG. 3 to obtain the magnetic flux. A phase error caused by non-ideal integration is completely compensated in the circuit F.

  The estimated error due to resistance is compensated by a rotor angle correction system described below in connection with FIG.

The stator angle corrector circuit of FIG. 1 is shown in detail in FIG. When the estimated rotor angle (FIG. 1) matches the actual rotor angle, the reference value of the reactive power Q input to the motor is
1.5 * We * (C_Lq * I * I + Flx_M * id + (C_Ld−C_Lq) * id * id)
Is equal to

  However, it should be noted that for permanent magnet surface mount (PMSM) motors, the magnetoresistance of the air gap is exactly the same in the d-axis and q-axis. Therefore, id = 0 and Ld = Lq. Therefore, the above-mentioned reference reactive power equation can be reduced to the following equation.

  1.5 * We * (C_Lq * I * I)

Next, the reactive power Q of an actual motor expressed only by voltage and current is
Q = 1.5 * (Vq * id−Vd * iq)
Calculated by
In the above formula,
C_Ld d-axis inductance C_Lq q-axis inductance I Stator current magnitude Flx_M Rotor magnet equivalent flux linkage number Q Terminal reactive power and We (omega e) Stator fundamental frequency.

  Since C_Ld = C_Lq, the rotor angle corrector can be realized with only one inductance parameter (Lq or Ld). Lq is used in this case. Of course, as will be appreciated by those skilled in the art, the present invention is also suitable for use with other motor types, such as embedded permanent magnet motors where Ld is not equal to Lq.

When the estimated rotor angle matches the actual rotor angle, the following relational expression is satisfied.
(Vq * id−Vd * iq) −We * C_Lq * I * I = 0

  Therefore, the reactive power error between Q and (We * C_Lq * I * I) (vertical axis in FIG. 5) can be used to zero the rotor angle error (horizontal axis in FIG. 5). Therefore, the maximum torque per ampere can be maintained even when there is an error in the resistance parameter used in the magnetic flux estimator (FIG. 3).

  The rotor angle estimation block of the second embodiment according to the present invention is shown in FIG. This system is simplified by deleting the upper block of the Mod-2π block in the system of FIG.

  A third embodiment is shown in FIG. In FIG. 7, the system of FIG. 6 is modified. In the first and second embodiments, two inputs of a PI (proportional integral) regulator are coupled together and both inputs receive the Pll_Err output from Demod-Flux. In this embodiment, the two inputs of the PI regulator receive their respective separated outputs of the startup block, and the two inputs of the startup block both receive Pll_Err from Demod-Flux.

The activation module and its interface to the PLL angle tracking module are shown in more detail in FIG. In FIG.
Rtr_Ang Estimated rotor angle Flx_A Estimated alpha magnetic flux Flx_B Estimated beta magnetic flux Pll_Err PLL error signal Trq Estimated motor torque We Inverter basic frequency.

  FIG. 6 shows the same PLL (FIG. 2) but with the block F moved to the PLL feedback path. This is done to provide the convenience of interfacing with the startup module. The movement of the block F does not affect the function of the PLL because the main function of the block F is to phase shift the estimated rotor angle Rtr_Ang. The PI block in FIG. 2 is expanded in FIG. 8 to show the P and I paths for interfacing with the activation module.

  FIG. 8 shows an interface between the activation module and the PLL module.

  When the motor BEMF is small (<10%), the fidelity of flux signals Flx_A and Flx_B (calculated from the estimated or measured motor voltage) is reduced, resulting in an inaccurate error correction signal Pll_Err. To address this problem, Pll_Err is generated by a simple motor model (load model in FIG. 8) at first start-up.

  The load model is only used for a short time during startup. When the motor frequency We reaches a certain threshold, the fidelity of Flx_A and Flx_B is improved and then Pl_Err is calculated from the quantity of Flx_A and Flx_B.

  In FIG. 8, the switches Sw1 and Sw2 are in the upper position during motor startup. The startup sequencer is in a stopped state and the input to the PI regulator is zero. In the stopped state, the initial motor angle is captured (by common techniques such as direct current injection) and initialized. After the stop is made, Sw2 is closed while Sw1 is still in the upper position (open loop mode).

  The integrator input of the PI regulator is given by a simple load model consisting of two gain multipliers Kt and Kf. The Kt path simulates the acceleration torque of the electric motor and is given by the load torque current feedback iq. The Kf path simulates the friction torque and is given by the frequency We.

  In some cases, the acceleration torque will be much greater than the friction torque during motor start-up, and in such cases, the Kf path can be dispensed with.

  When the motor starts to accelerate, the motor frequency We also increases. As soon as the absolute motor frequency exceeds a certain threshold (usually 10% of the rated frequency), the switch Sw1 is closed and the closed loop mode is activated.

  Although the invention has been described with reference to specific embodiments thereof, many other variations and modifications and other uses will be apparent to those skilled in the art. Accordingly, the present invention is not limited only by the specific disclosure herein.

1 is a block diagram showing a PMSM control system including a first embodiment of the present invention. FIG. It is a block diagram which shows the rotor angle estimator of FIG. 1 in detail. FIG. 3 is a circuit diagram of a rotor flux estimator associated with the block diagram of FIG. 2. It is a figure which shows the rotor angle corrector of FIG. 1 in detail. It is a graph which shows the relationship between a reactive power error and a rotor angle error which makes an electric motor rated output a unit. It is a block diagram which shows 2nd Embodiment of a rotor angle estimator. It is a block diagram which shows 3rd Embodiment of a rotor angle estimator. It is a figure which shows the rotor angle estimator of FIG. 7 in detail.

Explanation of symbols

F block, phase compensation circuit Flx_A Estimated alpha magnetic flux, magnetic flux signal Flx_B Estimated beta magnetic flux, magnetic flux signal Kt, Kf Gain multiplier Q Reactive power Rtr_Ang Estimated rotor angle Sw1, Sw2 Switch We Motor frequency iq Load torque current feedback

Claims (42)

A method for determining a rotor angle in drive control of an electric motor,
a) estimating the rotor angle based on the rotor flux in the motor;
b) correcting the estimated rotor angle based on the reactive power input to the electric motor,
The step (a) further includes the step of (a1) estimating the rotor angle according to a predetermined motor load model together with a startup sequencer when the motor is started.   The method of claim 1, wherein the load model is representative of motor acceleration torque.   The method of claim 2, wherein the model is responsive to load torque current feedback (iq).   The method of claim 2, wherein the load model is representative of friction torque.   The method of claim 4, wherein the model is adapted to be responsive to a motor frequency (We).   The method of claim 1, wherein step (a1) ends with an adjustable percentage of rated motor frequency.   The method of claim 6, wherein the adjustable percentage is about 10 percent.   The method of claim 1, wherein the step (a1) is performed in an open loop mode and ends upon transition from an open loop mode to a closed loop mode. A method for determining a rotor angle in drive control of an electric motor,
a) determining the rotor flux in the motor;
b) estimating the rotor angle based on the rotor magnetic flux in the motor and according to a predetermined motor load model together with the startup sequencer at motor startup;
c) correcting the estimated rotor angle based on reactive power input to an electric motor,
A method wherein step (a) includes non-ideal integration of stator voltage and current values.   The method of claim 9, wherein the load model is representative of motor acceleration torque.   The method of claim 10, wherein the model is adapted to respond to load torque current feedback (iq).   The method of claim 10, wherein the load model is representative of friction torque.   The method of claim 12, wherein the model is adapted to be responsive to a motor frequency (We).   The method of claim 9, wherein step (b) ends with an adjustable percentage of rated motor frequency.   The method of claim 14, wherein step (b) ends at about 10% of the rated motor frequency.   10. The method of claim 9, wherein step (b) is performed in an open loop mode and ends upon transition from an open loop mode to a closed loop mode.   The method of claim 9, wherein step (a) further comprises correcting a phase error caused by the non-ideal integration with a PLL circuit having phase compensation. In drive control of an electric motor, a system for determining a rotor angle,
A first circuit for estimating a rotor angle based on a rotor magnetic flux in the motor;
A second circuit for correcting the estimated rotor angle based on reactive power to the motor;
The system, wherein the first circuit further estimates the rotor angle according to a predetermined motor load model together with a startup sequencer when the motor is started.   The system of claim 18, wherein the load model is representative of motor acceleration torque.   The system of claim 19, wherein the model is adapted to respond to load torque current feedback (iq).   The system of claim 19, wherein the load model is representative of friction torque.   The system of claim 21, wherein the model is responsive to a motor frequency (We).   The system of claim 18, wherein the estimating step ends with an adjustable percentage of rated motor frequency.   24. The system of claim 23, wherein the estimating step ends at about 10% of a rated motor frequency.   The system of claim 18, wherein the estimating step is performed in an open loop mode and ends upon transition from an open loop mode to a closed loop mode. In drive control of an electric motor, a system for determining a rotor angle,
a) a first circuit for determining the rotor flux in the motor;
b) a second circuit for estimating the rotor angle based on the rotor magnetic flux in the motor and in accordance with a predetermined motor load model together with a startup sequencer at motor startup;
c) a third circuit for correcting the estimated rotor angle based on the reactive power input to the motor;
A system in which the first circuit is adapted to non-ideally integrate the stator voltage and current values.   27. The system of claim 26, wherein the load model is representative of motor acceleration torque.   28. The system of claim 27, wherein the model is adapted to respond to load torque current feedback (iq).   28. The system of claim 27, wherein the load model is adapted to represent friction torque.   30. The system of claim 29, wherein the model is responsive to a motor frequency (We).   27. The system of claim 26, wherein the estimating step ends with an adjustable percentage of rated motor frequency.   32. The system of claim 31, wherein the estimating step is adapted to end at about 10% of a rated motor frequency.   27. The system of claim 26, wherein the estimating step is performed in an open loop mode and ends upon transition from an open loop mode to a closed loop mode.   27. The system according to claim 26, wherein the second circuit is adapted to correct a phase error caused by the non-ideal integration by a PLL circuit (F) having phase compensation.   The correction step includes calculating a first reactive power input value and a second reactive power input value, determining a relationship between the first and second reactive power input values, and estimating the relationship The method of claim 1, wherein the method is performed by applying to an estimated rotor angle to obtain a corrected rotor angle.   The estimation step includes a step of non-ideal integration of a voltage value and a current value of a stator, and a step of correcting a phase error caused by the non-ideal integration by a PLL circuit (F) having phase compensation. Item 2. The method according to Item 1.   The step of correcting the estimated rotor angle based on the reactive power input to the electric motor calculates a first reactive power input value and a second reactive power input value, the first and first 10. The method of claim 9, wherein the method is performed by determining a relationship between two reactive power input values and applying the relationship to the rotor angle estimated in step (b) to obtain a corrected rotor angle. .   The method according to claim 9, wherein step (b) comprises the step of correcting a phase error caused by the non-ideal integration by means of a PLL circuit (F) having phase compensation.   The second circuit calculates a first reactive power input value and a second reactive power input value based on the estimated rotor angle based on the reactive power input to the electric motor; The system of claim 18, wherein the relationship is determined by determining a relationship between two reactive power input values and applying the relationship to the rotor angle estimated in the estimating step to obtain a corrected rotor angle. . The first circuit non-ideally integrates the voltage value and current value of the stator,
The system according to claim 18, wherein the second circuit corrects a phase error caused by the non-ideal integration by a PLL circuit (F) having phase compensation.   The estimated rotor angle is obtained by calculating a first reactive power input value and a second reactive power input value based on reactive power input to the electric motor, and the first and second reactive power input values. 27. The system of claim 26, wherein the system is corrected by determining the relationship and applying the relationship to the rotor angle estimated in step (b) to obtain a corrected rotor angle.   27. The system according to claim 26, wherein the second circuit corrects a phase error caused by the non-ideal integration by a PLL circuit (F) having phase compensation.

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