JP2007124714A - Oscillator and quadrature modem comprising the same - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To obtain a crystal quadrature output oscillator current consumption of which is made low, used for QPSK or QAM of digital communication. <P>SOLUTION: A crystal quadrature output oscillator comprises a crystal oscillation circuit including a transistor, a crystal vibrating chip, a capacitor and a resistor, a duplex mode crystal filter, a reactance element and an amplifier. The crystal quadrature output oscillator defines an output of the crystal oscillation circuit as a first output, applies the output to one terminal of the duplex mode crystal filter in which a common electrode is grounded, connects the reactance element in parallel between another terminal and a ground, and is configured to obtain a second output from the node through the amplifier. <P>COPYRIGHT: (C)2007,JPO&INPIT

Description

The present invention relates to a quadrature output oscillator used for digital communication, and more particularly to a crystal quadrature output oscillator that achieves low current consumption.

In recent years, high-speed and large-capacity systems that transmit and receive vast amounts of information at high speed have become mainstream in mobile communication systems, and the modulation method has high frequency utilization efficiency and is suitable for high-speed communication.
A modulation method using AM is widely used. The QPSK and QAM modulation circuits have 9
An orthogonal modulation method is used in which a carrier wave having a phase difference of 0 degrees is multiplied by a modulation signal. Regarding the quadrature modulation system, JP-A-8-223233 and JP-A-2001-217.
No. 886 and the like are disclosed in detail.
Figure 7 is a block diagram showing the configuration of a 90-degree phase shifter which is disclosed in JP 2001-217886, having an oscillator 21 for generating a local signal L _O, a phase difference from each other from the local signal L _O A phase shifter 22 that generates two local phase shift signals L _O I ′ and L _O Q ′ and a clock signal CLK having a frequency twice that of the local phase shift signals L _O I ′ and L _O Q ′ are generated. Oscillator 23
And a D flip-flop 24 that receives the local phase shift signal L _O Q ′ and the clock signal CLK and outputs a quadrature component L _O Q of the 90 ° phase shift signal, and is inverted to the local phase shift signal L _O I ′. And a D flip-flop 25 that receives the clock signal CLK and outputs the in-phase component L _O I of the 90-degree phase shift signal.

The local signal output from the oscillator 21 is input to the phase shifter 22, and two local phase shift signals L _O I ′ and L _O Q ′ having a phase difference are output from the phase shifter 22. Since the local phase shift signals L _O I ′ and L _O Q ′ include a phase shift error, the phase difference between them is not 90 degrees. The local phase shift signals L _O I ′ and L _O Q ′ are respectively supplied to the D flip-flops 24 and 25.
To enter. On the other hand, a clock signal CLK having a frequency f _CLK = 2 × f 1 ′ that is twice the frequency f 1 ′ of the local phase shift signals L _O I ′ and L _O Q ′ is non-inverted and input to the D flip-flop 24.
The D flip-flop 25 is inverting input. Therefore, the local phase shift signal L _O I ′ input to the D flip-flop 25 is shaped at the falling timing of the clock signal and output as the phase shift signal L _O I. The local phase shift signal L _O Q ′ input to the D flip-flop 24 is shaped at the rising timing of the clock signal and output as the phase shift signal L _O Q.

As a result, the phase shift signal L _O I and the phase shift signal L _O Q are signals having a phase difference of 1/2 cycle of the clock signal CLK (1/4 cycle of the phase shift signal), that is, 90 degrees each other. It is output synchronously as a signal having a phase difference. FIG. 8 is a timing chart of each part of the 90-degree phase shifter shown in FIG. Here, a is the clock signal CLK, b is local phase shifting signals L _O I 'phase component, c is the local phase shifting signals L _O Q' quadrature component L _O Q ', d is phase of 90 degree phase shifter of Component L _O I,
e indicates the quadrature component L _O Q of the 90-degree phase shifter.

JP-A-8-223233 JP 2001-217886 A

The problem to be solved is that the conventional phase shifter shown in FIG. 7 uses a double frequency clock oscillator and two D flip-flops to generate two local signals (carrier waves) that are 90 degrees out of phase. is doing. For example, when trying to obtain a 100 MHz local signal (carrier wave) from a 90 degree phase shifter, an oscillation circuit for a 200 MHz clock signal is required, and in order to continue oscillation in this high frequency oscillation circuit, a negative magnitude of a predetermined magnitude is required. It is necessary to generate a resistance, and a large current is required. Furthermore, since the current consumption of the frequency divider circuit is also added, there is a problem that the current cannot be reduced.

In order to reduce current consumption, the present invention includes a crystal oscillation circuit including a transistor, a crystal resonator, a capacitor and a resistor, a dual mode crystal filter, a reactance element, and an amplifier, and outputs the crystal oscillation circuit. The first output is added to one terminal of the dual-mode crystal filter whose common electrode is grounded, and the reactance element is connected in parallel between the other terminal and the ground, and an amplifier is connected from the connection point. The crystal quadrature output type oscillator is configured so as to obtain the second output.

The crystal quadrature output oscillator according to the present invention does not require a double frequency and does not require a flip-flop circuit, and thus has an advantage that the current consumption of the oscillation circuit can be reduced and the size can be reduced.

FIG. 1 is a schematic circuit diagram showing the configuration of an embodiment of a crystal quadrature output oscillator according to the present invention, in which a crystal resonator Y, a transistor Tr, capacitors C1, C2, C3, and resistors R _E , R1, R2
, R3 including Colpitts type oscillation circuit OSC, dual mode type monolithic crystal filter MCF (hereinafter referred to as MCF), reactance element Z, and amplifier AMP. One terminal of the crystal resonator Y is connected to the base of the transistor Tr of the Colpitts oscillation circuit OSC, and the other terminal is grounded. Then, the collector of the transistor Tr of the Colpitts type oscillation circuit OSC is connected to the first output terminal OUT1. Further, the collector of the transistor Tr and one electrode 1 of the MCF are connected, and the reactance element Z is connected in parallel between the other electrode 2 and the ground. Then, the MCF electrode 2 and the input of the amplifier AMP are connected, and the output of the amplifier AMP is connected to the second output terminal OUT2 to constitute a crystal orthogonal output type oscillator. Here, since the MCF is operated as a bandpass filter, the electrode 3 (
The common electrode is grounded.

The MCF element used here has two electrodes 1 and 2 arranged close to one surface of an AT-cut quartz substrate, and an electrode 3 (common electrode) facing the two electrodes 1 and 2 on the other surface. Is provided.

As is well known, the Colpitts type oscillation circuit is an oscillation circuit configured by connecting an inductive element between a collector and a base of a transistor, and a capacitive element between the base and emitter and between the collector and emitter. As an inductive element between the collector and the base of the transistor Tr, a crystal resonator Y is used between the base and the ground. In addition, a series connection circuit of capacitors C1 and C2 is connected between the base and ground, a resistor _RE is connected between the emitter and ground, and the midpoint of the series connection circuit of capacitors C1 and C2 is connected to the emitter. To do.

In the Colpitts type oscillation circuit, the power source Vcc and the ground are short-circuited at high frequency by the bypass capacitor C3, so that an inductive element of the crystal unit Y is inserted between the collector and the base in an equivalent circuit. Further, since the midpoint between the capacitors C1 and C2 is connected to the emitter, the capacitor C1 is inserted between the base and emitter of the transistor Tr, and the capacitor C2 is inserted between the collector and emitter. It will act as capacitive. Resistor R1,
R2 is a bleeder resistor and sets a base bias voltage.
Further, in the Colpitts type crystal oscillation circuit, the amplification degree seen from the both ends of the crystal unit Y, that is, the so-called negative resistance R (Ω) is inversely proportional to the capacitances C1 and C2 and the square of the frequency ω ² . It is known to be proportional to the collector current. That is, as the frequency increases, the negative resistance R (Ω
) Increases, reaches a peak value at a predetermined frequency, and then decreases as the frequency increases. In a normal Colpitts type crystal oscillator, the negative resistance R at the oscillation frequency is generally set to about 5 times the equivalent resistance of the crystal unit Y.

As is well known, in order to configure a bandpass filter using an MCF element, an equivalent circuit of the MCF element is considered as a lattice circuit, and when considered in terms of image parameters, as shown by a broken line in FIG. By making the antiresonance frequency (×) of the reactance curve P1 exhibited by the lattice arm and the resonance frequency (◯) of the reactance curve P2 exhibited by the series arm of the lattice circuit close to each other and terminating with an appropriate impedance, the bandpass filter is Can be configured.
From the resonance frequency (◯) of the reactance curve P1 to the anti-resonance frequency (×
) Is a pass band as shown by the alternate long and short dash line in FIG. 2, and the others are attenuation bands. And
As shown by the thick line in FIG. 2, the phase characteristic has a center frequency of 0 from π / 2 to −π / 2 in the passband.
Changes almost linearly.

Based on the above knowledge, the operation principle of the present invention will be described. First, the MCF is set so that the output frequency F1 obtained from the output terminal of the Colpitts-type crystal oscillator is located in the vicinity of the high-frequency cutoff of the passband. As described above, the relationship between the phase characteristics and the frequency of the MCF is as shown in FIG. 3 (extracted from the thick line in FIG. 2), and the frequency f in the vicinity of the high-frequency cutoff of the passband is higher than the center frequency. , Π / 2 (90 degrees) phase change. That is, the phase characteristics of the MCF are set so that the phase transition of the MCF becomes π / 2 (90 degrees) at the frequency F1 of the oscillator output, that is, f = F1. The MCF input electrode 1 is excited at the frequency F1, and the phase of the output signal obtained from the MCF output electrode 2 is 90 depending on the phase characteristics of the MCF.
The degree is changing. Therefore, the output signal of the oscillator and the output signal of the MCF have the same frequency, but the phases are different from each other by 90 degrees. That is, a crystal orthogonal oscillator can be realized by utilizing the phase characteristics of the MCF.
Similarly, even if the frequency f ′ in the vicinity of the low-frequency cutoff of the MCF passband is set to the output frequency F1 of the oscillator, a phase transition of π / 2 (90 degrees) can be provided, and the crystal orthogonal oscillator realizable.
Based on the principle described above, this circuit functions as a direct output type oscillator, but when a quadrature modulator / demodulator is configured using this circuit, the circuit of the next stage connected to each output terminal becomes a load. Works.
At this time, when a load is directly connected to the output terminal of the MCF, the phase characteristics of the MCF may change depending on the value of the load, and the phase difference from the signal supplied to the input electrode may be shifted from 90 degrees.
Therefore, in practice, as shown in FIG. 1, by inserting a predetermined reactance element Z between the output electrode of the MCF and the ground, a signal output from the output electrode is input regardless of the load of the next stage circuit. It operates so as to maintain a 90-degree phase difference from the signal supplied to the electrode side.
Therefore, in the circuit of FIG. 1, the oscillator output is the first output terminal OUT1, and the connection point between the output electrode of the MCF and the reactance element Z is the second output terminal OUT2. Further, if necessary, the reactance element Z may be varied to finely adjust the phase transition to 90 degrees accurately.
Furthermore, since the level of the output signal from the MCF tends to be lower than the oscillator output OUT1, it is desirable to adjust the level balance by inserting an amplifier AMP.

FIG. 4 is a schematic block diagram showing the configuration of the quartz quadrature output oscillator according to the second embodiment. The first and second electrodes 4 and 5 are placed close to each other on one surface of an AT-cut quartz substrate. The third electrode 6 is disposed with an interval that is not coupled to the two electrodes 4 and 5, and the first, second, and third electrodes are disposed.
A composite vibrator Y1 provided with a fourth electrode 7 so as to face the electrodes 4, 5, 6 of the first, a first amplifier AMP1, a second amplifier AMP2, a reactance element Z, and capacitors C1, C2. It has. A circuit in which the fourth electrode 7 of the composite vibrator Y1 is grounded, the third electrode 6 and the input of the first amplifier AMP1 are connected, and two capacitors C1 and C2 are connected in series between the input and the ground is connected in parallel. Connected, connected to the output of the first amplifier AMP1 and the first output terminal OUT1, and connected to the output of the first amplifier AMP1 and the midpoint of the capacitors C4 and C5 connected in series.
The output of MP1 is connected to the second electrode 5 of the composite vibrator Y1. A reactance element Z is connected in parallel between the first electrode 4 and the ground, and the first electrode 4 and the input of the second amplifier AMP2 are connected,
A crystal quadrature output type oscillator is configured by connecting the output of the second amplifier AMP2 to the second output terminal OUT2.

Since the third electrode 6 of the composite vibrator Y1 is spaced apart from the first and second electrodes 4 and 5 so as not to be acoustically coupled, the third electrode 6 and the fourth electrode 7 cause crystal vibration. A child Y2 is formed. on the other hand,
Since the first and second electrodes 4 and 5 are arranged close to each other, the MCF 1 is formed by the fourth electrode 7 provided facing the first and second electrodes 4 and 5. The operation of the quadrature output type oscillator shown in FIG. 4 will be described. A crystal oscillation circuit is constituted by the crystal resonator Y2 formed by the third electrode 6 and the fourth electrode 7, the first AMP1, and the capacitors C1 and C2. Then, the first output frequency F1 is output from the first output terminal OUT1. Since this output is connected to the second electrode 5 of the MCF 1 formed by the first, second and fourth electrodes 4, 5, 7, the MCF 1 is excited at the frequency F 1. The frequency f near the high-frequency cutoff of the passband of the MCF 1 has a phase change of π / 2 (90 degrees) compared to the center frequency. The phase characteristic of MCF1 is set so that the phase transition of MCF1 becomes π / 2 (90 degrees) at the frequency F1 of the oscillator output. The input electrode 5 of the MCF 1 is excited at the frequency F 1, and the phase of the output signal F 2 obtained from the output electrode 4 of the MCF changes by 90 degrees due to the phase characteristics of the MCF 1. Therefore, the output signal F1 of the oscillator and the output signal F2 of the MCF1 have the same frequency, but the phases are different from each other by 90 degrees.
The phase difference between the two frequencies F1 and F2 can be adjusted by the value of the reactance element Z.

FIG. 5 is a schematic block diagram showing the configuration of a crystal quadrature output oscillator according to the third embodiment, which is a Colpitts type comprising a transistor Tr, capacitors C1, C2, C3, C4 and resistors R _E , R4, R5. An oscillation circuit, an MCF, a first amplifier AMP1, a second amplifier AMP2,
A reactance element Z. The common electrode 3 of the MCF is grounded, and the electrode 2 of the MCF is connected to the base of the transistor Tr of the Colpitts type oscillation circuit via the capacitor C4.
A crystal oscillation circuit is formed by the resonance circuit composed of the CF electrode 2 and the common electrode 3 and the Colpitts oscillation circuit OSC. Take the output from across the resistor R _E connected between ground and the emitter of the transistor Tr, which was connected to the input of the first amplifier AMP1, connects the output of the first amplifier AMP1 to the first output terminal OUT1 . Further, the MCF electrode 1 is connected to the second amplifier AM.
The reactance element Z is connected in parallel between the input and the ground, and connected to the input of P2.
The output of the amplifier AMP2 is connected to the second output terminal OUT2 to constitute a crystal orthogonal output type oscillator.

The crystal resonance circuit formed by the electrodes 2 and 3 of the MCF and the Colpitts type oscillation circuit OSC form a Colpitts type crystal oscillation circuit, and the output frequency F1 is output from the first output terminal OUT1. At the same time, the MCF is also driven at this frequency F1. At a frequency f near the high-frequency cutoff of the MCF passband, the phase changes by π / 2 (90 degrees) compared to the center frequency. The phase characteristics of the MCF are set so that the phase transition of the MCF becomes π / 2 (90 degrees) at the frequency F1 of the oscillator output. The MCF input electrode 2 is excited at the frequency F1, and the MC
The phase of the output signal F2 obtained from the F output electrode 1 is shifted by 90 degrees due to the phase characteristics of the MCF. Therefore, the output signal F1 of the oscillator and the output signal F2 of the MCF have the same frequency, but the phases are different from each other by 90 degrees.
The phase difference between the two frequencies F1 and F2 can be adjusted by the value of the reactance element Z.

FIG. 6 is a perspective view of a fourth embodiment in which a crystal resonator element 9, an MCF element 10, and a reactance element 11 are formed on a crystal substrate 8. In FIG. By adding an oscillation circuit, an amplifier, etc. to this, it becomes possible to miniaturize the crystal orthogonal output type oscillator. In addition, the quartz substrate 8
The crystal resonator element 9, the MCF element 10 and the reactance element 11 formed on the I
Further reduction in size is possible by mounting a C oscillation circuit, an amplification circuit, and the like and housing a crystal substrate 8 in a package to form a crystal orthogonal output type oscillator.

In the above description of the crystal orthogonal output type oscillator, the Colpitts type oscillation circuit has been described. However, the present invention is not limited to the Colpitts type oscillation circuit. May be.
In the above description, a crystal quadrature output oscillator using a two-pole dual mode crystal filter and utilizing its phase characteristics has been described. However, a crystal quadrature output oscillator can be configured even if a dual mode crystal filter having two or more poles is used. There is no need to explain what you can do.
Furthermore, it can be easily assumed that a quadrature output type oscillator can be constructed using the phase characteristics of a surface acoustic wave filter instead of a dual mode quartz filter.

It is a circuit diagram which shows the structure of the orthogonal output type | mold oscillator which concerns on this invention. It is a figure which shows the reactance curve, filter characteristic, and phase characteristic of a dual mode quartz filter. It is a figure which shows the relationship between the frequency and phase characteristic of a dual mode quartz filter. It is a block diagram which shows the structure of the orthogonal output type | mold oscillator which is a 2nd Example. It is a circuit diagram which shows the structure of the orthogonal output type | mold oscillator which is a 3rd Example. It is a perspective view of what constituted a crystal oscillation element, a MCF element, and a reactance element on one quartz substrate which is the 4th example. It is a block diagram which shows the structure of the conventional orthogonal modulation apparatus. (A), (b) is a figure explaining the flip-flop circuit used for a quadrature modulation apparatus, and its operation | movement.

Explanation of symbols

1, 2, 3, 4, 5, 6, 7, 9, 10 ... electrodes, 8 ... quartz substrate, Z, 11 ... reactance element, Y, Y2 ... crystal resonator, Y1 ... compound resonator, MCF, MCF1 ... Double mode crystal filter, Tr ... Transistor, AMP, AMP1, AMP2 ... Amplifier, R _E , R1, R
2, R3... Resistance, C1, C2, C3, C4.

Claims (9)

A piezoelectric oscillation circuit having a piezoelectric vibrator as a frequency determining element and a monolithic piezoelectric filter are provided.
The output of the piezoelectric oscillation circuit is connected to the input terminal of the monolithic piezoelectric filter, the output from the piezoelectric oscillation circuit is the first output, and the output from the output terminal of the monolithic piezoelectric filter is the second output. An orthogonal output type oscillator characterized by An orthogonal output type oscillator, wherein a reactance element is inserted and connected between an output terminal of the monolithic piezoelectric filter and a ground. The composite vibration element in which the electrodes constituting the monolithic piezoelectric filter and the electrodes constituting the piezoelectric vibrator are arranged so as not to be acoustically coupled to each other on the same piezoelectric substrate. 2. An orthogonal output oscillator according to 2. The orthogonal output type oscillator according to any one of claims 1 to 3, wherein the reactance element is formed by an electrode pattern on a piezoelectric substrate constituting the monolithic piezoelectric filter or the piezoelectric vibrator. A monolithic piezoelectric filter, and an oscillation circuit configured with a frequency determining element on the output terminal side of the monolithic piezoelectric filter,
An orthogonal output type oscillator characterized in that an output from the oscillation circuit is a first output and an output from an output terminal of the monolithic piezoelectric filter is a second output. 6. The orthogonal output type oscillator according to claim 5, wherein a reactance element is inserted and connected between the output terminal of the monolithic piezoelectric filter and the ground. 7. The quadrature output oscillator according to claim 1, wherein an amplifying unit is provided in at least one of the first and second outputs. 8. The orthogonal output type oscillator according to claim 1, wherein an electronic component for constituting an oscillation circuit is mounted on the piezoelectric substrate constituting the monolithic piezoelectric filter or the piezoelectric vibrator. 9. The quadrature output oscillator according to claim 1, wherein at least one of the piezoelectric vibrator and the monolithic piezoelectric filter uses an AT-cut crystal.

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