JP2005261059A - Current bidirectional converter - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a current bidirectional converter which enables the improvement of its power efficiency, the reduction of its calorific value, and the high frequency action, by reducing the switching loss. <P>SOLUTION: An auxiliary circuit part 1 added to a step-up/down converter is equipped with capacitors C1 and C2 which are connected between the collectors and the emitters of transistors Q1 and Q2, and upper and lower auxiliary current paths are constituted between a junction X and a high-voltage power terminal T2 and a reference voltage terminal TS. Inductors L2 and Lr are common to the auxiliary current path. In the upper auxiliary current path, a path, which leads to the high-voltage power terminal T2 via a diode D3 and a transistor Q3 from the inductor Lr, is made, and in the lower auxiliary current path, a path, which leads to the reference voltage terminal TS via a transistor Q4 and a diode D4 from the inductor Lr, is made. The terminal voltage at switching of the transistors Q1 and Q2 becomes small by the capacitors C1 and C2 and the auxiliary current path, whereby this converter can reduce the switching loss. <P>COPYRIGHT: (C)2005,JPO&NCIPI

Description

  The present invention relates to a current bidirectional converter, and more particularly to loss reduction in a current bidirectional converter.

  As shown in FIG. 13, in the chopper circuit 260 disclosed in Patent Document 1, when the motor 110 is at a high output, the voltage of the battery 130 is boosted to supply power to the motor 110, and when the motor 110 is regenerated, the motor power generation voltage is increased. A circuit configuration for charging the battery 130 by lowering the voltage is disclosed. Here, the motor 110 is an induction motor used for a traveling motor such as an electric vehicle, and shows a case where it is driven by an inverter circuit 150.

  When rotating the motor 110 at a high rotational speed, the control circuit 420 turns on the transistor 280 and causes the current from the battery 130 to flow through the reactor 320 to accumulate electromagnetic energy. Thereafter, the transistor 280 is turned off, and the electromagnetic energy stored in the reactor 320 is stored in the capacitor 220 via the flywheel diode 290. The voltage between the terminals of the capacitor 220 becomes higher than the voltage of the battery 130, and the chopper circuit 260 functions as a step-up chopper.

  Moreover, when the rotation speed of the motor 110 falls, the motor 110 becomes regenerative braking. The control circuit 420 turns on the transistor 270 and charges the battery 130 with the regenerative current from the motor 110 via the reactor 320. Thereafter, the transistor 270 is turned off, and the electromagnetic energy stored in the reactor 320 is stored in the battery 130. The motor power generation voltage is stepped down to charge the battery 130. The chopper circuit 260 functions as a step-down chopper.

  In addition, there exists a DC-DC converter currently disclosed by patent document 2 as another related technique.

JP-A-8-214592 (paragraphs 0048 to 0051, FIG. 1) JP 2003-33013 A

  However, in the chopper circuit 260 of Patent Document 1, the transistors 270 and 280 are switched on / off from an off state in which a voltage is applied between the transistor terminals to an on state, and a current flows through the transistor. When switching from the on-state to the off-state, the transition of the conductive state of the transistors 270 and 280 takes a predetermined transition time. Therefore, in the transient state during the transition period, a voltage is applied between the terminals of the transistor. However, a so-called switching loss occurs in which current flows and power is consumed. This switching loss is dominant as a loss in switching control of transistors such as the chopper circuit 260 or the DC-DC converter.

  Due to the switching loss, the ratio of the driving power of the motor 110 to the input power from the battery 130, that is, the power efficiency is lowered, which is a problem. In addition, the power consumed unnecessarily due to the switching loss is converted into heat energy and causes heat generation of the device. When the amount of heat generated is limited due to the usage environment of the device, the cooling device such as a heat sink is increased in size. This is a problem because it may increase the size and weight of the equipment.

  The present invention has been made to solve at least one of the problems of the prior art, and the switching operation is performed with a small voltage applied between the terminals of the switching element, thereby reducing the switching loss. Thus, an object of the present invention is to provide a current bidirectional converter capable of improving power efficiency and reducing the amount of heat generation and capable of high-frequency operation.

  In order to achieve the above object, a current bidirectional converter according to claim 1 includes a first inductor having a first terminal connected to a low voltage power supply terminal, a second terminal of the first inductor, a reference voltage terminal, or a high voltage power supply terminal. A first switching element connected between one of the first switching element, a first rectifying element connected between a connection point between the first inductor and the first switching element and the other of the reference voltage terminal or the high-voltage power supply terminal. A bidirectional converter including: a capacitor connected in parallel between at least one of the first switching element and the first rectifying element; and an auxiliary current connected in parallel with the first switching element. The auxiliary current path is electromagnetically coupled to the first inductor, and a first terminal that induces an electromotive force having the same polarity as the first terminal of the first inductor is connected to the connection point. A second inductor that is, a second switching element, and a second rectifying element for preventing back flow, characterized in that it is connected in series.

  In the current bidirectional converter according to claim 1, when the first switching element is turned on, an input current flows through the first inductor and the first switching element, and electromagnetic energy is supplied from one of the low-voltage power supply terminal and the high-voltage power supply terminal to the first inductor. Is accumulated. The accumulated electromagnetic energy is released to the other of the low-voltage power supply terminal and the high-voltage power supply terminal by the conduction of the first rectifier element.

  When the first switching element is non-conductive, if the capacitor is connected in parallel between the current path terminals of the first switching element, the capacitor is in a discharged state, and the current is connected between the current path terminals of the first rectifying element. If the capacitors are connected in parallel, the capacitor is in a charged state, so that the voltage change at the connection point is performed gradually. Therefore, the first switching element is made non-conductive with a small voltage difference between the current path terminals. That is, zero volt switching (ZVS) is performed.

  Prior to the subsequent re-conduction of the first switching element, an auxiliary current path is formed by the conduction of the second switching element. Here, the conduction of the second switching element is performed by zero current switching (ZCS). The second inductor as an equivalent circuit due to the electromagnetic coupling of the first and second inductors according to the voltage at the connection point determined by the conducting first rectifier element and the electromotive force induced in the second inductor. Due to the voltage applied to the leakage inductor connected in series, the current flowing in the auxiliary current path increases with time. With time, more of the input current is bypassed, eventually the entire amount of input current is bypassed, no current flows through the first rectifier element, and the capacitor is charged and discharged. The voltage at the connection point is inverted, and the voltage difference between the current path terminals of the first switching element becomes small. In this state, the first switching element is turned on and zero volt switching (ZVS) is performed.

  After the voltage at the connection point is reversed, the voltage applied to the leakage inductor is reversed. Although the time gradient of the bypass current flowing in the auxiliary current path is reversed and the bypass current gradually decreases, no current flows back by the second rectifying element, and no current flows. In this state, the second switching element is turned off and zero current switching (ZCS) is performed.

  Here, the inductance value of the leakage inductor is considered to have a smaller value than the inductance value of the second inductor. A voltage higher than the voltage of the high-voltage power supply terminal applied between the connection point and the reference voltage terminal or the high-voltage power supply terminal is applied to both ends of the leakage inductor having a small inductance value. A current having a steep time gradient compared to the time gradient of the input current is bypassed to the auxiliary current path.

  Thereby, in the current bidirectional converter, the switching operation can be performed while the voltage difference between the current path terminals in the first switching element is reduced. Switching loss consumed by the first switching element during switching can be reduced.

  With the reduction of switching loss, it is possible to improve the power efficiency in the voltage conversion of the current bidirectional converter. Heat generation due to switching loss in the first switching element can be reduced, and a cooling device such as a heat sink can be reduced in size and weight.

  It is possible to increase the frequency of the switching operation in the current bidirectional converter, and to perform the switching operation at a frequency higher than the audible frequency band. The vibration of the first inductor or the like accompanying the electromagnetic energy during operation can be shifted from the audible frequency band, and abnormal noise during operation can be prevented.

  The time slope of the bypass current flowing in the auxiliary current path can be adjusted according to the leakage inductance value and the winding ratio of the first and second inductors. After the auxiliary current path is formed, the entire delay of the input current can be bypassed, the voltage at the connection point can be reversed, and the time delay until the first switching element can be made conductive by zero volt switching (ZVS) can be adjusted. . In addition, the voltage level applied to the second switching element when the second switching element is not conductive can be adjusted.

  In addition, after the voltage at the connection point is reversed and the current in the auxiliary current path decreases, the second rectifying element maintains a state in which no current flows in the auxiliary current path. Sufficient time can be secured.

  The bidirectional current converter according to claim 2 is the bidirectional current converter according to claim 1, wherein the auxiliary current path includes a third inductor connected in series to the second terminal of the second inductor. And

  In the current bidirectional converter according to claim 2, the time gradient of the current flowing through the auxiliary current path is determined by the third inductor instead of or together with the leakage inductor caused by the electromagnetic coupling of the first and second inductors. The Thereby, the time inclination of the current flowing through the auxiliary current path can be adjusted by adjusting the inductance value of the third inductor.

  Further, the current bidirectional converter according to claim 3 is the current bidirectional converter according to claim 1, wherein the direction opposite to the current direction of the input current is defined as the forward direction between the current path terminals of the first switching element. An anti-parallel diode connected is provided. Thereby, it can respond also at the time of the backflow of an electric current.

  The bidirectional current converter according to claim 4 is the bidirectional current converter according to claim 1, wherein the other terminal of the first switching element is connected to a reference voltage terminal, and the high voltage power supply terminal includes a low voltage power supply terminal. A voltage boosted with respect to the voltage is supplied. Thereby, a boost converter with reduced switching loss can be configured.

  The bidirectional current converter according to claim 5 is the bidirectional current converter according to claim 1, wherein the other terminal of the first switching element is connected to the high voltage power supply terminal, and the low voltage power supply terminal includes the high voltage power supply terminal. A voltage stepped down with respect to the voltage is supplied. Thereby, a step-down converter with reduced switching loss can be configured.

  The bidirectional current converter according to claim 6 is connected in series between a first inductor having a first terminal connected to the low voltage power supply terminal, and the high voltage power supply terminal and the reference voltage terminal. A bidirectional current converter including an upper switching element and a lower switching element to which a second terminal of one inductor is connected, a capacitor connected in parallel between at least one of the upper and lower switching elements, and a lower A lower auxiliary current path connected in parallel with the switching element and an upper auxiliary current path connected in parallel with the upper switching element, the lower auxiliary current path being electromagnetically coupled to the first inductor, and the first inductor The first terminal in which an electromotive force having the same polarity as that of the first terminal is induced passes through the second inductor connected to the connection point, and is used to prevent backflow. The rectifier element and the lower auxiliary switching element are connected in series, and the upper auxiliary current path is connected to the upper rectifier element for preventing backflow and the upper auxiliary switching element in series via the second inductor. And

  In the current bidirectional converter according to claim 6, when the input current flows from the low-voltage power supply terminal to the first inductor due to the conduction of the lower switching element, and the electromagnetic energy is accumulated, the accumulated electromagnetic energy is the conduction of the upper switching element. Is discharged to the high-voltage power supply terminal. The upper switching element may be turned on at the timing of performing a rectifying action, and may function as a so-called synchronous rectifying element.

  The non-conducting state of the lower switching element means that when the capacitor is connected in parallel between the current path terminals of the lower switching element, the capacitor is in a discharged state, and the capacitor is connected in parallel between the current path terminals of the upper switching element. If connected, the capacitor is in a charged state, so that the voltage change at the connection point is performed slowly. For this reason, the lower switching element is rendered non-conductive with a small voltage difference between the current path terminals. That is, zero volt switching (ZVS) is performed.

  The conduction of the upper switching element as the synchronous rectifying element is as long as a capacitor is connected in parallel between the current path terminals of the lower switching element by electromagnetic energy accumulated in the first inductor in response to the non-conduction of the lower switching element. The capacitor is charged, and if the capacitor is connected in parallel between the current path terminals of the upper switching element, the capacitor is discharged and then performed. The upper switching element conducts in a state where the voltage difference between the current path terminals is small. That is, zero volt switching (ZVS) is performed.

  After the upper switching element is turned off, the lower auxiliary current path is formed by the conduction of the lower auxiliary switching element prior to the re-conduction of the lower switching element. Here, the conduction of the lower auxiliary switching element is performed by zero current switching (ZCS). Thereafter, the capacitor is charged and discharged. The voltage at the connection point is inverted to a low voltage level, and the voltage difference between the current path terminals of the lower switching element becomes small. In this state, the lower switching element is turned on and zero volt switching (ZVS) is performed.

  After the voltage at the connection point is inverted to the low voltage level, the electromotive force induced in the second inductor is inverted, the first terminal becomes a positive voltage, and the voltage applied to the leakage inductor is inverted. Although the time slope of the bypass current flowing in the lower auxiliary current path is reversed and the bypass current gradually decreases, no reverse current flows due to the lower rectifying element, and no current flows. In this state, the lower auxiliary switching element is turned off and zero current switching (ZCS) is performed.

  In addition, when electromagnetic current is accumulated from the high-voltage power supply terminal to the first inductor due to the conduction of the upper switching element, the accumulated electromagnetic energy is released to the low-voltage power supply terminal according to the conduction of the lower switching element. The The lower switching element may be turned on at the timing of performing a rectifying action, and may function as a so-called synchronous rectifying element.

  When the upper switching element is non-conductive, if the capacitor is connected in parallel between the current path terminals of the upper switching element, the capacitor is in a discharged state, and the capacitor is connected in parallel between the current path terminals of the lower switching element. If connected, the capacitor is in a charged state, so that the voltage change at the connection point is performed slowly. For this reason, the upper switching element is rendered non-conductive with a small voltage difference between the current path terminals. That is, zero volt switching (ZVS) is performed.

  The conduction of the lower switching element as the synchronous rectifying element is as long as a capacitor is connected in parallel between the current path terminals of the upper switching element by electromagnetic energy accumulated in the first inductor in response to the non-conduction of the upper switching element. The capacitor is charged, and if the capacitor is connected in parallel between the current path terminals of the lower switching element, the capacitor is discharged. The lower switching element conducts in a state where the voltage difference between the current path terminals is small. That is, zero volt switching (ZVS) is performed.

  After the lower switching element is turned off, the upper auxiliary current path is formed by the conduction of the upper auxiliary switching element prior to the re-conduction of the upper switching element. Here, the conduction of the upper auxiliary switching element is performed by zero current switching (ZCS). Thereafter, the capacitor is charged and discharged. The voltage at the connection point is inverted to a high voltage level, and the voltage difference between the current path terminals of the upper switching element becomes small. In this state, the upper switching element is turned on and zero volt switching (ZVS) is performed.

  After the voltage at the connection point is inverted to the high voltage level, the electromotive force induced in the second inductor is inverted, the first terminal becomes a negative voltage, and the voltage applied to the leakage inductor is inverted. Although the time gradient of the bypass current flowing through the upper auxiliary current path is reversed and the bypass current gradually decreases, no reverse current flows due to the upper rectifying element, and no current flows. In this state, the upper auxiliary switching element is turned off and zero current switching (ZCS) is performed.

  Thereby, the buck-boost converter with reduced switching loss can be configured. Reduced switching loss in upper and lower switching elements, improved power efficiency in voltage conversion of current bidirectional converter, reduced heat generation in upper and lower switching elements, reduced equipment size and weight, higher switching operation frequency, transformer The adjustment of the conduction timing of the upper or lower switching element by means of the above has the same operation and effect as the first aspect.

  The time gradient of the bypass current flowing through the upper and lower auxiliary current paths can be adjusted according to the leakage inductance value and the winding ratio of the first and second inductors. After the formation of the upper and lower auxiliary current paths, the entire amount of input current can be bypassed to invert the voltage at the connection point, and the time delay until the upper and lower switching elements can be made conductive by ZVS can be adjusted. . Further, when the lower and upper switching elements operate as synchronous rectification elements, the non-conduction timing of the switching elements can be adjusted. Furthermore, the voltage level applied to the upper and lower auxiliary switching elements when the upper and lower auxiliary switching elements are non-conductive can be adjusted.

  A current bidirectional converter according to claim 7 is the current bidirectional converter according to claim 6, further comprising a third inductor connected in series to the second terminal of the second inductor and constituting upper and lower auxiliary current paths. It is characterized by providing.

  8. The bidirectional current converter according to claim 7, wherein the time gradient of the current flowing through the upper and lower auxiliary current paths is the third inductor instead of or together with the leakage inductor caused by electromagnetic coupling of the first and second inductors. Determined by. Thereby, the time inclination of the current flowing through the auxiliary current path can be adjusted by adjusting the inductance value of the third inductor.

  The current bidirectional converter according to claim 8 is the current bidirectional converter according to claim 6, wherein at least one of the current path terminals of the upper and lower switching elements includes a current direction of the input current. Is provided with an antiparallel diode connected with the reverse direction as the forward direction. Thereby, it can respond at the time of the backflow of an electric current. When the upper and lower switching elements are used as synchronous rectification elements, there is no need to adjust the conduction / non-conduction timing. Prior to the timing when the entire amount of input current is bypassed to the upper or lower auxiliary current path, the lower or upper switching element that is conducting as the synchronous rectifying element can be made non-conductive.

  According to the present invention, it is possible to provide a bidirectional current converter in which a switching operation can be performed with a small voltage applied between terminals of a switching element, and switching loss is reduced.

  DESCRIPTION OF THE PREFERRED EMBODIMENTS Embodiments of a current bidirectional converter according to the present invention will be described below in detail with reference to FIGS. In FIGS. 3 to 6 and FIGS. 8 to 11, the circuit elements in the non-conduction state are omitted for easy understanding of the conduction state.

  FIG. 1 is a circuit diagram of the current bidirectional converter of the first embodiment. It has a circuit configuration in which the auxiliary circuit unit 1 as an invention part is added to the buck-boost converter.

  The buck-boost converter has a voltage source V1 connected to the low voltage power supply terminal T1, boosts the voltage V1 and supplies it to the voltage source V2 connected to the high voltage power supply terminal T2, and the voltage source V2 is connected to the high voltage power supply terminal T2. Then, the voltage V2 is stepped down and supplied to the voltage source V1 connected to the low voltage power supply terminal T1. When a motor is connected as a load to the high-voltage power supply terminal T2, the voltage V2 that is the drive voltage of the motor is supplied by boosting the voltage V1, and the regenerative energy from the motor is recharged to the voltage source V1. Can be used. The buck-boost converter shown in FIG. 1 is a so-called non-insulated current bidirectional converter in which the reference voltage terminals TS of the voltage sources V1 and V2 are connected in common. In the transistors Q1 and Q2, the emitter terminal of the transistor Q1 and the collector terminal of the transistor Q2 are connected at the connection point X, the collector terminal of the transistor Q1 is connected to the high-voltage power supply terminal T2, and the emitter terminal of the transistor Q2 is connected to the reference voltage terminal TS. And is connected in series between the high voltage power supply terminal T2 and the reference voltage terminal TS. The base terminals of the transistors Q1 and Q2 are exclusively controlled by a controller (not shown). Antiparallel diodes D1 and D2 are connected to transistors Q1 and Q2 in the forward direction from the emitter terminal to the collector terminal. An inductor L1 is connected between the connection point X and the low-voltage power supply terminal T1. Capacitors C11 and C12 are connected in parallel with the voltage sources V1 and V2 between the low-voltage power supply terminal T1 and the high-voltage power supply terminal T2 and the reference voltage terminal TS. Here, the transistor Q1 is an upper switching element, and the transistor Q2 is a lower switching element.

  Here, the load connected to the high voltage power supply terminal T2 may be, for example, an induction motor driven via an inverter circuit or the like. An example is a case where the present invention is applied to a hybrid vehicle that travels by switching between a gasoline engine and motor drive, an electric vehicle that travels only by motor drive, and the like. For example, 200 V is supplied to the voltage V1, and 500 V is supplied to the voltage V2 to be supplied to the load.

  When operating as a boost converter that boosts the voltage V1 to the voltage V2, the electromagnetic energy accumulated in the inductor L1 due to the conduction of the transistor Q2 is supplied to the high-voltage power supply terminal T2 via the transistor Q1 and the antiparallel diode D1. Done. When operating as a step-down converter that steps down the voltage V2 to the voltage V1, the electromagnetic energy accumulated in the inductor L1 due to the conduction of the transistor Q1 is supplied to the low voltage power supply terminal T1 through the transistor Q2 and the antiparallel diode D2. Is done.

  Here, the capacitors C11 and C12 are smoothing capacitors. Transistors such as IGBT, MOS, and bipolar can be used as the transistors Q1 and Q2. In this case, the anti-parallel diodes D1 and D2 can be separately connected to a diode element in addition to the case where they are built in the transistors Q1 and Q2.

  The auxiliary circuit unit 1 includes capacitors C1 and C2 connected between the collectors and emitters of the transistors Q1 and Q2. Further, upper and lower auxiliary current paths are formed between the connection point X of the transistors Q1, Q2 and the high voltage power supply terminal T2 and the reference voltage terminal TS.

  The connection point X to the inductor L2 and the inductor Lr are common to the upper and lower auxiliary current paths. In the upper auxiliary current path, a path is formed from the inductor Lr to the high voltage power supply terminal T2 via the backflow preventing diode D3 and the transistor Q3. In the lower auxiliary current path, a path is formed from the inductor Lr to the reference voltage terminal TS through the transistor Q4 and the backflow prevention diode D4. The backflow prevention diode D3 is provided for the purpose of preventing backflow of the current in the upper auxiliary current path, and the input current that flows to the inductor L1 by flowing the current in the direction from the high voltage power supply terminal T2 toward the connection point X. Bypass. The backflow prevention diode D4 is provided for the purpose of preventing backflow of current in the lower auxiliary current path, and the input current that flows to the inductor L1 by flowing current in the direction from the connection point X to the reference voltage terminal TS. Bypass. The transistors Q3 and Q4 are upper and lower auxiliary transistors, and the backflow prevention diodes D3 and D4 are upper and lower rectifier elements.

  Here, the inductor L2 is electromagnetically coupled to the inductor L1. In addition, an electromotive force is induced between the first terminal of the inductor L1 connected to the low-voltage power supply terminal T1 and the first terminal of the inductor L2 connected to the connection point X with the same polarity. As the electromagnetic coupling, a case where a transformer is constituted by the inductor L1 and the inductor L2 can be considered.

  First, with reference to FIG. 2 and FIGS. 3 to 6, the boosting operation in the buck-boost converter of the first embodiment will be described. FIG. 2 shows a timing chart, and FIGS. 3 to 6 show circuit operation states in each operation. In the following description, the timing chart (FIG. 2) of the boosting operation will be described with reference to the operation state on the circuit (FIGS. 3 to 6) as appropriate. In FIG. 2, VGQ1, VGQ2, and VGQ4 are voltages applied to the base terminals GQ1, GQ2, and GQ4 of the transistors Q1, Q2, and Q4. IL1 and ILr indicate currents flowing through the inductors L1 and Lr with the current flowing from the voltage V1 to the connection point X and from the connection point X to the inductor Lr as positive directions. Among these, the inductor current IL1 is the input current. VQ2 indicates a voltage at the connection point X. VL1 represents a voltage between terminals when the terminal on the connection point X side of the inductor L1 is used as a reference, and VLr represents a voltage between terminals when the terminal on the transistors Q3 and Q4 side of the inductor Lr is used as a reference.

  In FIG. 2, (1), (2), and FIG. 3 are accumulation periods of electromagnetic energy in the inductor L1. In the period (1) in FIG. 2, electromagnetic energy is accumulated in the inductor L1. FIG. 3 (1) shows the operating state during this period. The gate voltage VGQ2 applied to the gate terminal GQ2 of the transistor Q2 is at a high level, and the transistor Q2 is conductive. A current path is established from the voltage source V1 to the reference voltage terminal TS via the inductor L1 and the transistor Q2. A voltage V1 is applied between the terminals of the inductor L1, and an inductor current IL1 having a predetermined positive time gradient flows in a direction from the voltage source V1 toward the connection point X (this direction is a positive direction). Inductor L1 stores electromagnetic energy corresponding to inductor current IL1.

  After a predetermined time elapses, the process proceeds to (2) in FIG. When the gate voltage VGQ2 applied to the gate terminal GQ2 transitions to a low level, the transistor Q2 becomes non-conductive. The voltage VQ2 at the connection point X at this time has a voltage value substantially equal to the reference voltage, which is the voltage at the reference voltage terminal TS, since the transistor Q2 is conductive until just before. For this reason, the capacitor C1 is in a charged state and the capacitor C2 is in a discharged state. Since the inductor current IL1 flowing through the inductor L1 after the transistor Q2 is turned off is consumed for discharging the capacitor C1 and charging C2 ((2) in FIG. 3), the voltage value of the voltage VQ2 at the connection point X is The increase is delayed after the non-conduction of the transistor Q2. Therefore, switching of the transistor Q2 to the non-conducting state is performed in a state where a slight voltage is applied between the collector and the emitter. Zero volt switching (ZVS) is performed, and the switching loss to the non-conducting state of the transistor Q2 can be reduced.

  In FIG. 2, (3), (4), and FIG. 4 are the discharge periods of electromagnetic energy from the inductor L1. In the period (3) in FIG. 2, the gate voltage VGQ1 applied to the gate terminal GQ1 of the transistor Q1 becomes high level, and the transistor Q1 becomes conductive. The transistor Q1 that has been turned on, together with the antiparallel diode D1, flows an inductor current IL1 from the inductor L1 toward the high-voltage power supply terminal T2. As a result, electromagnetic energy is discharged to the high-voltage power supply terminal T2, and the boosted voltage V2 is supplied to the voltage source V2 ((3) in FIG. 4). The connection point X is approximately equal to the voltage V2, and the voltage difference between the voltage V2 and the voltage V1 is between the terminals of the inductor L1 from the voltage source V1 toward the connection point X (this direction is a negative direction). And an inductor current IL1 having a predetermined negative time slope flows through the inductor L1. Note that when the gate voltage VGQ1 changes to a high level and the transistor Q1 changes to a conductive state, the capacitor C1 is in a discharged state and C2 is in a charged state. This is performed with a slight voltage applied between the emitters. Zero volt switching (ZVS) is performed, and the switching loss to the conduction state of the transistor Q1 can be reduced. Here, the transistor Q1 operates as a synchronous rectifier. Since the antiparallel diode D1 is connected in parallel, the diode D1 can also provide a rectifying action, and the transistor Q1 can be kept non-conductive in the boosting operation.

  From this state, the gate voltage VGQ1 is inverted to a low level to make the transistor Q1 nonconductive. Since the antiparallel diode D1 is connected in parallel, no voltage is applied to both ends of the transistor Q1, and zero volt switching (ZVS) can be performed. Thereafter, a high level gate voltage VGQ4 is applied to the gate terminal GQ4 of the transistor Q4 ((4) in FIG. 2). A lower auxiliary current path is formed from the connection point X to the reference voltage terminal TS through the inductors L2, Lr, the transistor Q4, and the backflow prevention diode D4, and the inductor current IL1 begins to be bypassed ((4) in FIG. 4). ). In addition, since the current due to the conduction is limited by the inductor Lr in the transition to the conduction state of the transistor Q4, the current rises after the conduction transition. Therefore, zero current switching (ZCS) is performed for switching the transistor Q4 to the conducting state, and switching loss to the conducting state of the transistor Q4 can be reduced.

  In the initial stage when the lower auxiliary current path is formed, the negative inter-terminal voltage VL2 is also applied to the inductor L2 in accordance with the negative inter-terminal voltage VL1 applied to the inductor L1. Thereby, a voltage obtained by adding the voltage VL2 between the terminals of the inductor L2 is applied to the terminal on the inductor L2 side in the inductor Lr when the voltage VQ2 at the connection point X is a high voltage. Since a substantially reference voltage is applied to the other terminal of the inductor Lr via the transistor Q4 and the diode D4, a high voltage level VH is applied to the inter-terminal voltage VLr of the inductor Lr.

  Here, if the voltage V1 = 200V, the voltage V2 = 500V, and the winding ratio of the inductors L1, L2 is 2: 1, the voltage VQ2 at the connection point X is approximately V2 (500V), and the terminal of the inductor L2 Since the inter-voltage VL2 is (V1−V2) / 2 ((200−500) / 2 = −150V), VQ2−VL2 is applied to the terminal on the inductor L2 side in the inductor Lr. That is, 500 − (− 150) = 650V is applied, and a voltage of about 650V is applied as the inter-terminal voltage VLr of the inductor Lr.

  2 and 4 (3) and (4), the voltage difference between the voltage V2 and the voltage V1 is applied in the negative direction of the inductor L1, and the inductor current IL1 is a current value corresponding to the accumulated electromagnetic energy. Decreases with a predetermined negative time slope. From this state, the gate voltage VGQ1 is inverted to a low level to make the transistor Q1 nonconductive. Since the antiparallel diode D1 is connected in parallel, no voltage is applied to both ends of the transistor Q1, and zero volt switching (ZVS) can be performed. Thereafter, the transistor Q4 is turned on to form a lower auxiliary current path. As a result, the inductor Lr having an inductance value smaller than the inductance values of the inductors L1 and L2 has a steep positive time gradient in the direction (positive direction) in which the inductor current IL1 is bypassed to the lower auxiliary current path. Thus, the terminal voltage VLr is applied in the direction of increasing the inductor current ILr. As time elapses, the inductor current ILr bypasses more of the inductor current IL1, and eventually the entire amount of the inductor current IL1 is bypassed.

  Since the diode D1 is connected in parallel with the transistor Q1, the transistor Q1 can be made non-conductive prior to the formation of the lower auxiliary current path. It is also possible to turn off transistor Q1 even after the formation of the lower auxiliary current path. Further, if the transistor Q1 is conductive even after the entire amount is bypassed, the voltage VQ2 at the connection point X is maintained substantially equal to the voltage V2. Therefore, if the lower auxiliary current path is formed, after the entire amount is bypassed. The transistor Q1 can be turned off at an appropriate timing. In either case, transistor Q1 can be zero volt switched (ZVS). In the first embodiment, zero volt switching (ZVS) of the transistor Q1 is performed in a state where the voltage VQ2 at the connection point X is substantially equal to the voltage V2, regardless of whether the inductor current IL1 is completely bypassed by the lower auxiliary current path. Switching loss can be reduced.

  Here, the inter-terminal voltage VLr applied between the terminals of the inductor Lr when the lower auxiliary current path is formed can be adjusted by adjusting the winding ratio of the inductors L1 and L2. The time gradient of increase in current flowing through the auxiliary current path can be adjusted, and the time delay until the transistor Q2 can be turned on by ZVS can be adjusted. Furthermore, the voltage level applied to the transistor Q4 in the non-conductive state can be adjusted.

  In FIG. 2, (5), (6), and FIG. 5 are periods in which the electromagnetic energy from the inductor L1 is shifted to the re-accumulation. As the bypass operation of the inductor current IL1 proceeds by the lower auxiliary current path ((4) in FIGS. 2 and 4), the current that has increased beyond the inductor current IL1 in the inductor current ILr is caused by the non-conduction of the transistor Q1. It is no longer supplied from the voltage source V2, and is covered by the capacitor C1 in the discharged state and the capacitor C2 in the charged state before that. The capacitor C1 is charged, the capacitor C2 is discharged, and the voltage value of the voltage VQ2 at the connection point X falls. In response to this, the terminal voltage VL1 of the inductor L1 is also inverted ((5) in FIGS. 2 and 5).

  After the voltage VQ2 at the node X drops to a voltage level near the reference voltage, the supply of the inductor current ILr to the lower auxiliary current path is continued through the antiparallel diode D2. Therefore, the voltage VQ2 at the connection point X is maintained at the reference voltage or a voltage that is dropped from the reference voltage by the forward voltage of the diode. Coupled with the induction of the positive terminal voltage VL2 in the inductor L2, the terminal voltage VLr of the inductor Lr is inverted and the inductor current ILr decreases with a negative time slope (FIG. 2). And (6) in FIG. 5).

  If the voltage V1 = 200V, the voltage V2 = 500V, and the winding ratio of the inductors L1 and L2 is 2: 1, the voltage VQ2 at the connection point X is substantially equal to the reference voltage (0V). As for VL2, from VL (200V), VL2 / 2 is applied to the terminal on the inductor L2 side in the inductor Lr. That is, 200/2 = 100V is applied, and the voltage VLr between the terminals of the inductor Lr is approximately 0-100 = -100V.

  The timing at which the transistor Q2 is turned on again after the transistor Q1 is turned off is shown in (7) of FIGS. As described above, when the transistor Q1 is turned off and the capacitors C1 and C2 are completely charged and discharged, the voltage VQ2 at the connection point X becomes substantially equal to the reference voltage. As a result, the voltage VL1 between the terminals of the inductor L1 rotates forward, and accumulation of electromagnetic energy in the inductor L1 is started. The inductor current IL1 flowing at this time is bypassed by the inductor current ILr flowing in the lower auxiliary current path in the initial stage, and the inductor current ILr at this time is a current that decreases with a negative time gradient. Therefore, if the transistor Q2 is turned on before the inductor current ILr falls below the inductor current IL1, the inductor current IL1 sequentially shifts from the bypass path to the transistor Q2, and returns to (1) in FIG. 2 and FIG. The step-up operation is performed by repeating the operation.

  Since the antiparallel diode D2 is conducting, the conduction transition of the transistor Q2 is performed in a state where a slight voltage is applied between the collector and emitter terminals. Zero volt switching (ZVS) is performed, and the switching loss to the conduction state of the transistor Q2 can be reduced.

  Since the non-conducting transition of the transistor Q4 is performed after the current does not flow in the lower auxiliary current path by the backflow preventing diode D4, zero current switching (ZCS) is performed and the switching loss can be reduced.

  Here, in the periods (4) to (7) (FIG. 2, FIG. 4 (4) to FIG. 6), the inductor current ILr having a steep time slope and decreasing after the increase also flows to the inductor L2. Thus, a back electromotive force acts on the inductor L1 based on the electromagnetic coupling between the inductors L1 and L2, and the inductor current IL1 increases after being decreased according to the inductor current ILr. The inductor current IL1 rapidly decreases due to the influence of current flowing through the inductor L2 that is electromagnetically coupled to the inductor L1. Here, in order for the lower auxiliary current path to completely bypass the inductor current IL1, the inductor current ILr needs to flow more than the inductor current IL1. However, the inductor current IL1 decreases rapidly, thereby reducing the peak of the inductor current ILr. be able to.

  Next, with reference to FIG. 7 and FIGS. 8 to 11, the step-down operation in the buck-boost converter of the first embodiment will be described. FIG. 7 shows a timing chart, and FIGS. 8 to 11 show circuit operation states in each operation. In the following description, the timing chart (FIG. 7) of the boosting operation will be described with reference to the operation state on the circuit (FIGS. 8 to 11) as appropriate. In FIG. 7, VGQ1, VGQ2, and VGQ3 are voltages applied to the base terminals GQ1, GQ2, and GQ3 of the transistors Q1, Q2, and Q3. IL1 and ILr indicate currents flowing through the inductors L1 and Lr with the current flowing from the voltage V1 to the connection point X and from the connection point X to the inductor Lr as positive directions. Among these, the inductor current IL1 is the input current. VQ2 indicates a voltage at the connection point X. VL1 represents a voltage between terminals when the terminal on the connection point X side of the inductor L1 is used as a reference, and VLr represents a voltage between terminals when the terminal on the transistors Q3 and Q4 side of the inductor Lr is used as a reference.

  In FIG. 7, (1), (2), and FIG. 8 are accumulation periods of electromagnetic energy in the inductor L1. In the period (1) in FIG. 7, electromagnetic energy is accumulated in the inductor L1. FIG. 8 (1) shows the operating state during this period. The gate voltage VGQ1 applied to the gate terminal GQ1 of the transistor Q1 is at a high level, and the transistor Q1 is conductive. A current path is established from the voltage source V2 to the voltage source V1 via the transistor Q1 and the inductor L1. A voltage V1-V2 is applied between the terminals of the inductor L1, and an inductor current IL1 having a predetermined negative time gradient flows in a direction from the connection point X toward the voltage source V1 (this direction is a negative direction). . Inductor L1 stores electromagnetic energy corresponding to inductor current IL1.

  After a lapse of a predetermined time, the process proceeds to (2) in FIG. When the gate voltage VGQ1 applied to the gate terminal GQ1 transitions to a low level, the transistor Q1 becomes non-conductive. At this time, the voltage VQ2 at the connection point X has a voltage value substantially equal to the voltage V2 that is the voltage of the high-voltage power supply terminal T2, since the transistor Q1 is conductive until just before. Therefore, the capacitor C1 is in a discharged state and the capacitor C2 is in a charged state. Since the inductor current IL1 flowing through the inductor L1 after the transistor Q1 is turned off is consumed for charging the capacitor C1 and discharging C2 ((2) in FIG. 8), the voltage value of the voltage VQ2 at the connection point X is The drop is performed after the non-conduction of the transistor Q1. Therefore, switching of the transistor Q1 to the non-conducting state is performed in a state where a slight voltage is applied between the collector and the emitter. Zero volt switching (ZVS) is performed, and the switching loss to the non-conducting state of the transistor Q1 can be reduced.

  In FIG. 7, (3), (4), and FIG. 9 are electromagnetic energy emission periods from the inductor L1. In the period (3) in FIG. 7, the gate voltage VGQ2 applied to the gate terminal GQ2 of the transistor Q2 becomes high level, and the transistor Q2 becomes conductive. The transistor Q2 that has become conductive flows the inductor current IL1 from the inductor L1 toward the low-voltage power supply terminal T1 together with the antiparallel diode D2. As a result, electromagnetic energy is discharged to the low-voltage power supply terminal T1, and the voltage V1 that is stepped down is supplied to the voltage source V1 ((3) in FIG. 9). The connection point X has a voltage substantially equal to the reference voltage, and a voltage difference from the voltage V1 is applied between the terminals of the inductor L1 in a direction from the connection point X to the voltage source V1 (this direction is a positive direction). The inductor current IL1 having a predetermined positive time gradient flows through the inductor L1. Note that when the gate voltage VGQ2 transitions to a high level and the transistor Q2 transitions to a conductive state, the capacitor C1 is in a charged state and C2 is in a discharged state. This is performed with a slight voltage applied between the emitters. Zero volt switching (ZVS) is performed, and the switching loss to the conduction state of the transistor Q2 can be reduced. Here, the transistor Q2 operates as a synchronous rectifier. Since the antiparallel diode D2 is connected in parallel, the diode D2 can also provide a rectifying action, and the transistor Q2 can be kept non-conductive in the step-down operation.

  From this state, the gate voltage VGQ2 is inverted to a low level to make the transistor Q2 non-conductive. Since the antiparallel diode D2 is connected in parallel, no voltage is applied to both ends of the transistor Q2, and zero volt switching (ZVS) can be performed. Thereafter, a high level gate voltage VGQ3 is applied to the gate terminal GQ3 of the transistor Q3 ((4) in FIG. 7). An upper auxiliary current path is formed from the high-voltage power supply terminal T2 to the connection point X through the transistor Q3, the backflow prevention diode D3, and the inductors Lr and L2, and the inductor current IL1 begins to be bypassed ((4) in FIG. 9). . Note that the transistor Q3 transitions to the conducting state, because the current due to the conduction is limited by the inductor Lr, the current rises after the conduction transition. Therefore, zero current switching (ZCS) is performed for switching the transistor Q3 to the conductive state, and switching loss to the conductive state of the transistor Q3 can be reduced.

  In the initial stage when the upper auxiliary current path is formed, the positive inter-terminal voltage VL2 is also applied to the inductor L2 in accordance with the positive inter-terminal voltage VL1 applied to the inductor L1. Thereby, a voltage obtained by adding the voltage VL2 between the terminals of the inductor L2 to the negative side is applied to the terminal on the inductor L2 side in the inductor Lr when the voltage VQ2 at the connection point X is low. Since substantially the voltage V2 is applied to the other terminal of the inductor Lr via the transistor Q3 and the diode D3, a large negative voltage VL is applied to the inter-terminal voltage VLr of the inductor Lr.

  Here, if the voltage V1 = 200V, the voltage V2 = 500V, and the winding ratio of the inductors L1 and L2 is 2: 1, the voltage VQ2 at the connection point X is approximately the reference voltage (0V). Since the inter-terminal voltage VL2 is V1 / 2 (200/2 = 100V), VQ2-VL2 is applied to the terminal on the inductor L2 side in the inductor Lr. That is, 0−100 = −100V is applied, and the voltage VLr between the terminals of the inductor Lr is approximately −600V.

  7 and 9 (3) and (4), the voltage V1 is applied in the positive direction of the inductor L1, and the inductor current IL1 has a predetermined positive time slope from the current value corresponding to the accumulated electromagnetic energy. To increase. From this state, the gate voltage VGQ2 is inverted to a low level to make the transistor Q2 non-conductive. Since the antiparallel diode D2 is connected in parallel, no voltage is applied to both ends of the transistor Q2, and zero volt switching (ZVS) can be performed. Thereafter, the transistor Q3 is turned on to form an upper auxiliary current path. Thus, the inductor Lr having an inductance value smaller than the inductance values of the inductors L1 and L2 has a steep positive time gradient in a direction (negative direction) in which the inductor current IL1 is bypassed to the upper auxiliary current path. Thus, the terminal voltage VLr is applied in the direction of increasing the inductor current ILr. As time elapses, the inductor current ILr bypasses more of the inductor current IL1, and eventually the entire amount of the inductor current IL1 is bypassed.

  Since the diode D2 is connected in parallel with the transistor Q2, it is possible to make the transistor Q2 non-conductive prior to the formation of the upper auxiliary current path. It is also possible to turn off the transistor Q2 even after the formation of the upper auxiliary current path. Further, if the transistor Q2 is conductive even after the full amount is bypassed, the voltage VQ2 at the connection point X is maintained substantially equal to the reference voltage. Therefore, if the upper auxiliary current path is formed, after the full amount is bypassed. The transistor Q2 can be turned off at an appropriate timing. In either case, transistor Q2 can be zero volt switched (ZVS). In the first embodiment, zero volt switching (ZVS) of the transistor Q2 is performed in a state where the voltage VQ2 at the connection point X is substantially equal to the reference voltage before and after bypassing the full amount of the inductor current IL1 by the upper auxiliary current path. Switching loss can be reduced.

  Here, the inter-terminal voltage VLr applied between the terminals of the inductor Lr when the upper auxiliary current path is formed can be adjusted by adjusting the winding ratio of the inductors L1 and L2. The time gradient of increase in current flowing through the auxiliary current path can be adjusted, and the time delay until the transistor Q1 can be turned on by ZVS can be adjusted. Furthermore, the voltage level applied to the transistor Q3 in the non-conductive state can be adjusted.

  In FIG. 7, (5), (6), and FIG. 10 are periods in which the transition from the release of electromagnetic energy from the inductor L1 to the re-accumulation is performed. As the bypass operation of the inductor current IL1 proceeds by the upper auxiliary current path ((4) in FIGS. 7 and 9), the current that has increased beyond the inductor current IL1 among the inductor current ILr is caused by the non-conduction of the transistor Q2. The current does not flow to the reference voltage terminal TS, and is covered by the charged capacitor C1 and the discharged capacitor C2 before that time. The capacitor C1 is discharged, the capacitor C2 is charged, and the voltage value of the voltage VQ2 at the connection point X rises. In response to this, the terminal voltage VL1 of the inductor L1 is also inverted ((5) in FIGS. 7 and 10).

  After the voltage VQ2 at the node X rises to a voltage level near the voltage V2, the inductor current ILr in the upper auxiliary current path is supplied via the anti-parallel diode D1. Therefore, the voltage VQ2 at the connection point X is maintained at the voltage V2 or a voltage that is increased from the voltage V2 by the forward voltage of the diode. Coupled with the induction of the negative terminal voltage VL2 in the inductor L2, the terminal voltage VLr of the inductor Lr is inverted, and the inductor current ILr decreases with a positive time slope (FIG. 2). And (6) in FIG. 5).

  If the voltage V1 = 200V and the voltage V2 = 500V and the winding ratio of the inductors L1 and L2 is 2: 1, the voltage VQ2 at the connection point X is approximately equal to the voltage V2 (500V). As for VL2, (V1-V2) / 2 is applied to the terminal on the inductor L2 side in the inductor Lr from V1-V2 (= 200-500 = -300V). That is, −300 / 2 = −150V is applied, and a voltage of approximately (500 − (− 150) −500 = 150V) is applied as the inter-terminal voltage VLr of the inductor Lr.

  The timing at which the transistor Q1 is turned on again after the transistor Q2 is turned off is shown in (7) of FIGS. As described above, when the transistor Q2 is turned off and charging and discharging of the capacitors C1 and C2 is completed, the voltage VQ2 at the connection point X becomes substantially equal to the voltage V2. As a result, the voltage VL1 between the terminals of the inductor L1 is inverted, and accumulation of electromagnetic energy in the inductor L1 is started. The inductor current IL1 flowing at this time is bypassed by the inductor current ILr flowing in the upper auxiliary current path in the initial stage, and the inductor current ILr at this time is a current that decreases with a positive time gradient. Therefore, if the transistor Q1 is turned on before the inductor current ILr falls below the inductor current IL1, the inductor current IL1 sequentially shifts from the bypass path to the transistor Q1, and returns to (1) in FIG. 7 and FIG. The step-down operation is performed by repeating the operation.

  Since the antiparallel diode D1 is conducting, the conducting transition of the transistor Q1 is performed in a state where a slight voltage is applied between the collector and emitter terminals. Zero volt switching (ZVS) is performed, and the switching loss to the conduction state of the transistor Q1 can be reduced.

  Since the non-conducting transition of the transistor Q3 is performed after the current does not flow in the upper auxiliary current path by the backflow preventing diode D3, zero current switching (ZCS) is performed and switching loss can be reduced.

Here, in the periods (4) to (7) (FIG. 7, FIG. 9 (4) to FIG. 11), the inductor current ILr having a steep time slope and increasing after the decrease also flows to the inductor L2. Thus, the back electromotive force acts on the inductor L1 based on the electromagnetic coupling between the inductors L1 and L2, and the inductor current IL1 decreases after increasing according to the inductor current ILr. The inductor current IL1 rapidly decreases (relative to the negative direction) due to the influence of current flowing through the inductor L2 that is electromagnetically coupled to the inductor L1. Here, in order for the upper auxiliary current path to bypass the entire amount of the inductor current IL1, the inductor current ILr needs to flow more than the inductor current IL1, but when the inductor current IL1 rapidly decreases, the peak of the inductor current ILr is reduced. be able to.

  FIG. 12 shows a circuit diagram in a case where the current bidirectional converter of the present invention is applied to a DC-DC converter in the second embodiment. FIG. 12A shows a case where it is applied to a boost converter, and FIG. 12B shows a case where it is applied to a step-down converter. In the buck-boost converter of the first embodiment, the configuration excluding the transistors Q1, Q3 and the diode D3 essential for the step-down operation is the boost converter of FIG. 12A, and the transistors Q2, Q4 and the diode D4 essential for the boost operation. The configuration excluding is the step-down converter of FIG. Since the circuit operations and effects of FIGS. 12A and 12B are the same as those in the step-up operation and the step-down operation in the first embodiment, description thereof is omitted here.

  As described above in detail, according to the current bidirectional converter according to the present embodiment, the collector-emitter voltage at the time of switching is applied to the transistors Q1 and Q2 that perform the switching operation when the electromagnetic energy is stored and discharged in the inductor L1. Reduction of switching loss can be realized by performing a zero volt switching (ZVS) operation with a slight difference.

  With the reduction of switching loss, it is possible to improve the power efficiency in the voltage conversion of the current bidirectional converter. Heat generation due to switching loss in the transistors Q1 and Q2 can be reduced, and a cooling device such as a heat sink can be reduced in size and weight.

  It is possible to increase the frequency of the switching operation in the current bidirectional converter, and to perform the switching operation at a frequency higher than the audible frequency band. Vibrations of the inductor L1 and the like accompanying electromagnetic energy during operation can be shifted from the audible frequency band, and noise during operation can be prevented.

  In addition, the time gradient of the bypass current flowing in the auxiliary current path is determined according to the inductance value of the leakage inductor and the inductor Lr or / and the winding ratio of the inductors L1 and L2 provided in the auxiliary current path that bypasses the inductor current IL1. Can be adjusted. After forming the auxiliary current path, the entire amount of input current is bypassed, the voltage at the connection point X is reversed, and the electromagnetic energy is stored in the inductor L1, the time delay until the transistors Q1 and Q2 are turned on by zero volt switching (ZVS) Can be adjusted. In addition, when the transistors Q1 and Q2 operate as synchronous rectification elements, the non-conduction timing of the transistors can be adjusted. Furthermore, the voltage level applied to the transistors Q3 and Q4 forming the auxiliary current path when the transistors Q3 and Q4 are not conductive can be adjusted.

The present invention is not limited to the above-described embodiment, and it goes without saying that various improvements and modifications can be made without departing from the spirit of the present invention.
For example, in the present embodiment, the case where the emitter terminal of the transistor Q4 is connected to the anode terminal of the backflow prevention diode D4 and the cathode terminal of the diode D4 is connected to the reference voltage terminal TS has been described. It is not limited to connection. The connection order of the transistor Q4 and the backflow prevention diode D4 may be reversed. That is, the anode terminal of the backflow prevention diode D4 may be connected to the terminal of the inductor Lr, the cathode terminal thereof is connected to the collector terminal of the transistor Q4, and the emitter terminal of the transistor Q4 may be connected to the reference voltage terminal TS. it can. With this connection, both the transistors Q2 and Q4 have the same potential at the emitter terminals, and conduction control can be performed by a common drive power supply.
In the present embodiment, the case where the transistor Q3 is kept non-conductive in the step-up operation (FIG. 2) and the transistor Q4 is kept non-conductive in the step-down operation (FIG. 7) has been described. It is not limited to this. In the step-up operation, even if the transistor Q3 is turned on and the transistor Q3 is turned on to form the upper auxiliary current path after the transistor Q2 is turned off, the voltage VQ2 at the connection point X is substantially equal to the voltage V2 at this time. Therefore, no voltage is applied between the terminals of the inductors L2 and Lr, and no unnecessary current flows. On the contrary, in the step-down operation, even if the transistor Q4 is turned off and the transistor Q4 is turned on and the lower auxiliary current path is formed after the transistor Q1 is turned off, the voltage VQ2 at the connection point X remains at the reference voltage at this time. Since the voltages are substantially equal, no voltage is applied between the terminals of the inductors L2 and Lr, and no unnecessary current flows. In other words, the transistor Q3 that is not originally required to be conductive in the step-up operation can be controlled to be conductive at the timing of the step-down operation, and the transistor Q4 that is not originally required to be conductive in the step-down operation is controlled to be conductive at the timing of the step-up operation. be able to. Common conduction control can be performed in the step-up operation and the step-down operation.
Further, the case where the backflow prevention diodes D3 and D4 are connected to the emitter terminal sides of the transistors Q3 and Q4, respectively, has been shown. However, the present invention is not limited to this, and any part of the auxiliary current path can be used. It is also possible to arrange them.
Further, the case where the transformer is configured as the electromagnetic coupling of the inductor L1 and the inductor L2 and the winding ratio is described as 2: 1 has been described, but the present invention is not limited to this, and the winding ratio is adjusted. be able to.
In this embodiment, the case where the capacitors C1 and C2 are connected in parallel between the current path terminals of the transistors Q1 and Q2 has been described as an example. However, the voltage at the connection point X determined by charging and discharging of the capacitors is described. In the case where the capacitance value of the capacitor can be secured so that the change gradually changes to the extent that zero-volt switching (ZVS) of the transistors Q1 and Q2 is possible, it is only necessary to provide either one of the capacitors C1 or C2.

It is a circuit diagram of the current bidirectional converter of the first embodiment. It is a timing chart which shows the step-up operation in the current bidirectional converter of the first embodiment. It is a figure which shows the accumulation | storage period of the electromagnetic energy to an inductor among pressure | voltage rise operations. It is a figure which shows the discharge | release period of the electromagnetic energy from an inductor among pressure | voltage rise operations. It is a figure which shows the period which changes from discharge | release of electromagnetic energy to accumulation | storage among boosting operations. It is a figure which shows conduction | electrical_connection of the transistor at the time of changing to the accumulation | storage state of electromagnetic energy among pressure | voltage rise operations. It is a timing chart which shows step-down operation in the current bidirectional converter of a 1st embodiment. It is a figure which shows the accumulation | storage period of the electromagnetic energy to an inductor among step-down operations. It is a figure which shows the discharge | release period of the electromagnetic energy from an inductor among step-down operations. It is a figure which shows the period which changes from discharge | release of electromagnetic energy to accumulation | storage in step-down operation. It is a figure which shows conduction | electrical_connection of the transistor at the time of changing to the accumulation | storage state of electromagnetic energy among step-down operations. It is a circuit diagram of the DC-DC converter of 2nd Embodiment. It is a circuit diagram of a motor drive device provided with a chopper circuit in the prior art.

Explanation of symbols

1, 1A, 1B Auxiliary circuit C1, C2 Capacitors D1, D2 Reverse parallel diodes D3, D4 Backflow prevention diodes L1, L2, Lr Inductors Q1 to Q4 Transistor T1 Low voltage power supply terminal T2 High voltage power supply terminal TS Reference voltage terminals V1, V2 Voltage source X Connection point IL1, ILr Inductor current

Claims (8)

A first inductor having a first terminal connected to the low voltage power supply terminal;
A first switching element connected between the second terminal of the first inductor and one of a reference voltage terminal and a high-voltage power supply terminal;
A bidirectional current converter comprising: a connection point between the first inductor and the first switching element; and a first rectifier element connected between the reference voltage terminal or the other of the high-voltage power supply terminals,
A capacitor connected in parallel between at least one of the first switching element and the first rectifying element;
An auxiliary current path connected in parallel with the first switching element,
The auxiliary current path is:
A first inductor that is electromagnetically coupled to the first inductor and induces an electromotive force having the same polarity as the first terminal of the first inductor; and a second inductor connected to the connection point;
A second switching element;
A bidirectional current converter characterized in that a second rectifying element for preventing backflow is connected in series.   The bidirectional current converter according to claim 1, wherein the auxiliary current path includes a third inductor connected in series to a second terminal of the second inductor.   2. The bidirectional current converter according to claim 1, further comprising an antiparallel diode connected between current path terminals of the first switching element with a direction opposite to a current direction of the input current as a forward direction. .   The other terminal of the first switching element is connected to the reference voltage terminal, and a voltage boosted with respect to the voltage of the low voltage power supply terminal is supplied to the high voltage power supply terminal. The current bidirectional converter described in 1.   The other terminal of the first switching element is connected to the high-voltage power supply terminal, and a voltage stepped down with respect to the voltage of the high-voltage power supply terminal is supplied to the low-voltage power supply terminal. The current bidirectional converter described in 1. A first inductor having a first terminal connected to the low voltage power supply terminal;
A bidirectional current converter comprising an upper switching element and a lower switching element connected in series between a high-voltage power supply terminal and a reference voltage terminal, and having the connection point connected to the second terminal of the first inductor,
A capacitor connected in parallel between at least one of the upper and lower switching elements;
A lower auxiliary current path connected in parallel with the lower switching element;
An upper auxiliary current path connected in parallel with the upper switching element,
The lower auxiliary current path is:
A first terminal that is electromagnetically coupled to the first inductor and induces an electromotive force having the same polarity as that of the first terminal of the first inductor passes through a second inductor connected to the connection point,
A downward rectifier for preventing backflow;
The lower auxiliary switching element is connected in series,
The upper auxiliary current path is:
Through the second inductor,
An upper rectifying element for backflow prevention;
A current bidirectional converter, wherein an upper auxiliary switching element is connected in series.   The current bidirectional converter according to claim 6, further comprising a third inductor connected in series to a second terminal of the second inductor and constituting the upper and lower auxiliary current paths.   The antiparallel diode connected to a current direction of the input current as a forward direction is provided at least one of the current path terminals of the upper and lower switching elements. 6. The current bidirectional converter according to 6.

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