JP2003033013A - Resonant bidirectional DC-DC converter and control method therefor - Google Patents

Abstract

(57) [Abstract] [Problem] To suppress generation of noise and heat by soft switching, and to boost DC voltage to DC voltage of different voltage,
Alternatively, a step-down resonant bidirectional DC-DC converter is provided. SOLUTION: MOSFETs Q1, Q2 connected in series
Is connected to a smoothing inductance L and a MOSFET Q
1 and Q2, buffer capacitors C1 and C2
In the resonance type bidirectional DC-DC converter, the MOSFET Q for flowing a resonance current to the buffer capacitors C1 and C2 across the smoothing inductance L.
An auxiliary circuit H1 having a bidirectional switch circuit using r1 and Qr2 and a resonance inductance Lr connected in series thereto is connected. In both the step-up and step-down operations of the DC-DC converter, the buffer capacitor C
1, the charge and discharge of C2 are controlled by the resonance current flowing through the resonance inductance Lr.
Soft switching is realized.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a DC-DC converter for converting a DC voltage into a DC voltage having a different voltage.

[0002]

2. Description of the Related Art In recent years, a 42V type vehicle power supply device has been proposed as a new power supply system with an increase in the load of a 12V type vehicle power supply device in a vehicle device such as an automobile. However, it will take some time for the vehicle power supply system to completely shift from the 12V system to the 42V system, including issues related to infrastructure development. Therefore, considering the diversion of the electric components for vehicles so far, etc. It is preferable that there are two power supply systems of 12V system and 42V system.
Further, when these 12V system and 42V system power supply devices for a vehicle are used together, when one of the batteries has a small remaining amount of power, energy is supplied from the 42V system to the 12V system,
On the contrary, it is preferable that energy can be supplied from the 42V system to the 12V system. Therefore, a power conversion device that mutually complements the power of the two power supply systems is required.

Conventionally, as a power converter (DC-DC converter) for converting such a DC voltage into a DC voltage of a different voltage, for example, there is one described in JP-A-8-80038. The DC-DC converter switches a DC voltage to intermittently supply it to an inductive load such as a coil or a transformer, rectifies the electric energy excited by this inductive load, and extracts it as a DC voltage having a different voltage. . By connecting two such DC-DC converters in parallel, the DC voltage can be stepped up to a DC voltage of a different voltage and the DC voltage can be stepped down like the above-described mutual complementation of the 12V system and the 42V system. A corresponding method can be considered,
There is much waste in terms of circuit scale and cost.

Therefore, as shown in FIG. 19, in order to mutually complement the 12V system and the 42V system, the non-insulation corresponding to both the step-up and step-down of the DC voltage to the DC voltage of different voltage is performed. A bidirectional DC-DC converter is used. This non-isolated bidirectional DC-DC converter
It is arranged between the 12V vehicle power supply device V1 connected to the 12V system load R1 and the 42V vehicle power supply device V2 connected to the 42V system load R2, and intermittently switches one electric energy. Supply to the other. Specifically, a MOSFET (Metal Oxide) that is a first and second main switching element in which a drain terminal DC1 and a source terminal SC2, which are turned on or off by switching control, are connected in series at both ends of a 42V vehicle power supply device V2. Semico
nductor Field Effect Transistor) Q1 and MOS
It is equipped with a FET Q2. In addition, MOSFETs Q1 and M
One end is connected to the connection point of OSFET Q2, 12V
A smoothing inductance L for storing electric energy used for voltage conversion is provided, the other end of which is connected to the vehicle power supply device V1.

Further, between the drain and source terminals of the MOSFETs Q1 and Q2 (between DC1 and SC1 or DC2).
And SC2), the parasitic diodes QD1 and QD2 included in the MOSFET exist in parallel. Also, 12V
A smoothing capacitor C3 for voltage smoothing is connected to both ends of the vehicle power supply device V1, and a smoothing capacitor C4 for voltage smoothing is connected to both ends of the 42V vehicle power supply device V2.

Next, the operation of the above-described non-isolated bidirectional DC-DC converter will be briefly described. First, non-isolated bidirectional D
In the step-up operation in which the C-DC converter supplies the voltage to the 42V system by using the electric energy of the 12V vehicle power supply device V1, the switching operation in which conduction and interruption between the drain terminal DC1 and the source terminal SC1 of the MOSFET Q1 are intermittently repeated. I do. Then, the voltage induced in the smoothing inductance L by this is rectified by the rectifying diode D2 and further smoothed by the smoothing capacitor C4.
It is used to charge the 2V vehicle power supply device V2 or is supplied to the 42V load R2.

In the step-down operation in which the non-insulating bidirectional DC-DC converter supplies the voltage to the 12V system by using the electric energy of the 42V vehicle power supply device V2, the MOSFE is used.
A switching operation is performed in which conduction and interruption between the drain terminal DC2 and the source terminal SC2 of TQ2 are intermittently repeated.
The voltage induced in the smoothing inductance L by this is rectified by the rectifying diode D1 and further smoothed by the smoothing capacitor C3 to be used for charging the 12V vehicle power supply device V1 or the 12V load R1. Or to supply.

The voltage ratio before and after the conversion in the step-up or step-down operation of the above-mentioned non-insulated bidirectional DC-DC converter is M by PWM (Pulse Width Modulation) control.
It is determined by the time ratio (duty ratio of the switching signal) of conduction and interruption between the drain terminal and the source terminal of the OSFET Q1 or the MOSFET Q2. In addition, FIG.
The diodes QD1 and QD2 shown in FIG.
It is a parasitic diode of FETQ1 and Q2, and is actually M
It is an element that is inside the OSFET element and does not exist as an actual component.

[0009]

However, in the above-mentioned non-isolated bidirectional DC-DC converter, the switching operation of the main switching element is hard switching in which a voltage is applied to the switching element and a current flows. Therefore, there is a problem that special measures against noise generated by switching such as surge voltage are indispensable. Further, since the ratio of the switching loss generated in the switching element is high, the heat sink cannot be downsized, and as a result, it is difficult to downsize the DC-DC converter.

The present invention has been made in view of the above problems, and it is a resonance type bidirectional DC-DC which suppresses generation of noise and heat by soft switching and boosts or lowers a DC voltage to a DC voltage of a different voltage. The purpose is to provide a converter.

[0011]

In order to solve the above-mentioned problems, the present invention according to claim 1 provides a first and a second main switching element (for example, MOSFET Q1 of the embodiment) in which diodes are connected in antiparallel. MOSFET Q2 or I
IGBTQ3 and diode D1 combination and IGBTQ
4 and the diode D2), and the first and second
And a smoothing inductance having one end connected to a connection point of the main switching element (for example, the smoothing inductance L of the embodiment) and a buffer connected in parallel with the first and second main switching elements. A capacitor (for example, the buffer capacitors C1 and C2 of the embodiment) and an auxiliary circuit (for example, the auxiliary circuit of the embodiment) connected in parallel to both ends of the smoothing inductance in order to flow a resonance current through the buffer capacitor. H1 or an auxiliary circuit H2), which is a resonant bidirectional DC-DC converter, wherein the auxiliary circuit is connected in series to the bidirectional switch circuit and the two bidirectional switch circuits. It is characterized by having a resonance inductance (for example, the resonance inductance Lr of the embodiment) that resonates with the buffer capacitor. With the above configuration, in both the step-up operation and the step-down operation of the DC-DC converter, charging and discharging of the two buffer capacitors connected in parallel to the first and second main switching elements are performed by the buffer capacitors. It is controlled by the resonance current flowing through the resonance inductance forming the resonance circuit, thereby realizing the switching operation (so-called "soft switching operation") at the zero voltage and the zero current of the first and second main switching elements. , DC-DC
It is possible to perform an efficient operation while suppressing the generation of loss in the switching of the converter.

According to a second aspect of the present invention, there is provided the method of controlling the resonant bidirectional DC-DC converter according to the first aspect, wherein all the currents flowing through the smoothing inductance are:
A diode provided in anti-parallel with the first or second main switching element (for example, the MOSFET of the embodiment).
In the state where the current flows through the parasitic diodes QD1 and QD2 of TQ1 and Q2 or the rectifying diodes D1 and D2), the resonance inductance of the auxiliary circuit is turned on, and all the current flowing in the smoothing inductance is , The resonance inductance of the auxiliary circuit is set to a non-conducting state while flowing through the first or second main switching element (eg, MOSFET Q1, Q2 or IGBT Q3, Q4 of the embodiment). It is characterized by According to the above control, the electric energy stored in the smoothing inductance by one power source is supplied to the other power source as a stepped up or down voltage, and then the electric energy is supplied to the smoothing inductance again by the above one power source. The soft switching using the auxiliary circuit of each switching element forming the DC-DC converter is realized until the state of accumulating is stored.

According to a third aspect of the present invention, the first and second main switching elements of the resonant bidirectional DC-DC converter according to the first aspect are turned on or off by switching control. Of the MOSFET (for example, MOSFETQ1 and MOSFETQ2 of the embodiment), and the diode is a parasitic diode included in the MOSFET (for example, the parasitic diodes QD1 and QD2 of the MOSFETQ1 and Q2 of the embodiment). A control method for controlling a DC-DC converter, wherein all currents flowing in the smoothing inductance are
When the current starts to flow via the parasitic diode included in the first MOSFET or the second MOSFET, the MOSFET including the parasitic diode through which a current flows is made conductive at the same time. With the above configuration and control, DC-D
In the bidirectional voltage conversion operation of the C converter, the MOSFET on the side that performs the rectification function is made conductive, so that M
The voltage conversion operation of the DC-DC converter with a small voltage drop is realized by utilizing the synchronous rectification by the OSFET.

According to a fourth aspect of the present invention, the bidirectional switch circuit of the resonant bidirectional DC-DC converter according to the first aspect has third and fourth MOSFETs in which source terminals or drain terminals are connected to each other. (For example, M of the embodiment
A control method for controlling the DC-DC converter when it is composed of an OSFET Qr1 and a MOSFET Qr2, in which the third and fourth MOSFETs are simultaneously turned on in order to pass a resonance current through the resonance inductance. Of the third and fourth MOSFETs, just before the resonance current flowing through the resonance inductance becomes zero,
One of the MOSFETs in which the resonance current flows from the source terminal to the drain terminal is cut off, and after the resonance current flowing in the resonance inductance becomes zero, the other MOSFET is turned off.
It is characterized by blocking T. With the above configuration and control, in the bidirectional switch circuit for flowing the resonance current, two MOSFETs of the bidirectional switch circuit are simultaneously turned on, so that the voltage drop at the diode is eliminated and the conduction loss is reduced. Further, even if the resonance current becomes zero, the current of the resonance inductance does not flow in the opposite direction due to the parasitic diode of the MOSFET that is cut off.

According to a fifth aspect of the present invention, there is provided a first bidirectional switch circuit of the resonance type bidirectional DC-DC converter according to the first aspect, wherein the bidirectional switch circuit is turned on or off by switching control and connected in a reverse direction. , The second auxiliary switching element (for example, the IGBTQr3 and the IGBT of the embodiment).
Qr4) and the first and second direction control diodes (for example, the direction control diode Dr1 and the direction control diode Dr2 of the embodiment) connected in antiparallel with the first and second auxiliary switching elements, respectively. ) And the control method for controlling the DC-DC converter, the direction of the resonance current flowing through the resonance inductance and the first direction for flowing the resonance current through the resonance inductance. , One of the first and second auxiliary switching elements is made conductive in accordance with the conducting direction of the second direction control diode, and the conductive auxiliary switching element is made to flow to the resonance inductance. It is characterized in that it is cut off after the resonance current becomes zero. With the above configuration and control, the soft switching operation of the DC-DC converter is performed by the alternate conduction / interruption operation of the auxiliary switching element corresponding to the direction of the resonance current that changes between the step-up operation and the step-down operation of the bidirectional DC-DC converter. To realize.

[0016]

BEST MODE FOR CARRYING OUT THE INVENTION (First Embodiment) An embodiment of the present invention will be described below with reference to the drawings. Figure 1
Is a resonance type bidirectional DC-D according to the first embodiment of the present invention.
It is a circuit diagram which shows a C converter. In FIG. 1, the first
The resonant bidirectional DC-DC converter of the embodiment of
It is arranged between the 12V vehicle power supply device V1 connected to the 12V system load R1 and the 42V vehicle power supply device V2 connected to the 42V system load R2, and intermittently switches one electric energy. Supply to the other. Specifically, a MOSFET Q1 and a MOSFET Q2, which are turned on or off by switching control, are provided at both ends of the 42V vehicle power supply device V2, and the drain terminal DC1 of the MOSFET Q1 and the source terminal S of the MOSFET Q2 are provided.
C2 is connected and source terminal SC1 of MOSFET Q1
And the drain terminal DC2 of the MOSFET Q2 are connected to both ends of the 42V vehicle power supply device V2. Further MOSF
One end is connected to the connection point of ETQ1 and MOSFET Q2, and the other end is connected to the 12V vehicle power supply device V1. The smoothing inductance L for storing electric energy used for voltage conversion is provided. In addition, MOSF
ETQ1 and Q2 form first and second main switching elements.

Further, the drain terminal D of the MOSFET Q1
A buffer capacitor C1 is connected in parallel between C1 and the source terminal SC1, and the drain terminal D of the MOSFET Q2 is connected.
A buffer capacitor C2 is connected in parallel between C2 and the source terminal SC2. Also, MOSFETs Q1 and Q
Between the drain and source terminals of 2 (between DC1 and SC1,
Parasitic diodes QD1 and QD2 included in the MOSFET exist in parallel between DC2 and SC2. Furthermore,
In order to pass a resonance current through the buffer capacitors C1 and C2,
The MOSFET Qr1 is connected to both ends of the smoothing inductance L.
And MOSFET Qr2 and resonance inductance Lr
Is connected in parallel. Here, the auxiliary circuit H1 has source terminals (SCr1, SC
r2) connected to two MOSFETs Qr1 and Qr2
In addition, the resonance inductance Lr is connected in series. The two MOSFETs Qr1 and Qr2 may be connected so that the parasitic diodes QDr1 and QDr2 are in opposite directions, and the drain terminals (DCr1 and DCr2) are connected to each other.
A configuration in which 2) is connected may be used.

A smoothing capacitor C3 for voltage smoothing is connected to both ends of the 12V vehicle power supply device V1.
A smoothing capacitor C4 for voltage smoothing is connected to both ends of the 2V vehicle power supply device V2. The diodes QD1, QD2, QDr1 and QDr2 shown in FIG.
These are parasitic diodes of the MOSFETs Q1, Q2, Qr1, and Qr2, respectively, which are elements that are actually inside the MOSFET elements and do not exist as actual parts.

Next, the operation of the above resonance type bidirectional DC-DC converter will be briefly described. First, non-isolated bidirectional D
In the step-up operation in which the C-DC converter uses the electric energy of the 12V vehicle power supply device V1 to supply a voltage to the 42V system, a switching operation in which conduction and interruption between the drain terminal and the source terminal of the MOSFET Q1 and MOSFET Q2 are alternately repeated. To do. The voltage induced in the smoothing inductance L by this is rectified by the MOSFET Q2 and further smoothed by the smoothing capacitor C4.
It is used to charge the 2V vehicle power supply device V2 or is supplied to the 42V load R2. At this time, especially MOSF
When switching the ETQ1 from the cutoff state to the conduction state, the buffer capacitors C1 and C2 and the resonance inductance Lr
The voltage applied to the MOSFET Q1 and the flowing current are controlled by the resonance currents of and to realize soft switching of the MOSFET Q1.

In the step-down operation in which the non-insulated bidirectional DC-DC converter supplies the voltage to the 12V system by using the electric energy of the 42V vehicle power supply device V2, the MOSFE is used.
A switching operation is performed in which conduction and interruption between the drain terminal and the source terminal of TQ1 and MOSFET Q2 are alternately repeated. The voltage induced in the smoothing inductance L by this is rectified by the MOSFET Q1, smoothed by the smoothing capacitor C3, and used for charging the 12V vehicle power supply device V1 or supplied to the 12V load R1. Or At this time, particularly when switching the MOSFET Q2 from the cutoff state to the conduction state, the buffer capacitor C
The voltage applied to the MOSFET Q2 and the flowing current are controlled by the resonance current of the inductors C1 and L2 and the resonance inductance Lr to realize the soft switching of the MOSFET Q2.

The voltage ratio before and after the conversion in the step-up or step-down operation of the above-described non-insulated bidirectional DC-DC converter is the MOSFET Q1 or MO controlled by PWM.
It is determined by the time ratio (duty ratio of the switching signal) of conduction and interruption between the drain terminal and the source terminal of the SFET Q2. In addition to MOSFETs, reverse blocking thyristors,
A GTO (Gate Turn Off Thyristor), a bipolar transistor, an IGBT, or the like may be used. An example using a reverse blocking thyristor, a GTO, a bipolar transistor, and an IGBT is the second embodiment, representatively when the IGBT is used. The details will be described later.

Next, the control procedure and operation of the resonance type bidirectional DC-DC converter of the first embodiment will be described in detail with reference to the drawings. In explaining the operation of the circuit,
The notation of voltage and current of each part in the circuit diagram of FIG. 1 will be defined first. First, the definition of the voltage is that the buffer capacitor C1
The voltage applied to both ends of the resonance inductance L is expressed as Vc1 in the positive direction, and the voltage applied to both ends of the buffer capacitor C2 to the 42V system vehicle power supply device V2 is expressed in the positive direction. The voltage thus set is Vc2. Further, the definition of the current is that the current flowing through the smoothing inductance L is IL. Here, since the direction in which the current IL flows is opposite between the step-up operation and the step-down operation, during the step-up operation, the 12V system vehicle power supply device V1 to the MOSFET Q
1 or the direction flowing toward the MOSFET Q2 is the positive direction, and the reverse direction is the positive direction during the step-down operation.

Similarly, the current flowing through the resonance inductance Lr is ILr. Here, since the direction in which the current ILr flows is opposite between the step-up operation and the step-down operation, during the step-up operation, the direction in which the current ILr flows from the MOSFET Q1 or the MOSFET Q2 to the 12V system vehicle power supply device V1 is the positive direction, When operating, the opposite is the forward direction.

In addition, the current flowing from the side connected to the resonance inductance L to the MOSFET Q1 is expressed as the positive direction, and the current flowing to the MOSFET Q2 from the side connected to the IQ1 and 42V system power supply device V2 is positive. The current expressed as a direction is IQ2.

Next, based on the notation of voltage and current of each part defined above, first, each state from (a) mode a-1 to (i) mode a-9 shown in FIG. 2 to FIG. 5 and the waveform diagram shown in FIG. 5, the boosting operation of the resonant bidirectional DC-DC converter of the first embodiment will be described. 2 to 4, the circuit elements in the non-conducting state are omitted in order to make it easier to understand the conducting state. In the waveform diagram of FIG. 5, the mode numbers shown at the bottom correspond to the above mode numbers, and each waveform shows a signal waveform corresponding to each of the above modes. First, (a) in mode a-1, the resonance type bidirectional DC of the first embodiment
-DC converter is in steady state of boosting operation
The V-system vehicle power supply device V1 stores electric energy in the smoothing inductance L. At this time, the MOS
The FET Q1 is in the conductive state and the MOSFET Q2 is in the cut-off state. Therefore, the current I flowing through the smoothing inductance L is
All L flows through the MOSFET Q1.

Next, in (b) mode a-2, MO
When the SFET Q1 is turned off, the current IL flowing through the smoothing inductance L charges the buffer capacitor C1 and discharges the buffer capacitor C2. This state is
Smoothing inductance L and buffer capacitors C1 and C2
And are in resonance. Further, at this time, the MOSFET Q1 remains in the ON state until the resonance state is reached, and the voltage Vc1 across the buffer capacitor C1 to which no voltage is applied rises. However, the voltage Vc1 at both ends cannot be rapidly increased because of the time constant given by the buffer capacitor C1, and the MOSFET Q1 is turned off in a state where the voltage at both ends of the buffer capacitor C1, that is, the voltage Vc1 at the end is “zero”. Zero voltage switching (ZVS: Ze
ro Voltage Switching). In the waveform diagram of FIG. 5, the part which becomes the zero voltage switching is “(a) ZVS”.
Is shown as.

When the voltage Vc2 across the buffer capacitor C2 reaches "zero" by turning off the MOSFET Q1, (c) in mode a-3, the MOS is turned on.
The parasitic diode QD2 of the FET Q2 becomes conductive,
All the current IL flowing through the smoothing inductance L comes to flow through the parasitic diode QD2, but when the parasitic diode QD2 becomes conductive, the MOSFET Q2 is turned on at the same time. Then, the smoothing inductance L
The current IL flowing through the source Q does not pass through the diode QD2, and the source terminal S of the MOSFET Q2 having a low on-resistance is
CV and the drain terminal DC2 can be made to flow, the conduction loss is reduced as compared with the case where the current is conducted to the diode QD2, and the electric power is efficiently supplied to the 42V vehicle power supply device V2
And synchronous rectification of the MOSFET is realized. Therefore, due to the rectifying action of this MOSFET Q2,
The electric energy boosted from the 12V vehicle power supply device V1 and stored in the smoothing inductance L is transmitted to the 42V vehicle power supply device V2, and the resonance type bidirectional DC-DC converter of the first embodiment is used. Boost operation.

Further, when the MOSFET Q2 is turned on, the voltage Vc2 across it is "zero", and IQ2 is "zero" because the current does not start flowing.
The turn-on of the FET Q2 is performed by zero voltage switching and zero current switching (ZCS: Zero current Switchin).
g). In the waveform diagram of FIG. 5, the part which is the zero voltage switching and the zero current switching is shown in “(b) ZV”.
S ”and“ (c) ZCS ”.

Next, the MOSFET Q2 is turned off.
Then, as a preparation for returning to the steady state (a) mode a-1,
(D) In mode a-4, the MOSFET Qr2 is turned on, and at the same time, the MOSFET Qr1 is turned on. Then, at both ends of the resonance inductance Lr, 4
Since the voltage of the difference between the voltage of the 2V system vehicle power supply device V2 and the voltage of the 12V system vehicle power supply device V1 is applied, the resonance current ILr flowing through the resonance inductance Lr.
Rises linearly. At this time, MOSFETQ
r1 and Qr2 turn on when no current is flowing
Therefore, zero current switching is performed. In the waveform diagram of FIG. 5, the part which becomes the zero current switching is indicated by “(d)
ZCS ”and“ (e) ZCS ”. At this time, the resonance current ILr can be made conductive by the parasitic diode QDr1 without turning on the MOSFET Qr1. However, since more conduction loss occurs in the parasitic diode QDr1, in order to avoid this, the on-resistance of the on-resistance is increased. By turning on the small MOSFET Qr1, the loss during conduction is reduced as compared with the case where the parasitic diode QDr1 is conducted.

In this state, when the resonance current ILr flowing through the resonance inductance Lr becomes larger than the current IL flowing through the smoothing inductance L, in the (e) mode a-5, the 42V system vehicle power supply device V2 is supplied to the MOSFET Q2.
A current in the direction of flowing into the MOSFET Q2 begins to flow from the side connected to. Next, in the (f) mode a-6, when the MOSFET Q2 is turned off, resonance occurs between the resonance inductance Lr and the buffer capacitors C1 and C2, and a part of the resonance current ILr flowing through the resonance inductance Lr becomes The buffer capacitor C1 is discharged and the buffer capacitor C2 is charged. Further, at this time, the voltage Vc2 across the buffer capacitor C2 cannot be rapidly increased because of the time constant given by the buffer capacitor C2.
The SFET Q2 is turned off with zero voltage switching when the voltage Vc2 across the SFET Q2 is "zero". In the waveform diagram of FIG. 5, the part that causes the zero voltage switching is shown as “(f) ZVS”.

In (g) mode a-7, when the voltage Vc1 across the buffer capacitor C1 becomes "zero", the parasitic diode QD1 of the MOSFET Q1 becomes conductive, but the transition timing to this state is When the MOSFET Q1 is turned on synchronously, no current flows through the parasitic diode QD1 and synchronous rectification is performed between the source terminal SC1 and the drain terminal DC1 of the MOSFET Q1. Although the parasitic diode QD1 can be rectified by the parasitic diode QD1 without turning on the MOSFET Q1 when the parasitic diode QD1 becomes conductive, the MOSFET Q1 having a small ON resistance is forcibly rectified.
By turning on, the loss during conduction can be reduced more than when conducting the current through the parasitic diode.
Further, when the MOSFET Q1 is turned on, the voltage Vc1 across the MOSFET Q1 is "zero", and IQ1 is "zero" before the current starts flowing. Therefore, the turn-on of the MOSFET Q1 is zero voltage switching and zero current switching. Become. In the waveform diagram of FIG. 5, the part which becomes the zero voltage switching and the zero current switching is represented by “(g) Z”.
VS ”and“ (h) ZCS ”.

At this time, the resonance inductance L
Since the voltage of the 12V system vehicle power supply device V1 is applied to r in the direction of decreasing the resonance current ILr, the resonance current ILr flowing through the resonance inductance Lr gradually decreases. In this state, when the resonance current ILr flowing through the resonance inductance Lr begins to become smaller than the current IL flowing through the smoothing inductance L, the MOSFET Q1 is in the (g) mode a-7 in the (h) mode a-8. On the contrary, the current in the direction of flowing to the MOSFET Q1 starts to flow from the side connected to the smoothing inductance L.
Further, the resonance current IL flowing through the resonance inductance Lr
When r decreases and approaches "zero", in the (i) mode a-9, the MOSFET Qr1 is turned off first.
F Then, the current flows through the parasitic diode QDr1 of the MOSFET Qr1.

When the resonance current ILr flowing through the resonance inductance Lr decreases to "zero", the MOSFET Qr is moved to the next steady state (a) mode a-1.
Turn off 2 At this time, MOSFET Qr2
Causes zero current switching because the current is turned off when the current is "zero". In the waveform diagram of FIG. 5, the part that causes the zero current switching is shown as “(i) ZCS”.

The turn-off of the MOSFET Qr2
Can be performed while the resonance current is not flowing, so turn off during the steady state (a) mode a-1.
It may be done. Further, even if the resonance current ILr decreases to zero, the parasitic diode Dr1 prevents the resonance current ILr from flowing in the opposite direction. As described above, one cycle of the boosting operation of the resonant bidirectional DC-DC converter of the first embodiment is completed. Also, DC
-The voltage ratio of the voltage conversion of the DC converter is the time ratio for maintaining the states of (a) mode a-1 and (c) mode a-3,
It is determined by controlling by PWM.

Next, (a) mode b shown in FIG. 6 to FIG.
-1 to (i) Modes b-9, and the waveform diagram shown in FIG. 9, the step-down operation of the resonant bidirectional DC-DC converter of the first embodiment will be described. To do. In the waveform diagram of FIG. 9, the mode numbers shown at the bottom correspond to the above mode numbers, and each waveform shows a signal waveform corresponding to each of the above modes. First, in (a) mode b-1, the resonant bidirectional DC-DC converter of the first embodiment is in a steady state of step-down operation,
The 42V system power supply device V2 for a vehicle has a smoothing inductance L
It is assumed that electric energy is being stored in the. At this time, the MOSFET Q1 is in the cut-off state and the MOSFET Q2 is in the conducting state. Therefore, all the current IL flowing through the smoothing inductance L flows through the MOSFET Q2.

Next, in (b) mode b-2, MO
When the SFET Q2 is turned off, the current IL flowing through the smoothing inductance L discharges the buffer capacitor C1 and charges the buffer capacitor C2. This state is
Smoothing inductance L and buffer capacitors C1 and C2
And are in resonance. Further, at this time, the MOSFET Q2 is in the ON state until the resonance state is reached, and the voltage Vc2 across the buffer capacitor C2, to which no voltage is applied, rises. However, the voltage Vc2 cannot rise rapidly due to the time constant given by the buffer capacitor C2,
The MOSFET Q2 has zero voltage switching in which the voltage across the buffer capacitor C2, that is, the voltage Vc2 across the MOSFET Q2 is turned off when the voltage is zero. In the waveform diagram of FIG. 9, the part that causes the zero voltage switching is shown as “(a) ZVS”.

When the voltage Vc1 across the buffer capacitor C1 reaches "zero" by turning off the MOSFET Q2, (c) in mode b-3, the MOS is turned on.
The parasitic diode QD1 of the FET Q1 becomes conductive,
All the current IL flowing through the smoothing inductance L comes to flow through the parasitic diode QD1, but when the parasitic diode QD1 becomes conductive, the MOSFET Q1 is turned on at the same time. Then, the smoothing inductance L
A current IL flowing through the MOSFET Q1 comes to flow, and the electric energy stored in the smoothing inductance L does not pass through the parasitic diode QD1 and has a low on resistance and a small conduction loss. The signal is transmitted to the 12V vehicle power supply device V1 through the space and the synchronous rectification of the MOSFET is realized. Therefore, due to the rectifying action of the MOSFET Q1, the electric energy stored in the smoothing inductance L after being stepped down from the power supply voltage of the 42V type vehicle power source device V2 is transmitted to the 12V type vehicle power source device V1.
This is the step-down operation of the resonant bidirectional DC-DC converter of the first embodiment.

When the MOSFET Q1 is turned on, the voltage Vc1 across the MOSFET Q1 is "zero" and IQ1 is "zero" because the current does not start flowing.
Turn-on of the FET Q1 results in zero voltage switching and zero current switching. In the waveform diagram of FIG. 9, the portions that are the zero voltage switching and the zero current switching are shown as “(b) ZVS” and “(c) ZCS”.

Next, the MOSFET Q1 is turned off.
Then, in preparation for returning to the steady state (a) mode b-1,
(D) In mode b-4, the MOSFET Qr1 is turned on, and at the same time, the MOSFET Qr2 is turned on. Then, 1 is provided at both ends of the resonance inductance Lr.
Since the voltage of the 2V system vehicle power supply device V1 is applied, the resonance current I flowing through the resonance inductance Lr
Lr rises linearly. At this time, the MOSF
The ETQr1 and Qr2 are turned on in the state where no current is flowing, so that zero current switching is performed. In the waveform diagram of FIG. 9, the part which becomes this zero current switching is shown as "(d) ZCS" and "(e) ZCS." At this time, the resonance current ILr can be made conductive by the parasitic diode QDr2 even if the MOSFET Qr2 is not turned on. However, a voltage drop occurs due to the conduction loss in the parasitic diode QDr2. By turning on the MOSFET Qr2 having a small value, the loss during conduction is reduced more than when the parasitic diode QDr2 is conducted.

In this state, when the resonance current ILr flowing through the resonance inductance Lr becomes larger than the current IL flowing through the smoothing inductance L, in the (e) mode b-5, from the side connected to the resonance inductance L in the MOSFET Q1. The current flowing in the MOSFET Q1 begins to flow. Next, in (f) mode b-6, M
When the OSFET Q1 is turned off, resonance occurs between the resonance inductance Lr and the buffer capacitors C1 and C2, and the resonance current IL flowing through the resonance inductance Lr.
Part of r charges the buffer capacitor C1 and discharges the buffer capacitor C2. At this time, the voltage Vc1 across the buffer capacitor C1 is equal to the buffer capacitor C1.
It cannot rise rapidly because of the time constant given by
TQ1 is turned off when the voltage Vc1 at both ends is "zero".
Zero voltage switching is performed. In the waveform diagram of FIG. 9, the part that causes the zero voltage switching is represented by “(f) Z
It is shown as "VS".

In (g) mode b-7, when the voltage Vc2 across the buffer capacitor C2 becomes "zero", the parasitic diode QD2 of the MOSFET Q2 becomes conductive, but the transition timing to this state is When the MOSFET Q2 is turned on synchronously, no current flows through the parasitic diode QD2, and synchronous rectification is performed between the source terminal SC2 and the drain terminal DC2 of the MOSFET Q2. When the parasitic diode QD2 becomes conductive, it can be rectified by the parasitic diode QD2 without turning on the MOSFET Q2, but the MOSFET Q2 having a small on-resistance is forced.
By turning on, the loss during conduction can be reduced more than when conducting the current through the parasitic diode.
Further, when turning on the MOSFET Q2, the voltage Vc2 across it is "zero", and IQ2 is "zero" because no current starts flowing, so turning on of the MOSFET Q2 results in zero voltage switching and zero current switching. Become. In the waveform diagram of FIG. 9, the part which becomes the zero voltage switching and the zero current switching is represented by “(g) Z”.
VS ”and“ (h) ZCS ”.

At this time, the resonance inductance L
r is the voltage of the 42V system vehicle power supply device V2 and the 12V system vehicle power supply device V in the direction of decreasing the resonance current ILr.
Since the voltage having the voltage difference of 1 is applied, the resonance current ILr flowing through the resonance inductance Lr gradually decreases.
In this state, when the resonance current ILr flowing through the resonance inductance Lr begins to become smaller than the current IL flowing through the smoothing inductance L, the (h) mode b-8 is different from that in the (g) mode b-7 in the MOSFET Q2. On the contrary, a current in the direction of flowing into the MOSFET Q2 starts to flow from the side connected to the 42V type vehicle power supply device V2. Further, when the resonance current ILr flowing through the resonance inductance Lr decreases and approaches "zero", in the (i) mode b-9, the MOSFET Qr2 is turned off first. Then, the current flows through the parasitic diode QDr2 of the MOSFET Qr2.

When the resonance current ILr flowing through the resonance inductance Lr decreases to "zero", the MOSFET Qr is moved to the next steady state (a) mode b-1.
Turn 1 off. At this time, MOSFET Qr1
Is turned off when the resonance current ILr is "zero", so that zero current switching is performed. In the waveform diagram of FIG. 9, the part which becomes the zero current switching is represented by “(i) Z
It is shown as "CS".

The turn-off of the MOSFET Qr1
Can be performed while the resonance current is not flowing, so turn-off is performed during the steady state (a) mode b-1.
It may be done. Even if the resonance current ILr decreases to zero, the parasitic diode Dr2 does not cause the resonance current ILr to flow in the reverse direction. From the above, the first
One cycle of the step-down operation of the resonant bidirectional DC-DC converter of the embodiment is completed. Further, the voltage ratio of the voltage conversion of the DC-DC converter is determined by controlling the time ratio for maintaining the states of (a) mode b-1 and (c) mode b-3 by PWM.

(Second Embodiment) Next, a second embodiment of the present invention will be described with reference to the drawings. Figure 10
Is a resonance type bidirectional DC-D according to the second embodiment of the present invention.
It is a circuit diagram which shows a C converter. Second, shown in FIG.
The resonant bidirectional DC-DC converter of the embodiment of
The MOSFETs Q1 and Q2 of the first embodiment have IGBTs Q3 and I in which a collector terminal and an emitter terminal, which are turned on or off by switching control, are connected in series.
Further, the rectifying diode D1 is connected in antiparallel between the collector terminal and the emitter terminal of the GBQQ4, and the rectifying diode D2 is connected in antiparallel between the collector terminal and the emitter terminal of the IGBTQ4.

Similarly, M of the first embodiment is
The auxiliary circuit H1 including the OSFET Qr1, the MOSFET Qr2, and the resonance inductance Lr is the IGBTQ.
r3, Qr4 and direction control diodes Dr1, Dr2,
And an auxiliary circuit H2 including a resonance inductance Lr. Here, in the auxiliary circuit H2, the direction control diodes Dr1 and Dr2 are respectively connected in antiparallel to the two IGBTs Qr3 and Qr4 whose emitter terminals are connected to each other, and the resonance inductance Lr is further connected to the two IGs.
It is configured by connecting BTQr3 and BTQr4 in series. In addition,
The two IGBTs Qr3 and Qr4 may have collector terminals connected to each other.

As described above, in the resonance type bidirectional DC-DC converter according to the second embodiment of the present invention shown in FIG. 10, the main switching element or the switching element of the auxiliary circuit is from the MOSFET to the IGBT and the rectifying diode. The other parts of the resonance type bidirectional DC-DC converter are the same as those of the resonance type bidirectional DC-DC converter of the first embodiment of the present invention shown in FIG. Is the resonance type bidirectional DC-D of the first embodiment.
It is no different from the C converter. Therefore, in boost operation,
In particular, when switching the IGBT Q3 from the cut-off state to the conductive state, the resonance current of the buffer capacitors C1 and C2 and the resonance inductance Lr controls the voltage applied to the IGBT Q3 and the flowing current, thereby realizing the soft switching of the IGBT Q3. To do. On the other hand, in the step-down operation, particularly when the IGBT Q4 is switched from the cutoff state to the conductive state, the voltage applied to the IGBT Q4 and the flowing current are controlled by the resonance current of the buffer capacitors C1 and C2 and the resonance inductance Lr. Implements soft switching of the IGBT Q4.

The voltage ratio before and after the conversion in the step-up or step-down operation of the above-described non-insulated bidirectional DC-DC converter is determined by the magnitude of the smoothing inductance L and the collector terminal and the emitter terminal of the IGBTQ3 or the IGBTQ4 by PWM control. It is determined by the time ratio of conduction and interruption (duty ratio of switching signal).

Next, the control procedure and details of the operation of the resonant bidirectional DC-DC converter of the second embodiment will be described with reference to the drawings. Resonant bidirectional DC of the second embodiment
-The control procedure and operation of the DC converter are basically the same as the control procedure and operation of the resonant bidirectional DC-DC converter of the first embodiment, so the description will focus on the different parts. . The other parts are the same as those in the first embodiment except that MOSFET Q1 is an IGBT Q3, parasitic diode QD1 is a rectifying diode D1, and MOSF.
The ETQ2 is replaced with the IGBTQ4, the parasitic diode QD2 is read as the rectifying diode D2, and the MOSFET Qr1 in the auxiliary circuit is the IGBTQr3, and the parasitic diode QDr1 is the direction controlling diode Dr1 and the MOSFET.
This can be explained by reading Qr2 as the IGBT Qr4 and the parasitic diode QDr2 as the direction control diode Dr2.

First, since the switching elements have been changed, in the circuit diagram of FIG. 10, the notation of voltage and current of each part different from that shown in the circuit diagram of FIG. 1 will be defined first. The definition of voltage is the same as that of the first embodiment, and nothing is changed. Also, the definition of the current is defined by the smoothing inductance L
The current flowing through the resonance inductance Lr does not change. On the other hand, IQ3, 4 is a current that is expressed as a positive direction from the side connected to the resonance inductance L to the parallel circuit of the IGBT Q3 and the rectifying diode D1.
From the side connected to the power supply device V2 for 2V vehicle, the IGBT
The current flowing in the parallel circuit of Q4 and the rectifying diode D2 as a positive direction is represented by IQ4.

Next, based on the notation of voltage and current of each part defined above, first, each state from (a) mode c-1 to (h) mode c-8 shown in FIG. 11 to FIG. And the waveform diagram shown in FIG. 14, the boosting operation of the resonant bidirectional DC-DC converter of the second embodiment will be described. In the waveform diagram of FIG. 14, the mode numbers shown at the bottom correspond to the above mode numbers, and each waveform shows a signal waveform corresponding to each of the above modes.

Here, in the control procedure and operation of the boosting operation of the resonant bidirectional DC-DC converter of the second embodiment, the boosting operation of the resonant bidirectional DC-DC converter of the first embodiment is described. The difference from the control procedure and operation is that in the (c) mode c-3, the rectifying diode D2 becomes conductive and the current IL flowing through the smoothing inductance L all flows through the rectifying diode D2. Even if the IGBT Q4 is turned on at the same time,
This means that the IGBT cannot perform synchronous rectification. Therefore, due to the rectifying action of the rectifying diode D2, the electric energy stored in the smoothing inductance L is
4 together with the voltage generated by the 12V vehicle power supply device V1
The voltage is transmitted to the 2V system vehicle power supply device V2, and the boost operation of the resonant bidirectional DC-DC converter of the second embodiment is performed.

Further, the IGBTQ4 is turned off, and preparation for returning to the steady state (a) mode c-1 is started (d).
In mode c-4, turn only IGBTQr4 O
It is not necessary to turn on the IGBT Qr3 at the same time. Because the current flowing through the auxiliary circuit H2 is IGB
When TQr4 is turned on, the direction control diode Dr
IG only when the IGBTQr3 is made conductive by only flowing 1
This is because the BTQr3 does not flow and the IGBT cannot perform synchronous rectification. Therefore, during the boosting operation, the IGBT
Qr3 is never turned on or turned off,
In order to allow the resonance current to flow through the resonance inductance Lr, the IGB corresponding to the direction of the resonance current flowing through the resonance inductance Lr and the conduction direction of the direction control diode Dr1.
The TQ4 is turned on to make it conductive, and after the resonance current becomes zero, the IGBT Q4 is turned off to cut it off.
The above two points are the resonance type bidirectional DC- of the second embodiment.
This is a part of the control procedure and operation of the boosting operation of the DC converter that is different from the control procedure and operation of the boosting operation of the resonant bidirectional DC-DC converter of the first embodiment.

Next, using the diagrams showing the respective states from (a) mode d-1 to (i) mode d-8 shown in FIGS. 15 to 17 and the waveform diagram shown in FIG. 18, the second The step-down operation of the resonant bidirectional DC-DC converter of the embodiment will be described. In the waveform diagram of FIG. 18, the mode numbers shown at the bottom correspond to the above mode numbers, and each waveform shows a signal waveform corresponding to each of the above modes.

Here, in the control procedure and operation of the step-down operation of the resonant bidirectional DC-DC converter of the second embodiment, the step-down operation of the resonant bidirectional DC-DC converter of the first embodiment is described. The difference from the control procedure and operation is that, in the (c) mode d-3, the rectifying diode D1 becomes conductive, and the current IL flowing through the smoothing inductance L all flows through the rectifying diode D1. Even if the IGBT Q3 is turned on at the same time,
This means that the IGBT cannot perform synchronous rectification. Therefore, due to the rectifying action of the rectifying diode D1, the electric energy stored in the smoothing inductance L is
The voltage is transmitted to the 12V system vehicle power supply device V1 and the resonant bidirectional DC-DC converter according to the second embodiment operates to step down.

Further, the IGBTQ3 is turned off, and preparation for returning to the steady state (a) mode d-1 is started (d).
In mode d-4, turn only IGBTQr3
It is not necessary to turn on the IGBT Qr4 at the same time. Because the current flowing through the auxiliary circuit H2 is IGB
When TQr3 is turned on, the direction control diode Dr
Even if the IGBT Qr4 is made conductive only by flowing through 2
This is because it does not flow through the BTQr4 and it does not have the same effect of reducing the conduction loss as the MOSFET. Therefore, the IGBTQr4 is not turned on or turned off during the step-down operation, and the resonance current flows through the resonance inductance Lr so that the resonance current flows through the resonance inductance Lr and the conduction direction of the direction control diode Dr2. Corresponding to, the IGBT Q3 is turned on to make it conductive, and after the resonance current becomes zero, the IGBT Q3 is turned off to cut it off.

The above two points are the same as the resonant bidirectional DC-DC of the first embodiment in the control procedure and operation of the step-down operation of the resonant bidirectional DC-DC converter of the second embodiment.
This part is different from the control procedure and operation of the step-down operation of the converter. As described above, in the resonance type bidirectional DC-DC converters of the first and second embodiments of the present invention, for example, one side is a 12V system and the other side is a 42V system. Can be replaced efficiently, so
Since it is not necessary to use two DC-DC converters, a step-up DC-DC converter and a step-down DC-DC converter, unnecessary parts are reduced and the size can be reduced. Therefore, it is possible to obtain the effect that the space required for installation is small when it is mounted on a vehicle device.

[0058]

As described above, according to the first aspect of the present invention, in both the step-up operation and step-down operation of the DC-DC converter, the first and second main switching elements are connected in parallel. The charging and discharging of the two buffer capacitors thus controlled are controlled by the resonance current flowing in the resonance inductance that forms a resonance circuit with the buffer capacitors, whereby soft switching of the first and second main switching elements is achieved. In addition, it is possible to realize efficient operation while suppressing the occurrence of loss in the switching of the DC-DC converter. Therefore, it is possible to obtain two types of voltages with one DC-DC converter without using the two DC-DC converters of the step-up DC-DC converter and the step-down DC-DC converter. By reducing the switching loss in the DC converter and suppressing the generation of heat, it is possible to obtain the effect that the circuit can be downsized. In addition, DC in which noise generated by switching in the DC-DC converter is reduced
-The effect that a DC converter can be realized is obtained.

According to the second aspect of the present invention, the electric energy stored in the smoothing inductance by one power source is supplied to the other power source as a voltage stepped up or stepped down, and then the above-mentioned power source is supplied. Soft switching using the auxiliary circuit of each switching element that constitutes the DC-DC converter is realized until the power is again stored in the smoothing inductance by one of the power supplies. Therefore, the effect that the soft switching of the DC-DC converter can be achieved by operating the auxiliary circuit only when necessary and performing control without waste is obtained.

According to the invention described in claim 3, D
In the bidirectional voltage conversion operation of the C-DC converter,
The voltage conversion operation of the DC-DC converter with a small voltage drop is realized by utilizing the synchronous rectification by the MOSFET by turning on the MOSFET on the side having the rectification function. Therefore, it is possible to further reduce the switching loss and noise generated in the DC-DC converter.

According to the invention described in claim 4, in the bidirectional switch circuit for flowing the resonance current, the MO
Since two SFETs are turned on at the same time, the conduction loss can be suppressed lower than when the parasitic diode is turned on. Therefore, even in the auxiliary circuit that realizes the soft switching of the DC-DC converter, it is possible to reduce the generated switching loss and noise.

According to the fifth aspect of the invention, the alternate conduction / interruption operation of the auxiliary switching element corresponding to the direction of the resonance current that changes between the step-up operation and the step-down operation of the bidirectional DC-DC converter is performed. , Realizes the soft switching operation of the DC-DC converter. Therefore, the effect that the DC-DC converter that operates efficiently can be realized by the simple control of the auxiliary switching element.

[Brief description of drawings]

FIG. 1 is a resonance type bidirectional D according to a first embodiment of the present invention.
It is a circuit diagram which shows a C-DC converter.

FIG. 2 is a diagram showing an operation for each mode in a boosting operation of the resonant bidirectional DC-DC converter of the same embodiment.

FIG. 3 is a diagram showing an operation for each mode in a boosting operation of the resonant bidirectional DC-DC converter of the same embodiment.

FIG. 4 is a diagram showing an operation for each mode in the boosting operation of the resonant bidirectional DC-DC converter of the same embodiment.

FIG. 5 is a waveform diagram showing a waveform of each portion that changes for each mode in the boosting operation of the resonant bidirectional DC-DC converter of the same embodiment.

FIG. 6 is a diagram showing an operation for each mode in the step-down operation of the resonant bidirectional DC-DC converter of the same embodiment.

FIG. 7 is a diagram showing an operation in each mode in the step-down operation of the resonant bidirectional DC-DC converter of the same embodiment.

FIG. 8 is a diagram showing an operation for each mode in the step-down operation of the resonant bidirectional DC-DC converter of the same embodiment.

FIG. 9 is a waveform diagram showing a waveform of each portion that changes for each mode in the step-down operation of the resonant bidirectional DC-DC converter of the same embodiment.

FIG. 10 is a circuit diagram showing a resonant bidirectional DC-DC converter according to a second embodiment of the present invention.

FIG. 11 is a diagram showing an operation for each mode in the boosting operation of the resonant bidirectional DC-DC converter of the same embodiment.

FIG. 12 is a diagram showing an operation for each mode in the boosting operation of the resonant bidirectional DC-DC converter of the same embodiment.

FIG. 13 is a diagram showing an operation for each mode in the boosting operation of the resonant bidirectional DC-DC converter of the same embodiment.

FIG. 14 is a waveform diagram showing a waveform of each portion that changes for each mode in the boosting operation of the resonant bidirectional DC-DC converter of the same embodiment.

FIG. 15 is a diagram showing an operation for each mode in the step-down operation of the resonant bidirectional DC-DC converter of the same embodiment.

FIG. 16 is a diagram showing an operation in each mode in the step-down operation of the resonant bidirectional DC-DC converter of the same embodiment.

FIG. 17 is a diagram showing an operation in each mode in the step-down operation of the resonant bidirectional DC-DC converter of the same embodiment.

FIG. 18 is a waveform diagram showing a waveform of each portion that changes for each mode in the step-down operation of the resonant bidirectional DC-DC converter of the same embodiment.

FIG. 19 is a circuit diagram showing a conventional non-isolated bidirectional DC-DC converter.

[Explanation of symbols]

V1 12V power supply device for vehicle V2 + 48V power supply device for vehicle Q1, Q2, Qr1, Qr2 MOSFET QD1, QD2, QDr1, QDr2 Parasitic diodes Q3, Q4, Qr3, Qr4 IGBT D1, D2 Rectification diode Dr1, Dr2 Direction control diode C1, C2 buffer capacitors C3, C4 smoothing capacitor L smoothing inductance Lr resonance inductances H1, H2 auxiliary circuit

Claims (5)

[Claims] 1. A first and second main switching element in which diodes are connected in antiparallel, and a smoothing inductance whose one end is connected to a connection point of the first and second main switching elements, A buffer capacitor connected in parallel to the first and second main switching elements, and an auxiliary circuit connected in parallel to both ends of the smoothing inductance for flowing a resonance current in the buffer capacitor. A resonance type bidirectional DC-DC converter comprising: the auxiliary circuit, a bidirectional switch circuit, and a resonance inductance that is connected in series to the bidirectional switch circuit and resonates with the two buffer capacitors. And a resonance type bidirectional DC-DC converter. 2. The resonance type bidirectional DC-D according to claim 1.
A method of controlling a C converter, wherein all the currents flowing through the smoothing inductance are flowing through a diode provided in antiparallel with the first or second main switching element,
When the resonance inductance of the auxiliary circuit is energized and all the current flowing in the smoothing inductance is flowing through the first or second main switching element, the resonance inductance of the auxiliary circuit A method of controlling a resonant bidirectional DC-DC converter, characterized in that the inductance is in a non-conducting state. 3. The resonance type bidirectional DC-D according to claim 1.
When the first and second main switching elements of the C converter are composed of first and second MOSFETs that are turned on or off by switching control, and the diode is composed of a parasitic diode included in the MOSFET, A control method for controlling a DC-DC converter, wherein all currents flowing through the smoothing inductance start flowing through a parasitic diode included in the first or second MOSFET. A method of controlling a resonant bidirectional DC-DC converter, characterized in that MOSFETs including a parasitic diode are simultaneously turned on. 4. The resonance type bidirectional DC-D according to claim 1.
If the bidirectional switch circuit of the C converter is composed of third and fourth MOSFETs in which source terminals or drain terminals are connected to each other, the D
A control method for controlling a C-DC converter, wherein the third and fourth MOSFETs are simultaneously turned on so that a resonance current flows through the resonance inductance, and the resonance current flowing through the resonance inductance becomes zero. Immediately before, one of the third and fourth MOSFETs in which a resonance current flows from the source terminal to the drain terminal
A method of controlling a resonance type bidirectional DC-DC converter, characterized in that the FET is cut off, and after the resonance current flowing through the resonance inductance becomes zero, the other MOSFET is cut off. 5. The resonance type bidirectional DC-D according to claim 1.
A bidirectional switch circuit of a C converter is first or second auxiliary switching element which is turned on or off by switching control and is connected in the opposite direction, and antiparallel to each of the first and second auxiliary switching elements. A first and a second direction control diode connected to, and a control method for controlling the DC-DC converter in the case of comprising: One of the first and second auxiliary switching elements is made conductive in accordance with the direction of the resonance current flowing in the resonance inductance and the conduction direction of the first and second direction control diodes, and the conduction is established. The resonance type dual switching device is characterized in that the auxiliary switching element is cut off after the resonance current flowing through the resonance inductance becomes zero. Toward DC-DC converter control method of.