JP2017508437A - Adaptive synchronous switching in resonant converters. - Google Patents

Abstract

One embodiment of a resonant converter includes a resonant circuit having an inductive element and a capacitive element configured to cause electrical resonance when an input voltage is applied, between at least a portion of the resonant circuit and the output of the resonant converter. And a synchronous rectifier connected to. The synchronous rectifier includes a diode and an electrical switch. The control circuit is configured to operate the electrical switch so that the electrical switch is on when there is substantially no voltage across the diode and the diode current flow is positive in the anode-to-cathode direction. .

Description

  [0001] Power electronics are widely used in various applications. Power adapters with power electronics are commonly used to convert forms of electrical energy, for example, from AC to DC conversion, from one voltage level to another voltage level, or in some other way. ing. Such devices can operate over a wide range of power levels, from milliwatt level mobile devices to hundreds of megawatt level high voltage transmission systems. Although power electronics conversion systems have made progress, in order to achieve high efficiency and increase the size, weight and complexity of power electronics equipment and its applications, it is necessary for advanced system architectures and methods of operation. Technology is needed.

  [0002] The present invention generally relates to power electronics converters. In particular, the present invention relates to a resonant converter and an adaptive control circuit. Techniques that may be used in the embodiments include (1) synchronous switching at the primary circuit of the resonance circuit, (2) synchronous switching at the output synchronous rectifier drive circuit, and (3) conditions from light load to heavy load. Resonant converter operation in “burst mode” to maintain zero voltage switching at (4), and (4) Minimize unnecessary energy clamps that can lead to increased component power loss and reduced power converter efficiency There are a variety of techniques including active voltage clamping to suppress

  [0003] One embodiment of a resonant converter according to the present disclosure includes a resonant circuit having an inductive element and a capacitive element configured to cause electrical resonance when an input voltage is applied, and resonant with at least a portion of the resonant circuit. A synchronous rectifier connected between the output of the converter. The synchronous rectifier includes a diode and an electrical switch. The control circuit is configured to operate the electrical switch so that the electrical switch is on when there is substantially no voltage across the diode and the diode current flow is positive in the anode-to-cathode direction. .

  [0004] One embodiment of a power conversion method according to the present disclosure provides a resonant converter comprising a resonant circuit having an inductive element and a capacitive element so as to cause electrical resonance when an input voltage is applied to the resonant circuit. Including that. The power conversion method further includes rectifying the output voltage of the resonant converter using a synchronous rectifier connected between at least a part of the resonant circuit and the output of the resonant converter. The synchronous rectifier includes a diode and an electrical switch. This power conversion method also operates the electrical switch so that the electrical switch is on when there is substantially no voltage across the diode and the current flow of the diode is positive in the direction from the anode to the cathode. Including that.

  [0005] Another embodiment of a resonant converter includes a resonant circuit having an inductive element and a capacitive element configured to cause electrical resonance when an input voltage is applied, a first synchronous rectifier and a second And a synchronous rectifier. The first synchronous rectifier and the second synchronous rectifier each include a diode and an electrical switch disposed in parallel with the diode. For each of the first synchronous rectifier and the second synchronous rectifier, when the diode current flow is positive in the anode-to-cathode direction, the first synchronous rectifier and the second synchronous rectifier are such that the electrical switch is turned on. A control circuit is configured to operate the rectifier.

  [0006] One embodiment of a non-isolated resonant converter according to the present disclosure includes a resonant circuit having an inductive element and a capacitive element configured to cause electrical resonance when an input voltage is applied. The non-insulated resonance converter includes a first electrical switch connected to the resonance circuit so as to flow a current of the resonance circuit, and a voltage that is connected to the resonance circuit and substantially no voltage is applied to the first electrical switch. A voltage monitor circuit configured to determine and a control circuit configured to receive the input from the voltage monitor circuit and operate the first electrical switch. The control circuit is configured to turn on the first electrical switch when it is detected that the voltage across the first electrical switch is substantially absent.

  [0007] One embodiment of a power conversion method according to the present disclosure provides a resonant converter comprising a resonant circuit having an inductive element and a capacitive element so as to cause electrical resonance when an input voltage is applied to the resonant circuit. Including that. This power conversion method uses a voltage monitor circuit to determine when there is substantially no voltage across the electrical switch connected to the resonant circuit, and detects that there is virtually no voltage across the electrical switch. And further comprising operating the electrical switch such that the electrical switch is turned on.

  [0008] One embodiment of a resonant converter according to the present disclosure is configured to receive an input voltage and includes an input stage comprising a first electrical switch connected in series with a primary winding of a transformer, and provides an output voltage And an output stage including a capacitive element connected to the secondary winding of the transformer so that the electrical resonance can occur when an input voltage is applied. The resonant converter is connected to the first electrical switch and is configured to determine when the voltage across the first electrical switch is substantially absent, and receives an input from the voltage monitor circuit, And a control circuit configured to operate one electrical switch. The control circuit is configured to turn on the first electrical switch when it is detected that the voltage across the first electrical switch is substantially absent.

  [0009] The present invention provides many advantages over the prior art. The method provided in this disclosure allows an AC to DC converter to operate efficiently while maintaining a desired output power level from light to heavy loads. Also, high efficiency is obtained, which reduces the thermal requirements of the power system and significantly increases the power density. Furthermore, the disclosed techniques can help maintain the integrity of the switching elements when operating at high voltages and / or high frequencies. The disclosed technique can be applied to both isolated and non-isolated resonant converters. These and other embodiments of the present invention, as well as the advantages and features of the present invention, are described in detail with reference to the following description and accompanying drawings.

1 is a schematic diagram illustrating a non-insulated resonant converter according to one embodiment. FIG. 2 is a waveform diagram of V _S , I _S1, and Drive 1 of the non-insulated resonant converter of FIG. 1. FIG. 6 is a schematic diagram illustrating types of feedback that can be used to notify a control circuit. FIG. 6 is a schematic diagram illustrating types of feedback that can be used to notify a control circuit. FIG. 5 shows a waveform for V _S and Drive 1 illustrating a burst mode according to one embodiment. FIG. FIG. 6 shows VS and Drive 1 waveforms showing burst mode according to one embodiment. FIG. And how the Drive 1 is to reduce the T _ON to reduce the output power, thereby indicating waveforms illustrating how the current may become insufficient to drive to return the V _S to zero. It is the schematic which shows the insulation type resonant converter by one Embodiment. FIG. 3 is a simplified diagram of a transformer that can cause leakage inductance, according to one embodiment. It is the schematic which shows embodiment of the output stage of an insulation type resonant converter. It is the schematic which shows embodiment of the output stage of an insulation type resonant converter. It is the schematic which shows embodiment of the output stage of an insulation type resonant converter. It is the schematic which shows embodiment of the output stage of an insulation type resonant converter. FIG. 2 is a schematic diagram illustrating an embodiment of a circuit for driving a synchronous rectifier such as that shown in the previous figures. FIG. 2 is a schematic diagram illustrating an embodiment of a circuit for driving a synchronous rectifier such as that shown in the previous figures. FIG. 2 is a schematic diagram illustrating an embodiment of a circuit for driving a synchronous rectifier such as that shown in the previous figures. It is a flowchart which shows embodiment of the power conversion method. It is a flowchart which shows embodiment of the power conversion method. According to one embodiment, it is a schematic diagram illustrating an exemplary control circuit for performing S ₁ control. It is the schematic which shows embodiment of an active clamp circuit. It is the schematic which shows embodiment of an active clamp circuit. It is the schematic which shows embodiment of an active clamp circuit. FIG. 6 is a schematic diagram illustrating the use of an active clamp in another embodiment.

  [0024] In the accompanying drawings, similar components and / or features may be given the same reference numerals. Also, various components of the same type may be distinguished by placing a dash or additional reference after the reference symbol to distinguish between similar components. If only the first reference number is used herein, the description can be applied to any similar component having the same first reference number, with or without additional reference numbers.

  [0025] The present invention relates generally to power electronics converters. In particular, the invention relates to adaptive resonant converters and corresponding control circuits. The disclosed embodiments adapt to internal and external changes while maintaining very high efficiency over a range from no load to heavy load, under various line parameters, load parameters, environmental parameters and component parameters. It has a function to operate in. Such power converters can be used in AC / DC power converters that supply power to any of a variety of electronic devices with very high power densities, such as laptop computers, USB power supplies, and the like. The techniques detailed herein can be applied to both isolated and non-isolated resonant converters.

[0026] In particular, in the embodiments disclosed below, four methods and techniques are applied. 1) A synchronous switching technique is used in the resonant circuit primary switch. By utilizing zero voltage switching, embodiments of the present invention can significantly reduce switching losses. 2) The synchronous switching technology is also used in the output synchronous rectifier drive circuit. The control circuit may monitor the voltage and / or current across a particular switch of the primary circuit and / or the secondary circuit to enable such efficient switching. 3) Embodiments may include operating the resonant converter in “burst mode” to maintain zero voltage switching from light to heavy load conditions. This function is automatically adjusted to compensate for line and load variations, as well as changes in environmental and / or component parameters. Further, the “burst mode” function may be programmable by the user to adapt to the application and maintain excellent efficiency. 4) An active voltage clamp circuit is used to minimize unnecessary energy clamps that result in increased component power loss and reduced power converter efficiency. This is particularly the case when the isolation transformer of the resonant converter is deliberately designed with high leakage inductance for integration and component reduction. The relationship between the primary resonance inductance (Lp) and the transformer leakage inductance (Llk) is expressed as follows when the transformation ratio Lp = Llk is 1: 1.

(Expression 1) Lp = (Llk.Np ² ) / Ns ² (1)

The leakage inductance is selected for maximum power transfer from the primary circuit to the secondary circuit.

  [0027] Advanced high density power electronics packaging may be applied to significantly reduce output stage loop inductance and switching losses, especially in high frequency resonant converters. Power switching wiring technology can have advantages in sensing, thermal management and EMI containment. An assembled structure may be used instead of the integrated structure produced by the integrated process flow.

  [0028] Also, some or all of the disclosed techniques may be applied to a power factor correction (PFC) or active rectifier circuit that supplies a resonant converter.

[0029] The circuits and techniques referred to above will now be described.
(Synchronous switching)

[0030] The control circuit may monitor the voltage and / or current across the circuit switch to allow synchronous zero voltage switching. There are many different ways to detect voltage or current. Embodiments are not specific to the particular way in which voltage or current sensing is achieved. For example, the current can be measured by a Hall effect sensor, a precision resistor or current transformer with or without an active circuit for isolation. Only the detection of the primary current may not be a good representation of the output resonant current for switching purposes. There is a significant phase shift between the primary and secondary currents that can vary with changes in load, temperature and component parameters.
In some cases, a single current sensor can be used to monitor resonant current and predict current flowing through two or more switches in isolated and non-insulated resonant circuits. Desired voltage and current feedback may be utilized for control to determine the best turn-on time for a particular switch.

  [0031] FIG. 1 is a schematic diagram illustrating a non-isolated resonant converter 100 according to one embodiment. The embodiment shown in FIG. 1 and other descriptions herein are given as non-limiting examples. Those skilled in the art will recognize various variations, modifications, and alternatives to the components presented herein.

[0032] In FIG. 1, the non-insulated type resonant converter 100, an AC may be a DC or rectified AC voltage source having an input voltage V _IN, the output voltage may be input voltage V _IN is greater than the voltage V _O Configured to supply. The operation of the non-isolated resonant converter 100 is partly due to the operation of Drive 1 that drives the electronic switch S ₁ to adjust the electronic switch S ₁ to meet the power requirements and to self-adapt the non-isolated resonant converter 100. It is determined. Inductive element _{L B} and _{L R} and the capacitive element _{C P} and _{C R} enables electrical resonance in non-isolated resonant converter 100. _CP is the lumped circuit capacity of node V, and any other electrically connected stray capacitance and L at that node, such as the parasitic capacity of semiconductor switch S ₁ , the capacity associated with L _B , _LR , etc. _May include circuit load of _R, CR network. Synchronous rectification of the output is given by the switches S _R and the diode D _B, the output power is supplied to the load resistor R _L. Hereinafter, synchronous rectification and Drive 2 will be described in more detail.

  [0033] Specialized components may be utilized so that the non-isolated resonant converter 100 can operate in high power applications at high switching frequencies. For example, in some embodiments, the transistors and diodes utilized in the switch can be devices based on high bandgap materials such as GaN or silicon carbide (SiC). As a result, the non-insulated resonant converter 100 can operate at a higher voltage, higher temperature, and higher frequency than when a conventional silicon-based device is used. However, any semiconductor device can be used.

  [0034] If an isolation transformer is used, functional magnet material and geometry may be used to improve the high frequency operation of the resonant circuit. If non-isolated and isolated topologies are utilized, advanced materials for other magnetic components may be used.

  [0035] The values of the various components utilized in the non-isolated resonant converter 100 may vary depending on the desired functionality, manufacturing considerations, and / or other personnel.

[0036] FIG. 2 is a diagram for explaining the operation of the non-isolated resonant converter 100, and shows the waveforms of V _S , I _S1 and Drive 1 of the non-isolated resonant converter 100 of FIG. It is. The Drive 1 waveform shows how Drive 1 switches to perform multiple on / off cycles with a switching period that includes a time T _{ON when the} switch is turned on and a time T _OFF when the switch is turned off. indicating capable of operating the S _1.

[0037] As shown, when the switch _{S 1} is on, the voltage at node _{V S} is reduced, the current _{I S1} initiates a rise, peak current I when the Drive 1 turns off the switch _{S 1 P} is reached. When the switch _{S 1} is turned _off, L B, the equivalent impedance of the resonance and _{L R} and _{C R} between _{C P} occurs. Current flows through the C _R, where, in some embodiments, may reach up to three times the voltage of the input voltage V _IN. However, after that, V _S drops to a voltage lower than V _IN due to the reverse direction of the current I _S1 . The resonant frequency is as follows:
(2)
Wherein, L and C are the Thevenin equivalent inductance and capacitance at node V _S.

[0038] wherein as the One advantage of the non-insulated resonant converter 100, the switching L _B of S _1, it is possible to zero-voltage switching of S ₁ in the voltage waveform caused by the resonance generated between L _R and C _P It is. That is, when the voltage applied to the S ₁ is at zero volts or near zero volts, that S ₁ is being switched, the switching loss of the capacitor C _P is greatly reduced. The voltage across S ₁ returns to zero voltage when I _S1 becomes sufficiently large and is zero voltage due to the action of parasitic or intentionally included S ₁ parallel diodes, or subsequent turn-on of S ₁ at the optimal time T1. To be kept.

[0039] In order to optimally turn on S ₁ at T1, the voltage V _S can be monitored directly or via a similar typical voltage waveform (eg, transformer winding). For example, a zero voltage detection circuit can be utilized such that when V _S approaches or reaches zero volts, the next T _ON transition of S ₁ is initiated.

[0040] efficiency increased from zero voltage switching of S ₁ node, minimal in each switching cycle, by the following equation, the energy of the capacitance C _P that should have disappeared at S ₁ if Dere originally It is obtained from restraining.
(3)

[0041] Turning on S ₁ immediately when the voltage is at or near zero volts reduces and / or eliminates conduction in diodes (parasitic or intentional) in parallel with S ₁ .

[0042] optimum turn-on of the S ₁ at zero voltage detector and T1 is resonant network (inductance, capacitance or resistance) due to changes in the dynamic value or initial value, with the shape of the resonance waveform changes This is also advantageous in that the optimum turn-on start time T1 also changes. However, the zero voltage detection circuit can adapt the switching waveform every switching cycle so that T1 is optimal, largely independent of other circuit variations.

[0043] Thus, an efficient power voltage from the input to the output (V _IN to V _O ) of the non-isolated resonant converter 100 and similar resonant converters allows S ₁ while maintaining resonant operation to obtain zero voltage switching. It is mainly controlled by controlling the on / off time. 3A and 3B are simplified schematic diagrams illustrating how feedback may be used in the control circuit driving S ₁ in various embodiments of the resonant converter.

[0044] FIG. 3A is a diagram illustrating providing feedback to a system around an output stage using voltage feedback. FIG. 3B is a diagram illustrating providing feedback to the system around the output stage using current feedback. However, as another configuration, a combination of both may be used. For example, in embodiments, output feedback (if P _O = V _O * I _O ) may be used to control the energy sent to the output. As shown, the feedback signal in some embodiments may be galvanically isolated (output from input) in an isolated controller scheme using a signal isolation circuit (such as a photocoupler or signal transformer).

The [0045] embodiment, as a method for modulating the S ₁ so as to achieve an output adjustment while maintaining zero voltage switching, a variety of different methods may be utilized. Such methods include so-called “burst modes”, such as three methods, frequency modulation, on-time (T _ON ) modulation and pulse density modulation. This technique includes a controlled burst mode that maintains zero voltage switching under varying internal and external conditions. Hereinafter, this feature will be described in more detail.
(Synchronous switching output stage)

  [0046] Synchronous switching may be utilized at the output stage of the resonant power converter to further reduce loss and increase efficiency. Synchronous rectification can provide advantages for topologies such as flyback. Examples that may be advantageous are: 1) the magnetizing current can be negative, discontinuous conduction mode is avoided, the output voltage is adjusted even under no-load conditions, and 2) zero voltage switching can be achieved. 3) The conduction loss of the rectifier may be significantly reduced especially at a constant voltage level.

  [0047] The following are examples of the case of the insulation type configuration and the case of the insulation type configuration.

  [0048] As shown in FIG. 1, the non-isolated resonant converter 100 is a simple embodiment of a converter that is not provided with isolation. However, as shown in FIG. 7 and subsequent figures, various modifications may be made to the simple design of FIG. 1, including the use of an isolated circuit. Good.

[0049] FIG. 7 is a schematic diagram illustrating an isolated resonant converter 700 according to one embodiment. In this embodiment, the transformer T ₁ is given insulation, may further result in a voltage change in accordance with the desired functionality. A dotted double-pointing arrow indicates another configuration. Thus, the secondary winding of the transformer T _1, as described in further detail below, for example, may be connected to one depending on the desired stage of operating a dielectric resonator converter 700. In some embodiments, L _B inductor of Figure 1 may be implemented by magnetizing inductance L _MAG of the transformer T _1, and / or may be replaced. Further, as illustrated, the resonant inductor _{L R} may be disposed in any of the input side _{(L RB)} or output side _{(L RA).} In other forms, some embodiments may include both of the above.

[0050] The value of the inductive element may vary depending on the input and / or output specifications of the converter. For example, if the output voltage is much lower than the input voltage, the value of the resonant inductance decreases in proportion to the square of the transformer turns ratio so that the inductance L _RA is much lower than L _RB. sell. Thus, for example, instead of the inductor which can lead to more inductance losses by providing a magnetic core in the position L _RB, loss is likely to use a very low air-core inductor.

[0051] Also, leakage inductance is added between the primary and secondary windings so that either or both of the input side (L _RB ) and output side (L _RA ) inductances can be included in the circuit. The leakage inductance increases with increasing physical separation between the primary and secondary windings.

  [0052] FIG. 8 is a simplified diagram of a transformer 800 that may provide such leakage inductance, according to one embodiment. According to this embodiment, instead of winding the primary and secondary windings around the same core leg (eg, one above the other), they are wound side by side on separate core legs to intentionally produce the desired amount of leakage inductance. You may make it introduce in. Here, a primary winding 820 is wound on one side of the magnetic core 810, and a secondary winding 830 is wound on the other side, and the magnetic core 810 transmits a magnetic flux 840 between these windings.

[0053] This method may have at least two significant advantages when applied to the topology of FIG. 8 (and other embodiments for providing isolation). First, the components L _RA and / or L _RB can be excluded as physical components. Furthermore, galvanic isolation between the primary winding 820 and the secondary winding 830 can be achieved very easily. This is the case when the winding is of a winding type or when the winding is embedded in a multilayer printed circuit board. Also, since the windings are not stacked one above the other, the number of winding layers is halved. This can greatly reduce the cost of the PCB in embodiments where a multilayer printed circuit board (PCB) is used to construct the windings. Further, even in an embodiment that does not use a PCB, winding of the winding is relatively easy and an insulating tape is not required for the transformer, so that the manufacturing cost can be reduced.

[0054] With further reference to the isolated resonant converter 700 of FIG. 7, the polarity of the secondary winding 830 of the transformer 800 may be in any direction, the direction of this polarity being L _R (L It may determine the _RA and / or _{L RB)} and _{C R} power transfer through the results _{S 1} phase.

  [0055] Special magnet materials and geometric shapes may be employed to improve the high frequency operation of the isolated resonant circuit. High performance materials may also be used for other magnetic components of the circuit.

[0056] In the insulation type resonant converter 700 in FIG. 7, the output diode D _O is output zero voltage reference instead of a positive voltage rail (i.e., the output GND) may be shifted to. That is, since the voltage reference of the driving signal Drive 2 may be zero volts, in addition to the diode rectifier D _O, or as an alternative, it is that is can be very easily with a semiconductor switch rectifier.

[0057] FIG. 9 illustrates one embodiment of an output stage 900 of an isolated resonant converter. This output stage can be, for example, a modification of the output stage of the isolated resonant converter 700 of FIG. Here, similarly to FIG. 7, the secondary winding of the transformer T _1, depending on the desired functionality, may be connected to either direction. In the output stage 900, D _O is located on the high side (positive voltage rail) in the same configuration as the non-insulated resonant converter. As described above, such a configuration is possible, but it may be more difficult to supply a drive waveform to the semiconductor switch.

  [0058] Conductivity of the switch can be reduced by paralleling the semiconductor switch in place of the diode position (not shown) or in parallel with the diode position (not shown), as known as a synchronous rectifier. Such synchronous rectifiers can be included in many applications where lower resistance is desired. Drive 2 will be described in detail below.

[0059] The position of the capacitor C _R 7 and 9, additionally or alternatively, may be connected in parallel across the position of the diode D _O. C _R is typically much smaller than the output capacitance, C _R forms an electrically equivalent circuit when it becomes a very large output capacitance and series. Therefore, the resonance circuit described in the above-described embodiment includes the parasitic capacitance of the diode D 2 _O.

[0060] FIG. 10 illustrates another embodiment of an output stage 1000 of an isolated resonant converter. Again, the secondary winding of the transformer T _1, depending on the desired functionality, may be connected to either direction. However, this output stage 1000 includes a variation in which L _R (which may be a leakage inductance, as described above) and C _R are connected in series rather than in parallel with the secondary side of the transformer.

[0061] The topology of output stage 1000 is such that the voltage of rectifiers D _O1 and D _O2 (and / or their semiconductor switch equivalents) is almost limited to the output voltage plus the switch voltage drop when conducting. It can be an advantage. In fact, for example, in the embodiment described above, V _S1 is 3 to 4 times lower than V _S. This can be an advantage because low rated voltage diodes and semiconductor switches can be used. These components are usually of low resistance and low conduction voltage drop, so that thermal suppression and high efficiency are achieved. For some applications, such as high output current applications, if these benefits are obtained, the increased complexity of the output stage 1000 may be worth it.

[0062] FIGS. 11 and 12 illustrate further embodiments of output stages 1100 and 1200, respectively. These configurations show modifications that are electrically equivalent to the output stage 1000 of FIG. FIG. 10 shows that S _R1 can be connected in series between the resonant circuit and the positive output rail of the resonant converter, whereas FIG. 11 shows that S _R1 is the negative of the resonant circuit and the resonant converter. It shows a state that it can be connected in series with the output rail. FIG. 12 shows that the output capacitor C _O can be split in two and the secondary winding of the transformer T ₁ can be coupled between the two new capacitors C _O1 and C _O2 .
(Control of synchronous rectifier)

  [0063] Synchronous switching can be used in the output side synchronous rectifier drive circuit. By utilizing zero voltage switching, embodiments of the present invention can significantly reduce switching losses. The control circuit may monitor the voltage and / or current across a particular switch of the primary circuit and / or the secondary circuit to enable such efficient switching.

[0064] For the output stage circuits shown in FIGS. 1, 7 and 9, the output stage resonant circuits are in parallel and there is a phase difference between the current waveform and the voltage waveform. Further, there is a phase difference also between S ₁ waveform and transformer waveform and the output rectifier (D _O) position. Therefore, S ₁ controller can not determine when to switch the synchronous rectifier (S _R).

[0065] If there are S _R parasitic or intended diode position, ideal situation to turn the switch S _R is the voltage of the diode is smallest (conducting state), the flow of current in the diode is positive Occurs when (from anode to cathode). S _R is turned on, the voltage is minimized, since it becomes difficult to determine when the low D _{O /} S _R voltage to be off the S _R, the only voltage information to operate the S _R is It may not be enough.

[0066] FIG. 13 is a schematic diagram showing a block-level solution to this problem. Here, the current transformer, the AND gate, an inverter amplifier and driver circuit, a voltage is minimized (e.g., near zero or zero), D O _/ S flow _R of the current is positive (the direction from the anode to the cathode) it is intended to prevent the S _R is turned on when it is. S _R, when the current flow is substantially zero, is turned off again. Here, the primary winding of the current transformer is connected to the current (eg, I _R , I _DB or I _DO ) of at least a portion of the resonant circuit, the inverter amplifier is connected to the node of the diode D _O , and the AND gate Is configured to perform a Boolean AND function using the output of the current transformer (eg, from the secondary winding) and the output of the inverter amplifier. This circuit can be used to provide a signal to Drive 2 of FIGS.

[0067] For the circuit shown in FIG. 12 from FIG. 10, the capacitor C _R, because they are connected to the D _O element in series, it may knowing when current flow in the rectifier is positive (cathode from the anode). Accordingly, the circuit of FIG. 14 shows a solution for determining (individually) the control drive of S _R1 and S _R2 .

[0068] In another embodiment, the current of C _R is detected since the position S _R1 and S _R2 can communicate in either the opposite phase of the two antiphase secondary winding, bidirectional single current transformer May be. An example of such a circuit is shown in FIG. In the figure, the current of the secondary resonance circuit is detected using the first secondary winding and the second secondary winding of the current transformer, and drives the switches S _R1 and S _R2 via the drive unit. To do. When the current is positive, one rectifier is turned on, and when the current is negative, the other rectifier is turned on. In other configurations, a single current transformer secondary winding (not shown) with positive and negative current detection may be utilized.

  [0069] FIG. 16 is a flow diagram illustrating a power conversion method according to one embodiment. Functionality may be provided in whole or in part by hardware and / or software that includes the circuits and other components described for FIGS. 1, 7, and 9-12.

  [0070] At block 1610, the resonant converter is provided with a resonant circuit having an inductive element and a capacitive element, causing electrical resonance when an input voltage is applied to the resonant circuit. The values of the inductive and capacitive elements can vary depending on the switching frequency, the desired functionality and / or other factors.

[0071] In block 1620, a voltage monitor circuit is used to determine when substantially no voltage is applied to the electrical switch connected to the resonant circuit. As shown in FIGS. 1-6, control of a switch (eg, switch S _{1 in} FIG. ₁ ) can be based on the voltage across the switch. Switching efficiency is optimized when the voltage is at or near zero. Accordingly, at block 1630, the electrical switch is actuated to turn on when it is detected that the electrical switch is substantially free of voltage.

  [0072] As an optional step, at block 1640, the electrical switch is operated in a mode that is turned off after the switch repeats the on / off cycle multiple times over a period of time. In “burst mode”, the resonant converter may be able to maintain output power while allowing zero voltage switching.

  [0073] FIG. 17 is a flow diagram illustrating a power conversion method according to another embodiment. Similar to FIG. 16, the functionality shown in FIG. 17 may be implemented in whole or in part in hardware and / or software including the circuits and other components described with respect to FIGS. 1, 7, and 9-12. Can be given.

  [0074] In block 1710, the resonant converter is provided with a resonant circuit having an inductive element and a capacitive element so that electrical resonance occurs when an input voltage is applied to the resonant circuit. Again, the value of the inductive element and the value of the capacitive element may vary depending on the switching frequency, the desired functionality and / or other factors.

[0075] At block 1720, the output voltage of the resonant converter is rectified using a synchronous rectifier comprising a diode and an electrical switch. Such commutation, for example, given in the above embodiment the switch S _R. As described above, the switching efficiency of the synchronous rectifier may be timed based on current as well as voltage. Thus, at block 1730, the electrical switch is actuated so that the electrical switch is turned on when substantially no voltage is applied to the diode and the current flow of the diode is positive in the direction from the anode to the cathode. Is done.

  [0076] It should be noted that the specific blocks shown in FIGS. 16 and 17 illustrate power conversion methods according to two specific embodiments. Other embodiments may include additional functionality and / or additional functionality. Embodiments may further include functionality not shown in FIGS. 16 and 17. Furthermore, steps may be added, removed and / or rearranged depending on the particular application. Those skilled in the art will recognize various variations, modifications, and alternatives.

  [0077] It should be noted that the examples and embodiments described for “zero voltage” switching may not operate the switch at a strictly zero voltage. Different tolerances of the components and materials used in the circuit can cause variations in, for example, the zero voltage detected by the zero voltage detector. However, such a detector is substantially zero voltage (ie substantially zero) if it is considered to be zero voltage for the purpose of use, even if voltage is present, if within tolerances. May be detected).

[0078] Also, the examples and embodiments described herein are for illustrative purposes only, and various modifications and changes made in consideration of these examples and embodiments will be presented to those skilled in the art. And is intended to be within the spirit and scope of the application and the appended claims.
(Control burst mode)

[0079] under the control includes a "burst mode", there may be a variety of different methods may be utilized embodiments for modulating the S ₁ in order to achieve the output adjustment while maintaining zero voltage switching. Hereinafter, an outline of such a modulation technique will be described.

[0080] For (frequency modulation) frequency modulation, as the frequency becomes higher, the value of I _P becomes low. In other words, the switching frequency can be used to adjust the output power. However, zero voltage detectors (which use zero voltage detection methods), which conflict with frequency modulation, are not commonly used in frequency adjustment. This is because the use of a zero voltage detector can lead to changes in T _OFF time (duty cycle) and consequently switching period (and hence frequency).

[0081] (Maximum T _ON modulation) In response to the maximum T _ON modulation, S ₁ may be modulated such that T _ON has a maximum value at a time proportional to 1 / V _IN . That is, the higher the input voltage _{V IN,} the length of the _{T ON} is shortened. Thus, the maximum power transfer in the circuit, since the I _P is closely related to T _ON, even when V _IN is varied becomes relatively stable. This may be the case for the maximum power voltage (maximum output load), but in order to adjust the output (voltage, current or power) to a lower output load / light output load, further _TON adjustment is required. It may be necessary. In the following, further details regarding optimal _TON modulation will be described.

[0082] (Control Burst _{Mode) T ON} maximum time it is sometimes preferred to be proportional to 1 _{/ V IN} also. However, where the switch S ₁ is driven to be switched burst interval not continuous,. In this way, the transmitted average power is reduced.

[0083] FIGS. 4 and 5 show the waveform of V _{S and} the waveform of Drive 1 showing the burst mode according to one embodiment. FIG. 4 shows a series (or “burst”) on / off cycle waveform for S ₁ . The output power can be maintained and / or adjusted by adjusting the burst frequency. FIG. 5 shows an example of how bursts are continuously applied to maintain a certain output power. Further, as described above, T _ON may be adjusted to maintain a constant output power and zero voltage switching in.

[0084] As shown in FIG. 4, it is also possible to make the ON / OFF cycle of the on-time T _ON in the burst mode is gradually lengthened. Have granted B1, B2, B3 and B4 in _{T ON} period such that the length is gradually increased. With the progress of Drive 1 from B1 to B4, by increasing the length of T _ON, resonant network, without causing overshoot, it is gradually possible to establish the resonance in each burst . If such does not use progressive modulation, the initial resonance peak of V _S following the B1, B2 and later is very high, there is a possibility that overshooting occurs, it can lead to adversely affect the situation in the switch S ₁ .

[0085] It should be noted that the waveforms of FIGS. 4 and 5 are shown for illustrative purposes only. In practice, the various features of the illustrated waveform, such as the number of switching periods in a single burst period, the magnitude of V _S and Drive 1, the duty cycle of each on / off cycle, can vary in configuration, power requirements and / or other Depending on the requirements, it can vary.

  [0086] The "burst mode" function can also be made user programmable to easily adapt to a particular application and maintain high efficiency. If zero voltage is not achieved in the resonant circuit, burst mode starting may be useful. This is especially true when the converter operates to minimize power loss at light loads to achieve high efficiency. Without a zero volt detection mechanism, the switch can be damaged at light loads when switching at very high frequencies.

According to [0087] some embodiments, the switch _S 1 is, MOSFET, can be a GaN transistor, such as MESFET. In such embodiments, the switch S ₁ may be modulated at a frequency higher than a similar silicon-based switches. By reducing the size of the magnetic component and the capacitive component by high-frequency switching, it is possible to reduce the overall size and cost of the power adapter. For example, in some embodiments, the order of the switching frequency is in megahertz, while the order of the burst frequency can be in the tens of kilohertz.

[0088] (T _ON modulation) As a variation of the burst mode described above, T _ON can be controlled for lower output loads to achieve the required set point and adjustment (although some In embodiments, _TON modulation may be used with other modulation techniques). _TON modulation works well in the output power range for most applications, but the smaller the output load, the shorter the _TON time. As the _TON time decreases, the current _IP can decrease. Furthermore, it may be inadequate as an energy back to zero volts V _S is a circular energy in resonant network. FIG. 6 is a diagram useful with such a dilemma to illustrate a solution that can be implemented according to some embodiments.

[0089] FIG. 6 shows how Drive 1 reduces T _ON to reduce output power using T _ON modulation, which causes current I _S1 (not shown) to return V _S to zero. In this way, a waveform showing how the current may be insufficient for driving is shown. Here, it recognizes when it is no longer happen that zero voltage switching, the likelihood of damage to the efficiency losses and a switch S ₁ which may occur by switching happens when V _S is not near zero or zero circuit prevention A zero voltage detection circuit can also be used to do this.

[0090] In the illustrated embodiment, a zero voltage detection signal may be monitored to determine when there is an inability state where V _S cannot return to zero. For example, if a zero volt detection signal is not received during multiple switching cycles, when the drive 1 is disabled for a period of time and the drive 1 is re-enabled, the circuit requests higher instantaneous power to Adjust the output to the required level. When power demand increases, T _ON time of Drive 1 that is long, during the next several cycles, it is possible to zero-voltage switching again.

  [0091] That is, when an inability to zero voltage switching of Vs is detected, the burst mode can be periodically initialized.

  [0092] The zero voltage detection method enables adaptive control that enhances burst mode operation and eliminates the possibility of switch damage while maintaining converter efficiency at all loads, particularly light loads.

[0093] FIG. 18, according to one embodiment, illustrates an exemplary control circuit for providing the S ₁ control. A 555 timer is used in stable mode to generate the resonant frequency of the circuit. R _A values, _{R B} values and C values to determine this frequency. After the 555 timer output frequency and zero volt detect signal are provided to the OR gate, the OR gate will trigger the timer output. The zero volt detection signal is necessary to set the timer output to a high level. The output of the 555 timer is supplied to the drive unit 1. Also, the output voltage is detected and supplied to a voltage adjustment comparator. Thus, the 555 timer is reset if the output voltage is higher than the reference or other protection signal is activated at a low level.
(Control active clamping)

  [0094] Peak resonant voltage is kept at a predetermined level to force zero volt switching to reduce the possibility of stress voltage damage to the switch and to eliminate unnecessary clipping that leads to excess loss and converter inefficiency As such, a controlled active clamping technique can be used. In an isolated converter, if the transformer peak reset voltage is significantly greater than the input voltage, the clamp circuit is activated at a predetermined peak voltage. Modulation can reduce the excess loss of the clamp circuit under varying load conditions. Modulation of the peak voltage allows for efficient power voltage and controllable output voltage adjustment.

  [0095] Common snubber and clamp circuits, such as resistor-capacitor-diode (RCD) circuits, are used in the switches to limit voltage spikes and reduce component stress. Thereby, further circuit heat dissipation can be obtained, so that labor saving can be realized. In such a circuit, when the switch is turned off and the current flow in the primary winding is suddenly stopped, a voltage spike is caused by the energy stored in the leakage inductance of the transformer of the isolated resonant circuit. The first step in reducing both voltage spikes and clamp losses is to design a transformer with minimum leakage inductance, which may not be ideal for a resonant converter. The resonance between the inductance and the parasitic capacitance of the switch causes a loss along with a large voltage stress, thus reducing the conversion efficiency. Although the clamp resistance can be high to further reduce the loss, doing so also increases the voltage spike. During the remainder of the switching cycle, the affected output voltage is applied to the clamp resistor, causing further losses. Using a higher voltage switch gives margin to the voltage spike and allows a larger resistor. However, when the rated voltage is large, the on-resistance is increased and the efficiency at a high load is lowered. When the controller is operating in burst mode, the clamp circuit discharges between ON states. If the clamp capacitor is too large, excess energy is stored and released during the OFF state. In some circumstances, the clamp capacitor may not discharge completely before the next ON state begins.

  [0096] In embodiments, an active clamping technique may be utilized rather than an RCD clamp circuit. The lossless LC and clamp switch circuit may, for example, cause transformer leakage inductance energy to oscillate at the input as reactive power and / or transfer energy to the load as active power. In either case, energy is not released at the resistor and losses are reduced. As an advantage to the active clamp circuit, it is possible to transmit energy under wide wiring and load fluctuation. Such a technique is suitable for a resonant circuit including a power factor correction (PFC) circuit. An active clamp circuit consisting of a switch and a capacitor handles the leakage inductance of the transformer to achieve transformer reset. The active clamp circuit acts as a controllable current source to adjust the power in response to load variations.

  [0097] Such an arrangement provides many advantages. For example, a duty cycle higher than 50% results in a high turns ratio, a low primary current, a low secondary voltage and a small output inductor. Also, the primary switch voltage stress remains relatively constant over the entire input voltage range, thus improving overall efficiency. Furthermore, since this method allows zero volt switching, the size can be further reduced by increasing the switching frequency.

  [0098] FIGS. 19-21 are schematic diagrams of examples of active clamp circuits that may be used in an isolated converter embodiment. The various components involved, configuration changes, and other change values can vary depending on the desired functionality, and such variations will be understood by those skilled in the art. . As shown, the circuits of FIGS. 19 and 20 utilize a comparator and driver to determine when the active clamp is on, as described above, depending on the application and desired functionality. Depending, it can occur at any of a variety of desired voltages (eg, 500V, 800V, etc.). On the other hand, the circuit of FIG. 20 shows how the clamp switch can be supplied by the windings of the transformer with the clamp switch at node 2 and node 3. Thus, the turn-on voltage of the clamp switch can be determined by the number of windings between the node 2 and the node 3. Thus, the circuit of FIG. 21 illustrates how active clamping can be performed with passive components.

  [0099] FIG. 22 illustrates a technique applied to a PFC circuit similar to the non-isolated circuit of FIG. In low power, low current applications, it may be more appropriate to use a small magnetizing inductance to achieve zero volt switching. The resonance between the leakage inductance and the clamp capacitance occurs when the transformer is reset. In order to provide zero volt switching at high input voltages and keep the transformer size and losses small, the magnetizing inductance is designed with the switching frequency.

[0100] If the peak reset voltage of the transformer is significantly greater than the input voltage, the clamp circuit can be operated at a predetermined peak voltage. Modulation can reduce the excess loss of the clamp circuit under varying load conditions. Modulation of the peak voltage allows for efficient power voltage and controllable output voltage adjustment. By providing the ability to control the clamping of the peak voltage level, the clamp switch is possible with zero-voltage switching of S _1. Clamp circuit and burst mode control provide zero volt switching under varying load conditions, especially under light loads.

[0101] the voltage across the S ₁ may be detected using techniques similar to the method shown in FIGS. 19 to 21. Other methods are also applicable to voltage detection. The detection signal is compared with a reference voltage by a comparator in a clamp circuit. At a predetermined OFF state voltage (ie, threshold voltage), the clamp switch is turned on and excess resonance energy is returned to _VBUS . Such circuit adaptability makes it possible to compensate for variations in load and environment to achieve high efficiency.

  [0102] While various embodiments of the present invention have been described above, the examples and embodiments described herein are for illustrative purposes only and various modifications made in light of such descriptions. Such changes and modifications are suggested to those skilled in the art and are within the spirit and scope of the present application and the appended claims.

Claims (45)

A resonant circuit having an inductive element and a capacitive element configured to cause electrical resonance when an input voltage is applied;
A synchronous rectifier comprising a diode and an electrical switch and connected between at least a portion of the resonant circuit and the output of the resonant converter;
Configured to operate the electrical switch to turn on the electrical switch when the voltage across the diode is substantially absent and the current flow of the diode is positive in the direction from the anode to the cathode. And a resonant converter.   The resonant converter of claim 1, wherein the electrical switch comprises a GaN transistor. The control circuit includes:
A current transformer connected to the current of at least a portion of the resonant circuit;
An inverter amplifier connected to the diode;
The resonant converter of claim 1, further comprising an AND gate configured to perform a Boolean AND function using the output of the current transformer and the output of the inverter amplifier.   The resonant converter according to claim 1, wherein the resonant converter is a non-insulated resonant converter. The resonant converter is an isolated resonant converter having a transformer that connects an input stage and an output stage,
The resonant converter of claim 1, wherein the output stage comprises the synchronous rectifier.   The output of the resonant converter has a positive rail and a negative rail, and the synchronous rectifier is connected in series between the resonant circuit and the positive rail of the output of the resonant converter. Resonant converter.   The output of the resonant converter has a positive rail and a negative rail, and the synchronous rectifier is connected in series between the resonant circuit and the negative rail of the output of the resonant converter. Resonant converter. Providing a resonant converter comprising a resonant circuit having an inductive element and a capacitive element so as to cause electrical resonance when an input voltage is applied to the resonant circuit;
Rectifying the output voltage of the resonant converter using a synchronous rectifier connected between at least a portion of the resonant circuit and the output of the resonant converter and comprising a diode and an electrical switch;
Power comprising operating the electrical switch to turn on when the voltage across the diode is substantially absent and the current flow of the diode is positive in the anode-to-cathode direction. Conversion method.   The power conversion method according to claim 8, wherein the electrical switch includes a GaN transistor. Operating the electrical switch
Using a current transformer to monitor the current of at least a portion of the resonant circuit;
Invert the voltage at the node of the diode,
9. The power conversion method of claim 8, comprising using a control circuit configured to utilize an AND that performs a Boolean AND function using the output of the current transformer and the inverted voltage.   The power conversion method according to claim 8, further comprising providing insulation to the resonant converter between the input voltage and the output voltage. A resonant circuit having an inductive element and a capacitive element configured to cause electrical resonance when an input voltage is applied;
A first synchronous rectifier and a second synchronous rectifier each comprising a diode and an electrical switch arranged in parallel with the diode;
For each of the first synchronous rectifier and the second synchronous rectifier, the first synchronous rectifier is such that the electrical switch is turned on when the current flow of the diode is positive in the direction from the anode to the cathode. And a control circuit configured to operate the second synchronous rectifier.   The resonant converter of claim 12, wherein the electrical switch of one or both of the first synchronous rectifier and the second synchronous rectifier comprises a GaN transistor. The control circuit includes:
A primary winding configured to flow current of at least a portion of the resonant circuit;
A first secondary winding configured to drive the electrical switch of the first synchronous rectifier;
A current transformer having a polarity opposite to that of the first secondary winding and having a second secondary winding configured to drive the electrical switch of the second synchronous rectifier The resonant converter of claim 12, further comprising: The first secondary winding is configured to drive the electrical switch of the first synchronous rectifier through a first drive;
The resonant converter of claim 14, wherein the second secondary winding is configured to drive the electrical switch of the second synchronous rectifier via a second drive.   The resonant converter of claim 14, wherein the second secondary winding is further connected to an output of the resonant converter.   The output of the resonant converter has a positive rail and a negative rail, and the first synchronous rectifier is connected in series between the resonant circuit and the negative rail of the output of the resonant converter. 13. A resonant converter according to item 12.   The output of the resonant converter has a positive rail and a negative rail, and the first synchronous rectifier is connected in series between the resonant circuit and the positive rail of the output of the resonant converter. 13. A resonant converter according to item 12. A resonant circuit having an inductive element and a capacitive element configured to cause electrical resonance when an input voltage is applied;
A first electrical switch connected to the resonant circuit to pass a current of the resonant circuit;
A voltage monitor circuit connected to the resonant circuit and configured to determine when a voltage across the first electrical switch is substantially absent;
A control circuit configured to receive an input from the voltage monitor circuit and to operate the first electrical switch;
A non-isolated resonant converter configured to turn on the first electrical switch when the control circuit detects that a voltage across the first electrical switch is substantially absent.   The non-insulated resonant converter of claim 19, wherein the first electrical switch comprises a GaN transistor.   The control circuit is further configured to operate the first electrical switch in a mode in which the first electrical switch is turned off after a plurality of on / off cycles over a period of time. 19. A non-insulated resonant converter according to 19.   The control circuit is configured so that, for each on / off cycle of the plurality of on / off cycles, the time during which the first electrical switch is turned on gradually increases with each successive on / off cycle. The non-isolated resonant converter of claim 21 further configured to operate the first electrical switch.   The non-isolated resonant converter of claim 21, wherein the control circuit is further configured to operate the mode periodically to maintain a certain output power.   The non-isolated resonant converter according to claim 19, wherein the control circuit includes a modulation circuit.   25. The non-isolated resonant converter of claim 24, wherein the modulation circuit is programmable.   The non-isolated resonant converter of claim 19, wherein the control circuit is further configured to receive voltage feedback from an output of the non-isolated resonant converter.   The non-isolated resonant converter of claim 19, wherein the control circuit is further configured to receive current feedback from an output of the non-isolated resonant converter. Further comprising a synchronous rectifier between the resonant circuit and the output of the non-isolated resonant converter, the synchronous rectifier,
A diode,
A second electrical switch arranged in parallel with the diode;
When the voltage across the diode is detected substantially absent and the current flow of the diode is positive in the anode-to-cathode direction, the second electrical switch is turned on so that the second electrical switch is turned on. 20. A non-insulated resonant converter according to claim 19, comprising a switching circuit configured to operate the electrical switch.   30. The switching circuit of claim 28, wherein the switching circuit is further configured to operate the second electrical switch such that the second electrical switch is turned off when the current flow is substantially zero. The non-insulated resonant converter described. Providing a resonant converter comprising a resonant circuit having an inductive element and a capacitive element so as to cause electrical resonance when an input voltage is applied to the resonant circuit;
Using a voltage monitor circuit to determine when there is substantially no voltage across an electrical switch connected to the resonant circuit;
A power conversion method comprising operating the electrical switch such that the electrical switch is turned on when it is detected that substantially no voltage is applied to the electrical switch.   31. The power conversion method according to claim 30, further comprising operating the electrical switch in a mode in which the electrical switch is turned off after passing through a plurality of on / off cycles over a certain period of time.   Operating the electrical switch so that the time for which the electrical switch is turned on gradually increases with each successive on / off cycle for each on / off cycle of the plurality of on / off cycles. The power conversion method according to claim 31, further comprising:   Further comprising operating a synchronous rectifier having a second electrical switch connected to the resonant circuit and the output of the resonant converter and connected in parallel to a diode, wherein operating the synchronous rectifier is applied to the diode Actuating the second electrical switch such that the second electrical switch is turned on when it is detected that substantially no voltage is present and the current flow of the diode is positive from the anode to the cathode The power conversion method according to claim 30, further comprising:   34. The power conversion of claim 33, further comprising operating the second electrical switch such that the second electrical switch is turned off when it is detected that the current flow is zero. Method. An input stage comprising a first electrical switch configured to receive an input voltage and connected in series with the primary winding of the transformer;
An output stage configured to supply an output voltage, comprising a capacitive element connected to a secondary winding of the transformer so that the electrical resonance can occur when the input voltage is applied;
A voltage monitor circuit connected to the first electrical switch and configured to determine when a voltage across the first electrical switch is substantially absent;
A control circuit configured to receive input from the voltage monitor circuit and operate the first electrical switch;
A resonant converter configured to turn on the first electrical switch when the control circuit detects that a voltage across the first electrical switch is substantially absent.   36. The resonant converter of claim 35, wherein one or both of the input stage and the output stage includes an inductive element configured to provide the electrical resonance with the capacitive element.   36. The resonant converter of claim 35, wherein the control circuit is further configured to operate the first electrical switch in a mode that is turned off after a plurality of on / off cycles over a period of time.   The control circuit is configured so that, for each on / off cycle of the plurality of on / off cycles, the time during which the first electrical switch is turned on gradually increases with each successive on / off cycle. 38. The resonant converter of claim 37, further configured to operate the first electrical switch. The output stage further comprises a synchronous rectifier connected to an output node of the resonant converter, the synchronous rectifier comprising:
A diode,
A second electrical switch arranged in parallel with the diode;
When the voltage across the diode is substantially absent and the current flow of the diode is positive in the anode-to-cathode direction, the second electrical switch is turned on so that the second electrical switch is on. 36. The resonant converter of claim 35, comprising a switching circuit configured to operate. The output stage further comprises a first synchronous rectifier and a second synchronous rectifier, wherein the first synchronous rectifier and the second synchronous rectifier are respectively
A diode,
An electrical switch arranged in parallel with the diode;
The resonant converter is such that, for each of the first synchronous rectifier and the second synchronous rectifier, the electrical switch is turned on when the current flow of the diode is positive in the direction from the anode to the cathode. 36. The resonant converter of claim 35, further comprising a switching circuit configured to operate each of the electrical switches of the first synchronous rectifier and the second synchronous rectifier.   36. The resonant converter of claim 35, further comprising a clamp circuit that controls a voltage across the first electrical switch.   42. The resonant converter of claim 41, wherein the clamp circuit comprises an active clamp circuit that turns on the clamp switch when the voltage across the first electrical switch reaches a threshold voltage. The clamp circuit is
A clamp capacitor;
An electrical clamp switch connected in series with the clamp capacitor;
A sensor configured to measure a voltage across the first electrical switch;
A comparison circuit connected to the output of the sensor and configured to compare a voltage across the first electrical switch with a reference voltage;
42. The resonant converter of claim 41, comprising: a drive unit connected to the output of the comparison circuit and configured to turn on the electrical clamp switch.   36. The resonant converter of claim 35, wherein the control circuit further comprises a modulation circuit.   45. The resonant converter of claim 44, wherein the modulation circuit is programmable.