JP2005151779A - High-frequency power supply unit - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a high-frequency power supply unit for supplying high-frequency to a load capable of protecting the load by meeting abrupt changes in the load impedance. <P>SOLUTION: A luminescent and wide-gap bipolar semiconductor control element, radiating light with intensity proportional to the energized current, is used for the switching element of an inverter generating high-frequency power to detect the radiated light. The increase in the energized current, when load impedance is decreased, is detected by an increase in the intensity of light, and a capacitor in a filter circuit is discharged by the detected signal, high-frequency power is made to decrease at a high speed. <P>COPYRIGHT: (C)2005,JPO&NCIPI

Description

  The present invention relates to a high frequency power supply apparatus that supplies high frequency power to a plasma processing apparatus or the like.

  A high-frequency power supply that supplies high-frequency power to a plasma processing apparatus or the like includes a DC power supply configured to control a DC output, and a DC output obtained from the DC power supply using an inverter to generate a high-frequency AC of a desired frequency. And a direct current / high frequency alternating current converter for converting the output. The direct current output of the direct current power supply unit is controlled so that the traveling wave power supplied from the direct current / high frequency alternating current conversion unit to the load is maintained at a set value.

  FIG. 5 is a circuit diagram showing a configuration of a conventional high-frequency power supply device 1 disclosed in Japanese Patent Laid-Open No. 2003-143861. In the figure, the high frequency power supply device 1 includes a DC power supply unit 1A, a DC / high frequency AC conversion unit 1B, a power detection unit PD, and a control unit CU including a microprocessor.

  The DC power supply unit 1A rectifies and smoothes the three-phase input of the commercial power supply AC, the inverter INV1 that converts the DC voltage obtained from the rectifying and smoothing circuit RS1 into an AC voltage, and the output of the inverter INV1 And rectifying / smoothing circuit RS2 for smoothing.

  The inverter INV1 has a bridge circuit of four semiconductor switch elements Su1, Sv1, Sx1, and Sy1. The control gates of the semiconductor switch elements Su1, Sv1, Sx1, and Sy1 are connected to the control unit CU by a wiring NS1 that collectively represents a plurality of wirings, and are controlled by an output signal of the control unit CU. . The rectifying / smoothing circuit RS2 includes a rectifier REC that rectifies the output of the inverter INV1, and a smoothing circuit including a reactor L and a smoothing capacitor C.

  In the inverter INV1, the control unit CU alternately generates a period in which the pair of switch elements Su1 and Sy1 at the diagonal positions are simultaneously turned on and a period in which the other pair of switch elements Sv1 and Sx1 are simultaneously in the on state. The direct current output of the rectifying / smoothing circuit RS1 is converted into alternating current. The rectifying / smoothing circuit RS2 rectifies and smoothes the AC output of the inverter INV1, and generates a DC voltage with a ripple component attenuated at both ends of the smoothing capacitor C.

  The inverter INV1 has a function of adjusting the output of the DC power supply unit 1A. The output frequency of the inverter INV1 is appropriately determined in consideration of the frequency characteristics of the switch element, the DC / AC conversion efficiency, and the like regardless of the output frequency of the high-frequency power supply device 1.

  As the switch elements Su1, Sv1, Sx1, and Sy1 constituting the inverter INV1, switch elements capable of on / off control such as FETs, bipolar transistors, and IGBTs are used. In the illustrated example, a MOSFET is used.

  The direct current / high frequency alternating current conversion unit 1B includes an inverter INV2 that converts a direct current voltage supplied from the direct current power supply unit 1A into a high frequency alternating voltage, and a filter circuit F.

  Similarly to the inverter INV1, the inverter INV2 has a bridge circuit of semiconductor switch elements Su2, Sv2, Sx2, and Sy2 that can be controlled on and off. The control gates of the semiconductor switch elements Su2, Sv2, Sx2, and Sy2 are connected to the control unit CU by a wiring NS3 that represents a plurality of wirings as one. In the semiconductor switch elements Su2 to Sy2, the pair of switch elements Sv2 and Sx2 in the other diagonal positions are simultaneously turned on during the period in which the pair of switch elements Su2 and Sy2 in the diagonal positions are simultaneously turned on by the control unit CU. The direct current voltage supplied from the direct current power supply unit 1A is converted into a high frequency alternating current voltage by being controlled so that the period for entering the state occurs alternately.

  The inverter INV2 is controlled by the control unit CU so that the output frequency of the high frequency power supply device 1 becomes a desired frequency. For example, when the output frequency of the high-frequency power supply device 1 is 1 MHz, the control unit CU outputs an AC voltage of 1 MHz from the inverter INV2. The switch elements Su2 to Sy2 of the inverter INV2 are on / off controlled in a state where the specific space Ton / Toff (or the duty ratio Ton / (Ton + Toff)) between the on period Ton and the off period Toff is kept constant.

  As the semiconductor switch elements Su2, Sv2, Sx2, and Sy2 constituting the inverter INV2, semiconductor switch elements that can perform both the switch operation and the operation in the active region are used. In the example shown in the figure, these switch elements are MOSFETs, but other switch elements such as bipolar transistors and IGBTs may be used.

  In each semiconductor switch element constituting each of the inverters INV1 and INV2, well-known feedback diodes are normally connected in parallel in the reverse direction to the forward direction of each semiconductor switch element. Illustration of the feedback diode is omitted.

  The filter circuit F includes an inductor and a capacitor, and attenuates harmonic components included in the output of the inverter INV2 to output a sinusoidal high-frequency AC voltage applied to the load 2.

  In the rectifying / smoothing circuit RS2 of the conventional example of FIG. 5, a series connection body of a current limiting resistor R having a sufficiently small resistance value and a capacitor discharging switch Sd capable of on / off control is provided in parallel at both ends of the smoothing capacitor C. The capacitor discharge circuit 1C is configured. The current limiting resistor R is provided to limit the discharge current of the capacitor C that flows through the switch Sd when the capacitor discharge switch Sd is turned on to the current capacity of the switch Sd or less. It is desirable to set the resistance value of the resistor R as small as possible within an allowable range.

  The capacitor discharge switch Sd is configured by a single semiconductor switch capable of on / off control, such as an FET, a bipolar transistor, or an IGBT, or a switch circuit configured by appropriately combining these semiconductor switch elements. In the illustrated example, the capacitor discharge switch Sd is composed of a single FET whose source is connected to the output terminal on the negative polarity side of the rectifying and smoothing circuit RS2, and its drain is connected to the positive electrode of the rectifying and smoothing circuit RS2 through the resistor R. It is connected to the output terminal on the sex side.

  The power detector PD is connected between the filter circuit F and the output terminal 1a. The traveling wave power supplied from the high frequency power supply device 1 to the load and the reflected wave reflected by the load and returned to the high frequency power supply device 1 side. Detect current. The power detector PD has a potential transformer (PT) for detecting voltage and a current transformer (CT) for detecting current, and a traveling wave power detection signal PF corresponding to the magnitude of the detected traveling wave current. And a reflected wave power detection signal PR corresponding to the magnitude of the detected reflected wave power is supplied to the control unit CU. The control unit CU having a microprocessor operates by a predetermined program (software) and performs various controls. FIG. 6 is a block diagram showing the operation of the control unit CU performed by the predetermined program, and blocks indicated as “means” such as the steady-state output control means 10 actually represent functions that operate by software.

  The control unit CU is provided with a traveling wave power setting signal PS that gives a setting value of the traveling wave power Pf supplied to the load 2 and a setting value of the reflected wave power Pr that is reflected by the load 2 and returns to the high frequency power supply device 1 side. The reflected wave power setting signal PRS is input. The control unit CU further provides a set time signal TS that gives a set value (set time) of a time for continuing the control operation for suppressing the output of the high frequency power supply device 1 when a sudden decrease in the impedance of the load 2 is detected; A suppression operation number setting signal NS that gives a set value of the number of times that the control operation for suppressing the output of the DC power supply unit 1A and the control operation for suppressing the output of the inverter INV2 are performed is input.

The control unit CU is a pair of switches that are in the on period of the inverter INV1 so that the DC power supply unit 1A outputs the DC power necessary to give a desired traveling wave power to the load 2 from the DC / high frequency AC conversion unit 1B. At least one of the elements is turned on / off at a predetermined duty ratio, and the output voltage of the inverter INV1 is PWM-controlled. At this time, the control unit CU compares the traveling wave power detection signal PF detected by the power detection unit PD with the traveling wave power setting signal PS, and calculates the deviation between them. The control unit CU controls the ratio Ton / Toff or the duty ratio Ton / (Ton + Toff) of the ON time Ton and the OFF time Toff of the switch element to be PWM controlled so that the calculated deviation becomes zero.
The control unit CU also controls at least one of the pair of switch elements in the ON period of the inverter INV2 by keeping the ratio Ton / Toff or the duty ratio Ton / (Ton + Toff) between the on time Ton and the off time Toff constant. To do.
In the program executed by the microprocessor built in the control unit CU, the traveling wave power detection signal PF and the traveling wave power setting signal PS are input as shown in FIG. A steady-state output control means 10 is also included for controlling the traveling wave power applied from the high frequency power supply device 1 to the load 2 so as to keep the set value set by the traveling wave power setting signal PS.

  The control unit CU also performs signal processing for detecting a change in the electrical change amount such as the impedance of the load 2 and a transient change in the direction of decreasing the impedance of the load 2 from the change in the electrical change amount. Load impedance change detecting means 11 for performing signal processing for detection is provided.

  In general, the high frequency power supply device 1 and the load 2 are connected by a coaxial cable having a characteristic impedance of 50Ω. When the impedance of the load 2 is 50Ω (pure resistance), all of the high frequency power (traveling wave power) output from the high frequency power supply device 1 is consumed by the load 2, reflected by the load 2, and reflected back to the high frequency power supply device 1. Wave power is not generated.

  However, if the impedance of the load 2 deviates from 50Ω (pure resistance), not all of the high frequency power generated by the high frequency power supply device 1 is consumed by the load 2. The power that has not been consumed is reflected by the load 2 and returned to the high frequency power supply device 1 as reflected wave power. Therefore, the most suitable load for the high frequency power supply device 1 is a pure resistance load having an impedance of 50Ω.

  In the conventional high frequency power supply device 1 shown in FIG. 5, the impedance of the load 2 changes in a decreasing direction while the traveling wave power supplied to the load is controlled to be maintained at a desired value, and the traveling wave power advances. When it is detected that the set value is larger than the set value by the wave power setting signal PS, the control unit CU controls the switch element of the inverter INV1 so that the duty ratio thereof is reduced, thereby reducing the output from the inverter INV1. Thereby, the traveling wave power supplied to the load can be returned to the set value.

  In the high frequency power supply device 1 shown in FIG. 5, for some reason, the impedance of the load 2 viewed from the output terminal 1a of the high frequency power supply device 1 instantaneously decreases from 50Ω (pure resistance) to about 5Ω (pure resistance), for example. Then, due to the control delay of the control unit CU, the current flowing through the power detection unit PD becomes slightly longer than when the current is 50Ω (pure resistance). Meanwhile, the traveling wave power detected by the power detection unit PD becomes larger than the set value given by the setting signal PS. The amount of increase in traveling wave power in this case (deviation between traveling wave power and set power PS) is determined by the circuit configuration and circuit constants of the filter circuit F.

  In particular, when the load 2 is a plasma processing apparatus such as plasma CVD, the impedance of the load 2 may suddenly decrease when abnormal discharge occurs in the plasma processing container. At that time, the traveling wave power output from the high-frequency power supply device 1 in a transient state when the impedance of the load 2 rapidly decreases may be significantly larger than the set power PS, and the object to be processed in the plasma processing container is damaged. There is a fear.

  Therefore, when the impedance viewed from the output terminal 1a of the high-frequency power supply device 1 to the load 2 side suddenly decreases from, for example, 50Ω (pure resistance) to 5Ω (pure resistance), the control unit CU lowers the DC voltage supplied from INV1 to the inverter INV2. Thus, the inverter INV1 is controlled. At this time, the control unit CU reduces the ratio of the on-time Ton to the off-time Toff of the switch elements Su1, Sy1, Sv1, and Sx1 in order to reduce the voltage charged in the smoothing capacitor C of the rectifying and smoothing circuit RS2. The switch elements Su1, Sy1 and Sv1, Sx1 are controlled.

  When the inverter INV1 is controlled as described above, the output voltage applied from the inverter INV1 to the capacitor C decreases. However, even if the output voltage of the inverter INV1 decreases, the electric charge charged in the smoothing capacitor C of the rectifying and smoothing circuit RS2 is discharged from the capacitor C through the circuit on the load side, and the terminal voltage decreases the traveling wave power. Until the value decreases, the output of the DC power supply unit 1A does not decrease. If the inverter INV1 is not provided with a discharge circuit for discharging the capacitor C, it takes a considerable time to discharge the capacitor C. The discharge time constant of the smoothing capacitor C is determined by the capacitance of the capacitor C and the impedance of the capacitor C viewed from the load 2 side. The time required for the voltage across the capacitor C to drop to a value that reduces the traveling wave power is affected by the load impedance and the capacitance value of the capacitor C, but generally takes several milliseconds. During this period, traveling wave power higher than the set power PS is output from the high frequency power supply device 1, and the load 2 may be adversely affected, or in the worst case, the load 2 may be damaged.

  Therefore, in the conventional high frequency power supply device 1 in FIG. 5, when the impedance of the load 2 undergoes a transient change in the decreasing direction, the smoothing capacitor C is discharged to set the output voltage of the DC power supply unit 1A. A capacitor discharge circuit 1C is provided as DC power supply output suppression means for reducing the voltage below the voltage. When the impedance of the load 2 suddenly decreases, the capacitor C is discharged by the capacitor discharge circuit 1C to forcibly reduce the output of the DC power supply unit 1A. As a result, the output of the high-frequency power supply device 1 can be reduced, and when the impedance of the load 2 rapidly decreases, it is possible to prevent the traveling wave power supplied to the load 2 from rapidly increasing and adversely affecting the load.

  The capacitor discharge circuit 1C includes a current limiting resistor R having a sufficiently small resistance value, and a capacitor discharge switch Sd connected in series to the resistor R, and this series connection is parallel to the smoothing capacitor C. It is connected to the. When the capacitor discharge switch Sd is turned on, the charge of the smoothing capacitor C is discharged almost instantaneously, and the terminal voltage of the smoothing capacitor C drops below the set voltage. In the capacitor discharge circuit 1C, when the load impedance change detection means 11 shown in FIG. 6 detects a transient change in the impedance of the load 2, the discharge switch control means 12 turns on the capacitor discharge switch Sd of the capacitor discharge circuit 1C. Put it in a state. When no transient change is detected in the impedance of the load 2, the discharge switch control means 12 turns off the capacitor discharge switch Sd.

  The load impedance change detecting means 11 is configured such that when at least one of traveling wave power, traveling wave voltage, and traveling wave current supplied to the load 2 from the direct current / high frequency alternating current conversion unit 1B exceeds a set value, It is configured to detect a transitional change in a decreasing direction. Further, the load impedance change detecting means 11 reflects when the load 2 reflects at least one of the reflected wave power, the reflected wave voltage, and the reflected wave current that returns to the DC / high-frequency AC conversion unit side 1B when the load exceeds the set value. It is configured to detect that a transitional change in the direction of decreasing impedance has occurred. The load impedance change detecting means 11 is a reflected wave power Pr that is reflected by the load 2 and returns to the DC / high-frequency AC converter 1B side, and a traveling wave power detection signal Pf supplied from the DC / high-frequency AC converter 1B to the load 2 When the ratio Pr / Pf exceeds the set value, or the reflected wave voltage Vr is reflected by the load and returns to the DC / high-frequency AC current conversion unit 1B side, toward the load 2 from the DC / high-frequency AC current conversion unit 1B side When the ratio Vr / Vf to the traveling wave voltage Vf exceeds the set value, or the reflected wave current Ir that is reflected by the load and returns to the DC / high-frequency AC conversion unit 1B and the traveling wave toward the load from the DC / high-frequency AC conversion unit When the ratio Ir / If with the current If exceeds a set value, it may be configured to detect that the impedance of the load 2 changes transiently in the decreasing direction.

  In the conventional high frequency power supply device 1 described above, the load impedance change detecting means 11 uses the traveling wave power detection signal PF and the reflected wave power detection signal PR as electrical variables, and these signals and the traveling wave power setting signal PS. , And the reflected wave power detection signal PRS, in one of the following four methods (a) to (d), it is configured to detect a transient change in the load impedance decreasing direction. Yes.

  (A) When the magnitude of the traveling wave power detection signal PF exceeds the magnitude of the traveling wave power setting signal PS, it is detected that a transient change in the direction of decreasing the load impedance has occurred.

  (B) When the magnitude of the traveling wave power detection signal PF exceeds the magnitude of the traveling wave power setting signal PS plus a constant value α, a transient change in the direction of decreasing the load impedance Detect that occurred.

  (C) When the magnitude of the reflected wave power detection signal PR exceeds the magnitude of the reflected wave power setting signal PRS, it is detected that a transitional change in the decreasing direction of the load impedance has occurred.

  (D) From the traveling wave power detection signal PF and the reflected wave power detection signal PR, the power ratio Pr / Pf between the reflected wave power Pr and the traveling wave power Pf corresponding to the square of the reflection coefficient of the voltage is calculated, When it is detected that the power ratio Pr / Pf exceeds the set value, it is detected that a transient change in the direction of decreasing the impedance of the load 2 has occurred.

The microprocessor of the control unit CU performs control to turn on the capacitor discharge switch Sd when it detects that the impedance of the load is abrupt by at least one of the methods (a) to (d). At the same time, a command for switching the operation of the semiconductor switches Su2, Sv2, Sx2, Sy2 of the inverter INV2 from the switching operation to the operation in the active region is given to a driver (not shown) of the inverter INV2, thereby suppressing the output of the high frequency power supply device 1 To do. A transient change in the impedance of the load 2 is detected by at least one of the methods (A) to (D), and the capacitor discharge switch Sd is turned off when a predetermined set time has elapsed from the detected time. To do. At the same time, the operation of the switching elements Su2 to Sy2 of the inverter INV2 is returned to the switching operation, the suppression of the output of the high-frequency power supply device 1 is released, and the steady-state control of the traveling wave power by the steady-state output control means 10 is resumed. .
It is desirable that the set time is set to a time slightly longer than the time required until the transient change in the impedance of the load 2 is resolved and the impedance of the load 2 is stabilized.

  The discharge switch control means 12 of the control unit CU turns on the capacitor discharge switch Sd when a transient change in the impedance of the load 2 is detected, and the transient change in the impedance of the load 2 is detected. When the set time elapses, the capacitor discharge switch Sd is turned off. That is, the discharge switch control means 12 and the capacitor discharge circuit 1C constitute means for suppressing the output of the DC power supply unit 1A.

  The inverter output suppression means 13 of the control unit CU performs the switching operation of the semiconductor switch elements Su2 to Sy2 constituting the inverter INV2 when a transient change in the impedance of the load 2 is not detected. When a transient change in the impedance of the load 2 is detected, the semiconductor switch elements Su2 to Sy2 constituting the inverter INV2 are operated in the active region. As a result, the inverter INV2 is controlled to suppress the output.

As described above, in the conventional high frequency power supply device 1 of FIG. 5, when the capacitor discharge switch Sd is turned on when a transient change occurs in the impedance of the load 2, the charge of the smoothing capacitor C is discharged almost instantaneously. To do. Therefore, the output voltage of the DC power supply unit 1 </ b> A decreases to a low value in a short time, and the traveling wave power supplied from the high frequency power supply device 1 to the load 2 is forcibly suppressed. Further, since the current flowing through the switch elements Su2 to Sy2 of the inverter INV2 is suppressed by switching the operation of the switch elements Su2 to Sy2 of the inverter INV2 from the switching operation to the active operation, the progress given to the load 2 from the high frequency power supply device 1 Wave power is forcibly suppressed.
JP 2003-143861 A JP 2002-325355 A JP 2002-325427 A

  The power detection unit PD of the conventional high-frequency power supply device 1 uses PT to detect voltage, and uses CT to detect current. The direct current / high frequency alternating current conversion unit 1B of the high frequency power supply device 1 operates at a high frequency of, for example, about 1 MHz, but cannot follow a rapid change in output power with a PT or CT having a relatively low response frequency. Therefore, when the impedance of the load 2 suddenly decreases for some reason, the capacitor discharge switch Sd cannot be turned on at high speed to discharge the smoothing capacitor C at the output section of the DC power supply section 1A at high speed. When the discharge is delayed, the traveling wave power supplied to the load 2 increases rapidly, although for a short time. As a result, the load 2 is adversely affected, and in the worst case, the load 2 may be damaged.

  The present invention provides a high-frequency power supply device having a small, lightweight, high-speed and low-loss power semiconductor element circuit that eliminates the above-mentioned drawbacks of the conventional example and can detect a sudden change in the impedance of a load at high speed. Objective.

The high frequency power supply device of the present invention is a DC power source in which a smoothing capacitor is connected between positive and negative output terminals, a capacitor discharging switch unit connected in parallel to the capacitor, and an output of the DC power source is input, and the output A predetermined number of luminescent wide-gap semiconductor switching elements that emit radiated light in accordance with an energized current, and the radiated light is received into the energized current of the semiconductor switching element. An inverter unit having a light receiving unit for outputting a corresponding electric signal, and an electric signal output from the light receiving unit of the inverter unit, and the output of the inverter unit rapidly increases to increase an energization current of the semiconductor switching element When it is detected, the output of the DC power supply is reduced and the capacitor discharge switch unit is driven to To discharge the service charge, and a control unit that performs control for reducing the output of the inverter unit.
According to the present invention, the radiated light that changes according to the energization current of the switching element of the luminescent wide gap semiconductor is detected by the light receiving unit, the output of the DC power supply is controlled based on the detection signal of the light receiving unit, and the smoothing Discharge the capacitor. Therefore, the response time from when the change of the energization current occurs until the output of the high frequency power supply is controlled is short. Moreover, since it detects using light, it is hard to receive the influence of an electrical noise.

  The high-frequency power supply device of the present invention uses a light-emitting wide gap semiconductor element that emits light according to an energization current as a switching element of an inverter, and detects the radiated light with a light-receiving element, thereby greatly changing the output current of the inverter. It can be detected at high speed. An increase in the output of the inverter that occurs when a transient change in the direction of decrease occurs in the impedance of the load connected to the high-frequency power supply is detected at high speed, and the smoothing capacitor at the output end of the DC power supply section is almost temporally Discharge without delay to suppress the output of the DC power supply. This prevents excessive high frequency power from being supplied to the load when the load impedance changes suddenly in the decreasing direction, thereby protecting the load.

  In particular, when the load is a plasma processing apparatus, if an abnormal discharge occurs in the plasma processing container for some reason, the impedance of the load may rapidly decrease. According to the present invention, it is possible to prevent an excessively high frequency power from being applied to the load from the high frequency power supply device when the impedance of the load changes in a decreasing direction. It can be prevented that the electrode is damaged or the life of the electrode is shortened.

  In addition, when the load impedance changes transiently in the decreasing direction, the voltage applied to the inverter component of the high-frequency power supply apparatus and the component of the power detection unit and the current flowing through these components become excessive. Can be prevented. As a result, the voltage rating and current rating of these components can be lowered from the ratings of components used in conventional power supply devices, and the cost can be reduced.

  A preferred embodiment of the high frequency power supply device of the present invention will be described below. As described in detail below, a wide-gap semiconductor material such as silicon carbide (SiC), gallium nitride (GaN), or diamond is suitable for the light-emitting wide-gap semiconductor element used as the switching element of the high-frequency power supply device of the present invention. ing. Wide-gap semiconductor materials have superior physical properties such that they have a higher dielectric breakdown electric field and thermal conductivity than silicon (Si) semiconductor materials, and operate at temperatures as high as 200 ° C. or higher. A wide gap semiconductor element formed of a wide gap semiconductor material has a high withstand voltage and a low loss, and the loss generated in the semiconductor element is small. Further, since the generated heat can be easily dissipated and the current can be increased to a considerably high temperature, it is suitable for a semiconductor element of a power semiconductor element circuit for controlling a large current.

  In the present invention, in an element using a wide gap semiconductor material, a carrier recombination center (an electron and a hole are recombined and disappears in one of an n-type layer and a p-type layer forming a junction). A place where an impurity atom that promotes the process or a complex of a plurality of impurity atoms exists). A light-emitting wide gap semiconductor element can be obtained because light is emitted when an electric current is passed through the junction.

  When a light emitting wide gap bipolar semiconductor control element is formed using a wide gap semiconductor material and a light emission window is provided in a part of the element, light can be emitted therefrom. From the detection output obtained by detecting this light with the light receiving element, the value of the energization current of the element and its change can be detected. The intensity of the emitted light is approximately proportional to the current flowing through the bipolar semiconductor control element. When this radiated light is detected using a light receiving element having photoelectric conversion characteristics with good linearity, the detection output of the light receiving element is proportional to the current flowing through the bipolar semiconductor control element. When the drive circuit of the bipolar semiconductor control element is controlled by the detection output of the light receiving element, the energization current of the bipolar semiconductor control element can be controlled. If the number of recombination centers is too large, the on-voltage of the semiconductor element increases and the power loss increases because recombination of electrons and holes more than the number necessary to secure the desired emitted light occurs. . It is desirable to optimize the doping amount of impurity atoms or the like that form the recombination center or to localize the doped region in a part of the semiconductor layer so that the power loss does not become so large. For example, the doping is limited to the region of the semiconductor layer facing the light emitting portion.

As the light receiving element, a Si semiconductor light receiving element, a wide gap semiconductor light receiving element, a photoconductive element, or the like is used. The light receiving element is, for example, a substantially square flat plate having a side of several mm and a thickness of about 1 mm, and is provided in a package of a light emitting wide gap bipolar semiconductor control element. An optical bipolar semiconductor control element is made by combining a bipolar semiconductor control element and a light receiving element. In the optical bipolar semiconductor control element, the light receiving element and the bipolar semiconductor control element are electrically insulated, and the light receiving surface of the light receiving element faces the light emitting portion of the bipolar semiconductor control element. The increase in the volume of the package due to the incorporation of the light receiving element and the bipolar semiconductor control element in one package is about 3 cm ^{3 for} an element with a withstand voltage of 6 kV, and the increase in weight is about 100 grams. The total transmission efficiency of the optical bipolar semiconductor control element is the light emission efficiency of the bipolar semiconductor control element, the light collection efficiency indicating the degree to which the light is collected on the light receiving element, and the light collected on the light receiving element is converted into electricity. It is expressed by the product of the photoelectric conversion efficiency. Although the total transmission efficiency varies greatly depending on the semiconductor material used, it is about 0.005 to 2% in the one used in this embodiment. For example, when the total transmission efficiency is 0.1%, when the current flowing through the bipolar semiconductor control element is 50 A, the photocurrent generated in the light receiving element is about 0.05 A. Since a voltage of 20 V to 30 V is normally applied to the light receiving element, the power loss generated in the light receiving element is about 2 to 3 W. Since the photocurrent of the light receiving element changes in proportion to the current flowing through the bipolar semiconductor control element, it can be detected even when the current changes slowly.

As another preferred embodiment of the present invention, there is a method in which light of a bipolar semiconductor control element using a light emitting wide gap semiconductor material is guided to a light receiving element via an optical fiber and detected. For example, the optical fiber is attached to the package of the bipolar semiconductor control element so that the incident end thereof faces the light emitting portion of the bipolar semiconductor control element. A light receiving element is attached to the output end of the optical fiber. In this configuration, in addition to the effects described above, the distance between the bipolar semiconductor control element and the light receiving element can be increased, so that the withstand voltage between them can be easily increased.
"Example"

Hereinafter, a preferred embodiment of the high frequency power supply device of the present invention will be described with reference to FIGS.
FIG. 1 is a circuit diagram of a high-frequency power supply device 10 of this embodiment, and the configuration thereof will be described below. The high frequency power supply device 10 of the present invention includes a DC power supply unit 1A, a DC / high frequency AC conversion unit 10B, and a control unit CU including a microprocessor. The DC power supply unit 1A includes a rectifying / smoothing circuit RS1 that rectifies and smoothes a three-phase AC input from the commercial power supply AC, an inverter INV1 that converts a DC voltage obtained from the rectifying / smoothing circuit RS1 into an AC voltage, and an inverter A rectifying / smoothing circuit RS2 for rectifying and smoothing the output of INV1 is provided.
The inverter INV1 has a bridge circuit of semiconductor switch elements Su1, Sv1, Sx1, and Sy1, and each control gate of the semiconductor switch elements Su1, Sv1, Sx1, and Sy1 is a wiring that represents a plurality of wirings in one. NS1 is connected to the control unit CU and is controlled by an output signal of the control unit CU. The rectifying / smoothing circuit RS2 includes a rectifier REC that rectifies the output of the inverter INV1, a smoothing circuit including a reactor L and a smoothing capacitor C.

In the inverter INV1, the control unit CU alternately generates a period in which the pair of switch elements Su1 and Sy1 at the diagonal positions are simultaneously turned on and a period in which the other pair of switch elements Sv1 and Sx1 are simultaneously in the on state. The direct current output of the rectifying / smoothing circuit RS1 is converted into alternating current. The rectifying / smoothing circuit RS2 rectifies and smoothes the AC output of the inverter INV1, and generates a DC voltage with a ripple component attenuated at both ends of the smoothing capacitor C.
The output frequency of the inverter INV1 is appropriately determined in consideration of the frequency characteristics of the switch element, the DC / AC conversion efficiency, and the like regardless of the output frequency of the high-frequency power supply device 1.
As the switch elements Su1, Sv1, Sx1, and Sy1 constituting the inverter INV1, switch elements capable of on / off control such as FETs, bipolar transistors, and IGBTs can be used. In the illustrated example, a MOSFET is used.
The direct current / high frequency alternating current conversion unit 10B includes an inverter INV10 that converts a direct current input from the direct current power supply unit 1A into a high frequency output of a desired frequency, and a filter circuit F provided between the output terminal and the load 2. .

The inverter INV10 has a bridge circuit of gate turn-off thyristors (hereinafter abbreviated as GTO) Su3, Sv3, Sx3, Sy3, which is a kind of optical bipolar semiconductor control element. Each control gate of GTOSu3, Sv3, Sx3, and Sy3 is connected to the control unit CU by a wiring NS10 that represents a plurality of wirings in one. The control unit CU alternates between a period in which the pair of switch elements Su3 and Sy3 at the diagonal positions are simultaneously turned on and a period in which the pair of switch elements Sv3 and Sx3 at the other diagonal positions are simultaneously in the on state. The switch elements Su3, Sy3, Sv3, and Sx3 are controlled so that the DC input from the DC power supply unit 1A is converted into a high-frequency AC output.
The inverter INV10 is controlled so that its output frequency is equal to the desired output frequency of the high-frequency power supply device 10. For example, when the output frequency of the high frequency power supply device 10 is 1 MHz, the control unit CU controls the inverter INV10 to output an AC voltage of 1 MHz. The switch elements GTOSu3, Sy3, Sv3, and Sx3 of the inverter INV10 are on / off controlled in a state in which the ratio Ton / Toff (or the duty ratio Ton / (Ton + Toff)) between the on period Ton and the off period Toff is kept constant. .

Each semiconductor switch element constituting each of the inverters INV1 and INV10 is normally connected in parallel with a known feedback diode in the reverse direction to the forward direction of each semiconductor switch element, but these feedback diodes are not shown. Has been.
The filter circuit F includes an inductor and a capacitor, attenuates harmonic components included in the output of the inverter INV10, and outputs a sine wave high-frequency AC voltage to the load 2.
A series connection body of a current limiting resistor R having a sufficiently small resistance value and a capacitor discharging switch Sd capable of on / off control is connected in parallel to both ends of the smoothing capacitor C of the rectifying and smoothing circuit RS2. A capacitor discharge circuit 1C is configured. The current limiting resistor R is provided to limit the discharge current of the capacitor C that flows through the switch Sd when the capacitor discharge switch Sd is turned on to be equal to or lower than the current capacity of the switch Sd. It is desirable to set the resistance value of the resistor R as small as possible within the allowable range.

The switch Sd is configured by a semiconductor switch element capable of on / off control such as an FET, a bipolar transistor, or an IGBT, or a switch circuit configured by appropriately combining these semiconductor switch elements. In the illustrated example, the capacitor discharge switch Sd is an FET whose source is connected to the output terminal on the negative polarity side of the rectifying and smoothing circuit RS2, and its drain is output through the resistor R on the positive polarity side of the rectifying and smoothing circuit RS2. Connected to the terminal.
In FIG. 1, the inverter INV10 of the direct current / high frequency alternating current conversion unit 10B includes GTOSu3, Sv3, Sx3, and Sy3 as semiconductor control elements, and is connected to the load 2 via the filter circuit F. GTO Su3, Sv3, Sx3, and Sy3 are light-emitting wide gap bipolar semiconductor control elements using light-emitting wide gap semiconductor material SiC. GTO Su3, Sv3, Sx3, and Sy3 emit light indicated by arrows 101a, 101b, 101c, and 101d, each having an intensity substantially proportional to the current flowing therethrough. The radiated light indicated by the arrows 101a, 101b, 101c, and 101d is received by each light receiving element 102a provided in each package of GTO Su3, Sv3, Sx3, and Sy3, as will be described in detail later with reference to FIG. , 102b, 102c, and 102d. The light receiving elements 102a, 102b, 102c, and 102d are connected to DC power supplies 104a, 104b, 104c, and 104d through resistors 103a, 103b, 103c, and 103d connected in series, respectively. Voltage probes 105a, 105b, 105c, and 105d for measuring photocurrents flowing through the respective light receiving elements 102a, 102b, 102c, and 102d are connected to both ends of the resistors 103a, 103b, 103c, and 103d.

  Next, the operation of the high frequency power supply device of this embodiment will be described. GTO Su3, Sv3, Sx3, and Sy3 are, for example, SiC-GTO thyristors having a withstand voltage of 6 kV and a current capacity of 200A. The light receiving elements 102a, 102b, 102c, and 102d use silicon photodiodes. The radiated light of GTO Su3, Sv3, Sx3, and Sy3 during energization is detected by the respective light receiving elements 102a, 102b, 102c, and 102d, and the photocurrent is indicated by arrows 106a, 106b, 106c, and 106d, respectively. , 102b, 102c, 102d, resistors 103a, 103b, 103c, 103d and power supplies 104a, 104b, 104c, 104d. At this time, respective voltages generated at both ends of the resistors 103a, 103b, 103c, and 103d are measured by the voltage probes 105a, 105b, 105c, and 105d, and signals of the obtained measurement values are sent to the control unit CU.

  When the impedance of the load 2 connected to the high-frequency power supply device 10 changes transiently in a decreasing direction, a larger current than normal flows in GTO Su3, Sv3, Sx3, and Sy3, and the intensity of each radiated light increases. . As a result, the photocurrents flowing from the power supplies 104a, 104b, 104c, and 104d to the respective light receiving elements 102a, 102b, 102c, and 102d increase, and the voltages at both ends of the resistors 103a, 103b, 103c, and 103d also increase. The increase in voltage is detected by the voltage probes 105a, 105b, 105c, and 105d, and a detection signal is sent to the control unit CU. When receiving the detection signal indicating the increase in voltage, the control unit CU applies the control signal NS1 to the inverter INV1 to reduce the output of INV1. At the same time, the control signal NS2 is applied to the gate of the semiconductor switch element, which is the capacitor discharge switch Sd of the capacitor discharge circuit 1C, to turn it on. As a result, both ends of the smoothing capacitor C are short-circuited via the resistor R, and the charge of the smoothing capacitor C is discharged. As a result, the output of the DC power supply unit 1A is suppressed, so that excessive high-frequency power is prevented from being supplied from the inverter INV10 to the load 2, and the load 2 can be protected.

Normally known snubber circuits are connected between the cathodes and anodes of GTO Su3, Sv3, Sx3, and Sy3, but the snubber circuit is not shown in FIG. The snubber circuit is preferably a combination of resistors, capacitors, diodes, and the like.
FIG. 2 is a cross-sectional view of an optical GTO element 200 having a withstand voltage of 6 kV and a current capacity of 200 A in which a GTO 201 and a light receiving element 202 are housed in one package as an optical bipolar semiconductor control element. GTO Su3, Sv3, Sx3, and Sy3 in FIG. 1 use GTO 201, and light receiving elements 102a, 102b, 102c, and 102d use light receiving element 202. In FIG. 2, the anode electrode 231 of the GTO 201 is fixed to the central portion of the metal base 203 connected to the anode terminal 214A. Each semiconductor layer 37 described in detail with reference to FIG. 4 is provided on the anode electrode 231. A light receiving element 202 is attached to the inner surface of a metal cap 204 attached to the metal base 203 via an insulating plate 202D with the light receiving part 202B facing the light emission window 219 of the GTO 201. The metal base 203 has two holes 217 and 218, and the cathode terminal 213 A is led out from the hole 217, and the gate terminal 216 is led out from the hole 218. The anode terminal 214A, the gate terminal 216, and the cathode terminal 213A are all about 3 cm in length. The cap 204 has two holes 210 and 211, and the anode terminal 202 </ b> A of the light receiving element 202 is led out from the hole 210, and the cathode terminal 209 </ b> B is led out from the hole 211. The holes 210, 211, 217, and 218 are all hermetically sealed with a known hermetic sealing material 17A.

  In the package of the optical GTO element 200, the cathode electrode 220 of the GTO 201 is connected to the cathode terminal 213A by two conducting wires 214B and 214C. The gate electrode 216 </ b> A of the GTO 201 is connected to the gate terminal 216 by a conducting wire 215. The number of conducting wires 214B and 214C may be increased or decreased according to the current value. The anode electrode 207A of the light receiving element 202 is connected to the anode terminal 202A by a conducting wire 206, and the cathode electrode 209A is connected to the cathode terminal 209B by a conducting wire 228. The GTO 201 and the light receiving element 202 are electrically insulated by an insulating plate 202D. The distance between the light emission window 219 of the GTO 201 and the light receiving portion 202B of the light receiving element 202 is about 1 cm. Light is emitted from the light radiant heat 219 as indicated by an arrow 1A and received by the light receiving portion 202B of the light receiving element 202. The In the configuration of FIG. 2, an optical fiber 243 is provided between the light emission window 219 and the light receiving unit 202B, and the light emitted from the light emission window 219 is guided to the light receiving unit 202B. The light receiving element 202 may be arranged so that light emitted from the radiation window 219 is received by the light receiving unit 202B.

  For example, the light receiving element 202 of a silicon photodiode is a square having a side of 3 mm and a thickness of about 0.5 mm. As described above, since the light receiving element 202 of the silicon photodiode is small, the size of the optical GTO 200 is relatively small. When this optical GTO 200 is compared with a SiC-GTO thyristor having a withstand voltage of 6 kV and a current capacity of 200 A, the optical GTO 200 is approximately 100 grams heavier and has a volume that is several percent larger. In the GTO 201 shown in FIG. 2, since a light emission window 219 is provided by removing a part of the pad for attaching the conductor of the anode electrode 220, the light emission efficiency is relatively low. Further, since the light emission window 219 and the light receiving portion 202B of the light receiving element 202 are separated by about 1 cm, the light collection efficiency is low. Accordingly, when the energization current instantaneously increases from 200 A to 1000 A, for example, when abnormal, the photocurrent of the light receiving element 202 increases to about 120 mA, but when the applied voltage of the light receiving element 202 is, for example, 10 V, the power loss is about 1. It is 2W and can be said to be an extremely low value.

  The light-emitting wide gap bipolar semiconductor control element such as the optical GTO 200 used in this embodiment will be described in detail below. In a conventional bipolar semiconductor control element such as a GTO, a recombination center that causes recombination of carriers in a p-type or n-type semiconductor layer forming a junction is provided in order to reduce the on-voltage and reduce the loss. It is configured not to include as much as possible. That is, the recombination center is not included in each semiconductor layer as much as possible. On the other hand, in the light-emitting wide gap bipolar semiconductor control element used in the present invention, contrary to the conventional bipolar semiconductor control element, at least one of a plurality of semiconductor layers forming the bipolar semiconductor control element is provided. It is configured so that some recombination centers exist. The recombination center is obtained by doping at least one SiC semiconductor layer with aluminum and nitrogen atoms. In this case, light is generated by recombination of holes captured at the impurity level formed by aluminum atoms and electrons captured at the impurity level formed by nitrogen atoms. When the semiconductor layer is doped with a large number of aluminum atoms and nitrogen atoms to form a large number of recombination centers, the intensity of the emitted light increases. However, since recombination inhibits the flow of electrons and holes, the on-resistance of the bipolar semiconductor control element is increased, and thus the on-voltage is also increased. As a result, the power loss of the bipolar semiconductor control element also increases.

Therefore, it is necessary to set the intensity of the emitted light and the magnitude of the on-resistance to desirable values in consideration of practicality. In SiC, which is the material of the light-emitting wide gap bipolar semiconductor control element of this embodiment, it is desirable that the number of aluminum atoms and nitrogen atoms be in the range of 1 × 10 ¹⁵ to 1 × 10 ¹⁹ atoms / cm ^3. . In the case of SiC, aluminum serves as a p-type impurity and nitrogen serves as an n-type impurity. Therefore, when the semiconductor layer having a recombination center is p-type, it is necessary to dope aluminum more than nitrogen. For example, aluminum may be increased to about 1 × 10 ²¹ atoms / cm ³ . Further, when the semiconductor layer having a recombination center is n-type, it is necessary to dope nitrogen more than aluminum. For example, nitrogen may be increased to about 1 × 10 ²¹ atoms / cm ³ .

  A detailed structure of the GTO 201 using a light-emitting wide gap semiconductor used in this embodiment will be described with reference to FIGS. FIG. 3 is a plan view of the GTO 201 in FIG. 2, and FIG. 4 is a sectional view taken along IV-IV in FIG. A GTO 201 in FIG. 2 shows a II-II cross section in FIG.

3 and FIG. 4, GTO201 is a _{n E-type} layer 232 having a thickness of about 100μm was formed on the cathode electrode 231, to form a _{p B-type} layer 233 having a thickness of about 70μm thereon. An n _B type layer 234 having a recombination center and having a thickness of about 3 μm is formed on the p _B type layer 233. In the figure of the n _B type layer 234, gate electrodes 216A are formed at both ends. The n _B type layer 234 is doped with aluminum atoms at a concentration of 3.5 × 10 ¹⁷ atoms / cm ³ and nitrogen atoms at a concentration of 8 × 10 ¹⁷ atoms / cm ³ . The _{p E-type} layer 235 is formed in the central portion of the n _B-type layer 234, and an anode electrode 220 provided thereon. When a current density of, for example, 100 A / cm ² was applied between the cathode electrode 231 and the anode electrode 220 of the GTO 201 configured as described above, the ON voltage was a relatively low value of 5.2V. The intensity of the emitted light in this energized state was about 16 milliwatts (mW), and the wavelength of the emitted light was about 470 nanometers (nm). As shown in FIG. 3, the GTO 201 has a known termination region 237 for relaxing the electric field in its outer peripheral region. The periphery of the light emission window 219 is surrounded by the anode electrode 220 so that a sufficient current flows through the p- _E layer without the anode electrode 220 facing the light emission window.

  As mentioned above, although the Example of the high frequency power supply device of the present invention was described, the present invention covers many more application ranges or derivative structures. For example, instead of GTO using a light emitting wide gap semiconductor material, other light emitting wide gap semiconductor bipolar control elements such as IGBT, emitter switch thyristor, electrostatic induction thyristor may be used. Furthermore, instead of the GTO, a light-emitting wide gap semiconductor bipolar control element formed of a plurality of wide gap semiconductor layers stacked on a Si substrate may be used. The GTO or IGBT may be a hybrid element in which a unipolar element such as a MOSFET formed of Si or a wide gap semiconductor material and a light-emitting wide gap semiconductor bipolar control element formed of a wide gap semiconductor material are combined. For example, an element having a hybrid structure in which a Si-MOSFET is connected between an emitter and a collector of a bipolar transistor using a wide gap semiconductor material may be used. In addition to SiC, gallium nitride (GaN) may be used as the wide gap semiconductor material. The light receiving element may be a phototransistor other than a Si photodiode, a CdS photoconductive element, or a light receiving element using a wide gap semiconductor material such as a SiC photodiode.

  The high frequency power supply device of the present invention can detect a change in the output current of the inverter due to a change in the impedance of the load at a high speed and control the output to the load. It is useful as a high frequency power supply device.

The circuit diagram which shows the structure of the high frequency power supply device of the Example of this invention Sectional drawing of the package of the optical bipolar semiconductor element used for the inverter of the high frequency power supply device of this invention Plan view of a light-emitting wide-gap bipolar semiconductor control element that constitutes an optical bipolar semiconductor element IV-IV sectional view of FIG. Circuit diagram of conventional high-frequency power supply Block diagram showing control software of a control unit of a conventional high frequency power supply device

Explanation of symbols

DESCRIPTION OF SYMBOLS 1,10 High frequency power supply device 1A DC power supply part 1B, 10B DC / high frequency AC conversion part 1C Capacitor discharge circuit 2 Load C Smoothing capacitor CU Control part 200 Optical GTO
201 GTO
202 Photodetector 202A Anode terminal 209A Cathode electrode 209B Cathode terminal 213A Cathode terminal 214A Anode terminal 216 Gate terminal 220 Cathode electrode 243 Optical fiber INV1 Inverter INV10 Inverter RS2 Rectification smoothing circuit Su3, Sv3, Sx3, Sy3 GTO
102a, 102b, 102c, 102d Light receiving element

Claims (6)

DC power supply with a smoothing capacitor connected between the positive and negative output terminals,
A switch unit for discharging the capacitor connected in parallel to the capacitor;
A predetermined number of light-emitting wide-gap semiconductor switching elements that emit radiation corresponding to an energization current for converting the output of the DC power source into AC output of a predetermined frequency, and the radiation An inverter unit having a light receiving unit that receives light and outputs an electrical signal corresponding to an energization current of the switching element; and an electrical signal output from the light receiving unit of the inverter unit is received, and the output of the inverter unit increases rapidly And reducing the output of the DC power supply when driving the energizing current of the switching element is detected, driving the capacitor discharging switch unit to discharge the charge of the capacitor, and the inverter unit A high-frequency power supply device having a control unit that performs control to reduce the output of. The light receiving unit of the inverter unit is
A light receiving element that receives the emitted light of the switching element of the light emitting wide gap semiconductor;
The high frequency power supply device according to claim 1, further comprising: a DC power supply for light receiving element and a resistor connected in series to the light receiving element; and a voltage probe for detecting a voltage across the resistor. The said switching element and at least the said light receiving element of the said light-receiving part are accommodated in one package so that the emitted light of the said switching element may be light-received by the said light receiving element. High frequency power supply. The said switching element and at least the said light receiving element of the said light-receiving part are connected by the optical fiber so that the radiated light of the said switching element may be light-received by the said light receiving element. High frequency power supply. 2. The high frequency power supply device according to claim 1, wherein the capacitor discharge switch section is configured by a series connection of a semiconductor switching element and a resistor. 3. The high frequency power supply device according to claim 1, wherein the light emitting wide gap semiconductor switching element is a gate turn-off thyristor.

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