JP2005130666A - Inverter control method and polyphase current supply circuit - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To configure a robust control system by improving the follow-up properties of an input current to its command value. <P>SOLUTION: A multiplier 641b multiplies a first current command value i<SB>m</SB><SP>*</SP>by a modulation factor r to create a second current command value i<SB>T</SB><SP>*</SP>. A multiplier 642c multiplies the basic frequency amplitude i<SB>in1</SB>of an input current i<SB>in</SB>by a modulation factor r to create the absolute command value ¾i<SB>in</SB><SP>*</SP>¾ of the input current. Based on the difference between the absolute command value ¾i<SB>in</SB><SP>*</SP>¾ of the input current and the absolute value ¾i<SB>in</SB>¾ of the input current i<SB>in</SB>, an input current compensating section 642e creates a compensated current command value i<SB>comp</SB><SP>*</SP>. The compensated current command value i<SB>comp</SB><SP>*</SP>is added to a second current command value i<SB>T</SB><SP>*</SP>to create a driving current command value i<SB>dq</SB><SP>*</SP>. Based on the driving current command value i<SB>dq</SB><SP>*</SP>and a current phase command value β<SP>*</SP>, q-axis current command value i<SB>q</SB><SP>*</SP>and d-axis current command value i<SB>d</SB><SP>*</SP>are created. The modulation factor r is set at (¾sinθ<SB>in</SB>¾-k<SB>1</SB>)×k<SB>2</SB>, for example. <P>COPYRIGHT: (C)2005,JPO&NCIPI

Description

  The present invention relates to inverter technology.

  For example, Patent Documents 1 and 2 and Non-Patent Document 1 disclose techniques for obtaining a multiphase AC current from a single-phase AC power supply without significantly reducing the capacity of the smoothing capacitor and using a power factor improving reactor. In this technology, a DC voltage applied to an inverter that outputs a multi-phase AC current is obtained by pulsating load power in synchronization with the phase of an input voltage supplied from a single-phase AC power source. It pulsates greatly at a frequency almost twice that of.

  By such pulsation, the conduction width of the input current supplied from the single-phase AC power supply can be expanded, and the power factor is improved. On the other hand, even if the DC voltage applied to the inverter has such pulsation, a multiphase AC current can be output by appropriately controlling the switching in the inverter. Such switching control is referred to herein as single-phase capacitorless inverter control.

  In the single-phase capacitorless inverter control, the smoothing capacitor can be reduced in size and a reactor is not required, so that the entire circuit including the rectifier circuit and the inverter is reduced in size, resulting in cost reduction.

FIG. 9 is a circuit diagram illustrating a system in which single-phase capacitorless inverter control is employed. AC power supply 1 applies a voltage v _in a sinusoidal to the diode bridge 2. The diode bridge 2 receives the input current i _in from the AC power source 1 and performs full-wave rectification. The capacitor 31 is a component of the smoothing circuit 3 that receives the output of the diode bridge 2. The inverter 4 receives the output of the smoothing circuit 3 and outputs a multiphase alternating current group _iz to the motor 5.

The current output from the diode bridge 2 branches into a current i _dir that flows directly to the inverter 4 and a current i _cha that charges the capacitor 31. On the other hand, the current i _inv input to the inverter 4 is the sum of the above-described current i _dir flowing directly from the diode bridge 2 to the inverter 4 and the current i _dis discharged from the capacitor 31. The voltage across the capacitor 31 is represented as v _dc .

FIG. 10 is a graph showing an ideal state of the voltage and current of each part in the circuit diagram shown in FIG. The subscript 0 indicates that each voltage and current is in an ideal state. The ideal end-to-end voltage v of the capacitor 31 _dc0 has a waveform which is full-wave rectified input voltage v _in. At this time, the diode bridge 2 is always in a conductive state, and the ideal input current i _in0 has a sine wave _{shape in} phase with the input voltage _vin . Further, when the both-end voltage v _dc is the ideal both-end voltage v _dc0 , the charging current i _cha0 and the discharging current i _dis0 of the capacitor 31 alternately _in a period of ¼ of the cycle of the input voltage _vin. It has a flowing waveform. These charging current i _{CHA 0,} the discharge current i _DIS0 is sufficiently small compared to the inverter current i _inv i.e. the input current i _in, since the charging current i _{CHA 0} in the input current i _in, the discharge current i _DIS0 negligible, the input current i _in Is an ideal input current i _in0 . In other words, if an absolute value waveform of a sine wave is adopted as the absolute value command value | i _in ^* | of the input current i _in , _in an ideal case, the both-end voltage v _dc is the ideal both-end voltage v _dc0 next, flowing ideal input current i _in0 as the input current i _in. Thus the absolute value of the command value of the input current i _in | i _in ^* | By setting can suppress harmonics of the input current i _in.

JP 2002-51589 A JP 2002-354826 A Jin Haga, Kazuo Saito, Isao Takahashi “Electrolytic Capacitor-less High Power Factor Inverter Control Method for Single-Phase Diode Rectifier Circuit”, Proceedings of National Conference of the Institute of Electrical Engineers of Japan 4-069 (March 2003), page 99

However, if the load on the capacitor 31 is reduced, it is not always possible to obtain a desired result by setting the command value | i _in ^* | as described above. 11A to 11C are graphs showing each voltage and current, and show a case where the load becomes lighter in this order. The lighter the load, the smaller the current i _dir flowing directly from the diode bridge 2 to the inverter 4, and the absolute value of the input current i _in and the current input to the inverter 4 as shown in FIGS. i _inv approaches the ideal charging current i _cha0 and discharging current i _dis0 , respectively. Further, if the load becomes lighter, the capacitor 31 is not completely discharged, and the voltage v _dc between both ends does not become zero as shown in FIG.

When the load is lightened in this way, the conduction width of the input current i _in is narrowed. Despite this, the absolute value of the command value of the input current i _in | i _in ^* | By adopting the absolute value of the waveform of the sine wave as, the input current i _in accordance to the charging current i _cha and the discharge current i _dis There is a problem that the harmonics of the input current i _in are increased due to an increase in distortion and a failure of the control system.

Further, since the inverter 4 controls the interphase voltage of the motor 5 using the voltage v _dc across the capacitor 31, the phase voltage (effective value) v _m of the motor with respect to the neutral point is v _dc / 6 ^1/2. It is as follows. However, since the speed is the phase voltage v _m increase in the motor 5 increases, weakening becomes impossible to sufficiently reduced to near zero phase voltage v _m by field control, the voltage across v _dc of the phase voltage v _m 6 ^Can rise up to ^1/2 times. Such a rise in the both-end voltage v _dc narrows the conduction width of the input current i _in and causes the above-mentioned problems.

  An object of the present invention is to improve the followability of an input current with respect to its command value and to construct a control system that is less likely to fail. If this is applied to the above problem, an increase in harmonics of the input current due to a failure of the control system or the like can be suppressed.

An inverter control method according to the present invention includes a diode bridge (2) that performs full-wave rectification by inputting an input voltage (v _in ) and an input current (i _in ) having a predetermined fundamental frequency from an AC power supply (1), A smoothing circuit (3) having a capacitor (31) that receives the output of the diode bridge, and an inverter (4) that receives the output of the smoothing circuit and outputs a polyphase alternating current (i _u , i _v , i _w ). Is used to control the switching of the inverter when the multiphase drive unit (5) is driven.

The first aspect includes (a) generating a command value (i _dq ^* ) of the driving current of the multiphase driving unit based on the phase angle (θ _in ) of the input voltage; sine value of a predetermined value with respect to the drive current (-β ^*) (-sinβ ^*) and the predetermined value of the cosine value by multiplying the (cos .beta ^*) and, respectively, d-axis current command value (i _d ^*) and q-axis A step of generating a current command value (i _q ^* ), and (c) a command value signal (T _u , T _v , _V) for switching operation of the inverter based on the d-axis current command value and the q-axis current command value. Generating T _w ). Here, the step (a) includes (a-1) a first current command value (i _m ^* ) based on a difference between the speed (ω _m ) of the drive unit and the command value (ω _m ^* ) of the speed. And (a-2) obtaining a non-negative modulation coefficient (r) that takes a value of zero in the first period and monotonously decreases after a monotonic increase in the second period, (a-3) A step of obtaining a second current command value (i _T ^* ) by modulating the first current command value using a modulation coefficient; and (a-4) the drive current based on the second current command value. Generating the command value. However, the first period and the second period are repeated with a period that is half the period of the input voltage.

A second aspect of the inverter control method according to the present invention is the first aspect of the inverter control method, wherein in the step (a-2), the phase angle (θ _in ) of the input voltage (v _in ) ) Is subtracted from a positive first constant (k ₁ ) smaller than 1 from the absolute value of the sine value (| sinθ _in |), and if the result is non-negative, the result is multiplied by a positive second constant (k ₂ ). If the result is negative, zero is adopted to obtain the modulation coefficient (r).

A third aspect of the inverter control method according to the present invention is the second aspect of the inverter control method, wherein the second constant (k ₂ ) is the modulation coefficient (r) and the input voltage (v _in ). The average value of the products of the phase angle (θ _in ) and the absolute value (| sin θ _in |) of the sine value is set to 0.5.

A fourth aspect of the inverter control method according to the present invention is the first to third aspects of the inverter control method, and in step (a-4), the amplitude (i _in1 ) of the fundamental wave of the input current. the absolute value of the command value of the input current obtained as the product of the modulation factor (r) and (| i _in ^* |) and the absolute value of the input current (| i _in |) compensation based on the difference between A result obtained by adding a current command value (i _comp ^* ) to the second current command value (i _T ^* ) is defined as the command value (i _dq ^* ) of the drive current.

According to a fifth aspect of the inverter control method of the present invention, (a) a command value (i _dq ^* ) of the drive current of the multiphase drive unit based on the phase angle (θ _in ) of the input voltage (v _in ). generating a, (b) sine value of a predetermined value with respect to the drive current (-β ^*) (-sinβ ^*) and the cosine value of the predetermined value (cos .beta ^*) by multiplying the respective d-axis current Generating a command value ( _id ^* ) and a q-axis current command value ( _iq ^* ); and (c) switching operation of the inverter based on the d-axis current command value and the q-axis current command value. Generating a command value signal (T _u , T _v , T _w ). The step (a) includes (a-1) a current command value (i _m ^* , i _T ^* ) based on a difference between the speed (ω _m ) of the drive unit and the command value (ω _m ^* ) of the speed. (A-2) A command value (| i _in ^*) of the absolute value of the input current that takes a value of zero in the first period, decreases monotonically after a monotonic increase in the second period, and is non-negative |) And (a-3) a compensation current command value (i _comp ^* ) based on a difference between the command value of the absolute value of the input current and the absolute value (| i _in |) of the input current ^. ) To the command value (i _dq ^* ) of the drive current as a result of adding the current command value to the current command value. However, the first period and the second period are repeated with a half period of the period of the input voltage.

A sixth aspect of the inverter control method according to the present invention is the fifth aspect of the inverter control method, wherein the step (a-2) includes: (a-2-1) the input voltage (v _in ) A positive first constant (k ₁ ) smaller than ₁ is subtracted from the absolute value (| sin θ _in |) of the sine value of the phase angle (θ _in ), and if the result is non-negative, the result is a positive second constant. Multiplying (k ₂ ) and adopting zero if the result is negative to obtain a modulation coefficient (r); (a-2-2) amplitude of the fundamental wave of the input current (i _in1 And calculating the command value (| i _in ^* |) of the absolute value of the input current as the product of the modulation coefficient (r).

A seventh aspect of the inverter control method according to the present invention is the sixth aspect of the inverter control method, wherein the step (a-1) is (a-1-1) the speed (ω _m ) of the drive unit. Calculating a first current command value (i _m ^* ) by performing a proportional / integral operation on the difference between the speed command value (ω _m ^* ) and (a-1-2) the modulation coefficient (r ) To obtain a second current command value (i _T ^* ) obtained by modulating the first current command value as the current command value.

An eighth aspect of the inverter control method according to the present invention is the seventh aspect of the inverter control method, wherein the second constant (k ₂ ) is the modulation coefficient (r) and the input voltage (v _in ). The average value of the product of the absolute value (| sin θ _in |) of the sine value of the phase angle (θ _in ) is 0.5.

A multi-phase current supply circuit according to the present invention is a diode bridge (2) that performs full-wave rectification by inputting an input voltage (v _in ) and an input current (i _in ) having a predetermined fundamental frequency from an AC power supply (1). And a smoothing circuit (3) having a capacitor (31) for receiving the output of the diode bridge, and receiving the output of the smoothing circuit and supplying multiphase alternating currents (i _u , i _v , i _w ) An inverter (4) for outputting to (5) and a control circuit (6) for controlling switching of the inverter are provided.

In the first aspect, the control circuit generates a command value (i _dq ^* ) of the drive current of the multiphase drive unit based on the phase angle (θ _in ) of the input voltage (v _in ), by multiplying a predetermined value sine of (-β ^*) (-sinβ ^*) and the cosine value of the predetermined value (cos .beta ^*) to said drive current, respectively d-axis current command value (i _d ^*) and q A command value current calculation unit (64) for generating a shaft current command value (i _q ^* ), and a command value signal (T for switching operation of the inverter based on the d-axis current command value and the q-axis current command value) command value signal generating means (60) for generating _u , T _v , T _w ). Further, the command value current calculation unit obtains a first current command value (i _m ^* ) generated based on a difference between the speed (ω _m ) of the drive unit and the command value (ω _m ^* ) of the speed. A current command value modulation unit (641) that obtains a second current command value (i _T ^* ) by modulation using the modulation coefficient (r), and the drive current based on the second current command value Drive current command value generation means (642) for generating a command value. Here, the modulation coefficient is non-negative, takes a value of zero in the first period, and decreases monotonically after increasing monotonously in the second period. However, the first period and the second period are repeated with a period that is half the period of the input voltage.

A second aspect of the multiphase current supply circuit according to the present invention is the first aspect of the multiphase current supply circuit, in which the current command value modulation section (641) is configured to adjust the phase of the input voltage (v _in ). The first positive constant (k ₁ ) less than ₁ is subtracted from the absolute value of the sine value of the angle (| sinθ _in |), and if the result is non-negative, the result is multiplied by a positive second constant (k ₂ ). Te, also employs a zero if the result is negative, the determining the modulation coefficient (r) modulation coefficient generating unit and (641a), the modulation factor and the first current command value (i _m ^*) And a multiplier (641b) for outputting the product of the above as the second current command value (i _T ^* ).

A third aspect of the multiphase current supply circuit according to the present invention is the second aspect of the multiphase current supply circuit, in which the modulation coefficient generator (641a) sets the second constant (k ₂ ) A memory (6415) is provided that stores an average value of the product of the modulation coefficient (r) and the absolute value (| sinθ _in |) of the sine value of the phase angle of the input voltage to be 0.5. .

A fourth aspect of the multiphase current supply circuit according to the present invention is the first to third aspects of the multiphase current supply circuit, wherein the drive current command value generating means (642) is configured to generate a fundamental wave of the input current. An average value (i _in1 ) for calculating the amplitude (i _in1 ) and a command value (| i of the absolute value of the input current as a product of the amplitude (i _in1 ) of the fundamental wave of the input current and the modulation coefficient (r) _in ^* |) and a compensation current command value (i _comp ^* ) based on the difference between the command value of the absolute value of the input current and the absolute value of the input current (| i _in |) ^. ) In addition to the second current command value (i _T ^* ), an adder (642f) for obtaining the command value (i _dq ^* ) of the drive current.

According to a fifth aspect of the multiphase current supply circuit of the present invention, the control circuit controls the command value of the drive current of the multiphase drive section (θ _in ) based on the phase angle (θ _in ) of the input voltage (v _in ). i _dq ^* ) is generated, and the drive current is multiplied by a sine value (−sin β ^* ) of a predetermined value (−β ^* ) and a cosine value (cos β ^* ) of the predetermined value, respectively. A command value current calculation unit (64) for generating a value ( _id ^* ) and a q-axis current command value ( _iq ^* ), and the inverter based on the d-axis current command value and the q-axis current command value. Command value signal generating means (60) for generating command value signals (T _u , T _v , T _w ) for the switching operation. Then, the command value current calculation unit takes a value of zero in the first period, monotonously decreases after a monotonic increase in the second period, and is a non-negative command value (| i _in ^* |) of the absolute value of the input current. The compensation current command value (i _comp ^* ) based on the difference between the command value of the absolute value of the input current and the absolute value of the input current (| i _in |) is _{calculated as} the speed ( ω _m ) and the current command value (i _m ^* , i _T ^* ) generated based on the difference between the speed command value (ω _m ^* ) and the command value ( drive current command value generation means (642) as i _dq ^* ). However, the first period and the second period are repeated with a half period of the period of the input voltage.

A sixth aspect of the multiphase current supply circuit according to the present invention is the fifth aspect of the multiphase current supply circuit, wherein the drive current command value generating means (642) is a sine of the phase angle of the input voltage. Subtract the positive first constant (k ₁ ) less than 1 from the absolute value of the value (| sinθ _in |), and if the result is non-negative, multiply the result by the positive second constant (k ₂ ) If the result is negative, zero is adopted to obtain a modulation coefficient (r), a modulation coefficient generation unit (641a), and an averaging unit (642a) for _obtaining the amplitude (i _in1 ) of the fundamental wave of the input current; A multiplier (642c) for obtaining the command value of the absolute value of the input current as a product of the amplitude of the fundamental wave of the input current and the modulation coefficient (r).

A seventh aspect of the multiphase current supply circuit according to the present invention is the sixth aspect of the multiphase current supply circuit, in which the command value current calculation unit (64) includes the speed (ω _m ) of the drive unit. A first current command value (i _m ^* ) obtained by performing a proportional / integral operation on the difference from the speed command value (ω _m ^* ) is modulated using the modulation coefficient (r) to obtain a first current command value (i _m ^* ). And a multiplier (641b) for obtaining a current command value (i _T ^* ) of 2 as the current command value.

An eighth aspect of the multiphase current supply circuit according to the present invention is the sixth or seventh aspect of the multiphase current supply circuit, wherein the modulation coefficient generation unit (641a) includes the second constant (k ₂ ) is set and stored so that the average value of the product of the modulation coefficient (r) and the absolute value of the sine value of the phase angle of the input voltage (| sinθ _in |) is 0.5 (6415) ).

  According to the first aspect of the inverter control method of the present invention, even when the decrease amount of the voltage across the capacitor is small and the non-conduction angle of the input current is widened, the failure of the control system is avoided, and the input current An increase in harmonics is suppressed.

  According to the second aspect of the inverter control method of the present invention, a modulation coefficient suitable for the first aspect of the inverter control method can be obtained.

  According to the third aspect of the inverter control method of the present invention, the average value of the input power can be made constant.

  According to the fourth aspect of the inverter control method of the present invention, it is possible to improve the followability of the input current with respect to the command value and to construct a control system that is unlikely to fail.

  According to the fifth aspect of the inverter control method of the present invention, it is possible to improve the followability of the input current with respect to the command value and to construct a control system that is less likely to fail.

  According to the sixth aspect of the inverter control method of the present invention, a modulation coefficient suitable for the fifth aspect of the inverter control method can be obtained.

  According to the seventh aspect of the inverter control method of the present invention, even when the decrease amount of the voltage across the capacitor is small and the non-conduction angle of the input current is widened, the failure of the control system is avoided, and the input current An increase in harmonics is suppressed.

  According to the eighth aspect of the inverter control method of the present invention, the average value of the input power can be made constant.

  According to the first aspect of the multiphase current supply circuit of the present invention, even when the decrease amount of the voltage across the capacitor is small and the non-conduction angle of the input current is widened, the failure of the control system is avoided, An increase in harmonics of the input current is suppressed.

  According to the second aspect of the multiphase current supply circuit of the present invention, a modulation coefficient suitable for the first aspect of the multiphase current supply circuit can be obtained.

  According to the third aspect of the multiphase current supply circuit of the present invention, the average value of the input power can be made constant.

  According to the fourth aspect of the multiphase current supply circuit of the present invention, it is possible to improve the followability of the input current with respect to the command value and to construct a control system that is unlikely to fail.

  According to the fifth aspect of the multiphase current supply circuit of the present invention, it is possible to improve the followability of the input current with respect to the command value and to construct a control system that is unlikely to fail.

  According to the sixth aspect of the multiphase current supply circuit of the present invention, a modulation coefficient suitable for the fifth aspect of the multiphase current supply circuit can be obtained.

  According to the seventh aspect of the multiphase current supply circuit of the present invention, even when the decrease amount of the voltage across the capacitor is small and the non-conduction angle of the input current is widened, the failure of the control system is avoided, An increase in harmonics of the input current is suppressed.

  According to the eighth aspect of the multiphase current supply circuit of the present invention, the average value of the input power can be made constant.

  FIG. 1 is a circuit diagram illustrating a drive device according to an embodiment of the invention. The drive device includes a motor 5 that is a multiphase drive unit, and a multiphase current supply circuit that supplies a multiphase current thereto.

The multiphase current supply circuit includes a diode bridge 2, a smoothing circuit 3, an inverter 4, and a control circuit 6. Power single-phase AC is connected to the diode bridge 2, the input current i _in of the input voltage v _in and single-phase AC of the single-phase AC is supplied.

Diode bridge 2 has a function of performing full-wave rectification, the input voltage v _in to a full-wave rectified input to the smoothing circuit 3. The smoothing circuit 3 has a capacitor 31. Specifically, the output of the diode bridge 2 is received between both ends of the capacitor 31, and the rectified voltage (voltage between both ends) v _dc generated at both ends of the capacitor 31 is output to the inverter 4.

The inverter 4 receives the output of the smoothing circuit 3, and the current i _inv is input to the inverter 4 from one end of the capacitor 31 (the cathode side of the diode constituting the diode bridge 2). Capacitance of the capacitor 31, the voltage across v _dc is set to pulsate at a frequency twice the frequency of the input voltage v _in. For example, the capacity of the capacitor 31 is set to about several tens of μF.

The inverter 4 supplies three-phase currents i _u , i _v , i _w to the motor 5. The currents i _u , i _v , and i _w correspond to the U phase, the V phase, and the W phase, respectively. The inverter 4 is connected to three transistors (upper arm side transistors) each having a collector connected to one end of the capacitor 31 and to the other end of the capacitor 31 (the anode side of the diode constituting the diode bridge 2). And three transistors (lower arm side transistors) having emitters.

Each of the upper arm side transistors is paired with each of the lower arm side transistors for each phase. The emitter of the upper arm side transistor forming the pair and the collector of the lower arm side transistor are connected in common, and currents i _u , i _v , i _w are output from the connection point. On / off switching of each of the upper arm side transistor and the lower arm side transistor is controlled based on the command value signals T _u , T _v , T _w of the switching operation from the control circuit 6. The command value signals T _u , T _v and T _w correspond to the U phase, V phase and W phase, respectively.

  In order to flow the regenerative current from the motor 5, a free wheel diode having an anode connected to the emitter and a cathode connected to the collector is provided for each of the upper arm side transistor and the lower arm side transistor. ing.

The control circuit 6 inputs the currents i _u , i _v , i _w , the rotation angle (mechanical angle) θ _m of the rotor of the motor 5, the phase angle θ _{in of the} input voltage, and the voltage v _dc across the capacitor 31. These quantities can be detected using known techniques. The control circuit 6 also receives a rotational angular velocity (mechanical angular velocity) command value ω _m ^* and a current phase command value β ^* which are the speeds of the motor 5. Based on these values, command value signals T _u , T _v , T _w are generated based on calculations described later.

  FIG. 2 is a block diagram illustrating the configuration of the control circuit 6. The control circuit 6 includes a position / speed calculator 61, a dq coordinate converter 62, a speed control calculator 63, a command value current calculator 64, a dq axis current controller 65, a PWM (Pulse Width Modulation) calculator 66, A PWM timer unit 67 is provided, and each has a function of executing the following calculation.

Based on the mechanical angle θ _m of the rotor of the motor 5, the position / speed calculating unit 61 and the rotational angle (electrical angle) θ _e of the rotor of the motor 5 and the rotational angular speed (the angular speed of the electrical angle) that is the speed of the motor 5 ) Ω _e and rotational angular velocity (mechanical angle) ω _m are obtained and output. The dq coordinate converter 62 obtains a so-called d-axis current i _d and q-axis current i _q from the currents i _u , i _v , i _w and the electrical angle θ _{e of the} motor 5 based on the equation (1).

The speed control calculation unit 63 includes a subtracter 63a and a PI calculation unit 63b. The subtractor 63a takes the difference between the command value ω _m ^* of the mechanical angle of the motor 5 and the angular speed ω _{m of the} mechanical angle, and the difference is subjected to proportional / integral calculation (PI calculation) by the PI calculation unit 63b. ^* the first current command value i _m is output.

The command value current calculation unit 64 includes a current command value modulation unit 641, a current command value compensation unit 642, and a dq current command value generation unit 643. The current command value modulation section 641 and modulated using a modulation coefficient r to be described later ^* first current command value i _m, and generates a ^* second current command value i _T. Based on the difference between the absolute value | i _in | of the input current i _in and the command value | i _in ^* |, the current command value compensation unit 642 generates a compensation current command value i _comp ^*, which will be described in detail later, A result added to the current command value i _T ^* is generated as a drive current command value i _dq ^* . In this sense, the current command value compensation unit 642 can be grasped as a drive current command value generation unit that generates a command value of the drive current.

The dq current command value generation unit 643 outputs the d-axis current command value i _d ^* and the q-axis current i _q ^* based on the equation (2) using the drive current command value i _dq ^* and the current phase command value β ^*. To do. That sine value of a predetermined value with respect to the drive current command value ^{_{i dq * (-β *) (}} -sinβ *) and the cosine value (cos .beta ^*) and multiplied by the respective d-axis current command value i _d ^* and the q-axis A current command value i _q ^* is generated.

The dq-axis current control unit 65 inputs the d-axis current i _d and the q-axis current i _q , the d-axis current command value i _d ^* and the q-axis current i _q ^* , and the angular velocity ω _{e of the} electrical angle, and the equation (3) The d-axis voltage command value v _d ^* and the q-axis voltage command value v _q ^* are output based on In Equation (3), K _d and K _q are proportional gains of d-axis and q-axis, L _d and L _q are motor inductances of d-axis and q-axis, respectively, and φ _a is a motor back electromotive voltage. It is a constant.

The rotation angle (electrical angle) θ _e of the rotor, the d-axis voltage command value v _d ^*, and the q-axis voltage command value v _q ^* are input to the PWM calculation unit 66, and each phase voltage is calculated based on Equation (4). Command values v _u ^* , v _v ^* , v _w ^* are generated.

Further, the both-end voltage v _dc is also input to the PWM calculation unit 66, and using this and each phase voltage command value v _u ^* , v _v ^* , v _w ^* , based on the equation (5), The on-time τ _j (j = u, v, w) of the upper arm side transistor is obtained. However, in the formula (5), a carrier cycle Tc is introduced. When the on-time τ _j exceeds the carrier period Tc, the value is forcibly set to Tc, and when the on-time τ _j is less than 0, the value is forcibly set to 0.

The PWM timer unit 67 stores on-time τ _u , τ _v , τ _w for each carrier cycle Tc, and command value signals T _u , T _v , T for turning on / off each phase transistor in response to the stored time. _{w is applied} to the inverter 4.

A group of the position / speed calculation unit 61, the dq coordinate conversion unit 62, the dq axis current control unit 65, the PWM calculation unit 66, and the PWM timer unit 67 includes a d axis current command value i _d ^* and a q axis current command value i. command value signal based on the _q ^* T _u, T _v, can be grasped as a command value signal generating means 60 for generating a T _w.

The current command value modulation unit 641 includes a modulation coefficient generation unit 641a and a multiplier 641b. Modulation coefficient generating unit 641a inputs the phase angle theta _in the voltage v _in to generate a modulation coefficient r. The multiplier 641b multiplies the first current command value i _m ^* by the modulation coefficient r to generate a second current command value i _T ^* .

If the ideal input current i _in0 is realized, the harmonic component of the input current i _in can be significantly suppressed. In this case, the input power P _in can be expressed by Equation (6).

Since the input power P _in such is proportional to the square of the sine wave phase theta _in the input voltage v _in, it varies at a frequency twice the frequency of the input voltage v _in.

Further, while the power P _m of the motor 5 as shown in equation (7) is the product of the rotational angular frequency omega _m and the torque tau _m of the motor 5, the efficiency eta ₄ of the inverter 4 to the input power P _in It can also be grasped as electric power multiplied by the efficiency η _{5 of the} motor 5.

The inertia occurs in the motor 5, the rotational angular frequency omega _m is kept substantially constant. Thus, varying at twice the frequency of the input voltage v _in the torque tau _m of the motor, from the viewpoint of controlling to the ideal input current i _in0 the input current i _in. Such control is introduced in, for example, Patent Document 1 described above.

The torque tau _m of the motor 5, when the phase beta of the current flowing through the motor (this command value is the current phase command value beta ^* at a) are constant, the drive current i _dq as shown in equation (8) ( This command value is proportional to the drive current command value i _dq ^* ).

Thus varying at twice the frequency of the input voltage v _in the drive current command value i _dq ^* is desirable from the viewpoint of control to the ideal input current i _in0 the input current i _in.

Therefore adopting a coefficient that varies at a frequency twice the frequency of the voltage v _in the modulation coefficient r. In an ideal case, the second current command value i _T ^* may be adopted as the drive current command value i _dq ^* . However, even if the drive current command value i _dq ^* is set in this way, the efficiency η ₄ , η ₅ varies depending on the variation of the operating point of the inverter 4 or the motor 5, or the charging current to the capacitor 31 varies. Accordingly, the input current i _in is distorted from the ideal input current i _in0 .

Therefore, the compensation current command value i _comp ^* is used as the drive current command value i _dq ^* in addition to the second current command value i _T ^* . Since the input power P _in is ideally proportional to the drive current i _dq , when the input current i _in is larger than the ideal input current i _in0 , the compensation current command value i _comp ^* is reduced, When the input current i _in is smaller than the ideal input current i _in0 , the compensation current command value i _comp ^* is increased.

In this embodiment, ideal to extend the in terms of phase theta _in the input voltage v _in, a period the input current i _in does not conduct (referred to as "non-conduction angle" in this case), as a command value of the input current It is preferable to use a value that fluctuates based on the modulation coefficient r rather than an input current i _in0 . A specific function form of the modulation coefficient will be described later.

The current command value compensation unit 642 generates a compensation current command value i _comp ^* based on the difference between the absolute value | i _in | of the input current i _in and its command value | i _in ^* | And an input current compensator 642e. An adder 642f is provided to add the compensation current command value i _comp ^* to the second current command value i _T ^* . The subtractor 642d subtracts the absolute value | i _in | of the input current i _in from the command value | i _in ^* |, and the input current compensator 642e performs PI control, for example, proportional control, on the result, and a compensation current command The value i _comp ^* is generated.

The current command value compensation unit 642 generates an absolute value | i _in | of the input current i _in and its command value | i _in ^* | in order to generate an averaging unit 642a, an absolute value generation unit 642b, and a multiplier 642c. Have. The averaging unit 642a averages the input current i _in to generate the amplitude i _in1 of the fundamental frequency component. The absolute value generation unit 642b generates an absolute value | i _in | of the input current i _in . The multiplier 642c generates a command value | i _in ^* | of the absolute value of the input current i _in as the product of the modulation coefficient r and the amplitude i _in1 of the fundamental frequency component. Therefore, the modulation coefficient generation unit 641a may be provided separately in the current command value compensation unit 642, or may be shared between the current command value modulation unit 641 and the current command value compensation unit 642.

FIG. 3 is a block diagram illustrating the configuration of the averaging unit 642a. The averaging unit 642a includes a sine value generation unit 6420, a multiplier 6421, and an integration calculation unit 6422. The amplitude i _in1 of the fundamental frequency component is calculated by equation (9).

Sine value generation section 6420 generates a sine value sin [theta _in the phase angle theta _in the input voltage v _in, the multiplier 6421 takes the product of the input current i _in and the sine value sin [theta _in, integration calculation portion 6422 is the product The result obtained by integrating the phase angle θ _{in from the} interval 0 to 2π is divided by π to obtain the amplitude i _in1 of the fundamental frequency component.

As described above, to obtain a drive current command value i _dq ^* by adding a compensation current command value i _comp ^* in ^* the second current command value i _T, by performing the switching control of the inverter 4 based on this , to improve the following capability of the input current i _in respect to the command value i _in ^* of the input current i _in, it is possible to construct a collapse less controllable system.

  Here, the modulation coefficient r is generated and output based on the equation (10).

In other words, the positive first constant k ₁ smaller than ₁ is subtracted from the absolute value | sinθ _in | of the sine value, and if the result is non-negative, the result is multiplied by the positive second constant k _2, and the result is negative. If so, zero is adopted to obtain the modulation coefficient r.

FIG. 4 is a graph illustrating the behavior of the modulation coefficient r with respect to time. For reference, (| sin θ _in | −k ₁ ) · k ₂ is also indicated by a broken line. Thus modulation coefficient r has at the period π in terms of period 2 sin ^-1 k ₁ of phase angle value becomes zero phase angle theta _in the input voltage v _in. In other words, the modulation coefficient r is non-negative, takes a value of zero in the first period, and takes a value that decreases monotonically after increasing monotonously in the second period. And the first period and the second period are repeated at half the period of the cycle of the input voltage v _in. In this embodiment, the modulation factor r, when adopting the portion of the sine wave in the second period and is illustrated, the length of the first period in terms of the phase angle theta _in the input voltage v _in 2 sin ⁻¹ k ₁

  FIG. 5 is a block diagram illustrating the configuration of the modulation coefficient generator 641a. The modulation coefficient generator 641a includes a sine value generator 6410, an absolute value generator 6411, an adder / subtractor 6412, a limiter 6413, a multiplier 6414, and a ROM 6415. However, the sine value generation unit 6410 can also be used as the sine value generation unit 6420 of the averaging unit 642a.

The absolute value generation unit 6411 takes and outputs the absolute value of the sine value sinθ _in given from the sine value generation unit 6410. The adder / subtractor 6412 obtains the first constant k ₁ from the ROM 6415 and subtracts it from the absolute value | sinθ _in |. The result is given to limiter 6413 to limit the negative side. That is, all negative values of (| sin θ _in | −k ₁ ) are replaced with zero. The multiplier 6414 obtains the second constant k ₂ from the ROM 6415 and multiplies it by the output from the limiter 6413 to obtain the modulation coefficient r. The contents of the ROM 6415 are preferably rewritable from the outside.

The modulation coefficient r as described above a non-negative, adopts the value zero in the first period, the second current command value i _T ^* also in terms of the phase angle theta _in the input voltage v _in 2 sin ^-1 k ₁ during the zero, and this period is repeated at half the period of the cycle of the input voltage v _in.

Modulation coefficient r so vary at twice the frequency of the input voltage v _in, you can realize a switching control for single-phase capacitorless inverter control. In addition, as described above, since the command value | i _in ^* | of the absolute value of the input current i _in is the product of the modulation coefficient r and the amplitude i _in1 of the fundamental frequency component, the second current command value i _T ^* and Similarly, the command value of the absolute value of the input current i _in | i _in ^* | even input voltage v 2 sin ^-1 between k ₁ becomes zero in terms of the phase angle theta _in the _in, this time the input voltage v _in It is repeated at half the cycle. Therefore, even if the voltage across v _dc is nonconductive angle generated in the input current i _in, due to such not sufficiently reduced, the command value i _in ^* of the input current i _in, therefore the absolute value of the input current i _in Since the non-conduction angle of the command value | i _in ^* | is widened, the control system is unlikely to fail.

In the present embodiment, when the second current command value i _T ^* is obtained, a coefficient by which the first current command value i _m ^{* is} multiplied and a command value | i _in ^* | of the absolute value of the input current i _in The modulation coefficient r is used as both of the coefficient by which the amplitude i _in1 of the fundamental frequency component is multiplied. However, this is not essential in the present embodiment, and either one may be different from this. For example, the coefficient for either one may be | sinθ _in | or may be sin ² θ _in . In this case, however, it is desirable that the coefficient fluctuates at twice the power supply frequency.

Or without employing a compensation current command value i _comp ^*, well be employed ^* second current command value i _T as the drive current command value i _dq ^*, or the first current command value i _m ^* The second current command value i _T ^* may be used as it is. It can be understood that the former corresponds to the case where the gain of the input current compensator 642e is very small, and the latter corresponds to the case where the gain is very large.

Next, specific examples of the first constant k ₁ and the second constant k ₂ will be described. Ideally, since the absolute value | i _in | of the input current i _in is equal to the command value | i _in ^* |, the input power Pin is expressed as shown _in Expression (11). Here V ₀ is the amplitude of the input voltage v _in.

And the amplitude V ₀ which input voltage v _in, the amplitude i _in1 of the fundamental frequency component of the input current i _in the small fluctuation, thus from the viewpoint of the average value of the input power P _in the constant r · | mean | sin [theta _in It is desirable to determine the modulation coefficient r so that the value does not change. When the absolute value | i _in | of the input current i _in does not have a non-conduction angle, it corresponds to the case of k ₁ = 0 and k ₂ = 1, so the average value of r · | sin θ _in | .5.

FIG. 6 is a graph plotting the first constant k ₁ and the second constant k ₂ where the average value of r · | sin θ _in | is 0.5. Therefore, it is desirable to adopt k ₁ and k ₂ listed in this graph.

FIG. 7 is a graph showing the results of experimentally determining the conditions under which the power supply harmonics are minimized when the first constant k ₁ is changed and the second constant k ₂ is selected according to FIG. The input power P _in the horizontal axis, the first constant k ₁ and the vertical axis, are taken, respectively. The first constant k ₁ appropriate decreases with an increase in input power P _in, therefore the second constant k ₂ suitable Referring to FIG. 6 is increased. This tendency does not depend much on the rotation speed when the load of the motor 5 is light. This is that there are charge energy of the capacitor 31 is at light load, because the voltage across v _dc is less likely to decrease, the first constant k ₁ is believed to be due to greatly depends on the input power P _in.

On the other hand, as explained in the problem to be solved by the invention, particularly when the load of the motor 5 is heavy due to the fact that the voltage v _dc does not decrease because the phase voltage v _m does not sufficiently decrease, The first constant k ₁ depends on the rotational speed. Phase voltage v _m the rotation speed is because increases.

FIG. 8 is a graph showing the results of experimentally determining the conditions under which the power supply harmonics are minimized when the first constant k ₁ is changed and the second constant k ₂ is selected according to FIG. The horizontal axis represents the rotation speed of the motor, and the vertical axis represents the first constant k ₁ . When the rotational speed is about 70 rps or less, k ₁ = 0 (and hence k ₂ = 1) is suitable. However, when the rotational speed exceeds 70 rps, the appropriate first constant k ₁ increases as the rotational speed increases. With reference to 6, the appropriate second constant k ₂ decreases.

Therefore, for example, when the motor rotates at a certain rotational speed under a certain input power, the larger one of the value obtained from FIG. 7 and the value obtained from FIG. 8 is adopted as the first constant k ₁ . be able to. In actual operation, it is unlikely that the electric power is small at high speed. Therefore, at a certain rotational speed, the value obtained from FIG. 7 is used below, and the value obtained from FIG. can do.

It is a circuit diagram which illustrates the drive device concerning an embodiment of the invention. 3 is a block diagram illustrating a configuration of a control circuit 6. FIG. It is a block diagram which illustrates the composition of averaging part 642a. It is a graph which illustrates the behavior with respect to time of the modulation coefficient r. It is a block diagram which illustrates the composition of modulation coefficient generating part 641a. The first constant k ₁ and the second constant k ₂ is a graph plotting. It is a graph showing the relationship between the input power P _in the first constant k ₁ and. It is a graph showing the relationship between the first constant k ₁ and the rotation speed. It is a circuit diagram which illustrates the system by which single phase capacitorless inverter control is adopted. FIG. 10 is a graph showing an ideal state of voltage and current at each part in the circuit diagram shown in FIG. 9. FIG. FIG. 10 is a graph showing the voltage and current of each part in the circuit diagram shown in FIG. 9.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 AC power supply 2 Diode bridge 3 Smoothing circuit 31 Capacitor 4 Inverter 5 Motor 6 Control circuit 60 Command value signal generation means 63 Speed control calculation part 64 Command value current calculation part 641 Current command value modulation part 642 Current command value compensation part (drive current) Command value generation means)
643 dq current command value generation unit

Claims (16)

A diode bridge (2) that inputs an input voltage (v _in ) and an input current (i _in ) having a predetermined fundamental frequency from an AC power supply (1) and performs full-wave rectification;
A smoothing circuit (3) having a capacitor (31) for receiving the output of the diode bridge;
Switching of the inverter when driving the multiphase drive unit (5) using the inverter (4) that receives the output of the smoothing circuit and outputs a multiphase alternating current (i _u , i _v , i _w ) A method of controlling
(A) generating a command value (i _dq ^* ) of the drive current of the multiphase drive unit based on the phase angle (θ _in ) of the input voltage;
(B) by multiplying a predetermined value sine of (-β ^*) (-sinβ ^*) and the cosine value of the predetermined value (cos .beta ^*) to said drive current, respectively d-axis current command value (i _d ^* ) And a q-axis current command value (i _q ^* ),
(C) generating a command value signal (T _u , T _v , T _w ) for switching operation of the inverter based on the d-axis current command value and the q-axis current command value;
The step (a) generates (a-1) a first current command value (i _m ^* ) based on a difference between the speed (ω _m ) of the drive unit and the command value (ω _m ^* ) of the speed. And steps to
(A-2) obtaining a non-negative modulation coefficient (r) that takes a value of zero in the first period and monotonously decreases after a monotonic increase in the second period;
(A-3) modulating the first current command value using the modulation coefficient to obtain a second current command value (i _T ^* );
(A-4) generating the command value of the drive current based on the second current command value;
The inverter control method, wherein the first period and the second period are repeated at a half cycle of the input voltage. In the step (a-2), a positive first constant (k ₁ ) smaller than 1 from the absolute value (| sin θ _in |) of the sine value of the phase angle (θ _in ) of the input voltage (v _in ). If the result is non-negative, the result is multiplied by a positive second constant (k ₂ ), and if the result is negative, zero is used to determine the modulation coefficient (r). The inverter control method according to claim 1. The second constant (k ₂ ) is an average value of the product of the modulation coefficient (r) and the absolute value (| sin θ _in |) of the sine value of the phase angle (θ _in ) of the input voltage (v _in ). The inverter control method according to claim 2, wherein the inverter control method is set to be 0.5. In the step (a-4), the command value (| i _in ^* |) of the absolute value of the input current obtained as the product of the amplitude (i _in1 ) of the fundamental wave of the input current and the modulation coefficient (r) And the compensation current command value (i _comp ^* ) based on the difference between the absolute value (| i _in |) of the input current and the second current command value (i _T ^* ). The inverter control method according to any one of claims 1 to 3, wherein the command value (i _dq ^* ) of the drive current is used. A diode bridge (2) that inputs an input voltage (v _in ) and an input current (i _in ) having a predetermined fundamental frequency from an AC power supply (1) and performs full-wave rectification;
A smoothing circuit (3) having a capacitor (31) for receiving the output of the diode bridge;
Switching of the inverter when driving the multiphase drive unit (5) using the inverter (4) that receives the output of the smoothing circuit and outputs a multiphase alternating current (i _u , i _v , i _w ) A method of controlling
(A) generating a command value (i _dq ^* ) of the drive current of the multiphase drive unit based on the phase angle (θ _in ) of the input voltage;
(B) by multiplying a predetermined value sine of (-β ^*) (-sinβ ^*) and the cosine value of the predetermined value (cos .beta ^*) to said drive current, respectively d-axis current command value (i _d ^* ) And a q-axis current command value (i _q ^* ),
(C) generating a command value signal (T _u , T _v , T _w ) for switching operation of the inverter based on the d-axis current command value and the q-axis current command value;
In step (a), (a-1) a current command value (i _m ^* , i _T ^* ) is calculated based on a difference between the speed (ω _m ) of the drive unit and the command value (ω _m ^* ) of the speed. Generating step;
(A-2) obtaining a command value (| i _in ^* |) of the absolute value of the input current that takes a value of zero in the first period, monotonously decreases after a monotonic increase in the second period, and is nonnegative; ,
(A-3) A compensation current command value (i _comp ^* ) based on a difference between the command value of the absolute value of the input current and the absolute value (| i _in |) of the input current is set to the current command value. And the step of setting the command value (i _dq ^* ) of the drive current as a result of adding to
The inverter control method, wherein the first period and the second period are repeated at a half cycle of the input voltage. The step (a-2)
(A-2-1) Subtract the positive first constant (k1) smaller than 1 from the absolute value (| sinθ _in |) of the sine value of the phase angle (θ _in ) of the input voltage (v _in ) Multiplying the result by a positive second constant (k2) if the result is non-negative and adopting zero if the result is negative to determine the modulation coefficient (r);
(A-2-2) _{obtaining the} command value (| i _in ^* |) of the absolute value of the input current as the product of the amplitude (i _in1 ) of the fundamental wave of the input current and the modulation coefficient (r) The inverter control method of Claim 5 provided with these. The step (a-1) includes: (a-1-1) performing a proportional / integral operation on the difference between the speed (ω _m ) of the driving unit and the command value (ω _m ^* ) of the speed, determining a command value (i _m ^*),
(A-1-2) obtaining a second current command value (i _T ^* ) obtained by modulating the first current command value using the modulation coefficient (r) as the current command value; The inverter control method of Claim 6 provided. The second constant (k ₂ ) is an average value of the product of the modulation coefficient (r) and the absolute value (| sinθ _in |) of the sine value of the phase angle (θ _in ) of the input voltage (v _in ). The inverter control method according to claim 7, wherein the inverter control method is set to be 0.5. A diode bridge (2) that inputs an input voltage (v _in ) and an input current (i _in ) having a predetermined fundamental frequency from an AC power supply (1) and performs full-wave rectification;
A smoothing circuit (3) having a capacitor (31) for receiving the output of the diode bridge;
An inverter (4) that receives the output of the smoothing circuit and outputs a polyphase alternating current (i _u , i _v , i _w ) to the polyphase drive unit (5);
A control circuit (6) for controlling the switching of the inverter,
The control circuit generates a command value (i _dq ^* ) of the drive current of the multiphase drive unit based on the phase angle (θ _in ) of the input voltage, and a predetermined value (−β ^* ) with respect to the drive current multiplied by the sine value (-sinβ ^*) and the cosine value of the predetermined value and (cosβ ^*), generates a d-axis current command value (i _d ^*) and q-axis current command value (i _q ^*), respectively A command value current calculation section (64);
Command value signal generating means (60) for generating command value signals (T _u , T _v , T _w ) for switching operation of the inverter based on the d-axis current command value and the q-axis current command value;
The command value current calculator is
A first current command value (i _m ^* ) generated based on a difference between the speed (ω _m ) of the driving unit and the command value (ω _m ^* ) of the speed is calculated using a modulation coefficient (r). A current command value modulating unit (641) for modulating and obtaining a second current command value (i _T ^* );
Drive current command value generating means (642) for generating the command value of the drive current based on the second current command value;
Have
The modulation factor is non-negative, takes a value of zero in the first period, monotonically decreases after a monotonic increase in the second period,
The multiphase current supply circuit, wherein the first period and the second period are repeated at a period that is half of the period of the input voltage. The current command value modulation section (641)
A positive first constant (k ₁ ) smaller than ₁ is subtracted from the absolute value (| sinθ _in |) of the sine value of the phase angle of the input voltage (v _in ), and if the result is non-negative, the result is positive. A modulation coefficient generator (641a) for multiplying the second constant (k ₂ ) and adopting zero if the result is negative to obtain the modulation coefficient (r);
The multiplicity (641b) according to claim 9, further comprising a multiplier (641b) that outputs a product of the modulation coefficient and the first current command value (i _m ^* ) as the second current command value (i _T ^* ). Phase current supply circuit. The modulation coefficient generation unit (641a) averages the product of the second constant (k ₂ ) and the modulation coefficient (r) and the absolute value (| sinθ _in |) of the sine value of the phase angle of the input voltage. The multiphase current supply circuit according to claim 10, further comprising a memory (6415) configured to store the value to be 0.5. The drive current command value generation means (642)
An averaging unit (642a) for _obtaining the amplitude (i _in1 ) of the fundamental wave of the input current;
A multiplier (642c) for obtaining a command value (| i _in ^* |) of the absolute value of the input current as a product of the amplitude (i _in1 ) of the fundamental wave of the input current and the modulation coefficient (r);
A compensation current command value (i _comp ^* ) based on the difference between the command value of the absolute value of the input current and the absolute value (| i _in |) of the input current is determined as the second current command value (i _T the command value of the driving current in addition to ^*) (i _dq ^*) to determine adder and a (642f), the multiphase current supplying circuit according to any one of claims 9 to 11. A diode bridge (2) that inputs an input voltage (v _in ) and an input current (i _in ) having a predetermined fundamental frequency from an AC power supply (1) and performs full-wave rectification;
A smoothing circuit (3) having a capacitor (31) for receiving the output of the diode bridge;
An inverter (4) that receives the output of the smoothing circuit and outputs a polyphase alternating current (i _u , i _v , i _w ) to the polyphase drive unit (5);
A control circuit (6) for controlling the switching of the inverter,
The control circuit generates a command value (i _dq ^* ) of the drive current of the multiphase drive unit based on the phase angle (θ _in ) of the input voltage, and a predetermined value (−β ^* ) with respect to the drive current multiplied by the sine value (-sinβ ^*) and the cosine value of the predetermined value and (cosβ ^*), generates a d-axis current command value (i _d ^*) and q-axis current command value (i _q ^*), respectively A command value current calculation section (64);
Command value signal generating means (60) for generating command value signals (T _u , T _v , T _w ) for switching operation of the inverter based on the d-axis current command value and the q-axis current command value;
The command value current calculator is
A command value (| i _in ^* |) of the absolute value of the input current that takes a value of zero in the first period, decreases monotonically after a monotonic increase in the second period, and is non-negative, and calculates the absolute value of the input current The compensation current command value (i _comp ^* ) based on the difference between the command value of the input current and the absolute value (| i _in |) of the input current is set as the speed (ω _m ) of the drive unit and the command value of the speed. (ω _m ^*) the current command value generated based on the difference between ^{_{(i m *, i T *}} ) the command value of the result of adding to said drive current (i _dq ^*) to drive current command Value generation means (642)
Have
The multiphase current supply circuit, wherein the first period and the second period are repeated with a period that is half the period of the input voltage. The drive current command value generation means (642)
If the positive first constant (k1) smaller than 1 is subtracted from the absolute value (| sinθ _in |) of the sine value of the phase angle of the input voltage and the result is non-negative, the positive second constant (k2) ), And if the result is negative, adopt zero if the result is a modulation coefficient generator (641a) to obtain the modulation coefficient (r);
An averaging unit (642a) for _obtaining the amplitude (i _in1 ) of the fundamental wave of the input current;
The multiphase current supply according to claim 13, further comprising a multiplier (642c) for obtaining the command value of the absolute value of the input current as a product of the amplitude of the fundamental wave of the input current and the modulation coefficient (r). circuit. The command value current calculation unit (64) is a first current command value obtained by performing a proportional / integral calculation on the difference between the speed (ω _m ) of the drive unit and the command value (ω _m ^* ) of the speed. A multiplier (641b) that modulates (i _m ^* ) using the modulation coefficient (r) to obtain a second current command value (i _T ^* ) as the current command value
The multiphase current supply circuit according to claim 14, further comprising: The modulation coefficient generation unit (641a) averages the product of the second constant (k ₂ ) and the modulation coefficient (r) and the absolute value (| sinθ _in |) of the sine value of the phase angle of the input voltage. The multiphase current supply circuit according to claim 14 or 15, further comprising a memory (6415) configured to store the value to be 0.5.

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