JP5338853B2 - Power converter - Google Patents

Description

The present invention relates to a power converter.

Conventionally, a system for driving a motor using one battery, a DCDC converter, and an inverter is well known. Conventionally, a configuration for driving a motor with high efficiency using a fuel cell as a main power source is disclosed in Japanese Patent Application Laid-Open No. 2002-118981 (refer to Patent Document 1). FIG. 1 shows the configuration of such a conventional motor drive system. In the system of FIG. 1, a battery is connected in parallel with a fuel cell via a DCDC converter, and the output voltage of the DCDC converter is controlled to improve the output efficiency of the power supply.

As an improvement on the above-described prior art, the applicant of the present application (Japanese Patent Application No. 2004-200545) does not use a DCDC converter, and is not limited to a combination of a fuel cell and a battery, but a plurality of power sources. We have developed a control method for power converters that can be used and distributed to reduce overall volume and loss. Further, the present applicant has developed a control method for selecting whether to flow a current flowing from an inductive load to a power supply bus in any of the other prior applications (Japanese Patent Application No. 2004-207031).

Japanese Patent Laying-Open No. 2002-118981 (paragraphs 0004-0006, FIG. 1)

However, since the DCDC converter is used in the above-described configuration, the volume of the entire system including the power source, the power conversion device, and the motor increases, and the battery passes through the DCDC converter in order to charge and discharge the battery. There was a problem of loss. In a prior application by the present applicant (Japanese Patent Application No. 2004-207031) that solves this problem and eliminates the need for a DCDC converter, a control method for selecting whether to flow the current flowing from the inductive load to the power supply bus to any of the buses However, when this selection is performed, there is a problem that a harmonic component is generated in the current if there is a difference in the voltage between the power supplies. However, the prior application has not shown a solution to the problem when such a voltage difference exists.

Therefore, the present invention follows the distribution ratio command value for the actual distribution of power supply power when there is a path in which unidirectional conduction between one of a plurality of DC power supplies and the AC motor is not possible. The purpose is to let you .

In order to solve the above-described problems, the power conversion device according to the first invention includes a plurality of DC power supplies ,
Connected to the DC power supply, and Rupa Luz generating and combining means generates and combining the pulses from a plurality of direct-current voltage output from the DC power supply, and a control unit for controlling said pulse generating and combining means, said pulse And an AC motor connected to the synthesizing / generating means and driven by the pulses output from the pulse synthesizing / generating means, and the control device determines the AC based on a torque command value and a motor rotational speed given from the outside. A current control unit that calculates a current command value of the motor, calculates a voltage command value based on the calculated current command value, a voltage command value calculated by the current control unit, a voltage value of the DC voltage, and the DC based on the distribution target value of the output power of the power supply calculates a voltage command value of the DC power sources, drive the pulse generating and combining means based on the voltage command values computed the DC power source If not the plurality of the DC power supply and drive pulse generating means for determining a conduction path between the AC motor, which determines the presence or absence of selection of the path of the return current of the AC motor, to implement the selection of the route, In the pulse generating / synthesizing means, when there is a path in which the reflux current cannot be conducted between any one of the plurality of DC power supplies and the AC motor, the pulse is generated according to an actual conduction path. Power distribution means for allocating output power of the generating / combining means to output power of a plurality of DC power supplies.

Further, in the power conversion device according to the second invention, the power distribution means includes at least a distribution ratio command value of the output power of the DC power supply, an output power command value, a motor current amplitude, according to an actual conduction path. The distribution target value is calculated from a plurality of DC power supply voltage command values, and the output voltage value of the power converter from each DC power supply is calculated from the output voltage value to the AC motor and the distribution target value. The output power of the plurality of direct-current power sources is adjusted by obtaining.

The power converter according to a third aspect of the invention is characterized in that the power distribution means calculates an output power to the AC motor from a current value of the motor and an output voltage command value to the motor.

According to a fourth aspect of the present invention, there is provided the power converter according to the present invention, wherein the power distribution means includes at least an output power distribution ratio command value of the DC power supply, the rotational speed of the AC motor, and the AC according to an actual conduction path. The distribution target value is calculated from the motor torque and a plurality of DC power supply voltage command values, and the power to be output from each power source from the output voltage command value to the AC motor and the distribution target value The output power of the plurality of DC power supplies is adjusted by obtaining the output voltage value of the converter.

The power converter according to the fifth invention is
The power distribution means is
When calculating the distribution target value, the calculation is performed with reference to a map created in advance.
It is characterized by that.

As described above, the solution of the present invention has been described as an apparatus. However, the present invention can be realized as a method substantially equivalent to these, a program to be executed by a computer, and a storage medium storing the program. It should be understood that these are included in the scope of the present invention.

As an example, when the first invention is realized as a method, the control method of the power converter according to the present invention is as follows.
A pulse generation / synthesis step for driving the AC motor by generating a drive voltage of the AC motor by generating / synthesizing pulses from the output voltages of the plurality of DC power supplies,
Generating a drive pulse for driving the pulse control / synthesis means for driving the motor;
In the pulse generation / synthesis step, when there is a path in which one-way conduction between one of the plurality of DC power supplies and the AC motor is impossible, the pulse is generated according to an actual conduction path. A power distribution step of distributing the output power of the generation / pulse control / synthesis means to the output power of a plurality of DC power sources .

According to the sixth aspect of the invention, the modulation factor command value is corrected from the voltage value of the power source without a path for charging from the motor and the voltage value of the power source with a path for charging from the motor. Thus, the pulse width can be corrected with high accuracy, and the generation of harmonic current can be suppressed.

According to the first invention, when there is a path in which unidirectional conduction between any one of the plurality of DC power supplies and the AC motor is present, the distribution of the actual power supply is determined as the distribution ratio command value. Can be followed.

According to the second invention, the distribution target value is calculated from the distribution ratio command value of the output power of the DC power supply, the output power value, the motor current amplitude, and the plurality of DC power supply voltage values. By calculating the voltage command value from each power source using the value, it is possible to realize power control that follows the distribution ratio command value.

According to the third invention, since the output power value is calculated from the current value of the motor and the output voltage value to the motor, the parameters of the general motor current controller are detected without detecting the output power. Can be used to estimate. Since it is not necessary to newly provide detection means such as a power sensor, the cost and volume do not increase.

According to the fourth aspect of the invention, the distribution target value is calculated by using the mechanical output such as the torque command value of the motor and the rotational speed as inputs, and the distribution target value is used. By calculating the voltage command value, it is possible to realize power control that follows the distribution ratio command value.

In addition, according to the fifth aspect , the calculation time can be shortened because the calculation of the distribution target value has only to refer to a map created in advance. Therefore, when calculating the control of the present invention, it is possible to use an inexpensive calculation means (typically, a microcomputer or the like) having a relatively low calculation capability by reducing the calculation time.

It is a figure which shows the structure of the conventional motor drive system. It is a figure which shows the structure of the motor control system of Example 1. FIG. It is a figure which shows the structure of the power converter in Example 1. FIG. It is a figure which shows the triangular wave used with the PWM pulse production | generation means of a 1st Example. It is a figure which shows the structure which extracted only the U phase from FIG. It is a wave form diagram showing pulse generation of A and E by triangular wave comparison. It is a wave form diagram showing pulse generation of D and C by triangular wave comparison. It is a wave form diagram which shows the example of the pulse generation to which the dead time was added. It is a figure which shows the structure of the power converter in Example 2. FIG. It is a figure which shows the structure of the power converter in Example 3. FIG. It is a figure which shows the structure of the modification of the power converter in Example 3. FIG. It is a block diagram which shows the detail of electric power control and a modulation factor calculation. It is a figure which shows the structure of the control system of Example 2. FIG. It is a figure which shows the experimental result (U-phase current) when not using a correction gain. It is a figure which shows the experimental result (U-phase current) at the time of using a correction gain. It is a figure which shows the experimental result (U-phase current) when not using path selection. It is a figure which shows the experimental result (U-phase current) at the time of using route selection. It is a flowchart which shows operation | movement of a correction | amendment gain calculator. 5 is a diagram showing details of current control means 42. FIG.

Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings.

First Embodiment FIG. 3 shows a circuit diagram of a power converter constituting a power conversion apparatus according to the present embodiment. The negative electrode of the power supply 10a and the negative electrode of the power supply 10b are connected to the common negative electrode bus 15. A pair of semiconductor switches 107a, 108a, 109a and diodes 107b, 108b, 109b is connected between the common negative electrode bus 15 and each phase terminal of the motor 20 in the same manner as a generally known lower arm of an inverter. . The positive electrode bus 14 of the power supply 10a and each phase terminal of the motor 20 are connected by semiconductor switches 101a / 101b, 102a / 102b, 103a / 103b, respectively, capable of controlling bidirectional conduction. Further, semiconductor switches 104a / 104b, 105a / 105b, and 106a / 106b that can control bidirectional conduction are also connected between the positive electrode bus 16 of the power supply 10b and each phase terminal of the motor 20. A smoothing capacitor 12 is provided between the positive electrode bus 14 and the common negative electrode bus 15 of the power source 10a, and a smoothing capacitor 13 is also provided between the positive electrode bus 16 and the common negative electrode bus 15 of the power source 10b.

The power converter 30 is a DC-AC power converter that generates a voltage to be applied to the motor based on the above three potentials, the common negative electrode bus, the positive electrode bus of the power source 10a, and the positive electrode bus of the power source 10b. The power converter 30 is composed of switch means 30U, 30V, 30W composed of a plurality of semiconductor switches and diodes for each of the U phase, V phase, and W phase. A semiconductor switch provided in each phase is a switch means for generating a voltage to be output to each phase of the AC motor, and is connected alternatively from these potentials, and the proportion of the connection time is changed. Then, the necessary voltage is supplied to the motor.

FIG. 2 is a diagram illustrating a configuration of the motor control system according to the first embodiment. As shown in the figure, the control device 40 includes torque control means 41, current control means 42, power control / modulation rate calculation means 43, PWM pulse generation means 44, three-phase / dq conversion means 45, dq / 3-phase coordinate conversion. Means 46, correction gain calculator 47, and corrected power distribution calculator 48 are provided. Based on the PWM pulse output from the control device 40, the power converter 30 drives the motor 20 using the power from the power source 10a and the power source 10b constituting the power source 10. Hereinafter, each element will be described in detail.

First, the torque control means 41 calculates the d-axis current command value id * and the q-axis current command value iq * of the AC motor from the torque command Te * and the motor rotation speed ω given from the outside. In the torque control means 41, a map with two elements of Te * and ω created in advance as axes (that is, a map including the following command values calculated in advance by substituting the numerical values of these two elements into a predetermined expression) And output id * and iq *.

The current control means 42 performs current control for matching the d-axis current command value id * and q-axis current command value iq * with the d-axis current value id and q-axis current value iq. By this control, voltage command values vu *, vv *, vw * for each phase of the three-phase AC are output.

Details of the current control means 42 will be described with reference to FIG. The current control means 42 includes a current control unit 201 and a dq / 3 phase converter. The current control unit 201 performs feedback control by PI control so that id and iq follow id * and iq *, respectively, to obtain a d-axis voltage command value vd * and a q-axis voltage command value vq *. id and iq are obtained from the U-phase current iu and the V-phase current iv by the three-phase / dq conversion means 45. Further, a dq / 3-phase converter 202 for converting a dq-axis voltage provided in the current control means 42 into a three-phase voltage command receives dq-axis voltage command values vd * and vq * as inputs, and a U-phase voltage command value vu. *, V phase voltage command value vv *, W phase voltage command value vw * are output.

Returning to the description of FIG. 2, the power control / modulation rate calculating means 43 performs power control using the distribution target values (rto_pa, rto_pb) of the power supplied from the power supplies 10 a and 10 b. The power distribution target value means the ratio of power between the power supply 10a and the power supply 10b when the correction voltage values vd_0 * and vq_0 * are 0. The power distribution target values rto_pa and rto_pb have the following relationship: .

rto_pa + rto_pb = 1
For this reason, if one power distribution target value is obtained, the other power distribution target value can be obtained from the above relationship. In FIG. 2, only rto_pa is shown as an input to the power control / modulation vagina calculation means 43, and rto_pb is calculated based on the above equation by calculation inside this means 43.

Details of the power control / modulation vagina calculation means 43 will be described with reference to FIG. The multiplier 203 multiplies vu *, vv *, and vw * by rto_pa to calculate the voltage command values vu_a, vv_a, and vw_a on the power supply 10a side (hereinafter, a voltage command generated from the power supply 10a is calculated). The power supply 10a voltage command and the voltage command generated from the power supply 10b are referred to as the power supply 10b voltage command).

vu_a = vu * ・ rto_pa
vv_a = vv * ・ rto_pa
vw_a = vw * ・ rto_pa
On the other hand, the voltage command value on the power supply 10b side is obtained from the voltage command values vu *, vv *, vw * obtained from the control voltage of the motor current control, and the voltage command value values vu_a *, vv_a *, vw_a * on the power supply 10a side. Is subtracted by a subtracter 206.

vu _b * = vu *-vu _a *
vv _b * = vv *-vv _a *
vw _b * = vw *-vw _a *
The following description of the modulation factor calculation and PWM pulse generation is performed only for the U phase, but the same operation is performed for the V phase and the W phase. 12 inputs the voltage Vdc_a of the power source 10a and the voltage Vdc_b of the power source 10b, respectively, and instantaneous modulation rate commands mu_a *, mu_b *, mv_a *, mv_b *, which are standardized voltage commands, respectively. Modulation rate calculation means for generating mw_a * and mw_b *.

Modulation rate calculation means 43
The modulation factor calculating means 43 shown in FIG. 12A is configured by multipliers 205 and 206. Here, the U-phase power supply 10a divided voltage command vu_a * and the power supply 10b divided voltage command vu_b * are normalized by half the value of each DC voltage, so that the power supply 10a instantaneous modulation rate command mu_a * and the power supply 10b instantaneous A modulation factor command mu_b * is obtained.

mu_a * = vu_a * / (Vdc_a / 2)
mu_b * = vu_b * / (Vdc_b / 2)
Modulation rate correction means 49
In the modulation rate correction means 49 shown in FIG. 12, in order to output the obtained modulation rate, the time width of the PWM cycle is allocated and the correction gain is multiplied to calculate the final modulation rate command value. Do. First, the modulation factor offset calculator 212 shown in FIG. 12B calculates the next modulation factor offsets ma_offset0 and mb_offset0 from the power supply voltages Vdc_a and Vdc_b and rto_pa. Here, rto_pb is calculated based on the above formula.

得られた変調率オフセットma_offset0, mb_offset0は、加算器209と210で、それぞれ電源10a分瞬時変調率指令ｍu_a*、電源10b分瞬時変調率指令ｍu_b*と加算する。電源10a分については、乗算器211で補正ゲインを乗じ最終的な変調率指令ｍu_a_c*を求める。電源10b分は、上記の加算から、変調率指令ｍu_b_c*を求める。これを式で表すと、以下のようになる。 rto_pb = 1 − rto_pa

The obtained modulation factor offsets ma_offset0 and mb_offset0 are added by the adders 209 and 210 to the instantaneous modulation factor command mu_a * for the power source 10a and the instantaneous modulation factor command mu_b * for the power source 10b, respectively. For the power supply 10a, the multiplier 211 multiplies the correction gain to obtain a final modulation rate command mu_a_c *. For the power supply 10b, the modulation rate command mu_b_c * is obtained from the above addition. This is expressed as follows.

mu_a_c * = (mu_a * + ma_offset *) ・ Gcmp_u_a
mu_b_c * = mu_b * + mb_offset *
PWM pulse generation means 44
In FIG. 4, a power supply 10a carrier is a triangular wave carrier for generating a PWM pulse for driving each switch in order to output a voltage pulse from the voltage Vdc_a of the power supply 10a. Similarly, a triangular wave carrier is used as the power supply 10b carrier. Is provided. These two triangular wave carriers have values of upper limit +1 and lower limit −1, and have a phase difference of 180 degrees. Here, the signals for driving the U-phase switches are set as follows based on FIG.

A: Switch drive signal conducting from the power supply 10a to the output terminal B: Switch drive signal conducting from the output terminal to the negative electrode C: Switch drive signal conducting from the output terminal to the power supply 10a D: Power supply Switch drive signal E conducting from 10b to output terminal: Switch drive signal conducting from output terminal to power supply 10b First, a pulse generation method when a voltage pulse is output from the power supply 10a will be described. When outputting a PWM pulse from the power supply 10a, it is necessary to turn on A. When there is a potential difference between the positive electrode and the positive electrode and Vdc_a> Vdc_b, when both A and E are turned on, a current that short-circuits between the positive electrodes flows. For example, when A is switched from on to off at the same time and E is switched from off to on, it takes time for A to completely turn off. A short-circuit current flows, and the amount of heat generated by the semiconductor switch installed in this path increases. In order to prevent such an increase in heat generation, A and E are switched from OFF to ON after a time during which both the drive signals A and E are turned off. In this way, pulse generation is performed by adding a short-circuit prevention time (dead time) to the drive signal. Similar to adding dead time to the drive signals of A and E, dead time is added to E and C, and in order to prevent a short circuit between the positive and negative electrodes, dead to A and B and E and B. Add time.

A method for adding dead time to the A and E drive signals will be described below with reference to FIG. In order to generate a drive signal with a dead time added, mu_a_c_up * and mu_a_c_down * offset from the mu_a_c * by the dead time are obtained as follows.

mu_a_c_up * = mu_a_c * + Hd
mu_a_c_down * = mu_a_c * − Hd
Here, Hd is obtained from the amplitude of the triangular wave (from the base to the apex) Htr, the period Ttr, and the dead time Td as follows.

Hd = 2Td · Htr / Ttr
The carrier and mu_a_c *, mu_a_c_up *, mu_a_c_down * are compared, and the drive signals for the A and E switches are obtained according to the following rules.

If mu_a_c_down * ≥ carrier for power supply 10a, A = ON
If mu_a_c * ≤ carrier for power supply 10a, A = OFF
If mu_a_c * ≧ carrier for power supply 10a, E = OFF
mu_a_c_up * ≤ Carrier for power supply 10a E = ON
Thus, by generating the drive signal, a dead time of Td can be provided between A and E, and a short circuit between the positive electrodes can be prevented.

The pulse generation method for outputting voltage pulses from the power supply 10b is the same as that of the power supply 10a, and the following mu_b_c_up * and mu_b_c_down * are obtained and compared with the carrier for the power supply 10b. (Fig. 9)
mu_b_c_up * = mu_b_c * + Hd
mu_b_c_down * = mu_b_c * − Hd
The drive signals for the D and C switches are obtained according to the following rules.

If mu_b_c_down * ≧ carrier for power supply 10b, D = ON
If mu_b_c * ≤ carrier for power supply 10b, D = OFF
If mu_b_c * ≧ carrier for power supply 10b, C = OFF
If mu_b_c_up * ≤ carrier for power supply 10b, C = ON
In this way, a dead time of Td can be provided between D and C, and a short circuit between the positive electrodes can be prevented. The drive signal B is generated from AND (logical product) of the generated drive signals E and C.

B = E ・ C
E is a drive signal with a dead time added to A, and C is a drive signal with a dead time added to D. For this reason, by generating B from the AND (logical product) of E and C, dead time can be generated for B and A and B and E as well. An example of pulse generation with a dead time added is shown in FIG. Based on the PWM pulse generated in this way, each switch of the power converter is driven on and off to generate an output voltage pulse. Taking the average of the voltage pulse generated from the voltage Vdc_a of the power supply 10a and the voltage pulse generated from the voltage Vdc_b of the power supply 10b for each period, the original three-phase voltage command values vu *, vv *, vw * Thus, a voltage pulse that realizes is generated.

Here, a method for selecting the route of the return current of the motor using root_sw will be described. The drive signals E and C are newly generated using the drive signals E and C calculated by the above-described pulse generation method and the root_sw signal.

When the root_sw signal is off (false), E and C output the drive signals E and C calculated by the above-described pulse generation method as they are.

When the root_sw signal is on (true), the drive signals E and C are replaced with E = on and C = off.

Thereby, if the root_sw signal is on (true), the semiconductor switches 101b, 102b, 103b connected to the positive bus of the power supply 10a are turned off, and the semiconductor switches 104b, 105b connected to the positive bus of the power supply 10b are turned off. 106b are turned on. When the motor phase current is negative, the phase current does not flow to the positive bus of the power supply 10a, but flows to the positive or negative bus of the power supply 10b. By using such path selection and the power distribution target value, it is possible to perform power distribution between the power supply 10a and the power supply 10b. In this power distribution, the power source 10b can be charged while the motor is powered by the power source 10a.

Furthermore, when charging from such a power source 10a to the power source 10b, if the route selection is not used, charging is performed by setting the power distribution target value of the power source 10b to be negative. The charging operation is possible even in the region where the power distribution target value of the power supply 10b is positive.

Based on the voltage command value divided using the power distribution target value of the power supply, voltage pulses are output alternately every PWM cycle, so if one power supply distribution target value is negative, Negative voltage pulses are alternately output from the respective power supplies, and a large ripple current having a frequency close to the PWM carrier frequency flows. On the other hand, when route selection is used, even when charging (regeneration), the charging operation can be performed while the power distribution target value on the charging side remains in the positive region. Since positive voltage pulses are alternately output from the respective power supplies, it becomes possible to reduce a ripple current having a frequency close to the PWM carrier frequency.

FIG. 16 shows the motor phase current when an experiment in which the power supply 10b is charged with the power distribution target value being negative is performed. FIG. 17 shows the motor phase current when the power supply 10b is charged with the same charging power as in FIG. 16 by using the path selection. In this way, by using the path selection, it is possible to reduce the ripple current having a frequency close to the PWM carrier frequency, and to reduce the motor loss and noise. In the case of performing such a current path selection, in this embodiment, the correction gain and the corrected power distribution target value are calculated. First, the correction gain calculation method and its effect will be described below. Three-phase alternating current command values iu *, iv * and iw * are obtained from id * and iq * by dq / 3-phase coordinate conversion 46 and input to correction gain calculator 47. Further, Vdc_a, Vdc_b, and root_sw signals are input to the correction gain calculator.

The U phase of the calculation by the correction gain calculator 47 will be described with reference to the flowchart of FIG. The same calculation is performed for the V phase and the W phase. First, using a route selection signal, it is determined whether root_sw = 1, that is, whether route selection is performed. When the condition is not satisfied, Gcmp_u_a = 1 is set and the calculation is terminated. When the condition of root_sw = 1 is satisfied, next, it is determined whether the U-phase current command value iu * is positive or negative. When the condition of iu * <0 is not satisfied, Gcmp_u_a = 1 is set and the calculation is terminated. When the condition of iu * <0 is satisfied, Gcmp_u_a = Vdc_a / Vdc_b is calculated, and the calculation of the correction gain calculator 47 is ended. The calculation of the correction gain calculator 47 is performed every PWM cycle.

In the calculation of the correction gain Gcmp_u_a, when the path is selected, if the current is negative, the current flows only through the semiconductor switch between the positive and negative electrodes of the power supply 10b. Therefore, the correction gain Gcmp_u_a is calculated using the voltage Vdc_a of the power supply 10a. The calculation is performed so as to correct the modulated rate to the voltage Vdc_b of the actually flowing path.

By multiplying the correction gain, it becomes possible to apply a voltage pulse corresponding to the voltage command value of the motor to the motor even when there is a difference between Vdc_a and Vdc_b when performing path selection. Control that suppresses the generation of harmonic current of the current becomes possible. FIG. 16 shows a U-phase current waveform in an experiment in which a path is selected without using a correction gain, and an error between a voltage command value and an actually applied voltage pulse due to a voltage difference between two power supply voltages. And a harmonic current is generated in the current. On the other hand, FIG. 17 shows a U-phase current waveform when the correction gain of the present embodiment is used. The harmonic current generated in FIG. 16 can be greatly reduced, and the torque ripple and loss of the motor can be reduced.

Next, the calculation method and effect of the corrected power distribution calculation 48 will be described. When performing route selection, as described above, the power supply 10b can be charged even when the power distribution target value remains positive. That is, a difference occurs between the power distribution target value and the actual power distribution. In the present invention, by using the modified power distribution calculation 48, the power distribution command is used as a command value for power distribution control, and the power distribution target value so far is replaced with the manipulated variable, so that the actual power distribution difference described above is obtained. The details will be described below.

The corrected power distribution calculation 48 performs the following calculation with id, iq, vd *, vq *, Vdc_a, Vdc_b, rto_pa ”, root_sw as inputs.

電力配分指令rto_pa”と電力配分目標値rto_paは次の関係になる。

ma_offset0は、前述のように、Ｖdc_a, Ｖdc_b ,rto_pa, rto_pbの関数であり、このrto_pbはrto_paで表すことができる。rto_pa”の、ma_offset0をＶdc_a, Ｖdc_b ,rto_pa で記述しなおすことで、上式はrto_paについて解くことができる。すなわち、rto_paをrto_pa”の関数Fとして記述することができる。

この演算を行って、図２における電力配分目標値rto_paを出力する。 Output power Pe when root_sw signal is on (true)
Pe = vd * ・ id + vq * ・ iq
Phase current amplitude Ipk

The power distribution command rto_pa ”and the power distribution target value rto_pa have the following relationship.

As described above, ma_offset0 is a function of Vdc_a, Vdc_b, rto_pa, rto_pb, and this rto_pb can be represented by rto_pa. By rewriting ma_offset0 of rto_pa by Vdc_a, Vdc_b, and rto_pa, the above equation can be solved for rto_pa. That is, rto_pa can be described as a function F of rto_pa ”.

This calculation is performed to output the power distribution target value rto_pa in FIG.

When the root_sw signal is off (false), the power distribution target value rto_pa is output as follows.

rto_pa = rto_pa ”
Note that the above-described problem does not occur when a route is not selected. Therefore, power distribution target value = distribution command. By the corrected power distribution calculation 48, the power Pa of the power source 10a and the power Pb of the power source 10b can be expressed by the following relationship between the power Pe and the power distribution command rto_pa ”.

Pa = rto_pa ”・ Pe
Pb = (1-rto_pa ”) ・ Pe
The effects of this embodiment are summarized as follows. When performing route selection, by determining the power distribution target value rto_pa using the power distribution command rto_pa ”, it becomes possible to follow the command value for the distribution of power supply power.

In the present embodiment, an example in which the route selection for selecting the power supply 10b side by root_sw is shown, but the power supply 10a side can also be selected. At this time, in the correction gain / corrected power distribution calculation shown in the present embodiment, each parameter may be replaced in consideration of the case where the power supply 10a and the power supply 10b are replaced.

Further, in this embodiment, the circuit configuration has a common negative electrode bus of the power supply, but may have a configuration having a common positive bus. Also in this case, the route of the power source can be selected, and the control according to the present invention can be applied.

In a fuel cell electric vehicle in which the power conversion device of the present invention is applied to drive a motor, the power source 10a is a fuel cell and the power source 10b is a battery. When the battery is being charged, the route is selected, and the command value for the charge energy is good by instructing the power distribution command to the corrected power distribution calculation 48 from the command value and the output power for the charge energy at this time. It is possible to perform power distribution control following the above. Further, since the harmonic current close to the carrier frequency can be reduced by route selection, motor loss and noise can be reduced. Even when there is a voltage difference between the battery voltage and the fuel cell, torque ripple and loss can be reduced without using a large harmonic current in the motor phase current by using the correction gain of the present invention.

Second Embodiment In the second embodiment, only the difference from the first embodiment is shown. FIG. 9 shows a circuit diagram of the power conversion device according to the present embodiment. In this embodiment, the power converter 30a is composed of switch means 30Ua, 30Va, and 30Wa formed of a plurality of semiconductor switches and diodes for each of the U-phase, V-phase, and W-phase. The semiconductor switch section capable of controlling bidirectional conduction in the first embodiment is constituted by a semiconductor switch and a diode as shown in FIG. The U-phase switch means 30Ua includes semiconductor switches 101as, 101bs, 104as, 104bs, 107a and diodes 101ad, 101bd, 104ad, 104bd, 107b. The V-phase switch means 3Va is composed of semiconductor switches 102as, 102bs, 105as, 105bs, 108a and diodes 102ad, 102bd, 105ad, 105bd, 108b. The W-phase switch means 30Wa includes semiconductor switches 103as, 103bs, 106s, 106bs, 109a and diodes 103as, 103bd, 106ad, 106bd, 109b. In such a configuration, for example, since the diode 101bd has resistance against a voltage in the reverse direction, a semiconductor element that does not have a sufficient reverse breakdown voltage can be used for the semiconductor switch 101as.

The configuration and operation of the control device 40a according to the second embodiment will be described with reference to FIG.

Correction Gain Calculation The correction gain calculator 47a receives the detected current values iu, iv as inputs, performs the following calculation, and estimates the W-phase current value iw.

iw =-iu-iv
The correction gain calculator 47a performs a calculation by replacing the current command value iu * in the flow of the correction gain calculator 47 in the first embodiment with iu, and outputs a correction gain. By multiplying this correction gain, it is possible to apply a voltage pulse corresponding to the voltage command value of the motor to the motor even when there is a difference between Vdc_a and Vdc_b when performing path selection. Control that suppresses generation of harmonic currents of the phase current is possible. In addition, since the detected current value is used, the follow-up to the command value by the current control is slow, and even when the difference between the actual motor phase current value and the command value is large, by performing the correction gain calculation of this embodiment, Similar effects can be obtained.

Next, the calculation method of the corrected power distribution calculation 48a and the effect thereof will be described. In the corrected power distribution calculation 48a, the following calculation is performed with Te *, ω, Vdc_a, Vdc_b, rto_pa ″, and root_sw as inputs.

When the root_sw signal is on (true)
From T *, ω, Vdc_a, Vdc_b, rto_pa ”, a map created in advance based on the experimental result is referred to, and a power distribution target value rto_pa is output.

When the root_sw signal is off (false), the power distribution target value rto_pa is output as follows.

rto_pa = rto_pa ”
When performing route selection, the power distribution target value rto_pa is obtained using the power distribution command rto_pa ”, so that the power supply power distribution can be made to follow the command value. It is only necessary to refer to the map, and the amount of calculation can be reduced, especially when the power converter is controlled using a microcomputer, because the calculation time can be reduced, so that it can be implemented with an inexpensive microcomputer. .

Third Embodiment In the third embodiment, only differences from the first embodiment and the second embodiment are shown. FIG. 10 shows a circuit diagram of the power converter in the present embodiment. In this embodiment, the power converter 30b includes switch means 30Ub, 30Vb, and 30Wb each formed of a plurality of semiconductor switches and diodes for each of the U phase, the V phase, and the W phase. This circuit has a configuration in which the number of semiconductor switches and diodes in FIG. 9 of the second embodiment is reduced. When there is no path connecting the motor terminal power source to the positive electrode bus 14 of the power source 10a and the motor current is negative, no current flows to the positive electrode bus 14 of the power source 10a. This represents a circuit configuration similar to that in the circuit configuration in the case of performing route selection with root_sw = 1 in the circuits of FIGS. The power converter having this circuit configuration is always controlled with root_sw = 1 using the controller 40 of the first embodiment or the controller 40a of the second embodiment.

According to this configuration, the number of semiconductor switches of the power converter can be reduced, and the control according to the present invention enables the control that suppresses the generation of the harmonic current of the motor phase current.

When the configuration of this embodiment is applied to a fuel cell electric vehicle, when the power source 10a is a fuel cell and the power source 10b is a battery, charging to the fuel cell is prevented by the diodes 101bd, 102bd, 103db, and the semiconductor With a circuit configuration in which the number of elements is reduced, it is possible to perform power distribution and control driving of the motor. Thereby, the cost of the power converter can be reduced, and the volume can be reduced.

FIG. 11 shows a modification of the second embodiment for the case where the power source 10a is a fuel cell. In this modification, the power converter 30c is provided for each of the U phase, the V phase, and the W phase. The switch means 30Uc, 30Vc, and 30Wc are composed of a plurality of semiconductor switches and diodes. In the circuit configuration of this modification, a diode 110 is provided instead of the diodes 101bd, 102bd, and 103db, and the circuit can be further simplified.

Although the present invention has been described based on the drawings and examples, it should be noted that those skilled in the art can easily make various modifications and corrections based on the present disclosure. Therefore, it should be noted that these variations and modifications are included in the scope of the present invention. In the embodiments and the like, the power conversion device using two power sources has been described, but the present invention can also be applied to a power conversion device including three or more power sources, and similar effects can be obtained. .

10,10a, 10b Power supply
12,13 Smoothing capacitor
14 Positive bus of power supply 10a
15 Common negative electrode bus
16 Power supply 10b positive bus
20 Motor
30 Power converter
40 Control unit
41 Torque control means
42 Current control means
43 Power control / modulation rate calculation means
44 PWM pulse generation means
45 Three-phase / dq conversion means
46 dq / 3-phase coordinate conversion means
47 Correction gain calculator
48 Modified power distribution calculation means
49 Modulation rate correction means
30U, 30V, 30W U phase, V phase, W phase switch means (Example 1)
30Ua, 30Va, 30Wa U phase, V phase, W phase switch means (Example 2)
30Ub, 30V, 30Wb U phase, V phase, W phase switch means (Example 3)
30Uc, 30Vc, 30Wc U-phase, V-phase, W-phase switch means (modified example of embodiment 3)
101a / 101b, 102a / 102b, 103a / 103b Semiconductor switch
104a / 104b, 105a / 105b, 106a / 106b semiconductor switch
107a, 108a, 109a Semiconductor switch
107b, 108b, 109b Diode
110 diodes
101as, 101bs, 104as, 104bs, 107a Diode
102as, 102bs, 105as, 105bs, 108a Semiconductor switch
102ad, 102bd, 105ad, 105bd, 108b Diode
103as, 103bs, 106s, 106bs, 109a Semiconductor switch
103as, 103bd, 106ad, 106bd, 109b Diode

Claims (5)

Multiple DC power supplies ,
Which is connected to a DC power source, a pulse Rupa pulse generating and combining means generates and synthesizing a plurality of DC voltage outputs of the DC power source,
A control device for controlling the pulse generation / synthesis means;
An AC motor connected to the pulse synthesizing / generating means and driven by the pulses output by the pulse synthesizing / generating means;
The controller is
A current control means for calculating a current command value of the AC motor from a torque command value and a motor rotation speed given from the outside, and calculating a voltage command value based on the calculated current command value;
The voltage command value of each DC power supply is calculated based on the voltage command value calculated by the current control means, the voltage value of the DC voltage, and the distribution target value of the output power of the DC power supply, and the calculated voltage of the DC power supply Drive pulse generating means for driving the pulse generating / synthesizing means based on a command value to determine a conduction path between the plurality of DC power supplies and the AC motor ;
In the case where the presence / absence of selection of the return current path of the AC motor is determined and the selection of the path is performed, in the pulse generation / synthesis unit, either one of the plurality of DC power supplies and the AC motor are selected. Power distribution means for allocating the output power of the pulse generating / combining means to the output power of a plurality of DC power sources according to the actual conduction path when there is a path in which the conduction of the reflux current between them exists. A power conversion device comprising: The power conversion device according to claim 1,
The power distribution means is
According to the actual conduction path, the distribution target value is calculated from at least the distribution ratio command value of the output power of the DC power supply, the output power value, the motor current amplitude, and the plurality of DC power supply voltage values, and the AC Adjusting the output power of the plurality of DC power sources by determining the output voltage command value of the power converter from each DC power source from the output voltage command value to the motor and the distribution target value. A power conversion device. The power conversion device according to claim 2,
The power distribution means is
The output power to the AC motor is calculated from the motor current value and the output voltage command value to the motor.
The power converter characterized by the above-mentioned. The power conversion device according to claim 1,
The power distribution means is
According to the actual conduction path, the distribution target value is determined from at least the distribution ratio command value of the output power of the DC power supply, the rotational speed of the AC motor, the torque of the AC motor, and the plurality of DC power supply voltage values. calculated, and the output voltage command value for the alternating-current motor from said distribution target value, by determining the output voltage command value of the power converter to be outputted from each of the power supply, outputs of the plurality of DC power supply Adjust power,
The power converter characterized by the above-mentioned. In the power converter device according to any one of claims 2 to 4,
The power distribution means is
When calculating the distribution target value, the calculation is performed with reference to a map created in advance.
The power converter characterized by the above-mentioned.

Images