JP2002171757A - Switching power circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To improve an image bend in a white peak condition. SOLUTION: This switching power circuit, comprising a self-excited or separately-excited oscillating switching frequency control mode voltage resonant converter on a primary side, and a complex resonant converter, including a voltage resonant circuit on a secondary side, is structured so that the inductance of a controlled winding NR is variably controlled by supplying the control signal, in response to the level and ripple voltages of a high-voltage DC output voltage to a control winding NC2 of an orthogonal control transformer PRT-2. The control winding NC2 of the orthogonal control transformer PRT-2 is formed by winding a plurality of wires NC21, NC22, NC23 in bundle for maintaining a magnetomotive force, thereby decreasing the time constant. Thus, high-speed transient response characteristics are improved.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a switching power supply circuit suitable for, for example, a television receiver.

[0002]

2. Description of the Related Art In a television receiver, a projector, or the like, a cathode ray tube (hereinafter referred to as C) is used as a display device.
RT (Cathode Ray Tube)) is widely used. For example, FIG. 6 shows an example of a switching power supply circuit that can be mounted on various display devices provided with a CRT, proposed as a prior art by the present applicant.

In the circuit of FIG. 6 as a prior art, first,
As a rectifying and smoothing circuit for receiving a commercial AC power supply (AC input voltage VAC) to obtain a DC input voltage, a full-wave rectifying and smoothing circuit including a bridge rectifying circuit Di and a smoothing capacitor Ci is provided. A rectified smoothed voltage (DC input voltage) Ei corresponding to the double level is generated.

[0004] The switching converter that receives the DC input voltage Ei and is intermittent is provided with a single switching element Q.
1 and a voltage resonance type converter that performs a switching operation in a so-called single-ended manner by a self-excited system. In this case, the switching element Q1
A high breakdown voltage bipolar transistor (BJT; junction transistor) is used.

The base of the switching element Q1 is connected to the positive electrode of the smoothing capacitor Ci via a current limiting resistor RB and a starting resistor RS, and the emitter is grounded to the primary side ground. Further, a series resonance circuit for self-excited oscillation driving, which is composed of a series connection circuit of a drive winding NB, a resonance capacitor CB, and a base current limiting resistor RB, is connected between the base of the switching element Q1 and the primary side ground. The clamp diode DD1 inserted between the base of the switching element Q1 and the negative electrode (primary ground) of the smoothing capacitor Ci forms a path for a clamp current flowing when the switching element Q1 is turned off. Switching element Q
The first collector is connected to one end of a primary winding N1 formed on the primary side of the isolated converter transformer PIT,
Its emitter is grounded.

A parallel resonance capacitor Cr1 is connected in parallel between the collector and the emitter of the switching element Q1. This parallel resonance capacitor Cr1 is
The primary parallel resonance circuit of the voltage resonance type converter is formed by its own capacitance and the leakage inductance L1 of the primary winding N1. Although not described in detail here, when the switching element Q1 is turned off, the voltage V1 across the resonance capacitor Cr1 due to the operation of the primary side parallel resonance circuit actually has a sinusoidal pulse waveform. Thus, a voltage resonance type operation is obtained.

The orthogonal control transformer PRT-1 is a saturable reactor in which a resonance current detecting winding ND, a driving winding NB, and a control winding NC1 are wound. The orthogonal control transformer PRT-1 is provided for driving the switching element Q1 and for controlling the constant voltage.

In this case, the orthogonal control transformer PRT-1 is used.
Is inserted in series between the positive electrode of the smoothing capacitor Ci and the primary winding N1, so that the switching output of the switching element Q1 is connected to the resonance current through the primary winding N1. It is transmitted to the detection winding ND. In the orthogonal control transformer PRT-1, the switching output obtained in the resonance current detection winding ND is induced in the drive winding NB via the transformer coupling, so that the drive winding NB has an alternating voltage as a drive voltage. Voltage is generated. This drive voltage is applied to a series resonance circuit (N
B, CB) is output to the base of the switching element Q1 as a drive current via the base current limiting resistor RB. As a result, the switching element Q1 performs a switching operation at a switching frequency determined by the resonance frequency of the series resonance circuit.

[0009] Insulation converter transformer (Power Isolation
Transformer) PIT transmits the switching output of switching element Q1 to the secondary side. The winding start end of the primary winding N1 of the insulating converter transformer PIT is connected to the collector of the switching element Q1, and the winding end is connected to the smoothing capacitor Ci through the series connection of the resonance current detection winding ND.
Is connected to the positive electrode. Also, on the secondary side, the first
, And a secondary winding N2
A tertiary winding N3 is provided as a second secondary winding formed by winding up the winding end of the winding. The secondary side parallel resonance capacitor C is connected to the secondary side winding (N2 + N3) including the secondary winding N2 and the tertiary winding N3.
2 are connected in parallel.

In this case, the winding start end of the secondary winding N2 is connected to the secondary side ground, and the winding end is connected to the rectifier diode D.
Connected to the anode of O1. A half-wave rectifying / smoothing circuit composed of the rectifying diode DO1 and the smoothing capacitor CO1 obtains a DC output voltage EO1 (for example, 135 V) for horizontal deflection of 110V to 140V.

A tap is provided at a required position of the secondary winding N2, and a rectifier diode DO2
The anode is connected. Then, a half-wave rectifying and smoothing circuit including the rectifying diode DO2 and the smoothing capacitor CO2 provides a DC output voltage EO2 for a signal system circuit.
(15V).

Further, on the secondary side of the insulated converter transformer PIT, the ending of the tertiary winding N3 wound around the secondary winding N2 is connected to the anode of the rectifying diode DO3, thereby connecting the rectifying diode DO3 to the rectifying diode DO3. Although a DC output voltage EO3 (200 V) for the video output circuit is obtained by a half-wave rectifier circuit including the smoothing capacitor CO3, the negative electrode of the smoothing capacitor CO3 is connected to the smoothing capacitor CO1.
Is connected to the positive electrode side of the capacitor, so that the smoothing capacitors CO1-CO3
The DC output voltage EO3 for the video output circuit is obtained from both ends of the series connection circuit. That is, in order to obtain the DC output voltage EO3 for the video output circuit, the DC output voltage EO1 generated across the smoothing capacitor CO1 is added to the smoothing capacitor C01.
The DC output voltage generated at both ends of O3 is accumulated, that is, the DC output voltage EO1 obtained from the secondary winding N2 and the DC output voltage obtained from the tertiary winding N3 are superimposed to obtain the DC output voltage EO3. ing. For this reason, the configuration of the rectifying and smoothing circuit including the tertiary winding N3, the rectifying diode DO3 and the smoothing capacitor CO3 includes a DC output voltage EO3.
(200 V), the DC output voltage EO1 (110 V to 14
0 V) is subtracted and a DC output voltage of 90 V to 60 V can be obtained.

A secondary side parallel resonance capacitor C2 is connected in parallel to a secondary side winding (N2 + N3) composed of a secondary winding N2 and a tertiary winding N3, so that a secondary side winding is provided. (N2 +
N3), the secondary side parallel resonance circuit is formed by the leakage inductance (L2 + L3) and the capacitance of the secondary side parallel resonance capacitor C2, and the alternating voltage induced in the secondary winding N2 and the tertiary winding N3 is the resonance voltage. The voltage resonance operation is obtained on the secondary side of the insulating converter transformer PIT.

That is, in the power supply circuit shown in FIG. 6, the primary side is provided with a parallel resonance circuit for performing a voltage resonance type switching operation, and the secondary side is provided with a parallel resonance circuit for obtaining a voltage resonance operation. Be provided. In the present specification, a switching converter having a configuration in which a resonance circuit is provided for the primary side and the secondary side and operates as described above is described below.
It is also referred to as a “composite resonance type switching converter”.

The DC output voltage EO1 is also branched and input to the control circuit 1. The control circuit 1 is constituted by, for example, an error amplifier or the like, and changes the orthogonal control transformer PRT-1 according to a change in the DC output voltage level EO1 output from the secondary side of the insulating converter transformer PIT.
, The inductance LB of the drive winding NB wound around the orthogonal control transformer PRT-1 is variably controlled by varying the level of the control current (DC current) supplied to the control winding NC1.
As a result, the resonance condition of the series resonance circuit in the self-excited oscillation drive circuit for the switching element Q1 formed including the inductance LB of the drive winding NB changes, and the switching frequency of the switching element Q1 is varied. Become. This operation stabilizes the DC output voltage output from the secondary side of the insulating converter transformer PIT.

The orthogonal control transformer PRT- which variably controls the inductance LB of the drive winding NB as described above.
When the switching element Q1 is provided, a period TOFF during which the switching element Q1 is turned off in varying the switching frequency.
Is kept constant, and the ON period TON is variably controlled. That is, in this power supply circuit, as a constant voltage control operation, the switching frequency is variably controlled to perform resonance impedance control on the switching output, and at the same time, the conduction angle control (PWM control) of the switching element Q1 in the switching cycle is also performed. You can see what you are doing. This complex control operation is realized by a set of control circuit systems. In the present specification, such complex control is also referred to as “complex control method”.

Further, in the power supply circuit shown in FIG. 6, a series resonance capacitor C3 and a primary winding N of the high voltage generating transformer HVT are connected in parallel with the secondary side parallel resonance capacitor C2.
4. A series resonance circuit comprising a series connection of the controlled winding (main winding) NR of the orthogonal control transformer PRT-2 is provided. That is, the secondary winding (N2) of the insulating converter transformer PIT is connected to the secondary parallel resonance capacitor C2.
+ N3) and the series resonance capacitor C3-the primary winding N4-
The series resonance circuits composed of the controlled windings NR are connected in parallel.

In the power supply circuit having such a configuration, the isolated converter transformer PIT operates as a complex resonance type switching converter, so that a resonance pulse voltage is generated across the secondary side parallel resonance capacitor C2. Then, a DC output voltage EO3 is obtained from a positive resonance pulse voltage generated in the insulating converter transformer PIT, and a resonance pulse voltage generated in the secondary-side parallel resonance capacitor C2 is converted to a primary voltage of the high-voltage generation transformer HVT through the series resonance capacitor C3. The input is made to the side winding N4. In this case, on the primary side of the high-voltage generating transformer HVT,
Series resonance capacitor C3-primary side winding N4-controlled winding N
Since the series resonance circuit composed of R is formed, the resonance pulse voltage V2 generated at both ends of the secondary parallel resonance capacitor C2 depends on the capacitance of the DC resonance capacitor C3 and the primary winding N4 of the high voltage generation transformer HVT. , And the current I4 flowing through the primary winding N4 of the high-voltage generating transformer HVT and the voltage V4 across the primary winding N4 by the series resonance operation of the inductance LR of the controlled winding NR of the orthogonal control transformer PRT-2. Both have substantially sinusoidal resonance waveforms.

The orthogonal control transformer PRT-2 is a saturable reactor in which the controlled winding NR and the control winding NC2 are wound, and outputs a high-voltage DC output voltage EHV output from a high-voltage generation circuit 4 described later. Provided for constant voltage control.
The structure of the orthogonal control transformer PRT-2 will be described later. The orthogonal control transformer PRT-2 includes a control winding N
According to the level of the control current (DC current) IC flowing through C2, the inductance LR of the controlled winding NR changes within a range of, for example, 50 μH to 10 μH. That is, the orthogonal control transformer PRT-2 is provided with a high-voltage generating transformer HV in order to make the high-voltage DC output voltage EHV a constant voltage.
The variable inductance of the controlled winding NR connected in series with the primary winding N4 of T controls the inductance of the series resonance circuit formed on the primary side of the high voltage generating transformer HVT. It is said.

High-voltage generating circuit 4 enclosed by a dashed line
Is composed of a high-voltage generating transformer HVT and a high-voltage rectifier circuit.
4 to raise the resonance voltage V4 to generate a high voltage corresponding to, for example, the anode voltage level of the CRT. Therefore, on the secondary side of the high-voltage generating transformer HVT, five sets of step-up windings NHV (NHV1 to NHV5) are divided and wound by slit winding or interlayer winding. In this case, the primary winding N4 and each boost winding NHV are wound so as to be tightly coupled, and are wound so that their polarities (winding directions) are opposite. Therefore, on the secondary side of the high voltage generating transformer HVT, the negative resonance voltage of the resonance voltage V4 generated in the primary winding N4 is inverted, and the winding ratio of the boost winding NHV and the primary winding N4 ( NHV / N4) to obtain a boosted voltage.

The high voltage rectifier diodes DHV1, DHV2, DHV3,
The anode sides of DHV4 and DHV5 are connected. And
The cathode of the high-voltage rectifier diode DHV1 is connected to the positive terminal of the smoothing capacitor COHV, and the remaining cathodes of the high-voltage rectifier diodes DHV2 to DHV5 are connected to the step-up windings NHV1 respectively.
To NHV4. That is, on the secondary side of the high-voltage generating transformer HVT, [boost winding NHV1, high-voltage rectifier diode DHV1], [boost winding NHV2, high-voltage rectifier diode DHV2], [boost winding NHV3, high-voltage rectifier diode DHV3], [ Boost winding NHV4, high voltage rectifier diode D
HV4], [Step-up winding NHV5, High-voltage rectifier diode DHV5]
That is, a so-called multisingle-type half-wave rectification circuit in which five sets of half-wave rectification circuits are connected in series is formed.

Therefore, on the secondary side of the high-voltage generating transformer HVT, five sets of half-wave rectifier circuits are provided by the boost windings NVH1 to NHH.
The operation of rectifying the current induced in V5 and charging the smoothing capacitor COHV is performed.
At both ends of the OHV, a high-voltage DC output voltage (anode voltage) EHV having a level corresponding to about five times the induced voltage induced in each of the boost windings NHV1 to NHV5 is obtained.

A capacitor COFV is inserted between the cathode of the high voltage rectifier diode DHV4 and the secondary side ground, and a high voltage DC output voltage having a voltage level lower than the anode voltage EHV obtained across the capacitor COFV. (Focus voltage) EFV is obtained.

Further, a series circuit including a resistor R1 and a resistor R2 is connected between the cathode of the high voltage rectifier diode DHV1 and the secondary side ground, and a voltage divided by the resistors R1 and R2 is used as a control circuit. 2 is input. The control circuit 2 is also formed of, for example, an error amplifier, and controls the control current (DC current) flowing through the control winding NC2 of the orthogonal control transformer PRT-2 in accordance with the level change of the voltage obtained from the high-voltage DC output voltage EHV. By varying the level IC,
The inductance LR of the controlled winding NR is variably controlled. The anode voltage EHV and the focus voltage EFV are made constant by varying the current I4 flowing through the primary winding N4 of the high voltage generating transformer HVT.

FIG. 7 shows operation waveforms of the power supply circuit. FIGS. 7A to 7F show that the AC input voltage VAC is 100, for example.
FIG. 7 (g) to FIG. 7 (l) are operating waveforms when the high voltage DC output voltage EHV = 30.5 KV and the high voltage current IHV = 4 mA.
Is an operation waveform when, for example, the AC input voltage VAC is 100 V, the high-voltage DC output voltage EHV is 30.5 KV, and the high-voltage current IHV is 0 mA.

When the high-voltage load of the high-voltage generating circuit 4 is set to the maximum load power, the switching frequency of the switching element Q1 is controlled so as to be, for example, 90.9 kHz, and the actual ON / OFF period TON / OFF of the switching element Q1 is controlled. TOFF
Is 7.4 μs / 3.6 μs. The resonance voltage V1 generated across the parallel resonance capacitor Cr1 by the on / off operation of the switching element Q1 is shown in FIG.
As shown in (a), in the period TOFF during which the switching element Q1 is turned off, a sine-wave-like waveform is obtained, indicating that the operation of the switching converter is of the voltage resonance type. At this time, the switching element Q1 has a collector current ICP as shown in FIG.
Flows. For example, when the switching element Q1 is turned on, the clamp diode DD1, the switching element Q1
When the damper current (negative direction) flows through the primary winding N1 through the base-collector of the first embodiment, and the damper period (for example, 1.4 .mu.s) in which the damper current flows ends, the collector current IC changes from the negative level to the positive level The level will rise rapidly.

By such an operation, the secondary winding (N2 + N3) of the insulated converter transformer PIT is connected as shown in FIG.
As shown in FIG. 7D, the resonance current I2 flows, and the voltage V2 generated in the secondary-side parallel resonance capacitor C2, as shown in FIG.
Due to the operation of the rectifier diodes DO1, DO3, the voltage level becomes a positive voltage level of 200 V, and during a period TOFF during which the switching element Q1 is turned off, a negative resonance pulse voltage whose peak voltage level is 500 Vp Become. The resonance pulse voltage generated in the secondary resonance capacitor C2 forms a series resonance circuit with the primary winding N4 of the high voltage generation transformer HVT and the controlled winding NR of the orthogonal control transformer PRT-2. The resonance voltage V4 generated at both ends of the primary winding N4 by being input to the primary winding N4 via the series resonance capacitor C3 has a peak voltage level of 400 as shown in FIG.
Vp and the primary winding N
As shown in FIG. 7F, the resonance current I4 flowing through 4 has a resonance current waveform having a peak value of 2 Ap.

On the other hand, when the high voltage load of the high voltage generation circuit 4 is set to the minimum load power (no load), the switching element Q1
Is controlled to be, for example, 100 kHz, and the actual ON / OFF period TON / TOFF of the switching element Q1 is 6.4 μs / 3.6 μs. In this case, the resonance voltage V1 generated at both ends of the parallel resonance capacitor Cr1 by the on / off operation of the switching element Q1 is a sinusoidal wave during the period TOFF when the switching element Q1 is off, as shown in FIG. Is obtained. At this time, the collector current ICP as shown in FIG. 7 (h) flows through the switching element Q1. In this case, the period of the damper current flowing when the switching element Q1 is turned on is set to about 2 μs. It is longer than the damper period (1.4 μs) at the time of the maximum load power shown in (b).

By such an operation, the secondary winding (N2 + N3) of the insulated converter transformer PIT is connected to FIG.
As shown in FIG. 7 (j), the resonance current I2 flows, and the voltage V2 generated across the secondary-side parallel resonance capacitor C2 is as shown in FIG.
As shown in (i). The negative resonance pulse generated in the secondary resonance capacitor C2 is input to the primary winding N4 of the high voltage generating transformer HVT via the series resonance capacitor C3, and the waveform of the voltage V4 across the primary winding N4 is 7 (k), and the current waveform of the current I4 flowing through the primary winding N4 becomes a resonance waveform as shown in FIG. 7 (l).

FIG. 8 shows the structure of the orthogonal control transformer PRT-2. FIG. 8A is an external perspective view for explaining the overall structure, and FIG. 8B is a cross-sectional perspective view for explaining the winding direction of the wound winding. As shown in FIG. 8A, the orthogonal control transformer PRT-2 is a three-dimensional core 20 in which two double U-shaped cores 21 and 22 are combined.
Is formed by One double U-shaped core 2
1 has four magnetic legs 21a, 21b, 21c and 21d as shown in FIGS. 8 (a) and 8 (b). The other double U-shaped core 22 also has four magnetic legs 22 as shown in FIGS. 8A and 8B, for example.
a, 22b, 22c, and 22d. The three-dimensional core 20 is formed by joining the ends of the magnetic legs 21a to 21d and 22a to 22d of the two double U-shaped cores 21 and 22.

In this case, as shown in FIG. 8B, for example, the two magnetic legs 21 of the double U-shaped core 21 are used.
A control winding NC2 is wound around b and 21c, and a controlled winding NR is wound around magnetic legs 22c and 22d of the double U-shaped core 22. That is, the orthogonal control transformer PRT-2 shown in FIG. 8 is configured as a saturable reactor in which the control winding NC2 is wound in a direction orthogonal to the controlled winding NR. As the control winding NC2 of the orthogonal control transformer PRT-2, for example, 100 μm
It is wound 1000T (turn) by a single wire of φ, the DC resistance Rc = 155Ω, and the inductance Lc = 375mH. The time constant of the control winding NC is Lc / Rc = 2.4 m
H / Ω.

[0032]

As described above, the switching power supply circuit shown in FIG. 6 is designed to make the high-voltage DC output voltage constant with respect to the fluctuation of the AC input voltage or the high-voltage load power.
Controlled winding NR using orthogonal control transformer PRT-2
Is controlled. When a high-voltage power supply for a television receiver is constituted by such a voltage-regulating circuit, the anode current of the cathode-ray tube becomes 3.
If it is about 0 mA or less, the level of the image bending due to the white peak signal is a level without any problem, but if it is 4 mA or more, the level of the image bending deteriorates. This is the orthogonal control transformer PRT-
This is because the high-voltage DC output voltage EHV is controlled by the DC control current of the control winding NC2. As described above, the control winding NC has a single wire of 100 μm wound 1000T. Therefore, the DC resistance of the winding becomes 155Ω and the inductance becomes 375 mH. When the high-voltage load changes rapidly, the time constant of the control current becomes longer due to the DC resistance and inductance of the winding, and the high-speed transient response of the high-voltage DC output voltage EHV The characteristics have deteriorated.

More specifically, when the high-voltage stabilizing circuit is used as a high-voltage power supply for a television receiver, the average cathode current (Ik) is 2.
Although the current is limited so as not to flow more than 15 mA, as shown in FIG. 9, in the case of a white peak signal having a small vertical amplitude and a black signal in the periphery, the cathode current Ik is not limited and 8 mA
Peak current flows. Therefore, for a white peak signal having a high degree of modulation, the high voltage load power is 30.5 KV × 8 mA = 2
44 W, the high-voltage DC output voltage EHV decreases, and the amplitude in the horizontal direction increases. Therefore, the high-voltage DC output voltage EHV
The control circuit 2 must supply the excessive high-voltage load power from the AC input power supply AC by lowering the inductance LR so that the voltage does not decrease. However, due to the time constant of the control winding NC2 of the orthogonal control transformer PRT-2,
As shown in FIG. 9, the control current Ic of the control winding NC2 cannot change rapidly during the transition between the fall and the rise, and the amplitude of the high-frequency sine wave voltage V4 of the primary winding N4 of the high-voltage generating transformer HVT also varies with time. It changes with a constant. For this reason, for example, as shown in FIG. 10, if a rectangular white peak image originally indicated by a solid line is displayed, the image is distorted so as to have a trapezoidal shape as indicated by a broken line. .

[0034]

SUMMARY OF THE INVENTION The present invention has been made in view of the above-mentioned problems, and has as its object to improve the image bending by constructing a switching power supply circuit as follows.

For this reason, as the switching power supply circuit, switching means formed with a switching element for intermittently outputting a DC input voltage, and a primary-side parallel resonance circuit which makes the operation of the switching means a voltage resonance type are provided. A primary parallel resonance capacitor provided as formed, and having a structure in which a required coupling coefficient that is loosely coupled between the primary side and the secondary side is obtained, and an output of the switching means obtained on the primary side And a secondary-side voltage resonance circuit formed by connecting a secondary-side parallel resonance capacitor to a secondary-side winding of the insulation-converter transformer. A low-voltage DC output voltage is formed by performing a rectifying operation on an alternating voltage obtained from the secondary winding, which is formed including a side voltage resonance circuit. A low DC output voltage generating means which,
A primary winding connected in parallel with a secondary winding of the insulation converter transformer via a secondary series resonance capacitor connected in series to form a secondary current resonance circuit; A high-voltage generating transformer, which is configured to transmit a resonance voltage obtained on a secondary winding of a transformer from a primary side to a secondary side to obtain a high voltage obtained by boosting the resonance voltage from the secondary side, By performing a rectifying operation on a high-voltage obtained on the secondary side of the high-voltage generating transformer, a high-voltage DC output voltage generating means configured to obtain a high-voltage DC output voltage, and a control signal based on the low-voltage DC output voltage are used. Controlling the switching frequency and the conduction angle of the switching element to make the low-voltage DC output voltage a constant voltage; A ripple voltage detecting means for detecting a pull voltage; a controlled winding connected in series to a primary winding of the high-voltage generating transformer forming the secondary current resonance circuit; An orthogonal control transformer comprising a control winding for controlling the inductance, and variably controlling the inductance of the controlled winding based on a control signal based on the level of the high-voltage DC output voltage and the ripple voltage, And a second constant voltage means for converting the high-voltage DC output voltage into a constant voltage. The control winding of the orthogonal control transformer is configured to bundle and wind a plurality of wires.

According to the above configuration, the switching power supply circuit is formed by the voltage resonance type converter and the high voltage DC output voltage circuit for receiving the switching output of the voltage resonance type converter and generating the high voltage DC output voltage. Will be done. The voltage resonance type converter has a parallel resonance circuit on the primary side of the insulating converter transformer for voltage resonance type operation, and a resonance circuit on the secondary side. Take. The constant voltage control of the high-voltage DC output voltage is performed by variably controlling the inductance of the controlled winding of the orthogonal control transformer according to the level of the high-voltage DC output voltage and the ripple voltage. Here, as for the control winding of the orthogonal control transformer, a plurality of wires are bundled and wound to reduce the time constant while maintaining the magnetomotive force.

[0037]

FIG. 1 is a circuit diagram showing the configuration of a switching power supply circuit according to an embodiment of the present invention. The power supply circuit shown in this figure has a configuration as a complex resonance type switching converter including a voltage resonance type converter on the primary side and a parallel resonance circuit on the secondary side. In the power supply circuit shown in FIG. 1, first, a full-wave rectifier comprising a bridge rectifier circuit Di and a smoothing capacitor Ci as a rectifying and smoothing circuit for receiving a commercial AC power supply (AC input voltage VAC) to obtain a DC input voltage. A smoothing circuit is provided to generate a rectified smoothed voltage (DC input voltage) Ei corresponding to a level that is one time higher than the AC input voltage VAC.

The switching converter which inputs and outputs the DC input voltage Ei intermittently has a single switching element Q
1 and a voltage resonance type converter that performs a switching operation in a so-called single-ended manner by a self-excited system. In this case, the switching element Q1
A high breakdown voltage bipolar transistor (BJT; junction transistor) is used.

The base of the switching element Q1 is connected to the positive electrode of the smoothing capacitor Ci via the current limiting resistor RB and the starting resistor RS, and the emitter is grounded to the primary side ground. Further, a series resonance circuit for driving self-excited oscillation composed of a series connection circuit of a drive winding NB, a resonance capacitor CB, and a base current limiting resistor RB is connected between the base of the switching element Q1 and the primary side ground. The clamp diode DD1 inserted between the base of the switching element Q1 and the negative electrode (primary ground) of the smoothing capacitor Ci forms a path for a clamp current flowing when the switching element Q1 is turned off. Switching element Q
The first collector is connected to one end of a primary winding N1 formed on the primary side of the isolated converter transformer PIT,
Its emitter is grounded.

A parallel resonance capacitor Cr1 is connected in parallel between the collector and the emitter of the switching element Q1. This parallel resonance capacitor Cr1 is
The primary parallel resonance circuit of the voltage resonance type converter is formed by its own capacitance and the leakage inductance L1 of the primary winding N1. When the switching element Q1 is turned off, the voltage V1 generated across the resonance capacitor Cr1 due to the operation of the primary side parallel resonance circuit is actually a sinusoidal pulse waveform, and a voltage resonance type operation is obtained. To be.

The orthogonal control transformer PRT-1 is a saturable reactor in which the resonance current detection winding ND, the drive winding NB, and the control winding NC1 are wound. The orthogonal control transformer PRT-1 is provided for driving the switching element Q1 and for controlling the constant voltage. Although the illustration of the structure of the orthogonal control transformer PRT-1 is omitted,
A three-dimensional core is formed by joining the ends of the magnetic legs of two double U-shaped cores having four magnetic legs. Then, the resonance current detecting winding ND and the driving winding NB are wound in the same winding direction with respect to two predetermined magnetic legs of the three-dimensional core.
And the control winding NC1 is wound in a direction orthogonal to the resonance current detection winding ND and the drive winding NB.

In this case, the orthogonal control transformer PRT-1
Is inserted in series between the positive electrode of the smoothing capacitor Ci and the primary winding N1, so that the switching output of the switching element Q1 is connected to the resonance current through the primary winding N1. It is transmitted to the detection winding ND. In the orthogonal control transformer PRT-1, the switching output obtained in the resonance current detection winding ND is induced in the drive winding NB through the transformer coupling, so that the drive winding NB has an alternating voltage as a drive voltage. Voltage is generated. This drive voltage is applied to a series resonance circuit (N
B, CB) is output to the base of the switching element Q1 as a drive current via the base current limiting resistor RB. As a result, the switching element Q1 performs a switching operation at a switching frequency determined by the resonance frequency of the series resonance circuit.

An isolated converter transformer (Power Isolation)
Transformer) PIT transmits the switching output of switching element Q1 to the secondary side. As shown in FIG. 2, the insulation converter transformer PIT has an EE-type core in which E-type cores CR1 and CR2 made of, for example, a ferrite material are combined such that their magnetic legs face each other.
The primary winding N1 and the secondary winding N2 are wound around the center magnetic leg of the EE type core using the split bobbin B, respectively. A gap G is formed for the center magnetic leg as shown in the figure. Thereby, loose coupling with a required coupling coefficient is obtained. Gap G is the center magnetic leg of E-shaped cores CR1 and CR2.
It can be formed by making it shorter than the outer magnetic leg of the book. The coupling coefficient k is, for example, 0.8 to 0.9.
A degree of loose coupling is obtained, so that a saturated state is hardly obtained.

The winding start end of the primary winding N1 of the insulating converter transformer PIT is connected to the collector of the switching element Q1 as shown in FIG. 1, and the winding end is connected in series with the resonance current detection winding ND. Connected to the positive electrode of the smoothing capacitor Ci. On the secondary side, a secondary winding N2 serving as a first secondary winding, and a second secondary side formed by winding up a winding end portion of the secondary winding N2. A tertiary winding N3 is provided as a winding. And
A secondary side parallel resonance capacitor C2 is connected to a secondary side winding (N2 + N3) including the secondary winding N2 and the tertiary winding N3.
Are connected in parallel.

In this case, the winding start end of the secondary winding N2 is connected to the secondary side ground, and the winding end is connected to the rectifier diode D2.
Connected to the anode of O1. Then, a low-voltage DC output voltage EO1 (for example, 135 V) for horizontal deflection of 110 V to 140 V is obtained by a half-wave rectification / smoothing circuit including the rectification diode DO1 and the smoothing capacitor CO1.

A tap is provided at a required position of the secondary winding N2, and a rectifier diode DO2
The anode is connected. Then, a low-voltage DC output voltage EO2 (15 V) for a signal system circuit is obtained by a half-wave rectifying / smoothing circuit including the rectifier diode DO2 and the smoothing capacitor CO2.

Further, on the secondary side of the insulated converter transformer PIT, the ending of the tertiary winding N3 wound around the secondary winding N2 is connected to the anode of the rectifying diode DO3, so that The low-voltage DC output voltage EO3 (200 V) for the video output circuit is obtained by the half-wave rectifier circuit including the smoothing capacitor CO3. In the power supply circuit shown in FIG.
Is connected to the positive electrode of the smoothing capacitor CO1, so that a low-voltage DC output voltage EO3 for the video output circuit is obtained from both ends of the series connection circuit of the smoothing capacitors CO1 to CO3. That is, in the power supply circuit shown in FIG. 1, in order to obtain a low-voltage DC output voltage EO3 for a video output circuit, a low-voltage DC output voltage EO1 generated across the smoothing capacitor CO1 and a DC output voltage generated across the smoothing capacitor CO3 are used. The low voltage DC output voltage EO3 is obtained by stacking the voltages, that is, by superimposing the low voltage DC output voltage EO1 obtained from the secondary winding N2 and the DC output voltage obtained from the tertiary winding N3. Therefore, as a configuration of the rectifying and smoothing circuit including the tertiary winding N3, the rectifying diode DO3, and the smoothing capacitor CO3, the low-voltage DC output voltage EO3 (200V) is subtracted from the low-voltage DC output voltage EO1 (110V to 140V) by 90V.
The configuration is such that a DC output voltage of up to 60 V can be obtained.

In this embodiment, the DC output voltages E01, E01
02 and E03 are particularly referred to as the "low-voltage DC output voltage" as described above, and this is the high-voltage DC output voltage EH output from the high-voltage generation circuit 4 described later in this specification.
V and EFV.

Although not shown, a DC output voltage (± 15 V) for the vertical deflection circuit and a DC output voltage (6.3 V) for the heater are obtained from the secondary side of the insulating converter transformer PIT. It is also possible to do.

On the secondary side of the insulating converter transformer PIT, a secondary side parallel resonance capacitor C2 is connected in parallel to a secondary side winding (N2 + N3) composed of a secondary winding N2 and a tertiary winding N3. Connected to. This allows the secondary winding (N2
+ N3) leakage inductance (L2 + L3),
A secondary parallel resonance circuit is formed by the capacitance of the secondary parallel resonance capacitor C2, and the alternating voltage induced in the secondary winding N2 and the tertiary winding N3 becomes a resonance voltage.
Voltage resonance operation is obtained on the secondary side of the insulating converter transformer PIT.

That is, in the power supply circuit shown in FIG. 1, a parallel resonance circuit for providing a voltage resonance type switching operation is provided on the primary side, and a parallel resonance circuit for obtaining the voltage resonance operation is provided on the secondary side. The configuration of a provided so-called composite resonance type switching converter is employed. Note that such a configuration as a composite resonance type switching converter is realized by forming a gap G in the insulating converter transformer PIT and loosely coupling with a required coupling coefficient, thereby obtaining a state that is more difficult to be saturated. Is what is done. For example, when the gap G is not provided for the insulating converter transformer PIT, there is a high possibility that the insulating converter transformer PIT becomes saturated during flyback operation and the operation becomes abnormal. It is difficult to hope that this is done properly.

The low-voltage DC output voltage EO1 is also branched and input to the control circuit 1. The control circuit 1 is composed of, for example, an error amplifier and the like, and, in accordance with a change in the low-voltage DC output voltage level EO1 output from the secondary side of the insulating converter transformer PIT, the orthogonal control transformer P
By varying the level of the control current (DC current) supplied to the control winding NC1 of the RT-1, the orthogonal control transformer PRT-1 is changed.
Variably controls the inductance LB of the drive winding NB wound around the motor. Thereby, the inductance LB of the drive winding NB is obtained.
, The resonance condition of the series resonance circuit in the self-excited oscillation drive circuit for the switching element Q1 is changed, and the switching frequency of the switching element Q1 is changed. This operation allows the isolation converter transformer P
The low-voltage DC output voltage output from the secondary side of the IT is stabilized. Note that the DC output voltage EO3 may be branched and input to the control circuit 1 to make the DC output voltage constant.

Here, when the orthogonal control transformer PRT-1 for variably controlling the inductance LB of the drive winding NB is provided, the period TOFF during which the switching element Q1 is turned off is made constant in changing the switching frequency. Thus, the ON period TON is variably controlled. That is, in the power supply circuit shown in FIG. 1, as a constant voltage control operation, the switching frequency is variably controlled to perform resonance impedance control on the switching output, and at the same time, the conduction angle control (PWM) of the switching element Q1 in the switching cycle. Control). The composite control operation is realized by a set of control circuit systems (composite control method).

Further, in the power supply circuit shown in FIG. 1, the series resonance capacitor C3 and the primary winding N of the high-voltage generating transformer HVT are connected in parallel with the secondary side parallel resonance capacitor C2.
4. A series resonance circuit comprising a series connection of the controlled winding (main winding) NR of the orthogonal control transformer PRT-2 is provided. That is, the power supply circuit shown in FIG.
T secondary winding (N2 + N3) and series resonance capacitor C
A series resonance circuit consisting of 3-primary winding N4-controlled winding NR is connected in parallel.

In the power supply circuit having such a configuration, when the insulating converter transformer PIT operates as a complex resonance type switching converter, a resonance pulse voltage is generated across the secondary side parallel resonance capacitor C2. Then, the DC output voltage EO3 is obtained from the positive resonance pulse voltage generated during the positive period in which the insulating converter transformer PIT performs the forward operation, and the resonance pulse voltage generated in the secondary parallel resonance capacitor C2 is converted into the series resonance capacitor C3. The voltage is input to the primary winding N4 of the high-voltage generating transformer HVT via the high-voltage generating transformer HVT. In this case, the series resonance capacitor C is provided on the primary side of the high voltage generation transformer HVT.
Since a series resonance circuit composed of 3-primary winding N4 and controlled winding NR is formed, the resonance pulse voltage V2 generated across the secondary parallel resonance capacitor C2 is equal to the capacitance of the DC resonance capacitor C3. And the inductance of the primary winding N4 of the high voltage generating transformer HVT and the inductance LR of the controlled winding NR of the orthogonal control transformer PRT-2, the series resonance operation causes the current flowing through the primary winding N4 of the high voltage generating transformer HVT. I4 and primary winding N
4 both have a substantially sinusoidal resonance waveform.

The orthogonal control transformer PRT-2 is a saturable reactor having the controlled winding NR and the control winding NC2 wound thereon, and outputs the high-voltage DC output voltage EHV output from the high-voltage generation circuit 4 described later. Provided for constant voltage control.
The structure of the orthogonal control transformer PRT-2 will be described later with reference to FIG. 4, but in the case of the present embodiment, the control winding NC2 includes, for example, three litz wires NC21 of 60 μmφ,
It is formed by being wound in a state where NC22 and NC23 are bundled.

In the orthogonal control transformer PRT-2, the inductance LR of the controlled winding NR is changed according to the level of the control current (DC current) IC flowing through the control winding NC2.
For example, it changes in the range of 50 μH to 10 μH. That is, the orthogonal control transformer PRT-2 is provided with an inductance of the controlled winding NR connected in series with the primary winding N4 of the high-voltage generating transformer HVT in order to make the high-voltage DC output voltage EHV constant. By variably controlling
The inductance control of the series resonance circuit formed on the primary side of the high voltage generating transformer HVT is performed.

High-voltage generating circuit 4 enclosed by a dashed line
Is composed of a high-voltage generating transformer HVT and a high-voltage rectifier circuit.
4 to raise the resonance voltage V4 to generate a high voltage corresponding to, for example, the anode voltage level of the CRT. For this reason, on the secondary side of the high-voltage generating transformer HVT, four to five sets of step-up windings NVH are divided and wound by slit winding or interlayer winding. In this case, the primary winding N
4 and the step-up winding NVH are wound so as to be tightly coupled, and are further wound so that their polarities (winding directions) are opposite. Therefore, on the secondary side of the high voltage generating transformer HVT, the negative resonance voltage of the resonance voltage V4 generated in the primary winding N4 is inverted, and the winding ratio of the boost winding NHV and the primary winding N4 ( NHV / N4) to obtain a boosted voltage.

FIG. 3 is a cross-sectional view of the high-voltage generating transformer HVT. Referring to FIG.
The structure of T will be described. The high-voltage generating transformer HVT shown in this figure has, for example, two U-shaped cores CR11 and CR.
The square core CR30 is formed by combining the twelve magnetic legs so as to face each other. A gap G is provided at a portion where the end of the U-shaped core CR11 and the end of the U-shaped core CR12 face each other. Further, as shown in the figure, by attaching a low-voltage winding bobbin LB and a high-voltage winding bobbin HB to one of the magnetic legs of the square core CR30, the low-voltage winding bobbin LB and the high-voltage winding bobbin HB are attached. The primary winding N4 and the boost winding NHV are respectively divided and wound. In this case, the primary winding N4 is wound around the low-voltage winding bobbin LB, and a plurality of boost windings NHV are wound around the high-voltage winding bobbin HB by inserting and winding up the interlayer film F. become.

In the power supply circuit shown in FIG. 1, the resonance voltage V4 input to the primary winding N4 of the high voltage generating transformer HVT is obtained from the secondary side of the insulating converter transformer PIT. The frequency is assumed to correspond to the switching frequency of the switching element Q1,
For example, the frequency is in the range of about several tens kHz to 200 kHz. For this reason, in the present embodiment, the high-voltage generating transformer H
A litz wire is used for the primary winding N4 of the VT,
An eddy current is prevented from being generated in the primary winding N4.

In this embodiment, the case where the boost winding NHV of the high-voltage generating transformer HVT is wound by interlayer winding is shown, but the winding of the boost winding NHV is limited to the interlayer winding. Although not shown, for example, the high-pressure bobbin HB may be divided into a plurality of regions, and the boost winding NHV may be wound around each divided region, that is, the high-voltage bobbin HB may be wound by so-called divided winding. . In other words, the structure of the high-voltage generating transformer HVT may be such that a plurality of boost windings NHV wound around the high-voltage bobbin HB are wound in a state in which they are insulated.

In the power supply circuit shown in FIG. 1, five sets of boost windings NVH1, NHV2,
NHV3, NHV4, and NHV5 are wound independently of each other, and high-voltage rectifier diodes DHV1, DHV2, DHV3, and DH are wound around the winding ends of the respective boost windings NHV1 to NHV5.
The anodes of V4 and DHV5 are connected. The cathode of the high-voltage rectifier diode DHV1 is connected to the smoothing capacitor CO.
High voltage rectifier diode DHV connected to positive terminal of HV
The cathodes of 2 to DHV5 are respectively boosted windings NHV1 to NH
Connected to V4 winding start end.

That is, on the secondary side of the high voltage generating transformer HVT, [boost winding NHV1, high voltage rectifying diode DHV1]
[Step-up winding NHV2, high-voltage rectifier diode DHV2], [Step-up winding NHV3, high-voltage rectifier diode DHV3], [Step-up winding N
HV4, high voltage rectifier diode DHV4], [boost winding NHV5,
Thus, a so-called multisingle-type half-wave rectifier circuit in which five sets of half-wave rectifier circuits called high-voltage rectifier diodes DHV5] are connected in series is formed.

Therefore, on the secondary side of the high-voltage generating transformer HVT, five sets of half-wave rectifier circuits are provided with the boost windings NHV1 to NHH.
The operation of rectifying the current induced in V5 and charging the smoothing capacitor COHV is performed.
At both ends of the OHV, a high-voltage DC output voltage (anode voltage) EHV having a level corresponding to about five times the induced voltage induced in each of the boost windings NHV1 to NHV5 is obtained.

A capacitor COFV is inserted between the cathode of the high-voltage rectifier diode DHV4 and the secondary-side ground, and a high-voltage DC output voltage having a voltage level lower than the anode voltage EHV obtained at both ends of the capacitor COFV. (Focus voltage) EFV is obtained.

Further, in the case of the present embodiment, the smoothing capacitor C0HV which can obtain the high-voltage DC output voltage EHV is:
A series connection circuit of voltage dividing resistors R1 and R2 is provided in parallel.
The voltage dividing point of the voltage dividing resistors R1 and R2 is determined by the control circuit 2
Connected to. That is, in the present embodiment,
A voltage level obtained by dividing the high-voltage DC output voltage EHV by the voltage dividing resistors R1 to R2 is input to the control circuit 2 as the detection voltage. This is the control circuit 2
Means to detect the fluctuation component (ΔEHV) of the high-voltage DC output voltage EHV.

As shown, a ripple detection circuit 3 is provided. The ripple detection circuit 3 detects a ripple voltage component of the high-voltage DC output voltage EHV from the smoothing capacitor C0HV and supplies it to the control circuit 2.

Therefore, the control circuit 2 outputs the high-voltage DC output voltage E
By changing the level of the control current (DC current) flowing through the control winding NC2 in accordance with the change in HV and the superimposed ripple voltage component, the controlled current wound on the orthogonal control transformer PRT-2 is changed. The inductance LR of the winding NR is variably controlled. Thereby, the primary winding N of the high-voltage generating transformer HVT
The current I4 flowing through the anode voltage E4 is varied, so that the anode voltage E4
The HV and the focus voltage EFV are made constant. The control circuit 2 varies the level of the control current IC flowing through the control winding NC2 based on the amount of change in the high-voltage DC output voltage EHV and the ripple voltage component;
The high-speed transient response characteristic is improved by the winding method 2 as described later.

FIG. 4 shows the structure of the orthogonal control transformer PRT-2. FIG. 4A is an external perspective view illustrating the overall structure, and FIG. 4B is a cross-sectional perspective view illustrating the winding direction of the wound winding. The orthogonal control transformer PRT-2 of this example is basically the same as the orthogonal control transformer PRT-2 described with reference to FIG. That is, the orthogonal control transformer PRT-2 is formed by the three-dimensional core 20 in which two double U-shaped cores 21 and 22 (ferrite cores) are combined.
One double U-shaped core 21 has four magnetic legs 2.
1a, 21b, 21c, 21d, and the other double U-shaped core 22 also has four magnetic legs 22a, 22b, 2
2c and 22d. Then, the two magnetic legs 21a to 21d, 2 of these two double U-shaped cores 21, 22 are formed.
The three-dimensional core 20 is formed by joining the ends of 2a to 22d. A gap G is provided at each joint of the double U-shaped cores 21 and 22.

In this case, for example, the double U-shaped core 21
The control winding NC2 is wound around the two magnetic legs 21b and 21c, and the controlled winding NR is wound around the magnetic legs 22c and 22d of the double U-shaped core 22. That is, the controlled winding NR
The control winding NC2 is configured as a saturable reactor wound in a direction perpendicular to the control winding NC2.

The orthogonal control transformer PRT-2
The control winding NC2 is, for example,
With the three litz wires NC21, NC22, and NC23 bundled,
It is wound 33 turns. As described above, in the case of the prior art shown in FIG. 8, assuming that the control winding NC2 is 1000T, the maximum magnetomotive force of the control winding NC2 of the orthogonal control transformer PRT is 1000T × 20mA = 20AT. .
Therefore, 33 litz wires NC21, NC22 and NC23 are bundled.
Considering the 3T control winding NC2, a control current Ic of about 60 mA may be applied to make the maximum magnetomotive force equal.

In this case, the DC resistance Rc of the control winding NC2 is 22Ω and the inductance Lc is 35mH.
The time constant Lc / Rc of the control winding NC2 is 1.60 mH / Ω. That is, the time constant can be reduced as compared with the case of the prior art.

In the present embodiment, the cathode current IK
FIG. 5 shows the control current Ic and the high-frequency sine wave voltage V4 of the primary winding N4 of the high-voltage generating transformer HVT when the white peak signal is generated. As can be clearly understood from comparison with FIG. 9, the control current Ic is applied to the white peak signal having a width of 1 ms.
Was about 3 ms, but in the case of this example, it was greatly improved to 1.8 ms. As a result, the image of the white peak was able to be in a state without image bending as shown by the solid line in FIG.

The fact that the number of turns of the control winding NC is reduced to 1/3 of that in the prior art also has the advantage that the manufacturing process of the orthogonal control transformer PRT can be shortened.

The control current Ic (DC current) is 20 m
A to 60 mA, but the control winding NC2
, The power loss of Rc, Ic, and Ic is almost equal, and the driving power of the control circuit 2 is increased by three times.

The embodiment has been described above, but the present invention is not limited to the configuration shown in the above embodiment. For example, high voltage generating transformer HV
The rectifier circuit on the secondary side of T is not limited to the above configuration, and any other circuit configuration may be employed as long as the level required for the high-voltage DC output voltage EHV can be obtained. is there. For example, a circuit configuration as one set of a high-voltage winding and a quadruple voltage rectification method may be adopted. Furthermore, although the configuration in which the secondary side of the insulating converter transformer PIT is a voltage resonance circuit has been described, a configuration as a current resonance circuit may be employed. In addition, although the example of the self-excited oscillation type switching drive system is given on the primary side, a separately excited oscillation type switching drive system may be used. In that case, a MOS-FET, an IGBT (insulated gate bipolar transistor), or the like can be employed as the switching element Q1. Further, the control winding NC2 of the orthogonal control transformer PRT-2 is
Although an example in which three litz wires are bundled is given, various other examples such as a wire having a length of, for example, 60 to 100 μmφ and a number of wires to be bundled, for example, four, can be considered.

[0077]

As described above, the present invention relates to a composite resonant converter in which the primary side is a self-excited oscillation type or separately excited oscillation type switching element and the secondary side is a voltage resonance circuit. In the switching power supply circuit, a control signal corresponding to the level of the high-voltage DC output voltage and the ripple voltage is supplied to the control winding of the orthogonal control transformer to stabilize the high-voltage DC output voltage, And the control winding of the orthogonal control transformer is bundled and wound with a plurality of wires to reduce the time constant while maintaining the magnetomotive force. This improves the fast transient response characteristics,
There is an effect that the image curve at the time of displaying a video signal of a white peak on a CRT can be improved. Also, in the control winding process at the time of manufacturing the orthogonal control transformer, the number of turns is 1 /
3 (in the case of bundling three wires) and the like, and there is also an advantage that the time of the winding step is greatly reduced and productivity can be improved.

[Brief description of the drawings]

FIG. 1 is a circuit diagram illustrating a configuration example of a switching power supply circuit according to an embodiment of the present invention.

FIG. 2 is a cross-sectional view illustrating a structural example of an insulating converter transformer.

FIG. 3 is a cross-sectional view illustrating a structural example of a high-voltage generating transformer.

FIG. 4 is a perspective view and a cross-sectional view showing a structural example of the orthogonal control transformer according to the embodiment;

FIG. 5 is a waveform diagram of a main part of the embodiment.

FIG. 6 is a circuit diagram showing a configuration example of a switching power supply circuit as a prior art.

7 is a waveform chart showing an operation of a main part in the switching power supply circuit of FIG. 6;

FIG. 8 is a perspective view and a cross-sectional view illustrating a structural example of a conventional orthogonal control transformer.

9 is a waveform diagram of a main part of the switching power supply circuit of FIG. 6;

FIG. 10 is an explanatory diagram showing how a white peak image displayed on a CRT is distorted.

[Explanation of symbols]

1, 2 control circuit, 3 ripple detection circuit, Ci smoothing capacitor, Q1 switching element, PIT isolation converter transformer, PRT-1 and PRT-2 orthogonal control transformer, Cr1 primary side parallel resonance capacitor, C2
Parallel resonance capacitor, C3 Series resonance capacitor, N1
Primary winding, N2 secondary winding, N3 boost transformer primary winding, NHV boost winding, NC2 control winding, NR controlled winding, DHV1 to DHV5 high voltage rectifier diode, C0HV smoothing capacitor

Claims (1)

[Claims] 1. A switching means formed with a switching element for intermittently outputting a DC input voltage, and a primary-side parallel resonance circuit having a voltage resonance type operation of the switching means is formed. A primary-side parallel resonant capacitor provided with a structure having a required coupling coefficient that is loosely coupled between the primary side and the secondary side, and the output of the switching means obtained on the primary side is output to the secondary side. An insulating converter transformer for transmitting, a secondary-side voltage resonance circuit formed by connecting a secondary-side parallel resonance capacitor to a secondary winding of the insulation converter transformer, and a secondary-side voltage resonance circuit. A low-voltage direct-current output voltage generating means formed so as to obtain a low-voltage direct-current output voltage by performing a rectifying operation on an alternating voltage obtained from the secondary winding; A primary-side winding connected in parallel with a secondary-side winding of the insulating converter transformer via a secondary-side series resonance capacitor connected in series to form a secondary-side current resonance circuit; A high-voltage generating transformer configured to transmit a resonance voltage obtained on a secondary winding of a converter transformer from a primary side to a secondary side to obtain a high-voltage obtained by boosting the resonance voltage from the secondary side; By performing a rectification operation on a high voltage obtained on the secondary side of the high voltage generating transformer, a high voltage DC output voltage generating means configured to obtain a high voltage DC output voltage, and a control signal based on the low voltage DC output voltage Controlling the switching frequency and the conduction angle of the switching element on the basis of the first low-voltage DC output voltage; Ripple voltage detection means for detecting the ripple voltage of the high-voltage generating transformer forming the secondary-side current resonance circuit, and a controlled winding connected in series to a primary winding of the high-voltage generating transformer; And a control winding that controls the inductance of the control winding, and variably controls the inductance of the controlled winding based on a control signal based on the level of the high-voltage DC output voltage and the ripple voltage. A second method for converting the high-voltage DC output voltage into a constant voltage.
A switching power supply circuit comprising: a constant voltage generating means; and the control winding of the orthogonal control transformer, wherein a plurality of wires are bundled and wound.