JP2002004717A - Power window drive control device - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a power window drive control device with a simple device composition, capable of conducting the highly accurate low-cost detection of pinching foreign matter. SOLUTION: In a reference circuit (QB, R66, CMP2, R61, C61, R62, and TR61) for generating a reference voltage (VDSB of Reference FETQB), a normal operation is performed in such a manner that the reference voltage is made to follow fluctuations in the drain/source voltage VDSA of main control FETQA even in the case of the above fluctuations. The following velocity of the reference voltage is set so as to make it impossible for the reference voltage to follow the rapid fluctuations in VDSA when the foreign matter is pinched in window glass. A control means (CMP1 and 111) detects that a differential between VDSA and the reference exceeds the first prescribed value in order to exert off-control on FETQA. On-control is exerted on FETQA when VDSA is increased due to the off-control to make the value of VDSA exceed the second prescribed value.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a power window drive control device for moving a window glass of a vehicle or the like up and down by a driving force of a motor. The present invention relates to a power window drive control device capable of detecting a foreign object with high accuracy at low cost and eliminating an adjustment operation.

[0002]

2. Description of the Related Art As a conventional power window drive control device, for example, there is one shown in FIG. The power window drive control device of this conventional example detects the entrapment of a foreign object by detecting the rotation speed of the motor with a pulse sensor, and stops or reverses the drive of the motor. In consideration of a foreign substance, damage to a window glass, or the like, it is desirable to reverse the motor drive after detecting the foreign substance.

[0003] In FIG. 6, a power window drive control device of the prior art includes a motor unit 501, a motor drive control unit 502, and an auto (automatic) switch 50.
3 and a manual (manual) switch 504. Here, the motor unit 501 includes a motor 511 that moves the window glass up and down, a pulse sensor 512 that generates a pulse corresponding to the number of rotations of the motor 511 to detect the rotation speed of the motor 511, and a pulse sensor 5.
12 is provided with a limit switch 513 for setting a dead zone.

[0004] The motor drive control unit 502 includes:
The control microcomputer 523 transmits operation instructions from the power supply circuit 521 and the auto switch 503 and the manual switch 504.
Interface unit 522 for transmitting a detection signal from the pulse sensor 512 to the control microcomputer 523, and a driving instruction for the motor 511 to the drive circuit 524 based on the operation instruction and the detection signal from the pulse sensor 512. Control microcomputer 523 to be provided, a relay switch unit 525 having a relay contact, and a relay switch unit 5 based on a driving instruction from the control microcomputer 523.
A drive circuit 524 that opens and closes 25 relay contacts and controls the rotation direction and speed of the motor 511 is provided.

In the auto switch 503, when the up-side contact or the down-side contact is turned on, the drive of the motor 511 is continued even if the hand is released, and the window glass moves until it is fully closed or fully opened. For example, when moving this window glass from the fully open state to the fully closed state, when the window glass is in the fully closed state,
Such movement is prevented by a window frame or the like, and the motor 511 is moved.
Rotation speed fluctuates. The amount of change in the rotation speed of the motor 511 is detected via the pulse sensor 512, and when the rotation speed exceeds a predetermined value, it is determined that the movement of the window glass has been completed, and the motor 511 is stopped. On the other hand, manual switch 5
Reference numeral 04 is for manually operating the vertical movement of the window glass. That is, the occupant moves the switch in the desired direction (up /
By turning on the (down) side, the window glass is moved in the direction in which it was turned on for the duration of the on,
The windowpane can be stopped at a predetermined position.

In such a conventional power window drive control device, detection of a state in which a foreign object is caught during the auto-up operation, that is, detection of a caught object is performed as follows. That is, the control microcomputer 523 detects both rising and falling edges of the detection pulse from the pulse sensor 512, calculates the rotation speed of the motor 511 when the power window rises, and detects pinching by a change in the rotation speed. . More specifically, when the rate of change of the rotation speed of the motor 511 in the decreasing direction falls below a preset threshold while the power window is rising, it is determined that a foreign object has been caught. Further, when the rotation speed of the motor 511 falls below a preset threshold value while the power window is rising, it is determined that a foreign object has already been caught. Note that the detection range of such entrapment detection is the operation range of the windowpane, but from the start of window rising to the input of 20 pulses, is set as a dead zone to absorb fluctuations in operation speed, In addition, the limit switch 513 is turned off even during the period from the fully closed state to the opened state of 4 [mm] of the window glass, so that a dead zone is set.

[0007]

However, in the above-described conventional power window drive control device, the pulse sensor 512 generally generates one pulse per one rotation of the motor 511 in order to reduce the cost. There is a problem that the accuracy of speed detection is low because a low resolution device is used. Also, in order to keep the reversal load after the detection of foreign object entrapment, the threshold value in entrapment determination is determined by setting the power supply voltage fluctuation, the ambient temperature fluctuation, the motor characteristic fluctuation, and the door standing fluctuation (initial, aging). ), The influence of wind pressure during high-speed running, the backlash of the window during running on rough roads, the wiring resistance of the harness, and other items need to be corrected. Further, it is necessary to include the control microcomputer 523, the pulse sensor 512, the drive circuit 524, the correction circuit, and the like, and the number of parts is large, the system configuration is complicated, and it is difficult to reduce the system scale and the apparatus cost. There were circumstances.

SUMMARY OF THE INVENTION An object of the present invention is to solve the above-mentioned conventional circumstances, and to simplify a device configuration by eliminating the need for a microcomputer or a sensor, and to detect a foreign object with high accuracy at low cost. It is an object of the present invention to provide a power window drive control device that can perform the adjustment work and can eliminate the adjustment work.

[0009]

In order to solve the above-mentioned object, a power window drive control device of the present invention is provided with a motor, and is a power window drive control device that opens and closes a window glass by a driving force of the motor. A switching circuit that is controlled in response to a control signal supplied to a control signal input terminal, controls a power supply from a power supply to the motor, and a reference circuit that generates a reference voltage,
In normal operation, even when the voltage between the terminals of the semiconductor switch fluctuates, the reference voltage operates so as to follow the fluctuation. A reference circuit that sets the following speed of the reference voltage so that it cannot follow a
When the difference between the voltage between the terminals of the semiconductor switch and the reference voltage exceeds a first predetermined value, the semiconductor switch is turned off. And control means for turning on the semiconductor switch when the predetermined value is exceeded.

A power window drive control device according to a second aspect of the present invention is a power window drive control device including a motor, which opens and closes a window glass by a driving force of the motor, and is supplied to a control signal input terminal. Switching control is performed in accordance with a control signal, a semiconductor switch that controls power supply from a power supply to the motor, and a reference circuit that generates a reference current.In a normal operation, a current flowing between terminals of the semiconductor switch is Even if it fluctuates, it operates so that the reference current follows the fluctuation equivalently, and equivalently follows a sudden change in the current flowing between the terminals of the semiconductor switch when the window glass sandwiches foreign matter. A reference circuit that sets an equivalent following speed of the reference current, a current flowing between terminals of the semiconductor switch, and a predetermined value of the reference current. When the difference from the equivalent reference current exceeds a third predetermined value, the semiconductor switch is turned off, and the current flowing between the terminals of the semiconductor switch is reduced by the off control so that the value becomes a fourth predetermined value. And control means for turning on the semiconductor switch when the voltage falls below the threshold.

According to a third aspect of the present invention, in the power window drive control device according to the first or second aspect, the reference circuit is connected in parallel to the semiconductor switch and the motor, and A circuit in which a second semiconductor switch and a second load, which are switching-controlled in accordance with a signal, are connected in series, and a voltage between terminals of the second semiconductor switch is used as the reference voltage, and flows between terminals of the second semiconductor switch. The current is used as the reference current.

According to a fourth aspect of the present invention, in the power window drive control device according to the first, second or third aspect, the semiconductor device has a characteristic that a reference voltage or a reference current of the reference circuit has. The characteristics are substantially equivalent to the voltage between the terminals of the switch or the current flowing between the terminals.

According to a fifth aspect of the present invention, in the power window drive control device according to the third or fourth aspect, the semiconductor switch and the second semiconductor switch transition from an off state to an on state. It has an equivalent characteristic with respect to the transient voltage characteristic of the voltage between terminals when performing.

The power window drive control device according to claim 6 is the power window drive control device according to claim 3, 4 or 5, wherein the current capacity of the second semiconductor switch is larger than the current capacity of the semiconductor switch. The resistance ratio of the motor and the second load is set so as to be as inversely proportional to the current capacity ratio of the semiconductor switch and the second semiconductor switch as possible.

According to a seventh aspect of the present invention, in the power window drive control device according to the sixth aspect, the resistance value of the second load is determined when the motor is rotating normally. The resistance value is set lower than a value obtained by multiplying the maximum resistance value of the motor by the resistance value ratio.

The power window drive control device according to claim 8 is the power window drive control device according to claim 3, 4, 5 or 6, wherein the second load is:
A first resistor, a circuit in which a second resistor and a third semiconductor switch connected in parallel to the first resistor are connected in series, and the reference circuit includes a terminal voltage on the motor side of the semiconductor switch; A comparator for comparing a terminal voltage of the second semiconductor switch on the second load side; and a delay circuit for delaying an output of the comparator for a predetermined time and supplying the output to a control signal input terminal of the third semiconductor switch. The response speed of the delay circuit is set to be lower than the change speed of the voltage between the terminals of the semiconductor switch or the change speed of the current flowing between the terminals of the semiconductor switch when the window glass sandwiches foreign matter. .

According to a ninth aspect of the present invention, in the power window drive control device according to the eighth aspect, the comparator is configured using an operational amplifier.

The power window drive control device according to claim 10 is a power window drive control device according to claims 1, 2, 3, 4, 5, 6, 7,
10. The power window drive control device according to 8 or 9, wherein the first predetermined value or the third predetermined value depends on a voltage of the power supply, and is set so as to decrease as the voltage of the power supply increases. Things.

The power window drive control device according to claim 11 is the power window drive control device according to claims 1, 2, 3, 4, 5, 6, 7,
11. The power window drive control device according to claim 8, 9 or 10, wherein when a current flowing between terminals of the semiconductor switch or a current flowing through the motor has a pulsating component due to a commutator action, the reference circuit follows the reference voltage of the reference circuit. Alternatively, the equivalent follow-up speed of the reference current of the reference circuit is set lower than the change speed of the pulsating component of the current so that the reference voltage or the reference current of the reference circuit does not follow the change due to the pulsating component of the current. And setting the value obtained by dividing the first predetermined value by the on-resistance of the semiconductor switch or the third predetermined value to a value equal to or more than half the amplitude of the pulsating component of the current.

A power window drive control device according to claim 12 is a power window drive control device according to claims 1, 2, 3, 4, 5, 6, 7,
12. The power window drive control device according to 8, 9, 10 or 11, wherein the control means is configured to determine whether a difference between a terminal voltage of the semiconductor switch and the reference voltage exceeds a first predetermined value, or When the difference between the current flowing between the terminals of the switch and the equivalent reference current that is a predetermined multiple of the reference current exceeds a third predetermined value, the semiconductor switch is turned on while the armature back electromotive force of the motor is large. The semiconductor switch is controlled so as to alternately repeat an on / off operation that repeats a transition between a state and an off state and a continuous on operation that operates continuously in an on state, and the motor flows during the on / off operation. Reducing the current, increasing the current flowing through the motor during the continuous on operation,
The current flowing through the motor is limited to a certain range, the armature back electromotive force of the motor decreases, the rotation speed of the motor decreases, and the semiconductor switch is turned on while the motor reaches a locked state. / Off operation only,
The range of the current flowing through the motor is limited.

According to a thirteenth aspect of the present invention, in the power window drive control device according to the twelfth aspect, when the semiconductor switch transitions from the continuous ON operation to the ON / OFF operation, The value of the current flowing between the terminals of the semiconductor switch or the upper limit of the current, and the value of the current flowing between the terminals of the semiconductor switch or the lower limit of the current when the semiconductor switch transits from on / off operation to continuous on operation The value is the first
It is set by a predetermined value or the third predetermined value, the first resistor, and the delay time of the delay circuit.

According to a fourteenth aspect of the present invention, in the power window drive control device according to the twelfth or thirteenth aspect, the control means measures a current limiting period for limiting a current flowing through the motor. A timer is provided, and when the timer has counted a predetermined time, the rotation direction of the motor is reversed.

In the power window drive control device according to a fifteenth aspect, in the power window drive control device according to the fourteenth aspect, the timer sets the semiconductor switch to the on / off operation during the current limiting period. A certain time is measured, and when the semiconductor switch makes a transition to the continuous ON operation, the time counted by the timer is reset.

A power window drive control device according to claim 16 is a power window drive control device according to claims 1, 2, 3, 4, 5, 6, 7,
The power window drive control device according to 8, 9, 10, 11, 12, 13, 14, or 15, further comprising overheat protection means for turning off the semiconductor switch to protect the semiconductor switch when the semiconductor switch is overheated. It is.

Further, the power window drive control device according to claim 17 is a power window drive control device according to claims 1, 2, 3, 4, 5, 6, 7,
8, 9, 10, 11, 12, 13, 14, 15, or 1
6. The power window drive control device according to claim 6, further comprising a mask unit that prohibits the control unit from turning on / off the semiconductor switch for a certain period after the semiconductor switch is turned on.

Further, the power window drive control device according to claim 18 is the power window drive control device according to claims 1, 2, 3, 4, 5, 6, 7,
The power window drive control device according to 8, 9, 10, 11, 12, 13, 14, 15, 16 or 17, wherein the semiconductor switch, the reference circuit, the control means, the timer, the overheat protection means, or The mask means is formed on the same chip.

Here, the semiconductor switch and the second semiconductor switch include a field effect transistor (FET: Fiel).
d-Effect Transistor) or static induction transistor (S
IT: Static Inducted Transistor) or MOS compound devices such as emitter-switched thyristor (EST), MOS-controlled thyristor (MCT) and IGBT
(Insulated Gate Bipolar Transistor) and other switching elements such as insulated gate power devices. These switching elements are n-channel type,
Any p-channel type may be used.

In the power window drive control device according to the first aspect of the present invention, in a reference circuit for generating a reference voltage, in a normal operation, the reference voltage follows the fluctuation even if the voltage between the terminals of the semiconductor switch fluctuates. The reference voltage is set so that it does not follow a rapid change in the voltage between the terminals of the semiconductor switch when the window glass sandwiches foreign matter. That is, the change rate of the load current (current flowing through the motor) due to the fluctuation of the driving mechanism sliding force in the normal operation is relatively small, and the fluctuation of the terminal voltage of the semiconductor switch due to the fluctuation of the load current in the normal operation is also relatively small. Therefore, the reference circuit operates so that the reference voltage always becomes equal to the inter-terminal voltage of the semiconductor switch following the fluctuation. On the other hand, when the window glass catches foreign matter, the rotation speed of the motor decreases and the back electromotive force decreases, so that the load current increases rapidly and the voltage between the terminals of the semiconductor switch also changes rapidly. Since the reference circuit cannot follow the fluctuation, the reference voltage cannot be made equal to the inter-terminal voltage of the semiconductor switch. In this case, when the difference between the terminal voltage of the semiconductor switch and the reference voltage exceeds a first predetermined value, the control means detects that the window glass has caught a foreign substance, and controls the semiconductor switch to be off. When the voltage between terminals of the semiconductor switch increases due to the OFF control and the value exceeds a second predetermined value, the semiconductor switch is controlled to be ON.

In a power window drive control device according to a second aspect of the present invention, in a reference circuit for generating a reference current, in a normal operation, even if a current flowing between terminals of a semiconductor switch fluctuates, the reference current is caused to fluctuate. Operates so as to follow equivalently, and sets the equivalent following speed of the reference current so that it cannot equally follow a sudden change in the current flowing between the terminals of the semiconductor switch when the window glass sandwiches foreign matter. Have been. That is, since the rate of change of the load current (current flowing through the motor) due to the fluctuation of the driving mechanism sliding force in the normal operation is relatively small, the reference circuit always flows the reference current between the terminals of the semiconductor switch following the fluctuation. It operates to be equivalent to the current. On the other hand, when the window glass catches foreign matter, the rotation speed of the motor decreases and the back electromotive force decreases, so the load current sharply increases, so that the reference circuit cannot follow the fluctuation and increases the reference current. It cannot be equivalently equal to the current flowing between the terminals of the semiconductor switch. In this case, the control means
When the difference between the current flowing between the terminals of the semiconductor switch and the equivalent reference current that is a predetermined multiple of the reference current exceeds a third predetermined value, it is detected that the window glass has caught a foreign substance, and the semiconductor switch is turned off. When the current flowing between the terminals of the semiconductor switch decreases due to the off control and the value of the current falls below a fourth predetermined value, the semiconductor switch is controlled to be on.

As described above, according to the present invention, during normal operation, the reference current follows the gradual change in the load current, and when pinching occurs, a sharp change in the load current with respect to the reference current is detected to detect pinching. The semiconductor switch is turned on / off to limit the load current, that is, the motor torque. In other words, the present invention directly detects the load current and detects the entrapment in comparison with the conventional pulse method in which the entrapment is detected based on the rotational speed or the rate of change of the rotational speed. And the detection accuracy can be improved.

After the jamming is detected, the semiconductor switch is turned on / off to limit the load current and to control the torque of the motor. When the load becomes normal during the OFF control, the operation can be returned to the normal operation. This allows
There is no need to set the judgment load high because of fear of malfunction, and the judgment load can be set low. As a result, highly accurate detection is possible. Furthermore, unlike the related art, since a microcomputer and a sensor are not required and the configuration is simple, the system scale of the power window drive control device can be reduced, and realization at low cost is possible. In addition, conventionally, it was necessary to correct the entrapment determination threshold value in order to keep the reversal load after the entrapment of foreign matter was detected. Becomes

Further, in the power window drive control device according to the third aspect, the reference circuit is connected in parallel to the semiconductor switch and the motor, and the second semiconductor switch and the second load, which are switching-controlled in accordance with a control signal, are connected in series. It is desirable that the circuit be provided with a connected circuit.
The voltage between the terminals of the semiconductor switch is the reference voltage, and the current flowing between the terminals of the second semiconductor switch is the reference current.

Further, in the power window drive control device according to the present invention, the characteristic of the reference voltage or the reference current of the reference circuit is made substantially equivalent to the voltage between the terminals of the semiconductor switch or the current flowing between the terminals. Preferably, in the power window drive control device according to claim 5, the semiconductor switch and the second semiconductor switch have equivalent characteristics with respect to a transient voltage characteristic of a terminal voltage when transitioning from an off state to an on state. It is desirable to have. Thereby, the detection accuracy can be further improved.

In the power window drive control device according to the present invention, the current capacity of the second semiconductor switch is smaller than the current capacity of the semiconductor switch, and the motor and the second power supply are controlled by the second switch.
It is preferable that the resistance ratio of the load is set so as to be as inversely proportional to the current capacity ratio of the semiconductor switch and the second semiconductor switch as possible.

Further, in the power window drive control device according to the present invention, the resistance value of the second load is multiplied by the resistance value ratio to the maximum resistance value of the motor when the motor is rotating normally. It is desirable to set it lower than the value set. In this case, the resistance value of the motor includes the armature back electromotive force equivalent resistance. With this setting, malfunction can be prevented.

In the power window drive control device according to the present invention, the second load is a circuit in which a first resistor, a second resistor connected in parallel with the first resistor, and a third semiconductor switch are connected in series. A reference circuit comprising: a comparator for comparing a terminal voltage of the semiconductor switch on the motor side with a terminal voltage of the second semiconductor switch on the second load side; and delaying an output of the comparator by a predetermined time. And a delay circuit for supplying the control signal input terminal of the third semiconductor switch to the control signal input terminal. In this case, the response speed of the delay circuit is determined by the voltage between the terminals of the semiconductor switch when the window glass sandwiches foreign matter. , Or the change speed of the current flowing between the terminals of the semiconductor switch.

In the power window drive control device according to the ninth aspect, it is preferable that the comparator is configured using an operational amplifier. This is because, as can be seen from the configuration in which the delay circuit is added to the output side of the comparator, it is desirable that the response of the comparator be slow. On the other hand, in a configuration in which the control means uses a comparator to detect that the difference between the voltage between the terminals of the semiconductor switch and the reference voltage exceeds the first predetermined value, the comparator has a faster response. Is required, and it is desirable to configure with a differential amplifier or the like.

In the power window drive control device according to the tenth aspect, the first predetermined value or the third predetermined value depends on the voltage of the power supply and is set so as to decrease as the voltage of the power supply increases. Is desirable. As the power supply voltage increases with respect to the same determination value, the inertia force of the window glass and the drive mechanism increases, and the load increases. This is because a determination value is set, and in a place where the power supply voltage at which the sandwiched load becomes small and the power supply voltage is low, the resistance to malfunction is given priority over the load.

In the power window drive control device according to the eleventh aspect, when the current flowing between the terminals of the semiconductor switch or the current flowing through the motor has a pulsating component due to the commutator action, the reference circuit follows the reference voltage of the reference circuit. An equivalent following speed of the reference current of the reference circuit is set to be lower than a changing speed of the pulsating component of the current, so that the reference voltage or the reference current of the reference circuit does not follow the change due to the pulsating component of the current, Is divided by the on-resistance of the semiconductor switch or a third predetermined value is preferably set to a value equal to or more than half the amplitude of the pulsating component of the current.

[0040] In the power window drive control device according to the twelfth aspect, the control means may determine whether the difference between the voltage between the terminals of the semiconductor switch and the reference voltage exceeds a first predetermined value or whether the difference between the terminals of the semiconductor switch is large. When the difference between the current flowing through the window glass and the equivalent reference current that is a predetermined multiple of the reference current exceeds a third predetermined value, it is detected that the window glass has caught a foreign object, that is, the clogging is detected. By the following control, the current flowing through the motor is limited to a certain range.
That is, while the armature back electromotive force of the motor is large, the on / off operation in which the semiconductor switch repeats the transition between the on state and the off state and the continuous on operation in which the semiconductor switch operates continuously in the on state are alternately repeated. The semiconductor switch is controlled to reduce the current flowing through the motor during the on / off operation, increase the current flowing through the motor during the continuous on operation, and limit the current flowing through the motor to a certain range. In addition, while the armature back electromotive force of the motor decreases, the number of revolutions of the motor decreases, and the motor reaches a locked state, the semiconductor switch is controlled to perform only on / off operation, and the motor is controlled. Limits the range of current flowing through.

Further, in the power window drive control device according to the thirteenth aspect, the value of the current flowing between the terminals of the semiconductor switch or the upper limit of the current when the semiconductor switch transitions from the continuous on operation to the on / off operation, A current value flowing between terminals of the semiconductor switch or a lower limit value of the current when the semiconductor switch transitions from an on / off operation to a continuous on operation is a first predetermined value or a third predetermined value;
It is desirable to set by the first resistor and the delay time of the delay circuit. The current flowing to the motor during the on / off operation of the semiconductor switch changes only slightly due to the large inductance of the motor. When an FET is used for a semiconductor switch, the ON / OFF operation is performed in a pinch-off region of the FET, so that the load current and the gate-source voltage correspond one-to-one, and one cycle in the ON / OFF operation. During this period, the gate-source voltage hardly changes. Further, the waveform of the motor-side terminal (source) voltage of the semiconductor switch (FET) is determined by the characteristics of the gate charge / discharge circuit. In this specification, FET
The terms "pinch-off region" and "ohmic region" are used in the device characteristics of "Analysis and Design of ANALOG INTEGR".
ATED CIRCUITS ”(Third Edition), PAUL R. GRAY, RO
See page 66 of BERT G MEYER.

Further, in the power window drive control device according to claim 14, the control means includes a timer for measuring a current limiting period for limiting the current flowing through the motor,
When the timer has counted a predetermined time, it is desirable to reverse the direction of rotation of the motor. It is desirable to measure the time during the on / off operation and to reset the timed content of the timer when the semiconductor switch transits to the continuous on operation. As described above, when the on / off operation of the semiconductor switch continues for a certain period of time, the motor reverses operation. In addition, these trade-offs are required for time setting.

Further, in the power window drive control device according to claim 16, when the semiconductor switch is overheated, the semiconductor switch is turned off to protect the semiconductor switch by the overheat protection means. This makes it possible to forcibly turn off the semiconductor switch and protect the semiconductor switch when the temperature of the semiconductor switch rises by the ON / OFF operation of the semiconductor switch and exceeds a predetermined temperature.

Further, in the power window drive control device according to the seventeenth aspect, the mask means inhibits on / off control of the semiconductor switch by the control means for a certain period after the semiconductor switch is turned on. I have. Normally, at the time of starting the load, an inrush current that is many times that of the steady state flows. However, there is a risk that the inrush current is detected by this control and the on / off control is performed by the control means. In the present invention, such a problem can be solved by the prohibition control of the mask means.

In the power window drive control device according to claim 18, it is desirable that the semiconductor switch, the reference circuit, the control means, the timer, the overheat protection means or the mask means be formed on the same chip. Such integration on the same chip can reduce the circuit configuration of the device, reduce the mounting space, and reduce the cost of the device. Further, by forming the semiconductor switch and the second semiconductor switch on the same chip, it is possible to remove (reduce) the common-mode error factor in current detection, that is, the influence of power supply voltage, temperature drift, and variation between lots.

[0046]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS An embodiment of a power window drive control device according to the present invention will be described below in detail with reference to FIGS.

[Embodiment] FIG. 1 is a configuration diagram of a power window drive control device according to an embodiment of the present invention.
The power window drive control device of the present embodiment moves a window glass of a vehicle or the like up and down by a driving force of a motor. In FIG. In this configuration, a drain D-source SA of a main control FET QA as a semiconductor switch is connected in series to a path to be supplied to the semiconductor switch 102. Here, the main control FET QA has a DMO
Although an nMOS type having an S structure is used, a pMOS type can also be realized.

In the power window drive control device of the present embodiment, as shown in FIG. 1, a reference FET QB, resistors RG, R2, R3, RV, R10, R61, R62, R
66, capacitor C61, current source IS1, zener diodes ZD1 to ZD3, diodes D4 to D8, pMO
SFET TR61, comparator CMP1, CMP
2, a charge pump 305, a drive circuit 111, and a switch SW1. Note that, although “R” and subsequent numbers and letters are used for the resistors as reference symbols, in the following description, they are used as reference symbols and also represent the resistance values of the resistors. Also, the capacitors are used as reference symbols and represent the capacitance values. Further, a portion 110 surrounded by a dotted line in FIG. 1 indicates a chip portion for analog integration, and T1 to T18 indicate input / output terminals thereof.

The power window drive control device according to the present embodiment detects "entrapment" in which a windowpane entraps a foreign substance in the same manner as in the conventional example, and prevents foreign substances and damage to the windowpane. The PW motor 102 to be controlled will be described. PW motor 10 of the present embodiment
2 also moves the window glass of the vehicle up and down by its driving force as in the conventional example. However, in addition to the PW motor, the conventional motor unit 501 (see FIG. 6) includes a pulse sensor 512 and a limit switch. However, in the present embodiment, since the drive control is performed based on the detection of the load current, there is a great difference in that no speed detecting means such as the pulse sensor 512 is required.

Next, the charge pump 305 and the drive circuit 111 as drive means of the main control FET QA will be briefly described. The drive circuit 111 includes a source transistor having a collector connected to the output of the charge pump 305 and a sink transistor having an emitter connected to the ground potential connected in series.
Based on a switching signal by switching off, the source transistor and the sink transistor are turned on / off to output a signal for driving and controlling the main control FET QA. The output voltage VB of the power supply 101 is, for example, 12
At [V], the output voltage of the charge pump 305 is, for example, VB + 10 [V].

The power window drive control device of the present embodiment includes an overheat cutoff protection circuit 306 as protection means for the main control FET QA. Overheat protection circuit 30
6 realizes an overheat protection function which is also added to a general temperature sensor built-in FET.
When the built-in temperature sensor detects that the temperature of the QA has risen to a temperature equal to or higher than the specified value, the detection information to that effect is held in the latch circuit, and the overheating connected between the gate and the source of the main control FET QA is performed. The main control FET QA is forcibly turned off by transitioning the cutoff FET to the on state. The information held by the latch circuit is output via the terminal T14 and can be used as diagnosis (diagnosis) information.

Further, the power window drive control device according to the present embodiment includes, as other components, an inrush current mask circuit 303 for preventing the on / off operation from occurring due to the inrush current and limiting the starting current, and an on / off operation. An on / off counting and integrating circuit 304 for performing cutoff control by integrating the number of times of turning on / off is provided.

Next, the characteristic configuration of the power window drive control device of this embodiment, that is, the reference circuit and the control means will be described. The reference circuit is a reference FET Q
B, resistors R61, R62, R66, capacitor C61,
It includes a pMOSFET (hereinafter referred to as a transistor) TR61 and a comparator CMP2, and generates a drain-source voltage VDSB of the reference FET QB as a reference voltage. In normal operation, even if the voltage VDSA between the drain and source of the main control FET QA fluctuates, the reference voltage follows the fluctuation, and the operation between the drain and source of the main control FET QA when the window glass traps foreign matter. The feature is that the reference voltage following speed is set so as not to follow a sudden change in the voltage VDSA.

The control means includes resistors R2, R3, R
V, current source IS1, zener diodes ZD2, ZD
3, including the diodes D6 to D8, the comparator CMP1, and the driving circuit 111,
In P1, when the difference between the drain-source voltage VDSA of the main control FET QA and the reference voltage exceeds a first predetermined value,
The main control FET QA is turned off by the drive circuit 111, and the main control FET QA is turned on when the drain-source voltage VDSA of the main control FET QA increases and exceeds a second predetermined value due to the off control. It is the characteristic. For the detailed circuit configuration on the input side of the comparator CMP1, see FIG.

First, the circuit configuration of the control means will be described with reference to FIG.
This will be described in detail with reference to FIG. First, the comparator CMP1 will be described. Comparator CMP1
Is supplied with the source voltage VSA of the main control FET QA via the diode D5 and the resistor RV. The source voltage VSB of the reference FET QB is supplied to the "+" input terminal of the comparator CMP1 via the diode D4. That is, when the potential supplied to the “+” input terminal is larger than the potential supplied to the “−” input terminal, the output becomes “H” level, and the drive circuit 111 controls the main control FET QA to be turned off.
When the potential supplied to the “+” input terminal is smaller than the potential supplied to the “−” input terminal, the output becomes “L” level, and the drive circuit 111 controls the main control FET QA to be turned on.

The resistor R2, diode D6 and zener diode ZD2 added to the "+" input terminal side of the comparator CMP1 constitute a diode clamp circuit. Assuming that the potential at the connection point between the resistor R2 and the Zener diode ZD2 is VE (Zener voltage of the Zener diode ZD2), this diode clamp circuit
When the source voltage VSB of the reference FET QB is VE-0.
Even if it falls below 7 [V], it will be clamped to the potential of VE-0.7 [V]. Note that 0.7 [V] is a forward voltage drop of the diode D6.

A resistor R3 and diodes D7 and D7 added to the "-" input terminal side of the comparator CMP1.
8 and the Zener diode ZD3 are also diode clamp circuits. Assuming that the potential at the connection point between the resistor R3 and the diode D8 is VC (sum of the Zener voltage of the Zener diode ZD3 and the forward voltage drop of the diode D8), this diode clamp circuit causes the source voltage VSA of the main control FET QA to reach VC-0. 0.7 V
It will be clamped to the potential of C-0.7 [V].
Further, the two clamp potentials have a relationship of VC> VE. The reason why the Zener diodes ZD2 and ZD3 are used is to eliminate the influence of the fluctuation of the power supply voltage VB.

With such a diode clamp circuit, even when the main control FET QA and the reference FET QB transit to the off state and the source voltage VSA and the source voltage VSB decrease, the "+" level of the comparator CMP1 is maintained.
The input terminal and the "-" input terminal are VE-0.
7 [V] and a potential of VC-0.7 [V], and since VC> VE, the comparator CMP1
Can be set to the “L” level, and the source voltage V
Regardless of the magnitude relation between SA and source voltage VSB, the main control F
ETQA can be reliably turned on. Further, since the “+” input terminal and the “−” input terminal of the comparator CMP1 do not drop to a potential below a certain value,
The withstand voltage of the “+” input terminal and the “−” input terminal can be improved.

Further, the resistor RV and the current source IS1 in FIG. 1, that is, the resistors RV and R25 to R2 in FIG.
7. Zener diode ZD4 and transistor TR
6, TR7 is an adjusting circuit for adjusting the offset voltage of the "-" input terminal of the comparator CMP1 to which the source voltage VSA of the main control FET QA is supplied.

As can be seen from the circuit configuration shown in FIG. 2, the current generated by the current source IS1 flows through the resistor R27 and a constant current based on the Zener voltage of the Zener diode ZD4, and flows through the resistor R25 to supply the power supply voltage VB. And a power supply voltage-dependent current component based on the Therefore, the voltage drop generated in the resistor RV also depends on the power supply voltage VB, and the circuit configuration is such that the offset voltage of the "-" input terminal of the comparator CMP1 can be adjusted according to the fluctuation of the power supply voltage.

That is, the offset voltage of the "-" input terminal of the comparator CMP1 due to the resistance RV increases due to the rise of the power supply voltage VB.
Is dependent on the power supply voltage VB, and is set to decrease as the power supply voltage VB increases. This is because, as the power supply voltage VB increases with respect to the same determination value (first predetermined value) of the entrapment, the inertial force of the window glass and the drive mechanism increases and the entrapment load increases. A determination value (first predetermined value) is set so as to satisfy the load regulation at the maximum value of the operation range,
This is because where the power supply voltage VB at which the load becomes small is low, the resistance to malfunction is given priority over the load.

In FIG. 2, the transistor T is connected between the diode D4 and the "+" input terminal of the comparator CMP1.
An emitter follower by R1 and a resistor R17 is added, and an emitter follower by a transistor TR2 and a resistor R18 is added between the diode D5 and the resistor RV. Further, a resistor R added between the diode D4 and the “+” input terminal of the comparator CMP1
23, R24, diodes D10 and D11, and transistor TR5 are additional circuits for speeding up the fall of the signal and preventing malfunction even when the capacitance on the "+" input terminal side of the comparator CMP1 is large.

Next, the circuit configuration of the reference circuit will be described with reference to FIG.
This will be described in detail with reference to FIG. The reference circuit is connected in parallel to the main control FET QA and the PW motor 102, and connects in series a reference FET QB as a second semiconductor switch and a second load, which are switching-controlled in accordance with the same control signal as that for the main control FET QA. It has a circuit. Here, the reference FET QB
Is the reference voltage.

In order to make the characteristic of the reference voltage (drain-source voltage VDSB) of the reference circuit substantially equivalent to the drain-source voltage VDSA of the main control FET QA, in this embodiment, the reference FET QB The main control FET QA is the same chip (11
0) The one formed above is used. The method of detecting the load current in the present embodiment is based on the drain-source voltage VDSA of the main control FET QA by the comparator CMP1.
The detection is performed by detecting the difference between the reference FET QB and the main control FE on the same chip.
By forming the TQA, it is possible to remove (reduce) the common-mode error factor in current detection, that is, the influence of power supply voltage, temperature drift, and variation between lots.

Further, the ratio of the number of transistors connected in parallel constituting each FET is set such that the current capacity of the reference FET QB is smaller than the current capacity of the main control FET QA. Of transistors).

Further, the resistance ratio of the PW motor 102 when the PW motor 102 is rotating normally to the resistance value of the second load is set so as to be as inversely proportional to the current capacity ratio of the main control FET QA and the reference FET QB as possible. When the PW motor 102 is rotating normally, the resistance value of the two loads
The resistance must be set lower than the maximum resistance value of the W motor 102 multiplied by the resistance value ratio. Specifically,
The second load has a configuration including a first resistor R66, and a circuit in which a second resistor R62 connected in parallel to the first resistor R66 and a transistor (third semiconductor switch) TR61 are connected in series.

The reference circuit includes a comparator CMP2 for comparing the source voltage VSA of the main control FET QA with the source voltage VSB of the reference FET QB,
A resistor R61 and a capacitor C61 are provided as a delay circuit for delaying the output of the comparator CMP2 for a predetermined time and supplying the output to the gate of the transistor TR61.

Here, the comparator CMP2 uses an operational amplifier to reduce the response. Further, the response speed of the delay circuit, that is, the resistor R61 and the capacitor C
The time constant of 61 is set so as to be slower than the changing speed of the source voltage VSA of the main control FET QA when the window glass sandwiches foreign matter. Thus, in normal operation, even if the source voltage VSA of the main control FET QA fluctuates relatively slowly, the reference voltage (reference F
The source voltage VSB of the ETQB can follow the fluctuation, and a sudden change in the source voltage VSA of the main control FET QA when the window glass interposes a foreign substance is not affected by the reference voltage (the source of the reference FET QB). Voltage V
SB) cannot be followed.

That is, the change rate of the load current (the current flowing through the PW motor 102) due to the fluctuation of the driving mechanism sliding force in the normal operation is relatively small, and the source voltage VSA of the main control FET QA due to the load current fluctuation in the normal operation is reduced. Since the fluctuation is relatively small, the reference circuit operates so that the reference voltage (the source voltage VSB of the reference FET QB) always becomes equal to the source voltage VSA of the main control FET QA following the fluctuation.

On the other hand, when foreign matter is caught in the window glass, the number of rotations of the PW motor 102 is reduced and the back electromotive force is reduced, so that the load current is rapidly increased and the source voltage VSA of the main control FET QA is increased. Changes rapidly, so that the reference circuit cannot follow the fluctuation and changes the reference voltage (source voltage VSB of the reference FET QB) to the main control FET.
It cannot be made equal to the source voltage VSA of QA.

At this time, the comparator CMP1 of the control means
Since the "-" input terminal of the comparator CMP1 (the source voltage VSA of the main control FET QA-the offset voltage of the adjustment circuit) is smaller than the "+" input terminal (the source voltage VSB of the reference FET QB), the output of the comparator CMP1 is It becomes “H” level, and the driving circuit 11
1 controls the main control FET QA to be turned off. That is, when the difference between the source voltage VSA of the main control FET QA and the reference voltage exceeds the first predetermined value, the entrapment is detected and the main control FET QA is turned off.

Next, the operation of the power supply control device of the present embodiment will be described based on the circuit configuration of the power supply control device of the present embodiment described above. In the following, FIGS. 1 and 2
A description will be given of a prototype circuit that specifically realizes the above circuit, and the operation thereof will be considered.

First, regarding the resistors R20 and R21, the source voltage VSA of the main control FET QA and the reference FET
When the source voltage VSB of QB is equal, the diode D
4, so that the currents flowing through D5 are equal. As a result, the voltage drops of the diodes D4 and D5 become equal. "When the source voltage VSA of the main control FET QA and the source voltage VSB of the reference FET QB are equal, the voltages of the"-"input terminal and the"-"input terminal of the comparator CMP2 are changed. Are equal "and vice versa.

The comparator CMP2 controls a reference current (a current flowing between the drain and the source of the reference FET QB) so that the voltage of the “−” input terminal becomes equal to the voltage of the “+” input terminal. In addition, the comparator CM
The voltage at the "+" input terminal of P2 is determined by the magnitude of the load current, and changes when the load current changes. Further, the voltage of the "-" input terminal of the comparator CMP2 follows the voltage of the "+" input terminal, and the followability is determined by the time constant of the delay circuit of the reference circuit.

That is, the time constant Tc of the delay circuit of the reference circuit
61 is determined by “C61 × R61”, and in this prototype circuit,
Tc61 = 0.94 μF × 510 kΩ = 480 [ms]
Set to. Gate voltage VG6 of transistor TR61
1 is about 10 when the power supply voltage VB = 12.5 [V].
[V] (measured value). Further, since the range of the output voltage of the comparator CMP2 used is 2 to 12 [V], when rising, VG61 = 2V × ｛1−Exp (−t
/ Tc61)｝, d / dt (VG61) = 2V / Tc6
1 × Exp (−t / Tc61).
G61 = 8V × Exp (−t / Tc61) + 2V, d /
dt (VG61) = − 8V / Tc61 × Exp (−t /
Tc61). Therefore, the rate of change at t = 0 is 2 V / 0.48 s = 4.2 [V / s] when rising.
-8 V / 0.48 s = -16.7 when falling
[V / s]. In other words, the output of the comparator CMP2 is four times faster at the time of falling than at the time of ascending, so that the output of the comparator CMP2 is shifted to the "H" level side.

Assuming that the reference current is Iref, the resistance R62
= 550 [Ω], the rate of change of the reference current Iref when falling is d / dt (Iref) = d / dt (VG6
1) /550=0.03 [A / s]. Main control FE
The current capacity ratio of TQA and reference FET QB is set to 7
When converted to the load side as 80, 780 × d / dt
(Iref) = 23.7 [A / s] (1) Therefore, if the increase rate of the load current is equal to or less than (1), the voltage of the “−” input terminal of the comparator CMP2 becomes the voltage of the “+” input terminal. The source voltage VSA of the main control FET QA and the source voltage VSB of the reference FET QB become equal. According to the actual measurement of the prototype circuit, the power supply voltage VB = 12.5
At [V], the current flowing through the resistor R62 during normal operation is small. At the power supply voltage VB = 14.5 [V], the output of the comparator CMP2 sticks to the “H” level side, and the current flows through the resistor R62. Did not.

By the way, the load current flowing through the PW motor 102 pulsates due to the presence of the commutator even when the main control FET QA is continuously ON. The characteristics (characteristics) of this pulsating current are as follows. First, the cycle of the pulsation is proportional to the rotation speed of the PW motor 102, and the amplitude of the pulsation is proportional to the magnitude of the load current (average value). (In the PW motor 102 used in the prototype circuit, the pulsation amplitude was 10 to 14% of the average value of the load current.) The ratio between the amplitude of the pulsation and the magnitude of the load current (average value) was PW Motor 10
It is a motor unique value determined by the number of poles and the structure (winding resistance, winding layout, field side magnetic flux density distribution).

Next, the PW motor 10 used in the prototype circuit was used.
In No. 2, the total amplitude (peak to peak) of the pulsating current is about 0.52 [A] when the power supply voltage VB = 12.5 [V], and the cycle is about 1.3 [ms]. At this time, the current change rate is 0.52 A / 0.8 ms = 650 [A / s]. On the other hand, since the response speed of the reference circuit is 23.7 [A / s], the increase rate of the pulsating current is about 27 times faster. Therefore, the increase amount of the pulsating current is determined by the entrapment determination current value ΔI
If it exceeds DLim, it will be erroneously determined as “sandwich” even during normal operation.

Therefore, in this embodiment, the comparator C
The pinch determination value is adjusted by an adjustment circuit on the input side of MP1. The judgment value for the voltage difference between the source voltage VSA of the main control FET QA and the source voltage VSB of the reference FET QB is ΔVSlim, and the judgment value for the current difference ID-Iref (ID is the drain current of the main control FET QA) is ΔI.
Dlim, Ron is the on-resistance of the main control FET QA,
If Iref is a reference equivalent current (converted value on the load side), ΔVSLim = ΔIDLim × Ron. When the voltage drop of the resistor VR is VVR, ΔVSLim = 410 mV−VVR. Here, 410 [mV] is the comparator CM
This is the threshold value of P1, which is the input potential difference when the main control FET QA transitions from the on state to the off state.

The current flowing through the resistor VR is the current source IS
It is made up of one transistor TR6, TR7 and consists of a constant current component and a power supply voltage dependent component. VVR = VR × ｛(6.72V−0.62V) / 15k
+ (VB−0.62V) /91k+0.013 (−terminal current)｝ = VR × (0.413mA + VB / 91k) If VR = 648 [Ω], the voltage determination value ΔVSLi
m, current determination value ΔIDLim = ΔVSLim / Ron
Are as follows, respectively. The values in parentheses are actually measured values. That is, when the power supply voltage VB is 14.5 [V], VVR is 371 [mV] (369 [mV]), the voltage determination value ΔVSLim is 39 [mV], and the current determination value ΔID
Lim is 0.78 [A], and the power supply voltage VB is 12.
At 5 [V], VVR is 357 [mV] (352 [m
V]), the voltage judgment value ΔVSLim is 53 [mV], the current judgment value ΔIDLim is 1.06 [A], and when the power supply voltage VB is 10.0 [V], VVR is 339 [mV].
(327 [mV]), and the voltage judgment value ΔVSLim is 71
[MV], and the current determination value ΔIDLim is 1.42 [A].

As can be seen from these results, the voltage judgment value ΔVSLim and the current judgment value ΔIDLim
It is set to increase as B decreases. This is because, as the power supply voltage VB increases with respect to the same determination value, the inertia force of the window glass and the driving mechanism increases and the load increases, so that the load is defined by the maximum value of the operating range of the power supply voltage VB. This is because the determination value is set so as to satisfy the following condition, and in places where the power supply voltage VB where the sandwiched load becomes small is low, the resistance to malfunction is given priority over the load.

The mechanism of the entrapment determination can be explained as follows. That is, when the load current (average value) changes slowly, the reference current Iref becomes the drain current ID.
Follow the average of That is, the reference current Iref matches the current corresponding to the middle of the pulsating current amplitude. Therefore, the amount of increase in the pulsating current with respect to the reference current Iref is not a full amplitude but a single amplitude. For example, the power supply voltage VB =
At 14.5 [V], the current determination value ΔIDLim = 0.7
8 [A] is larger than pulsating current piece amplitude = 0.26 [A]. Also, the power supply voltage VB = 12.5 [V]
At this time, the increase rate of the ID average value immediately before the pinch determination is 2 A / 100 ms = 20 [A / s] at a glance. With this increase rate, the change rate of the reference current Iref is 23.7.
Since it is [A / s], it can be followed.

Further, since the output of the comparator CMP2 is stuck at the "L" level, the capacitor C61 is discharged even when the current pulsation decreases.
This means that they have not been able to follow. 23.7
Consider why the reference current Iref cannot follow the change rate of [A / s]. First, when the average ID value increases, the reference current Iref cannot be applied to the pulsating current, so that the reference current Iref shifts to one side (close to the lower end of the pulsating current). But,
Since the average value can sufficiently follow, the reference current Iref follows near the lower end of the pulsating current amplitude. Next, while the reference current Iref relatively moves from the upper end to the lower end of the pulsating current amplitude, the output of the comparator CMP2 repeats the “H” level and the “L” level.
The movement time is longer than in the case where the outputs of No. 2 are continuously at the "L" level. That is, the reference current Iref falls within the pulsation amplitude range.
When there is, the reference current Iref hardly changes. Furthermore, when the reference current Iref is stuck to the lower end of the pulsating current amplitude, the pulsating current increase amount is not a one-sided amplitude but a full amplitude (peak to peal).
ak).

The total amplitude of the pulsating current is 0.52 [A],
Further, the amplitude increases as the average value of the load current increases. Eventually, the current exceeds the current determination value ΔIDLim and the on / off operation starts. In the case of the power supply voltage VB = 14.5 [V] having a small current determination value ΔIDLim, the current rise until the start of the on / off operation is small, and the power supply voltage VB = the current determination value ΔIDLim is set large.
At 12.5 [V] or 10 [V], the current determination value ΔIDlim is obtained simply by moving the reference current to the pulsating current lower end value.
, The amplitude of the pulsating current needs to be increased. Therefore, the amount of current rise before the determination of the entrapment becomes large.

When the ambient temperature is -30 [° C.], the rise of the current is small even immediately before the judgment of the entrapment. In particular, the power supply voltage VB
At = 14.5 [V], the current hardly rises. This means that the load current average value is larger and the pulsating current amplitude is larger than at room temperature, so that the reference current only approaches the lower end and approaches or exceeds the current determination value ΔIDLim.

In summary, it can be said that the entrapment determination method in this embodiment is amplitude detection (current change detection) using a pulsating current component of the current flowing through the PW motor 102. The judgment is not based on the current change rate.
However, there is a restriction on the reference current change rate, which is a change rate that does not follow the pulsating current, that is, a change rate of 1/27 or less of the pulsating current, which is sufficient for a change in the load current due to a load change of the drive mechanism. It is a necessary condition that the rate of change is such that the rate of change can be followed. Further, the time from the start of the pinching to the stop of the motor is usually about 120 [ms], and it is necessary that the circuit be such that the reference current does not change during this time.

To make a direct determination based on the current change rate, the change rate of the reference current must be faster than the pulsating current change rate. In this case, the change rate (increase rate) of the load current average value due to pinching is much higher. The response becomes fast, and it sufficiently follows the current change caused by the "sandwich", so that it becomes impossible to detect the "sandwich". If the reference current change rate is set slower than the pulsating current change rate,
The pulsating current is always detected. That is, the PW motor 10
As long as the current flowing through 2 has a pulsating component having a fast current change rate, the current change rate alone cannot cope with the pulsating component, and the current change amount needs to be detected.

Next, the pulsating current will be considered. First,
The presence of the pulsating current is not a necessary condition but a sufficient condition in the pinch detection of the present embodiment. If there is no pulsating current, the pinch determination values ΔVSLim and ΔIDLim can be made smaller. Since there is a pulsating current, the current judgment value Δ
IDLim must be set in consideration of the amplitude of the pulsating current. That is, unless the pulsating current amplitude is set to at least 1/2 or more, a malfunction occurs after the normal operation. Actually, it should be set to a value equal to or more than 1/2 + α in consideration of accidental fluctuation of the load current.

On the other hand, the presence of the pulsating current has the following advantages. First, since the amplitude of the pulsating current is proportional to the ID average value, if the drain current ID increases due to pinching, the pulsating current amplitude also increases, and the drain current I
As D increases, the detection becomes easier. Secondly, when the reference current Iref is in the pulsation amplitude range, the reference current Iref hardly changes, so that a change in the drain current ID can be easily detected. (Within the pulsation amplitude range, the output of the comparator CMP2 becomes “H” level, “L” level,
Outside the range, the level becomes “H” level or “L” level. )

Third, during normal operation, since the reference current Iref is within the pulsation amplitude, the output of the comparator CMP2 repeats the “H” level and the “L” level in each pulsation cycle, and the deviation from the drain current ID. Has the effect of reducing This increases the probability that the reference current Iref will stay near the middle of the pulsation amplitude, and has the following effects. First, the reference current Iref with respect to the drain current ID
, The erroneous operation is reduced, and the variation in determination is reduced. Next, when the driving torque increases due to the low temperature or the adhesion of water, dust, or the like, and the drain current ID during normal operation increases, the rate of increase of the drain ID when pinching occurs decreases. At this time, the pulsation current amplitude increases, so that only the relative movement of the reference current Iref within the pulsation amplitude (offset toward the lower end) exceeds the current determination value ΔIDLim, and the rate of increase of the drain current ID is low. Even if it is, the pinch can be reliably detected.

As described above, in the present embodiment, the pulsating current is handled as follows. First, the reference current Ire
The rate of change of f is set smaller than the rate of change of the pulsating current so that the reference current Iref does not follow the pulsating current.
Next, ΔIDLim is set so that the current determination value satisfies ΔIDLim> 1/2 (pulsating current amplitude) + α.

After detecting the entrapment, the main control FET Q
A performs an on / off operation and consequently limits the load current. The outline will be described with reference to FIG. FIG. 3 is an explanatory diagram for explaining the operating characteristics of the main control FET QA and the reference FET QB in the present embodiment. The vertical axis represents the drain current ID, and the horizontal axis represents the drain-source voltage DDS. In the figure, LD1 is a load line of a resistance R66 of the reference circuit, LD2 is a load line of a combined resistance Rr (a parallel combined resistance of the resistors R66 and R62 and the resistance of the transistor TR61) of the reference circuit, and LD3.
Is the combined resistance of the main control FET QA (armature resistance Ra + armature back electromotive force equivalent resistance + DC inductance equivalent resistance)
Load line, LD4 is the main control FET in the locked state
This is the load line (of the armature resistance Ra) of QA.

First, after the entrapment is detected, the main control FE
TQA alternately repeats on / off operation and continuous on operation. During this time, it can be considered that the capacitor C61 of the reference circuit maintains a constant value. If the reference current or the resistance of the reference circuit does not change, the load current when transitioning from the continuous on operation to the off state is always the same, and the operation curve draws the same loop as shown in FIG. Therefore, the drain current I
The upper and lower limits of D are fixed, and the value of the drain current ID is limited to a certain range. (The operation is easy to understand by comparing the resistance of the reference circuit with the load resistance.)

Here, the reason why the value of the capacitor C61 becomes constant will be described. First, the rate of change of the reference current (converted value on the load side) is 23.7 [A / s] = 23.7.
[MA / ms], and the duration of both the on / off operation and the continuous on operation is about 1 [ms] or less. When the change on the reference circuit side during this time is converted to the load side, it is 23.7.
[MA]. On the other hand, the load current is about 3
[A] changes, so 23.7 [mA]
Is a negligible amount. Second, during the continuous ON operation, the first half of the output of the comparator CMP2 is at the “H” level and the second half is at the “L” level. During the subsequent ON / OFF operation, the level becomes “L” level. When the continuous ON operation + ON / OFF operation is regarded as one cycle, the period between the “H” level and the “L” level becomes almost 1: 1, and the continuous “ The change in the charge amount of the capacitor C61 is smaller than that at the H level or the continuous "L" level.

Third, during the on / off operation, the voltage at the gate of the transistor TR61 oscillates at a lower level than during the continuous on operation. The average value at this time is about 3 [V] (the average value of the source voltage VSB of the reference FET QB is 5).
(The difference between [V] and the source-gate voltage 2V of the transistor TR61), the difference between the output of the comparator CMP2 at the "L" level and about 2 [V] is only 1 [V]. On the other hand, the gate voltage of TR61 during the continuous ON operation is 9 [V] or less, and the difference from 12 [V] when the output of the comparator CMP2 is at the "H" level is 3 [V] or more. Therefore, when the output of the comparator CMP2 is "H" level period: "L" level period = 1: 1, the capacitor C61 moves to the discharge side, and the current upper limit value tends to decrease.

Next, an outline of the on / off operation will be described. First, assuming that the reference current is Iref and the load current is ID, when Iref + ΔIDLim <ID (2) is satisfied, the main control FET QA transitions to the off state.
At this time, the source voltage VSA of the main control FET QA decreases toward the ground potential (GND). Reference F
The source voltage VSB of the ETQB is also pulled down by the source follower to the source voltage VSA of the main control FET QA and decreases. At this time, the relationship is VSA <VSB.

In the load side circuit, the load current (drain current) ID hardly changes due to inductance. On the other hand, the reference current Iref is equal to the resistance R66 (2.4 [k
Ω]) flowing through the source voltage V of the reference FET QB.
The reference current Iref is reduced when the source voltage VSB of the reference FET QB is reduced, which is composed of the proportional portion of SB and the constant current flowing through the resistor R62 (see FIG. 3). Further, assuming that the cathode potential of the diode D7 of the clamp diode circuit is VD7, when VSB <VD7 + 0.7V, the main control FET QA transitions to the ON state. After that, the main control F
The source voltage VSA of the ETQA and the source voltage VSB of the reference FET QB start to rise, and VSB> VD7 + 0.7
When the voltage becomes V, the main control FET QA transitions to the off state. This operation is repeated to perform the on / off operation.

Next, the restriction due to the inductance during the on / off operation will be considered. The load current flowing through the PW motor 102 during the on / off operation changes only slightly due to a large inductance. It may be considered that it does not change macroscopically. ON / OFF operation is performed by the main control FET QA
, The load current and the gate-source voltage VGSA correspond one-to-one, and the gate-source voltage VGSA hardly changes during one cycle of the on / off operation.

The gate-source voltage VGSA is constant despite the presence of charges entering and leaving the gate G through the gate circuit of the main control FET QA. In other words, the charge between the gate and the source is constant, including the course of the charge. Further, the above-mentioned electric charge is canceled including the intermediate progress due to the charge / discharge amount of the gate-drain capacitance CGD due to the change of the gate-drain voltage VGD. Since the gate-source voltage VGSA can be regarded as constant, ΔVGD ≒ ΔVDSA
Becomes Further, the gate-drain capacitance CGD depends on the gate-drain voltage VGD, but the dependency does not change in each cycle.

When the above conditions are satisfied, the following results are obtained. First, when the load has a large inductance, the source voltage VSA of the main control FET QA in the on / off operation is high.
Is determined by the gate charge / discharge circuit. Main control FET QA
If the amplitude of the source voltage VSA is the same, the waveform of each cycle is the same and the cycle is constant. The short-term operation of one cycle is not related to the load condition (the armature resistance Ra, the rotation speed of the PW motor 102, and the like). (However, long-term operation over multiple cycles is affected by load conditions.)

Second, the source voltage V of the main control FET QA
When the peak of SA gradually increases, the gate charge is discharged, the gate-source voltage VGS decreases, and the drain current ID decreases. Conversely, if the drain current ID gradually decreases, the peak of the source voltage VS must increase with each cycle. When the peak of the source voltage VS becomes large, the dip of the source voltage VS becomes shallow. This is because if the peak of the source voltage VS becomes large and the dip becomes deep, the gate-source voltage VGS cannot be changed. Third, the gate-source voltage VGSA of the main control FET QA during ON / OFF operation
And therefore the drain current ID depends on the gate charge / discharge circuit (charge / discharge drive voltage, gate series resistance RG, gate-drain capacitance CGD, gate-source capacitance CG
S) and the combined resistance Rr of the reference circuit.

Fourth, the DC load resistance is the armature resistance R
a, armature back electromotive force equivalent resistance, DC inductance equivalent resistance L × d (IDD) / dt / IDD. Fifth
In addition, the power relationship of the elements that determine the on / off operation is as follows. Gate charge / discharge circuit> Load circuit inductance (Drain current ID, VGSA
> VSA (VDS)>VD7> On or off delay time of main control FET QA

Next, the waveform of the source voltage VSA of the main control FET QA during the ON / OFF operation will be considered. Ra is the armature resistance, N is the rotation speed of the PW motor 102, φ is the stator magnetic flux, K is the proportionality constant, IDD is the component of the drain current ID close to DC, and IDA is the drain current ID.
, The source voltage VSA is given by the following equation. VSA = Ra.times. (IDA + IDD) + L.times.d (IDA + IDD) /dt+KN.phi.=Ra.times.IDA+L.times.IDA/dt+Ra.times.IDD + L.times.dIDD / dt + KN.phi. Divided into VSA = VSAC + VSDC VSAC = Ra × IDA + L × dIDA / dt VSDC = Ra × IDA + L × dIDA / dt + KNφ = IDD × (Ra + KNφ / IDD + L × dIDD / dt / IDD) = IDD × RDC where RDC is a DC load. RDC = Ra + KNφ / IDD + L × dIDD / dt /
IDD

The AC component (VSAC) is a waveform determined by the gate charge / discharge circuit, and is not affected by IDD and KNφ.
The DC component (VSDC) and the component IDD of the drain current ID close to the DC are on the DC load line LD3 determined by the armature resistance Ra, the armature back electromotive force equivalent resistance, and the DC inductance equivalent resistance. The load line is L × dIDD / dt =
An attempt is made to converge to a state of 0, that is, a load line determined only by the armature resistance Ra and the armature back electromotive force equivalent resistance. L × dI
It seems that the state of DD / dt = 0 is most stable in terms of energy.

Next, after the on / off operation is started, PW
The process until the motor 102 stops is as follows. First, when the on / off operation is started, the VSAC (= AC load line) swings horizontally with reference to the operating point on the DC load. The amplitude of VSAC swings substantially symmetrically around the operating point on the DC load line.

The DC load line is determined by DC load resistance = RDC = Ra + KNφ / IDD + L × dIDD / dt / IDD. Immediately after entering the on / off operation from the continuous on operation, the drain current ID is turned off.
/ Dt <0. Then, the component IDD close to DC is L
× dIDD / dt / IDD = 0. That is, the DC inductance equivalent resistance changes from the negative state to zero.

DC load resistance RDC and component I close to DC
The DC component VSDC is determined from DD. The peak of the source voltage VSA corresponding to this DC component VSDC, that is, VSA = VSDC
When R (= VSA / ID) becomes larger than the value of the combined resistance Rr of the reference circuit corresponding to the peak of + VSAC, the main control FE
TQA transitions to a continuous ON state. Immediately before the transition to the continuous ON operation, L × dIDD / dt is not yet zero, and L × dIDD / dt <0. Note that Rr
Is a parallel combined resistance of the resistor R66 (2.4 kΩ) and the constant current circuit. Therefore, the resistance value differs depending on the source voltage VSB. See LD2 in FIG. 3 for the load line of Rr.

In the continuous ON operation state, the component IDD (= ID) close to the direct current increases the target at the operating point determined by the load line of Ra + KNφ / IDD. At this time, L × d
IDD / dt is positive, and the absolute value increases as the distance to the target operating point increases. That is, the rate of increase of the component IDD close to DC increases. Further, the target operating point is determined by the number of revolutions N, and as N decreases, the distance to the target operating point increases. Therefore, as the number of revolutions N decreases, the period of the continuous ON operation gradually decreases, and finally, Disappears.

When the decreasing speed of the component IDD close to the direct current exceeds the decreasing speed of the motor rotation speed, that is, d / dt (K
If Nφ) <L × dIDD / dt, the on / off operation and the continuous on operation are alternately repeated. PW motor 10
2, the decrease speed of the rotation speed N is increased, and d / dt (KN
When φ) ≧ L × dIDD / dt, R cannot be larger than Rr, and only ON / OFF operation is performed.

The motor rotation speed N at the ID dip point of the drain current ID waveform is obtained. ID, R,
When Rr is IDmin, Rx, and Rrx, respectively, L ×
dIDD / dt = 0, and Rx = VSA / IDmin = Ra + L × d (IDA) / dt / IDmin + KNφ / IDmin <Rrx / nRa + L × d (IDA) / dt / IDmin + KNφ / IDmin <Rrx / nKNφ <(Rrx / n−Ra) × IDmin−L × d (IDA) / dt The ID dip point N increases as IDmin increases.

When the number of revolutions N further decreases, the component IDD close to DC starts to increase. L × dIDD when N = 0
/ Dt = 0, and RDC = Ra. ID at this time
The magnitude of D is determined by Ra, the waveform (amplitude) of VSAC, and the dummy voltages VD7, Rr as follows. First, Rr
Has a maximum value of 2.4 [kΩ]. Next, VSB−VSA = I
DD × (2.4 kΩ / 780-Ra) = IDD × (3.
08Ω-Ra) When the source voltage VSB of the reference FET QB is VD7 +
When the delay time Td (gate drive circuit delay time + FET delay time) elapses after exceeding 0.7 V, the main control FET
Since QA transitions to the off state, VD7 + 0.7V + β-IDD × (3.08Ω-Ra) becomes the peak of VSAC. Here, β is an increase amount of VSAC during the delay time Td.

Next, the waveform of VSAC indicates an AC load line,
The AC load line becomes horizontal, and the amplitude is determined by the gate charge / discharge characteristics of the main control FET QA. AC load line is armature resistance Ra
Are arranged so as to be equally divided by the DC load line.
That is, the IDD when the PW motor 102 stops is determined by the intersection of the AC load line and the DC load line.

Next, the delay time in the on / off operation will be considered. The on-state and off-state transition timings are determined by the source voltage VSB of the reference FET QB and the dummy voltage VB.
Although determined by the magnitude relationship of D7, there are the following two types of delay times before the main control FET QA actually transitions to the ON state or the OFF state.

The first is a delay time due to the gate drive circuit. The drive circuit 111 for driving the gate includes a source transistor and a sink transistor. These are in such a relationship that one turns off and the other turns on in order to prevent a through current. Therefore, the driving circuit 111 has at least a delay time until the driving transistor is turned off. This delay time is constant and does not change.

The second is the delay time of the FET. In the prototype circuit, rise time tr = 59 [μs], hall time tf = 38 [μs] (VDD = 30 V, ID = 1)
7A, RG = 18Ω, RD = 1.7Ω). In the operation example of this prototype circuit, since the gate-source voltage VGSA hardly moves, a large delay time such as a catalog value does not occur, but it is considered that there is a transmission time and the load-side condition changes. .

The ON / OFF delay time of the FET is Ra + KN
Relative distance between the DC load line determined by φ / IDD, VSA at the start of gate on / off, and ID operating point (= L × dIDD /
dt). When L × dIDD / dt is large, it becomes shorter, and when L × dIDD / dt becomes smaller, it becomes longer. That is, the delay time of the FET is not a constant value but changes under the condition on the load side. The ON / OFF timing required for the VSA waveform determined by the gate charge / discharge characteristics and the ON / OFF determined by VSB to VD7.
That is, it works to fill the gap of the off start timing. It can be said that the ON / OFF operation is achieved thanks to the adjustment function.

As described above, in the power window drive control device of the present embodiment, in the reference circuit for generating the reference voltage (the drain-source voltage VDDS of the reference FET QB), the main control FET QA
Even if the drain-source voltage VDSA changes, the reference voltage operates so as to follow the fluctuation, and when the window glass sandwiches a foreign substance, the sudden change in the drain-source voltage VDSA of the main control FET QA is The tracking speed of the reference voltage is set so as not to follow, and when the window glass catches a foreign substance, the control means controls the drain of the main control FET QA.
Detecting that the difference between the source-to-source voltage VDSA and the reference voltage exceeds a first predetermined value, turning off the main control FET QA,
The off control causes the main control FET QA to be turned on when the drain-source voltage VDSA of the main control FET QA increases and the value exceeds a second predetermined value.

Note that the power window drive control device of the present embodiment can be described in terms of “current”. That is, in a normal operation of a reference circuit for generating a reference current (a drain current of the reference FET QB),
Even if the drain current ID of the main control FET QA fluctuates, it operates so as to make the reference current follow the fluctuation equivalently. The equivalent follow-up speed of the reference current is set so as not to follow equivalently, and when the window glass catches a foreign substance, the control means sets the main control FET Q
When the difference between the drain current ID of A and the equivalent reference current that is a predetermined multiple of the reference current exceeds a third predetermined value, it is detected that the window glass has caught foreign matter, and the main control FET QA is turned off, The main control FET QA is turned on when the drain current ID of the main control FET QA decreases and falls below a fourth predetermined value by the off control.

As described above, in the power window drive control device according to the present embodiment, the reference current follows the gradual change in the load current during normal operation, and the amount of sudden change in the load current with respect to the reference current is reduced when pinching occurs. Detects the entrapment and controls the main control FET QA on / off,
The load current ID, that is, the torque of the PW motor 102 is limited. That is, in the present embodiment, the load current ID is directly detected to detect the entrapment, as compared with the conventional pulse type in which the entrapment is detected based on the rotational speed or the rate of change of the rotational speed. Can be directly detected, and the detection accuracy can be improved.

After the entrapment is detected, the main control FET Q
A can be turned on / off to limit the load current ID, thereby controlling the torque of the PW motor 102.
Further, even after the trapping is once detected, the normal operation can be resumed when the load becomes normal during the on / off control. Thus, it is not necessary to set the determination load high due to fear of malfunction, and the determination load can be set low. As a result, highly accurate detection is possible.

Further, unlike the related art, since a microcomputer and a sensor are not required and the configuration is simple, the system scale of the power window drive control device can be reduced, and realization at low cost is possible. In addition, conventionally, it was necessary to correct the entrapment determination threshold value in order to keep the reversal load after the entrapment of foreign matter was detected. Becomes

Further, by integrating the main control FET QA, the reference circuit, the control means and the like on the same chip 110, the circuit configuration of the device can be reduced, the mounting space can be reduced, and the device cost can be reduced. Also, the main control FET QA and the reference FET are mounted on the same chip.
By forming the QB, it is possible to eliminate (reduce) the influence of common-mode error factors in current detection, that is, the effects of power supply voltage, temperature drift, and variation between lots.

[Modification] In a first modification, the PW motor 102 is reversed when the on / off operation after the entrapment determination continues for a predetermined time. FIG. 4 is a circuit diagram of a timer used for that purpose. Show. It should be noted that the circuit of FIG.
And connected to the circuit of FIG.
In addition, the timer controls the main control FET during the current limit period.
The QA measures the time during the ON / OFF operation, and is about 100 to 150 [ms].

In FIG. 4, the timer includes resistors R31 to R31.
46, a capacitor C21, a Zener diode ZD6,
Diodes D15-D18 and transistors Q11-
Q19 and a switch SW2.

The base of the transistor Q12 is connected to the resistor R3
6 and a diode D16 are connected to the gate circuit of the main control FET QA. When the potential of the connection point G is lower than the power supply voltage VB by 1.4 [V] or more, the transistor Q
12 changes to the ON state. This condition is satisfied when the driving circuit 111 is turned off (the source transistor is turned off and the sink transistor is turned on), and the driving circuit 1
11 is turned on (the source transistor is turned on and the sink transistor is turned off).
This condition may be satisfied during the ON / OFF operation of the TQA.

The transistors Q14 and Q13 are connected to the main control F
When the source voltage VSA of the ETQA is at least 0.7 [V] with respect to the ground potential (GND), the ETQA transitions to the ON state. Therefore, when the main control FET QA is performing the on / off operation, the transistor Q12 is in the on state, and the main control FET QA is in the on state.
The condition that the source voltage VSA of A is 0.7 [V] or more is satisfied, and the capacitor C21 is charged through the resistor R35. When the main control FET QA is continuously in the on state or the off state, either the transistor Q12 or the transistor Q13 is turned off, and the capacitor C21 is not charged but is discharged through the diode D15 and the resistor R40. The charging time constant is 0.94 μF × 300 kΩ = 28
2 [ms] and the discharge time constant is 0.94 μF × 10
kΩ = 9.4 [ms].

When the potential of the capacitor C21 becomes higher than the sum of the emitter-base voltage 0.7V of the transistor Q11 and the Zener potential 4.7V of the zener diode ZD6, the transistor Q11 is turned on. Transitions to. As a result, the transistor Q10 transitions to the ON state, and subsequently, the transistor Q19 transitions to the ON state. As a result, the voltage difference between the two inputs of the comparator CMP1 becomes 410 [mV] or more, and the driving circuit 11
1 performs off control, and the main control FET QA is shut off. Note that transistors Q11 and Q10 form a latch circuit, and once transited to the ON state, power supply voltage VB
As long as is not lost, keep the ON state.

Next, in a second modification, the PW motor 102
Even with the inrush current at the time of start-up, the on / off operation of the main control FET QA occurs and the start-up current is limited. Therefore, a mask time (about 200 [ms]) is provided by a mask circuit.
This is to avoid the ON / OFF operation of the main control FET QA when the PW motor 102 is started. FIG. 5 shows a circuit diagram of the mask circuit. It should be noted that the circuit of FIG.
Is connected to the circuit of FIG.

In FIG. 5, the mask circuit includes a resistor R63
~ R65, capacitor C62, Zener diode ZD
8, diodes D21 to D24 and transistor Q2
0.

When the switch SW1 is turned on, the resistance R63
The capacitor C62 is charged through the transistor Q20, the charging current flows to the base of the transistor Q20, and the transistor TR20
62 is turned on. Due to the transition of the transistor TR61 to the ON state, the resistor R62 is grounded, and the resistance of the reference circuit becomes a parallel combined resistance of R62 and R66 (0.447 [kΩ] in the prototype circuit).

The load resistance equivalent to the combined resistance of this reference circuit is represented by a current capacity ratio (current sensing ratio) = 780.
Therefore, 0.447 kΩ / 780 = 0.6 [Ω], and when the power supply voltage VB = 12.5 [V], it is possible to flow up to the drain current ID = 12.5 V / 0.6Ω = 20.8 [A]. Become. Since the peak of the rush current of the PW motor 102 is 17 [A], even if the rush power flows, the main control FET
QA will not operate on / off.

On the other hand, when the transistor Q20 is turned on, the resistor R62 is grounded, and the lower potential of the capacitor C61 (that is, the gate voltage of the transistor TR61) is reduced to a voltage obtained by dividing the power supply voltage VB by the resistors R64 and R65. The voltage drops to a voltage obtained by adding a forward voltage drop of 0.7 V to D23. The upper potential of the capacitor C61 is the source voltage VSB of the reference FET, which is almost equal to the power supply voltage VB. Therefore, the capacitor C61 is charged to a potential difference of approximately VB / 2.

The plus side potential of the capacitor C62 is 8.2.
When the voltage reaches [V], the charging current of the capacitor C62 stops, and the transistor Q20 transitions to the off state. Since there is about 150 [ms] during this time, the inrush current of the PW motor 102 has almost ended. At this time, the resistors R62 and R64
Of the transistor TR6 is connected to the resistor R62.
A current determined by one gate voltage (= approximately VB / 2) flows. Since the equivalent resistance on the reference circuit side is smaller than the resistance on the load side, the source voltage VSB of the reference FET QB <the source voltage VSA of the main control FET QA, and the load current maintains a continuous ON state. The output of the comparator CMP2 becomes "H" level, the capacitor C61 is discharged, and the gate voltage of the transistor TR61 increases toward the source voltage VSB of the reference FET QB = the source voltage VSA of the main control FET QA.

Source voltage VSB of reference FET QB
= When the source voltage VSA of the main control FET QA is reached, the comparator CMP2 outputs an “H” level or an “L” level so that the source voltage VSB of the reference FET QB becomes equal to the source voltage VSA of the main control FET QA. At this point, the mask period ends.

[0135]

As described above, according to the power window drive control device of the present invention, the reference current follows the gradual change of the load current during the normal operation, and the load current sharply increases with respect to the reference current when the pinch occurs. The amount of change is detected to detect entrapment, and the semiconductor switch is turned on / off to limit the load current, that is, the motor torque. Compared to one that detects entrapment based on the rate,
In the present invention, since the load current is directly detected to detect the entrapment, the torque of the motor can be directly detected,
Detection accuracy can be improved.

After the entrapment is detected, the semiconductor switch is turned on / off to limit the load current, thereby controlling the motor torque. Even after the entrapment is detected, the on / off control is performed. When the load becomes normal during the OFF control, the normal operation can be restored. This allows
There is no need to set the judgment load high because of fear of malfunction, and the judgment load can be set low. As a result, highly accurate detection is possible. Furthermore, unlike the related art, since a microcomputer and a sensor are not required and the configuration is simple, the system scale of the power window drive control device can be reduced, and realization at low cost is possible. In addition, conventionally, it was necessary to correct the entrapment determination threshold value in order to keep the reversal load after the entrapment of foreign matter was detected. Becomes

[Brief description of the drawings]

FIG. 1 is a configuration diagram of a power window drive control device according to an embodiment of the present invention.

FIG. 2 is a detailed circuit configuration diagram of an input side of a comparator CMP1 of the embodiment.

FIG. 3 is an explanatory diagram illustrating operation characteristics of a main control FET and a reference FET in the embodiment.

FIG. 4 is a circuit diagram of a timer according to a first modified example.

FIG. 5 is a circuit diagram of a mask circuit according to a second modification.

FIG. 6 is a configuration diagram of a conventional power window drive control device.

[Explanation of symbols]

 101 Power supply VB Power supply voltage GND Ground potential 102 Load (PW motor) SW1, SW2 Switch 111 Drive circuit QA Main control FET (semiconductor switch) QB Reference FET RG, R2 to R66 Resistance C21, C61, C62 Capacitor IS1 Current source ZD1 to ZD8 Zener diode D4 to D24 Diode TR61 Transistor (pMOSFET) CMP1, CMP2 Comparator 303 Inrush current mask circuit 304 On / off frequency integration circuit 305 Charge pump 306 Overheat cutoff protection circuit 110 Chip portion to be analog integrated T1 to T18 Input / output terminals

Claims (18)

[Claims] 1. A power window drive control device comprising a motor, for opening and closing a window glass by a driving force of the motor, wherein the power window drive control device is switching-controlled in accordance with a control signal supplied to a control signal input terminal. A semiconductor switch for controlling power supply to a motor, and a reference circuit for generating a reference voltage. In a normal operation, even if a voltage between terminals of the semiconductor switch fluctuates, the reference voltage follows the fluctuation. A reference circuit that sets the following speed of the reference voltage so that it cannot follow a sudden change in the voltage between the terminals of the semiconductor switch when the window glass sandwiches foreign matter; and a terminal of the semiconductor switch. When the difference between the intermediate voltage and the reference voltage exceeds a first predetermined value, the semiconductor switch is turned off, and the terminal of the semiconductor switch is controlled by the off control A power window drive control device, comprising: control means for turning on the semiconductor switch when the inter-voltage increases and the value exceeds a second predetermined value. 2. A power window drive control device, comprising a motor, for opening and closing a window glass by a driving force of the motor, wherein the power window drive control device is switching-controlled in accordance with a control signal supplied to a control signal input terminal. A semiconductor switch for controlling power supply to a motor, and a reference circuit for generating a reference current. In a normal operation, even if a current flowing between terminals of the semiconductor switch fluctuates, the reference current is equivalent to the fluctuation. Operates so that the reference current does not follow an abrupt change in the current flowing between the terminals of the semiconductor switch when the window glass sandwiches a foreign substance. And a difference between a current flowing between terminals of the semiconductor switch and an equivalent reference current that is a predetermined multiple of the reference current exceeds a third predetermined value. Control means for controlling the semiconductor switch to be off, and for controlling the semiconductor switch to be on when the current flowing between the terminals of the semiconductor switch is reduced by the off control and the value thereof becomes lower than a fourth predetermined value. Power window drive control device characterized by the above-mentioned. 3. The reference circuit includes a circuit connected in parallel to the semiconductor switch and the motor, and a circuit in which a second semiconductor switch and a second load that are switching-controlled in accordance with the control signal are connected in series with each other. 3. The power window drive control device according to claim 1, wherein a voltage between terminals of the two semiconductor switches is set as the reference voltage, and a current flowing between terminals of the second semiconductor switch is set as the reference current. 4. The semiconductor device according to claim 1, wherein a characteristic of a reference voltage or a reference current of the reference circuit is substantially equivalent to a voltage between terminals of the semiconductor switch or a current flowing between terminals. 4. The power window drive control device according to 3. 5. The semiconductor switch according to claim 3, wherein the semiconductor switch and the second semiconductor switch have equivalent characteristics with respect to a transient voltage characteristic of a voltage between terminals when transitioning from an off state to an on state. 5. The power window drive control device according to 4. 6. The current capacity of the second semiconductor switch is smaller than the current capacity of the semiconductor switch, and the resistance ratio between the motor and the second load is as small as possible with the current capacity ratio of the semiconductor switch and the second semiconductor switch. 6. The method according to claim 3, wherein the setting is made in inverse proportion.
4. The power window drive control device according to 1. 7. The resistance value of the second load is set lower than a value obtained by multiplying the maximum resistance value of the motor when the motor is rotating normally by the resistance value ratio. The power window drive control device according to claim 6, wherein: 8. The method according to claim 1, wherein the second load includes a first resistor and the first resistor.
A circuit in which a second resistor connected in parallel with a resistor and a third semiconductor switch are connected in series. The reference circuit includes a terminal voltage of the semiconductor switch on the motor side, and a terminal voltage of the second semiconductor switch. (2) a comparator for comparing the terminal voltage on the load side with a delay, and a delay circuit for delaying an output of the comparator for a predetermined time and supplying the output to a control signal input terminal of the third semiconductor switch; The speed is set to be lower than a changing speed of a voltage between terminals of the semiconductor switch or a changing speed of a current flowing between terminals of the semiconductor switch when the window glass sandwiches a foreign substance.
7. The power window drive control device according to 4, 5, or 6. 9. The power window drive control device according to claim 8, wherein the comparator is configured using an operational amplifier. 10. The method according to claim 1, wherein the first predetermined value or the third predetermined value depends on a voltage of the power supply, and is set to decrease as the voltage of the power supply increases. The power window drive control device according to 1, 2, 3, 4, 5, 6, 7, 8, or 9. 11. When a current flowing between terminals of the semiconductor switch or a current flowing through the motor has a pulsation component due to a commutator action, a following speed of a reference voltage of the reference circuit or an equivalent of a reference current of the reference circuit. Following speed,
The current is set to be slower than the change rate of the pulsating component so that the reference voltage or the reference current of the reference circuit does not follow the change due to the pulsating component of the current, and the first predetermined value is set by the on-resistance of the semiconductor switch. 4. The method according to claim 1, wherein the divided value or the third predetermined value is set to a value equal to or more than half the amplitude of the pulsating component of the current.
The power window drive control device according to 4, 5, 6, 7, 8, 9, or 10. 12. The semiconductor device according to claim 11, wherein a difference between a voltage between terminals of the semiconductor switch and the reference voltage exceeds a first predetermined value, or a difference between a current flowing between terminals of the semiconductor switch and the reference current. When the difference from the equivalent reference current, which is a predetermined multiple, exceeds a third predetermined value, while the armature back electromotive force of the motor is large, an on / off operation in which the semiconductor switch repeats a transition between an on state and an off state. And the semiconductor switch is controlled so as to alternately repeat a continuous ON operation that operates continuously in an ON state, so that the current flowing through the motor is reduced during the ON / OFF operation, and the motor is controlled during the continuous ON operation. The current flowing through the motor is increased, the current flowing through the motor is limited to a certain range, the armature back electromotive force of the motor is reduced, the rotation speed of the motor is reduced, and the motor is locked. During the state, the semiconductor switch is controlled to perform only the on / off operation,
The range of a current flowing through the motor is limited.
Or a power window drive control device according to item 11. 13. A current value flowing between terminals of the semiconductor switch when the semiconductor switch transitions from the continuous ON operation to the ON / OFF operation, or an upper limit value of the current,
And a value of a current flowing between terminals of the semiconductor switch or a lower limit value of the current when the semiconductor switch transits from an on / off operation to a continuous on operation is the first predetermined value or the third predetermined value, 13. The power window drive control device according to claim 12, wherein the power window drive control device is set by a first resistor and a delay time of the delay circuit. 14. The control device according to claim 1, further comprising a timer for measuring a current limiting period for limiting a current flowing through the motor, wherein when the timer has counted a predetermined time, the rotation direction of the motor is reversed. Claim 12 or 1
4. The power window drive control device according to 3. 15. The timer measures the time during which the semiconductor switch is in the on / off operation during the current limit period, and resets the timed content of the timer when the semiconductor switch transits to the continuous on operation. The power window drive control device according to claim 14, wherein 16. The semiconductor device according to claim 1, further comprising overheat protection means for turning off the semiconductor switch to protect the semiconductor switch when the semiconductor switch is overheated.
The power window drive control device according to 6, 7, 8, 9, 10, 11, 12, 13, 14, or 15. 17. A semiconductor device according to claim 1, further comprising mask means for inhibiting on / off control of said semiconductor switch by said control means for a certain period after said semiconductor switch is turned on. , 4, 5, 6, 7, 8,
The power window drive control device according to 9, 10, 11, 12, 13, 14, 15, or 16. 18. The semiconductor switch, the reference circuit,
2. The method according to claim 1, wherein the control unit, the timer, the overheat protection unit, or the mask unit is formed on a same chip.
The power window drive control device described in 0, 11, 12, 13, 14, 15, 16 or 17.