JP2001327167A - Switching power circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To attain an increase in power conversion efficiency and a reduction in the size and weight of a circuit. SOLUTION: An active clamp circuit is provided on the primary side of a compound resonant switching converter having a voltage resonant converter on its primary side and a parallel resonant circuit on its secondary side to clamp a parallel resonant voltage pulse generated on both sides of the primary parallel resonant capacitor and suppress its level. A low breakdown voltage article can be selected concerning breakdown voltage of respective devices such as a switching device provided on a power circuit and primary parallel resonant capacitor. An auxiliary switching device Q2 on the active clamp circuit has a structure driven by a self-exciting oscillation circuit formed by winding the primary winding on an insulating converter transformer by 1T, thereby simplifying a riving circuit system of the active clamp circuit.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a switching power supply circuit provided as a power supply for various electronic devices.

[0002]

2. Description of the Related Art As a switching power supply circuit, a circuit employing a switching converter of a type such as a flyback converter or a forward converter is widely known. Since these switching converters have a rectangular switching operation waveform, there is a limit in suppressing switching noise. Also, due to its operating characteristics,
It has been found that there is a limit in improving the power conversion efficiency. Therefore, the present applicant has previously proposed various switching power supply circuits using various resonance type converters.
The resonance type converter can easily obtain high power conversion efficiency and realize low noise because the switching operation waveform is sinusoidal. It also has the advantage that it can be configured with a relatively small number of parts.

FIG. 7 is a circuit diagram showing an example of a prior art switching power supply circuit which can be constructed based on the invention proposed by the present applicant. In the power supply circuit shown in this figure, first, as a rectifying and smoothing circuit for receiving a commercial AC power supply (AC input voltage VAC) to obtain a DC input voltage, a full-wave rectifying circuit including a bridge rectifying circuit Di and a smoothing capacitor Ci. And generates a rectified smoothed voltage Ei corresponding to a level that is one time the AC input voltage VAC.

As a switching converter which receives the rectified smoothed voltage Ei (DC input voltage) and is intermittent, the following is known.
A voltage resonance type converter including a stone switching element Q1 and performing a so-called single-ended switching operation is provided. The voltage resonance type converter here has a separately excited configuration, and for example, a MOS-FET is used as the switching element Q1. The drain of the switching element Q1 is connected to the positive electrode of the smoothing capacitor Ci via the primary winding N1 of the insulating converter transformer PIT, and the source is connected to the primary side ground.

The drain of the switching element Q1
A parallel resonance capacitor Cr is connected in parallel between the sources. The capacitance of the parallel resonance capacitor Cr and the primary winding N1 of the insulation converter transformer PIT
The primary side parallel resonance circuit is formed by the leakage inductance obtained as described above. The resonance operation of the parallel resonance circuit is obtained in accordance with the switching operation of the switching element Q1, so that the switching operation of the switching element Q1 is a voltage resonance type.

The drain of the switching element Q1
Since a clamp diode DD formed by a so-called body diode is connected in parallel between the sources, a path for a clamp current flowing during a period in which the switching element is turned off is formed. Further, in this case, the drain of the switching element Q1 is connected to an oscillating circuit 11 in the switching drive unit 10 described below.
The output of the drain input to the oscillation circuit 11 is used for variably controlling the ON period of switching at the time of switching frequency control as described later.

The switching element Q1 includes an oscillation circuit 1
The switching drive unit 10 integrally includes the drive circuit 1 and the drive circuit 12. The switching drive is performed, and the switching frequency is variably controlled for constant voltage control. In this case, the switching drive unit 10 includes:
For example, it is provided as one integrated circuit (IC). The switching drive unit 10 is connected to the line of the rectified and smoothed voltage Ei via the starting resistor Rs. For example, at the time of starting the power supply, the switching driving unit 10 is started by applying the power supply voltage via the starting resistor Rs. It has been like that.

Oscillation circuit 11 in switching drive unit 10
Then, an oscillating operation is performed to generate and output an oscillating signal.
Then, the drive circuit 12 converts this oscillation signal into a drive voltage and outputs it to the gate of the switching element Q1. Thereby, the switching element Q1
Performs a switching operation based on an oscillation signal generated by the oscillation circuit 11. Therefore, the switching frequency of the switching element Q1 and the duty of the ON / OFF period within one switching cycle are determined by the oscillation circuit 4
Is determined depending on the oscillating signal generated at.

Here, the oscillation circuit 11 varies the oscillation signal frequency (switching frequency fs) based on the level of the secondary side DC output voltage EO input via the photocoupler 2 as described later. Is to do. In addition, while changing the switching frequency fs, the switching element Q1 is turned off during the period TO.
The FF is kept constant, and the oscillation signal waveform is controlled so that the period TON (conduction angle) during which the switching element Q1 is on is varied. The variable control of the period TON is performed based on the level of the switching resonance pulse voltage V1 obtained at both ends of the parallel resonance capacitor Cr. The operation of the oscillation circuit 12 stabilizes the secondary-side DC output voltage EO as described later.

The insulated converter transformer PIT transmits the switching output of the switching element Q1 to the secondary side. As shown in FIG.
For example, an EE-type core in which E-type cores CR1 and CR2 made of a ferrite material are combined so that their magnetic legs are opposed to each other is provided. The wire N1 and the secondary winding N2 are wound separately. A gap G is formed for the center magnetic leg as shown in the figure.
As a result, loose coupling with a required coupling coefficient can be obtained. Gap G is E type core CR1, CR
The two center magnetic legs can be formed by making them shorter than the two outer magnetic legs. The coupling coefficient k is set to a loosely coupled state of, for example, k ≒ 0.85, so that a saturated state is hardly obtained.

The end of the primary winding N1 of the insulating converter transformer PIT is connected to the drain of the switching element Q1 as shown in FIG. 7, and the start of the winding is connected to the positive electrode of the smoothing capacitor Ci (rectified smoothing voltage Ei). ) And connected. Accordingly, by supplying the switching output of the switching element Q1 to the primary winding N1, an alternating voltage having a cycle corresponding to the switching frequency is generated.

On the secondary side of the insulating converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. In this case, the secondary side parallel resonance capacitor C2 is connected in parallel to the secondary winding N2, so that the leakage inductance L2 of the secondary side winding N2 and the capacitance of the secondary side parallel resonance capacitor C2 are determined. A parallel resonance circuit is formed. With this parallel resonance circuit,
The alternating voltage induced in the secondary winding N2 becomes a resonance voltage.
That is, a voltage resonance operation is obtained on the secondary side.

That is, in this power supply circuit, a primary side is provided with a parallel resonance circuit for performing a voltage resonance type switching operation, and a secondary side is provided with a parallel resonance circuit for obtaining a voltage resonance operation. In the present specification, such a switching converter configured to operate with a resonance circuit provided on the primary side and the secondary side is also referred to as a “composite resonance type switching converter”.

On the secondary side of the power supply circuit formed as described above, a rectifying and smoothing circuit comprising a bridge rectifying circuit DBR and a smoothing capacitor CO is provided to obtain a secondary side DC output voltage EO. ing. That is, in this configuration, a full-wave rectification operation is obtained by the bridge rectification circuit DBR on the secondary side. In this case, the bridge rectifier circuit D
BR receives the resonance voltage supplied from the secondary-side parallel resonance circuit and generates a secondary-side DC output voltage EO corresponding to the alternating voltage induced in the secondary winding N2 at substantially the same level as the alternating voltage. Also, the secondary side DC output voltage EO is
In a state where the primary side and the secondary side are DC-insulated by passing through the oscillation circuit 1, the oscillation circuit 1 in the switching drive unit 10 on the primary side is connected.
1 is input.

Incidentally, the insulation converter transformer PIT
On the secondary side, the connection relationship between the polarity (winding direction) of the primary winding N1 and the secondary winding N2 and the rectifier diode DO (DO1, DO2), and the alternating voltage excited by the secondary winding N2 , The mutual inductance M between the inductance L1 of the primary winding N1 and the inductance L2 of the secondary winding N2.
May be + M or -M.
For example, when the circuit is equivalent to the circuit shown in FIG. 10A, the mutual inductance is + M, and when the circuit is equivalent to the circuit shown in FIG. 10B, the mutual inductance is -M. If this is made to correspond to the operation on the secondary side shown in FIG. 7, when the alternating voltage obtained in the secondary winding N2 has a positive polarity, the operation in which the rectified current flows through the bridge rectifier circuit DBR is the + M operation mode. (Forward operation). Conversely, when the alternating voltage obtained in the secondary winding N2 has a negative polarity, the operation in which a rectified current flows through the bridge rectifier diode DBR is the -M operation mode (flyback operation). Can be seen. Each time the alternating voltage obtained on the secondary winding N2 becomes positive / negative, the mutual inductance operates in the mode of + M / -M.

In such a configuration, the power increased by the action of the primary side parallel resonance circuit and the secondary side parallel resonance circuit is supplied to the load side, and the power supplied to the load side also increases accordingly, and the maximum load The rate of increase in power also increases. this is,
As described above with reference to FIG. 9, the gap G is formed in the insulating converter transformer PIT and loose coupling is performed with a required coupling coefficient, thereby achieving a state in which a saturation state is hardly obtained. Things. For example,
If the gap G is not provided for the insulating converter transformer PIT, there is a high possibility that the insulating converter transformer PIT will be in a saturated state during flyback operation and the operation will be abnormal. It is difficult to want to be done.

The stabilizing operation in the circuit shown in FIG. 7 is as follows. Primary side switching drive unit 1
As described above, the secondary-side DC output voltage EO is input to the oscillation circuit 11 within 0 through the photocoupler 30. In the oscillation circuit 11, the frequency of the oscillation signal is varied and output according to the level change of the input secondary-side DC output voltage EO. This means that the switching frequency of the switching element Q1 is changed. As a result, the resonance impedance between the primary-side voltage resonance type converter and the insulation converter transformer PIT changes, and the transmission impedance to the secondary side of the insulation converter transformer PIT is changed. The energy that is given will also change.
As a result, the secondary side DC output voltage EO is controlled to be constant at a required level. That is,
The power supply is stabilized.

In the power supply circuit shown in FIG. 7, when the switching frequency is varied in the oscillation circuit 11, as described above, the period TOFF during which the switching element Q1 is turned off is set to be constant. Thus, the ON period TON is variably controlled. That is,
In this power supply circuit, as a constant voltage control operation, an operation is performed to variably control a switching frequency, thereby performing resonance impedance control on a switching output, and at the same time, controlling a conduction angle of a switching element in a switching cycle (PWM control). Is something that is also to be done. This complex control operation is realized by a set of control circuit systems. In this specification,
Such a complex control is also called a “complex control method”.

FIG. 8 shows another example of a power supply circuit configured based on the content proposed by the present applicant. In this figure, the same parts as those in FIG. On the primary side of the power supply circuit shown in FIG.
A self-excited configuration is shown as a voltage resonance type converter circuit that performs single-ended operation by one switching element Q1. In this case, a high withstand voltage bipolar transistor (BJT; junction transistor) is employed as the switching element Q1.

The base of the switching element Q1 is connected to the positive side of the smoothing capacitor Ci (rectified smoothing voltage Ei) via the base current limiting resistor RB-starting resistor RS to obtain a base current at the time of starting from the rectifying smoothing line. Like that. Further, a series resonance circuit for self-excited oscillation driving, which is composed of a series connection circuit of a drive winding NB, a resonance capacitor CB, and a base current limiting resistor RB, is connected between the base of the switching element Q1 and the primary side ground. A clamp diode DD inserted between the base of the switching element Q1 and the negative electrode (primary ground) of the smoothing capacitor Ci forms a path for a clamp current flowing when the switching element Q1 is turned off. , The collector of switching element Q1 is connected to one end of primary winding N1 of insulating converter transformer PIT, and the emitter is grounded.

A parallel resonance capacitor Cr is connected in parallel between the collector and the emitter of the switching element Q1. In this case as well, the primary parallel resonance circuit of the voltage resonance type converter is formed by the capacitance of the parallel resonance capacitor Cr itself and the leakage inductance L1 on the primary winding N1 side of the insulating converter transformer PIT.

The orthogonal control transformer PRT shown in FIG.
Is a saturable reactor on which a resonance current detection winding ND, a drive winding NB, and a control winding NC are wound. This orthogonal control transformer PRT drives the switching element Q1 and is provided for constant voltage control. As the structure of the orthogonal control transformer PRT, although not shown, a three-dimensional core is formed by joining the ends of the two magnetic legs of two double U-shaped cores having four magnetic legs. . Then, a resonance current detection winding ND and a drive winding NB are wound around the predetermined two magnetic legs of the three-dimensional core in the same winding direction, and the control winding NC is connected to the resonance current detection winding. It is constructed by winding in a direction orthogonal to the winding ND and the driving winding NB.

In this case, the resonance current detecting winding ND of the orthogonal control transformer PRT is inserted in series between the positive electrode of the smoothing capacitor Ci and the primary winding N1 of the insulating converter transformer PIT, so that the switching element Q1 Is transmitted to the resonance current detection winding ND via the primary winding N1. In the orthogonal control transformer PRT, the switching output obtained in the resonance current detection winding ND is induced in the drive winding NB via the transformer coupling, so that an alternating voltage as a drive voltage is applied to the drive winding NB. appear. This drive voltage is output from the series resonance circuit (NB, CB) forming the self-excited oscillation drive circuit to the base of the switching element Q1 as a drive current via the base current limiting resistor RB. As a result, the switching element Q1 performs a switching operation at a switching frequency determined by the resonance frequency of the series resonance circuit. The switching output obtained at the collector is transmitted to the primary winding N1 of the isolated converter transformer PIT.

The insulation converter transformer PIT provided in the circuit shown in FIG. 8 also has the same structure as that described above with reference to FIG. 9, so that the primary side and the secondary side are A loosely coupled state is obtained.

The secondary parallel resonance capacitor C2 is also connected in parallel to the secondary winding N2 on the secondary side of the insulating converter transformer PIT in the circuit shown in FIG. A circuit is formed and therefore
This power supply circuit also has a configuration as a complex resonance type switching converter.

On the secondary side of the power supply circuit, one diode DO and a smoothing capacitor C are connected to the secondary winding N2.
With the provision of the half-wave rectifier circuit made of O, the secondary-side DC output voltage EO can be obtained by the half-wave rectification operation of only the forward operation. In this case, the secondary DC output voltage EO is also branched and input to the control circuit 1, and the control circuit 1 uses the DC output voltage EO as a detection voltage.

In the control circuit 1, the level of the control current (DC current) flowing through the control winding NC is varied in accordance with the change in the DC output voltage level EO on the secondary side, so that the orthogonal control transformer PRT is wound. The inductance LB of the driven winding NB is variably controlled. Thereby, the resonance condition of the series resonance circuit in the self-excited oscillation drive circuit for the switching element Q1 formed including the inductance LB of the drive winding NB changes. This changes the switching frequency of the switching element Q1, thereby stabilizing the DC output voltage on the secondary side. Further, even in the configuration of the constant voltage control having such an orthogonal control transformer PRT, since the primary-side switching converter is of the voltage resonance type, the switching control in the switching cycle is performed simultaneously with the variable control of the switching frequency. An operation is performed as a complex control system that also performs element conduction angle control (PWM control).

FIG. 11 is a waveform chart showing the operation of the primary side voltage resonance type converter in the power supply circuit shown in FIGS. 7 and 8. FIGS. 11A, 11B and 11C show the case where the AC input voltage VAC = 100 V and the maximum load power Pomax, respectively.
= 200 W, and FIG. 11 (d) (e) (f)
Shows the operation at the time of no load where the AC input voltage VAC is 100 V and the minimum load power Pomin is 0 W.

When the switching element Q1 performs a switching operation, a period TOFF during which the switching element Q1 is turned off.
In, the resonance operation of the primary-side parallel resonance circuit is obtained. As a result, the parallel resonance voltage V1 obtained at both ends of the parallel resonance capacitor Cr is as shown in FIG.
As shown in (d), the waveform becomes a waveform in which a sinusoidal resonance pulse appears in the period TOFF. Further, since the parallel resonance operation is obtained during the period TOFF, the parallel resonance current Icr flowing through the parallel resonance capacitor Cr is as shown in FIG.
As shown in (c) and (f), during this period TOFF, the current flows so as to transition from the positive direction to the negative direction in a substantially sinusoidal shape.

Here, as can be seen by comparing FIGS. 11A and 11D, the switching frequency fs is controlled to increase as the load power Po decreases, and the period TOFF Is constant, the switching frequency fs (switching cycle) is varied by varying the period TON during which the switching element Q1 is turned on. That is, the operation as the above-described composite control method is shown. Further, in the configuration of the voltage resonance type converter shown in FIGS. 7 and 8, the level of the parallel resonance voltage V1 changes according to the load power fluctuation. For example, when the maximum load power Pomax = 200W, it becomes 550Vp, When the power Pomin = 0 W, the voltage is 300 Vp. That is, the parallel resonance voltage V1 tends to increase as the load power increases.

The switching output current IQ1 flowing to the drain or the collector of the switching element Q1 is shown in FIG.
As shown in 1 (b) and (e), the signal flows at the 0 level during the period TOFF and flows according to the waveform shown in the period TON. The level of the switching output current IQ1 also tends to increase as the load power Po increases. For example, according to this figure, the maximum load power Pomax =
At 200 W, it becomes 3.8 A, and the minimum load power Pomin
When = 0W, it becomes 1A.

As a characteristic of the power supply circuit shown in FIGS. 7 and 8, the switching frequency f with respect to the AC input voltage VAC when the maximum load power Pomax is 200 W is shown.
FIG. 12 shows s, the period TOFF and the period TON in the switching cycle, and the fluctuation characteristics of the parallel resonance voltage V1.

As shown in FIG. 12, first, as the switching frequency fs, the AC input voltage VAC = 90V-
Fs = 110 KHz to 14 for 140 V fluctuation range
It is shown that it changes in the range of about 0 KHz. This means that the operation of stabilizing the fluctuation of the secondary side DC output voltage EO according to the DC input voltage fluctuation is performed. With respect to the fluctuation of the AC input voltage VAC, control is performed so as to increase the switching frequency as the level of the AC input voltage VAC increases.

As to the period TOFF and the period TON within one switching cycle, the period TOFF is constant with respect to the switching frequency fs, and the period TON decreases in a quadratic curve as the switching frequency fs increases. It is also shown here that the operation is based on the complex control system as the switching frequency control.

The parallel resonance voltage V1 is also assumed to change in accordance with the fluctuation of the commercial AC power supply VAC. As shown in the figure, the parallel resonance voltage V1 changes so that its level increases as the AC input voltage VAC increases. I do.

[0036]

For example, as shown in FIGS. 7 and 8, in a power supply circuit having a configuration for stabilizing a secondary-side DC output voltage by a complex control system, the power supply circuit shown in FIG.
As shown in FIGS. 12A and 12B and FIG. 12, the peak level of the parallel resonance voltage V1 changes according to the load condition and the fluctuation of the AC input voltage VAC. In particular, in a heavy load state close to the maximum load power, for example, the level of the AC input voltage VAC as the 100 V commercial AC power supply AC is 140 V
, The parallel resonance voltage V1 rises to 700 Vp at the maximum as shown in FIG.

For this reason, as for the parallel resonance capacitor Cr and the switching element Q1 to which the parallel resonance voltage V1 is applied, a product with a withstand voltage of 800 V is selected in order to support a commercial AC power supply of 100 V AC. A
In order to support the C200V system, it is necessary to select a 1200V withstand voltage product. As a result, both the parallel resonance capacitor Cr and the switching element Q1 become large and the cost increases.

The characteristics of the switching element are such that the higher the breakdown voltage of the switching element, the lower its characteristics. Therefore, by selecting a switching element Q1 having a high withstand voltage as described above, the power loss due to the switching operation increases and the power conversion efficiency decreases.

[0039]

SUMMARY OF THE INVENTION In view of the above problems, the present invention provides a switching power supply circuit having a configuration as a complex resonance type switching converter to improve the power conversion efficiency and reduce the size and weight. With the goal.

Therefore, according to the present invention, the switching power supply circuit is configured as follows. In other words, a switching unit formed with a main switching element for intermittently outputting the input DC input voltage and a primary parallel resonance circuit that makes the operation of the switching unit a voltage resonance type are formed. A gap is formed so as to obtain a required coupling coefficient that is loosely coupled between the primary side and the secondary side, and the output of the switching means obtained on the primary side winding. And an insulated converter transformer for transmitting to the secondary side. In addition, a secondary parallel resonance circuit formed by connecting a secondary parallel resonance capacitor in parallel to a secondary winding of the insulating converter transformer and a secondary winding of the insulating converter transformer are obtained. DC output voltage generating means configured to obtain a secondary DC output voltage by inputting an alternating voltage and performing a rectifying operation, and a drive winding formed by winding up a primary winding of an insulating converter transformer. A self-excited drive circuit comprising at least a line, and an auxiliary switching element which is switched and driven by the self-excited drive circuit, and a primary parallel circuit generated at both ends of the primary parallel resonant capacitor during a period when the main switching element is turned off. An active clamp means provided to clamp the resonance voltage; and the main clamp means according to the level of the secondary DC output voltage. Switching driving means for variably controlling the switching frequency of the switching element and performing constant voltage control by switching the main switching element so as to vary the on / off period within the switching cycle. Things.

According to the above configuration, a primary side parallel resonance circuit for forming a voltage resonance type converter is provided on the primary side, and a secondary side winding and a secondary side parallel resonance capacitor are provided on the secondary side. The configuration of a so-called composite resonance type switching converter provided with the formed secondary-side parallel resonance circuit is obtained. The primary side is provided with active clamp means for clamping a parallel resonance voltage generated when the main switching element is turned off, so that the level of the parallel resonance voltage is suppressed. The active clamp means is driven by a self-excited oscillation drive circuit having a drive winding formed by winding up a primary winding of an insulating converter transformer. This means that, for example, even with a simple drive circuit configuration, with on / off timing synchronized with the main switching element,
This means that the auxiliary switching element can be driven for switching. Further, the constant voltage control is performed by variably controlling the duty ratio of the ON / OFF period within one switching cycle together with the switching frequency.

[0042]

FIG. 1 shows a configuration of a switching power supply circuit according to a first embodiment of the present invention.
In this figure, the same parts as those in FIGS. 7 and 8 are denoted by the same reference numerals, and the description is omitted. The power supply circuit shown in FIG. 1 also employs a configuration as a complex resonance type switching converter having a voltage resonance type converter on the primary side and a parallel resonance circuit on the secondary side. Insulated converter transformer PI having the structure shown in FIG.
T is provided. The same applies to a power supply circuit according to another embodiment described later.

The overall configuration of the primary side of the power supply circuit shown in FIG.
1 and a voltage resonance type converter which basically performs a switching operation as a single-ended system by a separately excited system is provided. In addition, an active clamp circuit 20 for clamping the parallel resonance voltage V1 obtained at both ends of the parallel resonance capacitor Cr is provided as described later. In this case, both the main switching element Q1 and the auxiliary switching element Q2 have MO
An S-FET is used.

Switching drive unit 10 of the present embodiment
Employs, for example, a configuration similar to that of FIG. That is, the oscillation circuit 11 includes the oscillation circuit 11 and the drive circuit 12. The oscillation circuit 11 changes the oscillation frequency (switching frequency fs) in accordance with the change in the secondary-side DC output voltage EO, and outputs the pulse of the primary-side parallel resonance voltage V1. The main switching element Q1 is driven by varying the duty ratio of the waveform within one switching cycle according to the level and outputting the generated transmission signal to the drive circuit 12. As a result, the switching frequency of the main switching element Q1 changes according to the load fluctuation and the AC input voltage fluctuation,
The on / off period within one switching cycle is changed. In particular, in the present embodiment, as will be described later, the auxiliary switching element Q2 of the active clamp circuit 20 is driven by a self-excited drive circuit having a drive winding Ng wound around the insulating converter transformer PIT. The off period as well as the on period within one switching cycle is variably controlled. That is, in the present embodiment, as a composite control method, constant voltage control is performed by changing three parameters of the switching frequency of the main switching element Q1, the ON period and the OFF period within one switching cycle.

In this case, the active clamp circuit 20
Is the auxiliary switching element Q2 and the clamp capacitor CC
L, formed with a clamp diode DD2. For example, a so-called body diode is selected as the clamp diode DD2. In addition, the auxiliary switching element Q2
Is provided with a drive winding Ng, a capacitor Cg, and resistors R1 and R2.

A clamp diode DD2 is connected in parallel between the drain and source of the auxiliary switching element Q2. Here, the anode of the clamp diode DD2 is connected to the source, and the cathode is connected to the drain. The drain of the auxiliary switching element Q2 is connected to one terminal of the clamp capacitor CCL, and the other terminal is connected to a connection point between the line of the rectified smoothing voltage Ei and the winding start end of the primary winding N1. Connected. Further, the source of the auxiliary switching element Q2 is connected to the winding end of the primary winding N1. That is, as the active clamp circuit 20 of the present embodiment, the auxiliary switching element Q2 // clamp diode DD is used.
A clamp capacitor CCL is connected in series to the two parallel connection circuits. The circuit thus formed is connected in parallel with the primary winding N1 of the insulating converter transformer PIT.

As a drive circuit system for the auxiliary switching element Q2, a series connection circuit of a resistor R1, a capacitor Cg, and a driving winding Ng is connected to the gate of the auxiliary switching element Q2, as shown in the figure. This series connection circuit forms a self-excited oscillation drive circuit for the auxiliary switching element Q2. Here, the drive winding Ng is formed so as to wind up the winding end end side of the primary winding N1 in the insulating converter transformer PIT, and the number of turns in this case is, for example, 1T (turn). As a result, a voltage excited by the alternating voltage obtained in the primary winding N1 is generated in the driving winding Ng. In this case, a voltage having a polarity opposite to that of the primary winding N1 and the driving winding Ng is obtained from the relationship in the winding direction. In practice, if the number of turns of the drive winding Ng is 1T, the operation is guaranteed, but the present invention is not limited to this. The resistor R2 is connected to the primary winding N1 of the insulation converter transformer PIT.
And the connection point of the drive winding Ng.

On the secondary side of the power supply circuit, a secondary parallel resonance circuit including a secondary winding N2 and a secondary parallel resonance capacitor C2 is provided.
A half-wave rectifier circuit composed of one rectifier diode Do and a smoothing capacitor Co connected to the winding start end side of the secondary winding N2 is formed, and the secondary-side DC output voltage Eo is obtained by the half-wave rectifier circuit. It has been like that.

The waveform diagram of FIG. 2 mainly shows the switching operation of the primary side as the operation of the circuit shown in FIG. That is, the operation as a voltage resonance type converter provided with the active clamp circuit 20 is shown. The operation shown in FIG. 2 is obtained when the circuit shown in FIG. 1 has a configuration corresponding to the AC 100 V system, and FIGS. 2A to 2H show an AC input voltage VAC = 100 V, Maximum load power Pomax = 200
The operation of each unit under the condition of W is shown in FIG.
2 (a) to 2 (a) to 2 (a) under the conditions of AC input voltage VAC = 100V and minimum load power Pomin = 20W.
The operation of the same part as (h) is shown.

First, the operation when the maximum load power Pomax = 200 W shown in FIGS. 2A to 2H will be described. In this figure, five operation modes from mode to mode are shown for the operation modes within one switching cycle. It is during the period TON that the main switching element Q1 is controlled to be turned on.
In the ON state, operation as a mode is obtained. In addition,
The auxiliary switching element Q2 is controlled to be in the off state during this period TON.

In the mode (period TON), FIG.
According to the waveform shown in FIG.
A switching output current I1 flows through the drain of the main switching element Q1 via a leakage inductance L1 obtained in the primary winding N1 of the isolated converter transformer PIT.
It flows to. At this time, the switching output current I1 has a waveform that initially reverses from a negative direction to a positive direction, as shown in a period TON of FIG. 2B. Here, during the period in which the switching output current I1 flows in the negative direction, the discharge of the parallel resonance capacitor Cr ends at the end of the immediately preceding period td2, whereby the clamp diode DD1 conducts, and the clamp diode DD1 → the primary winding N1 is connected. When the switching output current IQ1 flows through the power supply mode, a mode in which power is regenerated on the power supply side is set. Then, at the timing when the switching output current I1 (FIG. 2B) reverses from the negative direction to the positive direction, the main switching element Q1 is turned on by ZVS (Zero Volt Switching) and ZCS (Zero Current Switching). .

Then, in the next period td1, the operation is performed as a mode. During this period, the main switching element Q1 is turned off, so that the current flowing through the primary winding N1 flows through the parallel resonance capacitor Cr. At this time, the current Icr flowing through the parallel resonance capacitor Cr has a positive polarity pulse-like waveform as shown in FIG. This is an operation as a partial resonance mode. At this time, since the parallel resonance capacitor Cr is connected in parallel with the main switching element Q1, the main switching element Q1 is turned off by ZVS.

Subsequently, a period in which the auxiliary switching element Q2 is controlled to be in the on state and the main switching element Q1 is controlled to be in the off state is a period, which is shown in FIG. This corresponds to a period TON2 in which the voltage V2 across the auxiliary switching element Q2 is at the 0 level. This period TON2 is an operation period of the active clamp circuit 20. First, the operation as a mode is performed, and then the operation as a mode is performed.

In the operation in the previous mode, the parallel resonant capacitor Cr is charged by the current flowing from the primary winding N1, and as a result, in the mode operation, the voltage level obtained in the primary winding N1 is reduced. At the initial stage (at the start of the period TON2), the potential is equal to or higher than the voltage level across the clamp capacitor CCL1. As a result, when the conduction condition of the clamp diode DD2 connected in parallel to the auxiliary switching element Q2 is satisfied and the conduction is performed, a current flows through a path from the clamp diode DD2 to the clamp capacitor CCL1, and the clamp current ICL is After the start of the period TON2 in FIG. 2 (e), a saw-tooth waveform approaching the 0 level as time elapses from the negative direction is obtained. Here, for example, if the capacitance of the clamp capacitor CCL1 is selected to be 50 times or more the capacitance of the parallel resonance capacitor Cr, depending on the operation in this mode,
Most of the current flows as the clamp current ICL to the clamp capacitor CCL1, and hardly flows to the parallel resonance capacitor Cr. As a result, during the period TON2, the slope of the parallel resonance voltage V1 applied to the main switching element Q1 is made gentle, and as a result, as shown in FIG. The corners will widen. That is, a clamping operation for the parallel resonance voltage V1 is obtained. On the other hand, the circuit according to the prior art (the circuits in FIGS. 7 and 8)
Is a pulse waveform having a level of 550 Vp.

Then, when the above mode is completed in the period TON2, the operation subsequently proceeds to the mode. At the start of this mode, the clamp current ICL shown in FIG. 2D is switched from a negative direction to a positive direction. At this timing, the auxiliary switching element Q
2 is turned on by ZVS and ZCS at the timing when the clamp current ICL reverses from the negative direction to the positive direction. In the state where the auxiliary switching element Q2 is turned on in this manner, the resonance action of the primary side parallel resonance circuit obtained at this time causes the primary winding N1 → the drain → source of the auxiliary switching element Q2 via the clamp capacitor CCL. 2D, a waveform increasing in the positive direction is obtained as shown in FIG.

Here, although not shown, the voltage applied to the gate of the auxiliary switching element Q2 is
g, which is a pulse voltage having a rectangular waveform. The gate inflow current Ig flowing to the gate of the auxiliary switching element Q2 is determined by the capacitor Cg and the resistance R
2 forms a differentiated waveform as shown in FIG. 2D and flows immediately after the end of the period td1 and in the period td2. Period t
The period d1 and the period td2 are threshold periods during which both the main switching element Q1 and the auxiliary switching element Q2 are turned off. The threshold period is maintained by the flow of the gate inflow current Ig.

The operation in the above-described mode is as follows.
The operation is terminated at the timing when the voltage V2 across the auxiliary switching element Q2, which has been set to the 0 level in the FF, starts rising, and then the operation shifts to the mode operation in the period td2. In mode,
An operation is obtained in which the parallel resonance capacitor Cr supplies a discharge current to the primary winding N1. That is, a partial resonance operation is obtained. At this time, the parallel resonance voltage V1 applied to the main switching element Q1 has a large slope due to the small capacitance of the parallel resonance capacitor Cr as described above, and as shown in FIG. hand,
It falls so as to rapidly descend to the 0 level. Then, the auxiliary switching element Q2 starts turning off at the timing when the mode ends and the mode is started. At this time, the parallel resonance voltage V1 falls with a certain slope as described above. Then, a turn-off operation by ZVS is performed. Further, the voltage generated when the auxiliary switching element Q2 is turned off is prevented from rising sharply by discharging the parallel resonance capacitor Cr as described above. This operation is performed, for example, as shown as the voltage V2 across the auxiliary switching element Q2 in FIG.
During the period td2 (in the mode), 0 with a certain inclination
It is shown as a waveform that transitions from level to peak level. The voltage V2 across the auxiliary switching element Q2 has, for example, 300 Vp in a period TOFF2 in which the auxiliary switching element Q2 is turned off, and has a zero level during a period td1 (in a mode) which is a start period of the period TOFF2. , And the waveform changes from the 0 level to 300 Vp in the period td2 (in the mode), which is the end period, as described above. And after that,
The operations of modes 1 to 4 are repeated every switching cycle.

FIG. 2 (g) shows the operation on the secondary side.
Shows a secondary alternating voltage V3 obtained at both ends of the secondary winding N2 (secondary parallel resonance capacitor C2), and FIG. 2 (h) shows a secondary rectified current Io flowing through the secondary rectifier diode DO.
Is shown. The secondary side alternating voltage V3 is clamped at the level of the secondary side DC voltage EO1 during the period DON during which the secondary side rectifier diode DO1 conducts and turns on, and peaks in the direction of negative polarity during the period DOFF during the off period. Thus, a waveform on a sine wave is obtained. The secondary side rectified current Io is at the 0 level during the period DOFF, and flows according to the waveform shown in the period DON.

The operation waveforms of the respective parts shown in FIGS. 2A to 2H are, for example, minimum load power Pomin = 20 W
Under the condition that the load power is reduced to, they change as shown in FIGS.

Here, as can be seen by comparing the primary side parallel resonance voltage V1 of FIG. 2A and FIG.
The waveform shown in (i) is the main switching element Q
The period TON during which 1 is turned on is significantly shorter, so that the switching frequency is higher than at the time of the maximum load power shown in FIG. In practice,
A slight change is also obtained in the period TOFF during which the main switching element Q1 is turned on. This is because the main switching element Q1 on the primary side is controlled so that the switching frequency becomes higher as the load power becomes smaller and the secondary side output voltage EO1 rises. This shows that the ON / OFF period is variably controlled. In other words, it indicates that the constant voltage operation by the complex control method that variably controls the three parameters (the switching frequency fs and the periods TON and TOFF) is obtained.

On the other hand, the auxiliary switching element Q2 is driven at a timing according to the voltage waveform obtained on the drive winding Ng. The voltage obtained on the drive winding Ng is the alternating voltage generated on the primary winding N1. Is excited by Accordingly, the auxiliary switching element Q2 has the on-time TON2 and the off-time TOFF2 variable so that the switching frequency is also changed in synchronization with the control of the switching operation of the main switching element Q1 as described above. Variable control. That is, in the present embodiment, regardless of whether the auxiliary switching element Q2 side is driven in a self-excited manner, the auxiliary switching element Q2 responds to a change in the ON / OFF period of the main switching element Q1.
The on / off period of the auxiliary switching element Q2 is simultaneously controlled. This is because the drive voltage level of the auxiliary switching element Q2 changes because the induced voltage of the drive winding Ng changes according to the load change and the change of the rectified smoothed voltage Ei and the like.

Even under such a light load condition, the operations in the modes 1 to 3 are performed at the timings shown in FIGS. 2 (i) to 2 (p), so that the primary side parallel resonance voltage V1
And the peak level of the voltage V2 across the auxiliary switching element Q2 is, for example, 240 Vp.
It is suppressed to the extent. In particular, the primary side parallel resonance voltage V1
Is suppressed to 150 Vp at the time of the minimum load power.

For reference, the values selected for the main elements in the power supply circuit shown in FIG. 1 when the experimental results shown in FIG. 2 are obtained are shown. First, a 400V / 10A low on-resistance product is selected for the main switching element Q1, and 400V / 10A for the auxiliary switching element Q2.
/ 3A low on-resistance product was selected. The following values were selected for the remaining elements. Parallel resonance capacitor Cr = 3300 pF Clamp capacitor CCL = 0.1 μF Secondary parallel resonance capacitor C2 = 0.01 μF Primary winding N1 = Secondary winding N2 = 43T Capacitor Cg = 0.33 μF Resistance R1 = 22Ω Resistance R2 = 100Ω The variable range of the switching frequency fs to be set when these elements are selected is, for example, 100
KHz to 150 KHz.

FIG. 3 shows the characteristics of the power supply circuit shown in FIG. 1 as the power conversion efficiency ηDC →
The relationship between DC, the switching frequency fs, and the periods TON and TOFF of the main switching element Q1 is shown. However, the characteristic shown in this figure is a characteristic corresponding to the condition of the AC 200 V system, and is measured under the condition that the rectified and smoothed voltage Ei is constant at 250 V. As shown in this figure, as the load power increases, the PWM control (conduction angle control) is performed so that the period TON during which the main switching element Q1 is turned on increases. It can be seen that the PWM control is performed so that the OFF period TOFF is also longer, though more gently. At the same time, the switching frequency fs is controlled so as to decrease. It can be seen that the primary-side parallel resonance voltage V1 slightly increases as the load increases, but is suppressed in the range of about 600V. The power conversion efficiency is expressed as load power Po = 5.
When the system load is about 0W, it is about 92%,
A good result that 94% or more is maintained when the load power Po is about 100 W or more is obtained.

FIG. 4 shows, as characteristics of the power supply circuit shown in FIG. 1, power conversion efficiency ηDC → DC, switching frequency fs, and rectified smoothed voltage Ei (DC input voltage).
The relationship between the periods TON and TOFF of the main switching element Q1 is shown. This characteristic is assumed to be obtained under the condition of load power Po = 200 W.

Also in this case, the main switching element Q
The period TON during which 1 is on is controlled so as to be shorter in accordance with the rise of the rectified smoothing voltage Ei, and at the same time, the period TOFF during which it is off becomes longer with a gentle slope. In general, the switching frequency fs is controlled so as to increase as the rectified smoothed voltage Ei increases. Further, although the primary side parallel resonance voltage V1 increases as the rectified smoothed voltage Ei increases,
For example, even when Ei = 400V, V1 is suppressed to less than about 800V. Regarding the power conversion efficiency, 94% or more is maintained regardless of the change in the rectified smoothed voltage Ei.

As can be understood from the above description, in the circuit shown in FIG. 1, the parallel resonance voltage V1 which is generated when the main switching element Q1 is turned off is clamped, and the level thereof is suppressed. . Then, for example, even when the AC voltage is increased to about VAC = 144 V in the AC 100 V system under the maximum load condition, the parallel resonance voltage V1 can be suppressed to about 300 Vp. AC2
Even in the case of the 00V system, the maximum value of the peak level of the parallel resonance voltage V1 can be suppressed to about 600 V during normal operation. Therefore, the circuit shown in FIG.
In the case of supporting a V system, a withstand voltage product of 400 V should be selected, and in the case of supporting an AC 200 V system, a withstand voltage product of 800 V should be selected. That is, for example, FIG.
In addition, a constant voltage product can be selected as compared with the circuit shown in FIG. A similar withstand voltage product may be selected for the auxiliary switching element Q2.

As a result, the characteristics of the switching element of the circuit shown in FIG. 1 are more improved than those of the circuits shown in FIGS. For example, if the switching element is MOS-F
In the case of ET, the on-resistance decreases. Thereby, the power conversion efficiency is improved.
For example, as an experimental result, the circuit shown in FIGS. 7 and 8 has a power conversion efficiency of 92%, while the circuit shown in FIG. 1 has a result of 93% (when the AC input voltage VAC = 100 V). Have been. Further, in the present embodiment, as described above, an operation as a complex control method in which the switching frequency fs and the three elements of the ON period and the OFF period of the switching element are variably controlled can be obtained. The control range for the constant voltage control is also expanded.

Also, by selecting a low withstand voltage product for the switching element, the size of the switching element can be reduced. For example, the switching element used in the circuits shown in FIGS. 7 and 8 requires a withstand voltage product of 1000 V or more, so that the size of the package becomes relatively large. As each of the switching elements Q1 and Q2, a smaller packaged product can be used. Further, since the level of the parallel resonance voltage V1 is suppressed, in the circuit shown in FIG. 1, it is possible to adopt a low withstand voltage product for the parallel resonance capacitor Cr as compared with the circuits shown in FIGS. Therefore, the size of the parallel resonance capacitor Cr can be reduced. Further, in the power supply circuit shown in this figure, since the switching frequency of the primary-side switching converter is variably controlled according to the fluctuation of the load power (secondary output voltage), the switching frequency is reduced when the load is short-circuited. Works like that. For example, in the circuits illustrated in FIGS. 7 and 8, the voltage applied to the switching element and the parallel resonance capacitor increases because the ON period becomes longer due to a decrease in the switching frequency when the load is short-circuited. For this reason, there has been a need for a protection circuit for protecting a switching element by limiting a high-level voltage and current generated when a load is short-circuited. On the other hand, in the power supply circuit according to the present embodiment, since the change in the parallel resonance voltage V1 with respect to the load change is small, even if the switching frequency decreases when the load is short-circuited, the increase in the parallel resonance voltage V1 is suppressed. In addition, it is possible to omit a protection circuit corresponding to a load short circuit.

In the present embodiment, in particular, in order to switch the auxiliary switching element Q2 in the active clamp circuit 20, the driving winding Ng,
One feature is that a self-excited drive circuit system including a capacitor Cg and resistors R1 and R2 is provided. For example, as another configuration for switchingly driving the auxiliary switching element Q2, the switching drive unit 10 may additionally include a separately-excited drive circuit system for driving the auxiliary switching element Q2 by combined control. . That is, the main switching element Q1
At the same time, the auxiliary switching element Q2 is driven by a circuit such as a separately excited IC. However, in this configuration,
It is necessary to provide a separately-excited circuit system for simultaneously performing the switching frequency control and the PWM control for the main switching element Q1, and a separately-excited circuit system for simultaneously performing the switching frequency control and the PWM control for the auxiliary switching element Q2. There is. Accordingly, the circuit configuration becomes more complicated and the number of parts increases, which hinders the reduction in size and weight of the circuit. On the other hand, if the above-described configuration is adopted in the present embodiment, the driving circuit system for the auxiliary switching element Q2 includes only a 1T winding wound around the insulating converter transformer PIT and two resistances. , And one capacitor, which has a very simple circuit configuration, and can realize the same operation as that of the separately excited type.

FIG. 5 is a circuit diagram showing a configuration of a power supply circuit according to a second embodiment of the present invention. This power supply circuit is a composite resonance type switching converter including a voltage resonance type converter on the primary side and a parallel resonance circuit on the secondary side. The primary-side voltage resonance converter employs a single-ended configuration including one switching element. In the power supply circuit shown in FIG.
7 and 8 are denoted by the same reference numerals, and description thereof is omitted.

The power supply circuit shown in FIG. 5 has the self-excited single-ended voltage resonance type converter shown in FIG. 8 as the primary side configuration. Further, in order to perform constant voltage control by the complex control method, a configuration including an orthogonal control transformer PRT is employed. And for this configuration,
An active clamp circuit 20A that operates in a self-excited manner is provided.

The active clamp circuit 20A includes an auxiliary switching element Q2 which is a BJT in this case. The collector of the auxiliary switching element Q2 is connected to the winding start end of the primary winding N1 via a clamp capacitor CCL. In this case, the winding start end of the primary winding N1 is connected to the positive terminal of the smoothing capacitor Ci via the current detection winding ND. The emitter of auxiliary switching element Q2 is connected to the collector of switching element Q1. Further, a drive circuit for self-excited type comprising a series connection circuit of a resistor R1-capacitor Cg-drive winding Ng is connected to the base of the auxiliary switching element Q2.
Also in this case, the drive winding Ng is formed so as to wind up the winding end end of the primary winding N1 of the insulating converter transformer PIT by 1T. Further, a clamp diode forming a path of a clamp current at the time of turn-on is connected in parallel between the base and the emitter of the auxiliary switching element Q2. In this case, the resistor R2 provided in the circuit shown in FIG. 1 is omitted.

The secondary side rectifier circuit of the power supply circuit has the same configuration as that shown in FIG. That is, the bridge rectifier circuit DBR is connected to the secondary parallel resonance circuit (secondary winding N2 // secondary parallel resonance capacitor C2). In this case, the secondary side DC output voltage E0 is branched and input as a detection voltage of the control circuit 1. The constant voltage control operation as a composite control method using the control circuit 1 and the orthogonal control transformer PRT is as described with reference to FIG. However, also in this case, as the complex control method, the control range is expanded by operating to variably control the switching frequency fs and the three elements of the ON period and the OFF period of the switching element.

In such a configuration, for example, the same operation as that described with reference to the waveform diagram of FIG. 2 can be obtained, and the suppression of the primary side parallel resonance voltage V1, the switching element and various capacitors And cost reduction.

FIG. 6 shows a configuration example of a power supply circuit according to the third embodiment. In this figure, FIG.
5, 7, and 8 are denoted by the same reference numerals, and description thereof is omitted. On the primary side of the power supply circuit shown in FIG.
Main switching element Q1 and auxiliary switching element Q2
Regarding the above, IGBT (insulated gate bipolar transistor) is selected. The configuration on the primary side other than this is the same as the circuit shown in FIG. Even with such a configuration, the same operation and effect as those of the above embodiments can be obtained, and, for example, higher power conversion efficiency can be obtained by selecting the IGBT.

On the secondary side of the power supply circuit shown in this figure, two rectifying diodes DO1 are connected to the secondary side parallel resonance circuit (secondary winding N2 // secondary side parallel resonance capacitor C2). , DO2 and two smoothing capacitors CO1, O2
Are connected by the connection form shown to form a voltage doubler rectifier circuit. As a result, as a voltage obtained between the positive terminal of the smoothing capacitor CO1 and the secondary-side ground, that is, a secondary DC voltage Eo, a level corresponding to twice the alternating voltage level obtained in the secondary winding N2 is obtained. Will be done. Accordingly, for example, when it is sufficient that the level of the secondary side DC voltage Eo is equivalent to that obtained when a normal equal-voltage rectifier circuit is connected, the secondary winding N
It is possible to reduce the number of turns of 2 to 、, and for example, it is possible to reduce the size and weight of the insulated converter transformer PIT.

The embodiments of the present invention are not limited to the configurations shown in the drawings. For example, in the above embodiment, the main switching element and the auxiliary switching element are MOS-FET, BJT,
Although an IGBT or the like is employed, other elements such as an SIT (static induction thyristor) may be employed. In addition, the configuration of the switching drive unit for driving the main switching element Q1 by separately-excited driving need not be limited to the one shown in each drawing, and may be changed to an appropriate circuit configuration as appropriate. Also, the secondary-side rectifier circuit formed including the secondary-side resonance circuit is not limited to the configurations shown in the drawings as the embodiments, and other circuit configurations may be adopted. Not something.

[0079]

As described above, in the switching power supply circuit according to the present invention, a voltage-resonant converter is provided on the primary side and a parallel-resonant circuit is provided on the secondary side. By providing the active clamp circuit on the primary side, a parallel resonance voltage pulse generated at both ends of the primary side parallel resonance capacitor is clamped and its level is suppressed. As a result, the withstand voltage of each element such as the switching element and the primary-side parallel resonance capacitor provided in the power supply circuit can be selected to be lower withstand voltage than before.

The active clamp circuit of the present invention includes a self-excited drive circuit having a drive winding formed by winding up a primary winding of an insulating converter transformer PIT, thereby simplifying a drive circuit system. Therefore, the number of components can be reduced, which can greatly contribute to the reduction in size and weight of the circuit. Further, in such a configuration of the drive circuit system, since the ON period and the OFF period of the switching elements (main and auxiliary) are simultaneously controlled as a variable operation within one switching period accompanying the switching frequency control, This also has the effect of expanding the constant voltage control range.

[Brief description of the drawings]

FIG. 1 is a circuit diagram illustrating a configuration example of a switching power supply circuit according to a first embodiment of the present invention.

FIG. 2 is a waveform chart showing an operation of a main part in the switching power supply circuit of the present embodiment.

FIG. 3 is a characteristic diagram showing, as characteristics of the switching power supply circuit of the present embodiment, power conversion efficiency, switching frequency, and period TON / TOFF with respect to load power.

FIG. 4 is a characteristic diagram showing, as characteristics of the switching power supply circuit of the present embodiment, a power conversion efficiency, a switching frequency, and a period TON / TOFF with respect to a DC input voltage.

FIG. 5 is a circuit diagram illustrating a configuration example of a switching power supply circuit according to a second embodiment of the present invention;

FIG. 6 is a circuit diagram illustrating a configuration example of a switching power supply circuit according to a third embodiment of the present invention;

FIG. 7 is a circuit diagram showing a configuration example of a switching power supply circuit as a prior art.

FIG. 8 is a circuit diagram showing another configuration example of a switching power supply circuit as a prior art.

FIG. 9 is a cross-sectional view illustrating a configuration of an insulating converter transformer.

FIG. 10 is an equivalent circuit diagram showing each operation when the mutual inductance is + M / −M.

FIG. 11 is a waveform chart showing an operation of the switching power supply circuit shown in FIGS. 7 and 8;

FIG. 12 is an explanatory diagram illustrating characteristics of the switching power supply circuit illustrated in FIGS. 7 and 8 with respect to an AC input voltage.

[Explanation of symbols]

10 switching drive unit, 11 oscillation circuit, 12 drive circuit, 20, 20A active clamp circuit,
30 Photocoupler, Q1 main switching element,
Q2 Auxiliary switching element, PIT isolation converter transformer, Cr primary side parallel resonance capacitor, C2 secondary side parallel resonance capacitor, CCL clamp capacitor, drive winding Ng, capacitor Cg, R1, R2 resistance

Claims (1)

[Claims] 1. A switching means comprising a main switching element for intermittently outputting an input DC input voltage, and a primary-side parallel resonance circuit which makes the operation of the switching means a voltage resonance type. And a primary side parallel resonant capacitor provided in such a manner that a gap is formed so as to obtain a required coupling coefficient that is loosely coupled between the primary side and the secondary side, and the switching obtained in the primary side winding is formed. An insulated converter transformer for transmitting the output of the means to the secondary side, and a secondary side parallel resonance circuit formed by connecting a secondary side parallel resonance capacitor in parallel to the secondary side winding of the insulation converter transformer And a rectifying operation by inputting an alternating voltage obtained in a secondary winding of the insulating converter transformer to obtain a secondary DC output voltage. A self-excited drive circuit comprising at least a DC output voltage generating means, a drive winding formed by winding up a primary winding of the insulating converter transformer, and an auxiliary switched by the self-excited drive circuit. An active clamp means provided with a switching element, and provided to clamp a primary-side parallel resonance voltage generated at both ends of the primary-side parallel resonance capacitor during a period in which the main switching element is turned off, and the secondary-side DC output voltage The switching frequency of the main switching element is variably controlled in accordance with the level of the main switching element, and the constant voltage control is performed by switching the main switching element so as to vary the ON / OFF period within the switching cycle. Switching driving means, and A switching power supply circuit comprising: