KR100609192B1 - Switching mode power supply implemented by quasi-resonance - Google Patents

Abstract

The present invention relates to a pseudo-resonant power supply device, and aims to reduce the size and weight of a transformer and a capacitor by increasing the switching frequency in a pseudo-resonant switching mode power supply device.

To this end, the quasi-resonant feeder includes a first switching device connected to a transformer to perform a switching function, a second switching device connected to the transformer to perform a switching function and an active clamp function, and a second switching device. A clamping capacitor configured to be connected to an active clamp function, a voltage detector connected to the input side winding of the transformer to measure a voltage for turning on a first switching element, and a pseudo resonance control circuit configured to form a first switching element. And a turn-on / off-control circuit unit for turning on or off the first switching element by controlling the drive unit by configuring a driving unit for driving the control unit and a pseudo resonance control circuit.

Therefore, according to the present invention, since the switching loss is reduced, the transformer and the capacitor can be reduced in size and weight by increasing the switching frequency, and the loss can be reduced, thereby minimizing the parallel panel, thereby reducing the volume of the power supply device.

Pseudo resonance, Partial resonance, SMPS, Power supply, Switching mode

Description

Switching mode power supply implemented by quasi-resonance}

1 is a circuit diagram showing a conventional pseudo resonance power supply.

Figure 2 is a circuit diagram showing a pseudo resonance power supply according to the present invention.

Figure 3a is a voltage input waveform diagram according to a conventional pseudo resonance power supply, Figure 3b is a current input waveform diagram.

Figure 4a is a voltage input waveform diagram according to the pseudo resonance power supply device according to the present invention, Figure 4b is a current input waveform diagram.

5 is a waveform diagram showing an output current waveform according to the present invention.

******** Description of the symbols for the main parts of the drawings **********

220: transformer, 221: reference point

222: first switching element, 223: ground

224: second switching element, 225: clamp capacitor

226: diode, 227: voltage smoothing capacitor

232: middle tab

The present invention relates to a pseudo-resonant power supply device, and more particularly, to increase the switching frequency in a switching mode power supply (SMPS) using a pseudo-resonant type to reduce the size and weight of a transformer and a capacitor. It is about a power supply that maximizes efficiency.

Generally, the power supply has a switching mode and a linear mode power supply. However, the switching mode power supply is a method of providing power by repeatedly performing on / off. A diagram showing a circuit example of such a switched mode power supply device is shown in FIG. To explain this, the rectifying smoothing circuit 11 inputs an alternating current (AC) supply voltage, full-wave rectifies it through a diode bridge, smoothes it through a capacitor, and finally, converts the obtained direct current (DC) voltage into a transformer (13). Output to one end of the primary winding (L1).

The other end of the primary winding L1 of the transformer 13 is connected to the drain of the switching element Q1. The source of the switching element Q1 is connected to the ground which is also connected to the ground portion of the rectification smoothing circuit 11. The capacitor C1 is connected in parallel between the source and the drain of the switching element Q1.

The magnetic energy accumulated in the primary winding L1 of the transformer 13 is sequentially induced in the secondary winding by the switching operation of the switching element Q1 controlled by the ON-OFF operation of the controller 25 described later. do. Half-wave rectification is then performed on the magnetic energy induced in the secondary winding by a diode D1 connected to one end of the secondary winding L2, smoothed by the capacitor C3, and then smoothed DC voltage. This is output to the load 17 and to the output DC voltage detection circuit 19.

The output DC voltage detection circuit 19 converts the output DC voltage supplied to the load 17 into a feedback signal and outputs it to the ON period control circuit 29 mounted in the controller 25.

The diode D3 of the output smoothing circuit 21 half-wave rectifies with respect to the flyback voltage generated in the auxiliary winding L3 of the transformer 13 and the capacitor C5 smoothes the voltage obtained from the diode D3. Let's do it. The control unit 25 inputs the smoothed voltage Vcc from the capacitor C5.

When the starting voltage of the predetermined voltage or more is supplied to the starting resistor R1, the control unit 25 starts oscillation.

When the switching element Q1 is OFF, ringing occurs in the primary winding L1 of the transformer 13. The resonance frequency f is as follows.

f = 1 / (2∏√L1xC1)

At the same timing, ringing also occurs in the auxiliary winding L3. In the ring generating circuit 23, ringing is divided by the resistors R3 and R5 after detection through the diode D5. The ringing signal from which the high frequency component is removed through the resistor R3 and the capacitor C7 is output to the comparator circuit 27 mounted on the controller 25. The comparator circuit 27 compares the ringing signal input to the comparator COMP1 with the reference voltage Vref1, and outputs a high level signal when the ringing signal is greater than the reference voltage Vref1.

The ON period control circuit 29 generates an ON period control signal for stabilizing the output DC voltage supplied to the load 17 by adjusting the ON period by the feedback signal from the output DC voltage detection circuit 19 to generate a frequency control circuit. Output what was generated by (31).

The frequency control circuit 31 controls the ON period by the ON period control signal from the ON period control circuit 29, for example, oscillates a fixed frequency defined by the time constants of the capacitor and the resistor, thereby driving the drive circuit ( 33) to output the control signal.

In the inverter INV1 of the driving circuit 33, both the control signal V3 from the comparator circuit 27 and the control signal V18 from the frequency control signal 31 for one end of the OR gate OR1 are simultaneously low. When the level is reached, the high level drive signal V119 is output to the switching element Q1.

In the conventional partial resonant power supply, the energy is accumulated in the transformer during the turn-on period of the switching element, and then the energy is released during the turn-off period.

The problem at this time is that the high resonant frequency at turn-off causes high voltage to be applied to L1, which requires high breakdown voltage of the switching element. Also, the duty (duty, ratio of period and turn-on period) is determined by the input / output voltage and the primary and secondary winding ratios of the transformer, and the voltage when it is turned on soon. At this time, the turn-on voltage becomes turn-on voltage = input voltage-output voltage x primary and secondary winding ratios (n1 / n2).

In addition, the importance of turn-on voltage is that the energy charged in the C1 capacitor is turned on by the switching element, which causes loss and radiation noise (noise standard: EMI).

In addition, in the conventional partially resonant power supply, a condenser and a transformer are enlarged to store energy during the turn-on of the switching device. In addition, since the switching element resonates in combination with the leakage inductance of the main transformer and the parasitic capacitors in each region during the turn-off period, a sizzling waveform is formed at a high voltage. Therefore, there is a problem that the power generation is not efficient.

The present invention has been proposed to solve the problems posed by the prior art, and aims to reduce the size and weight of the transformer and capacitor by increasing the switching frequency in the pseudo-resonant switching mode power supply.

In addition, the present invention aims to make efficient use of stored energy by selectively adding or subtracting back EMF generated in a transformer during a turn-off period of a switching device.

In addition, it is an object of the present invention to reduce loss of heat, reduce radiated noise, and reduce the breakdown voltage of the switching element Q1 221 and the rectifying element D 226 by performing a loss of reliable zero voltage switching.

In order to achieve the above object of the present invention, a capacitor and a switching element for an active clamp are connected in parallel to the input side of a transformer so as to be generated by coupling resonance with the leakage inductance of the main transformer and parasitic capacitors at each turn-off of the first switching element. The present invention provides a switching power supply for implementing a pseudo resonance type for converting a distortion voltage into a smooth voltage.

To this end, the quasi-resonant feeder includes a transformer consisting of an input side winding in which magnetic energy is accumulated before transformation and a secondary winding for output after transformation by receiving energy, and a first connected to the transformer to perform a switching function. A switching device, a second switching device connected to the transformer to perform a switching function and an active clamp function, a clamp capacitor connected to the second switching device to perform an active clamp function, and connected to the input side winding of the transformer And a voltage detector for measuring a voltage for turning on the first switching element, a driver configured to drive the first switching element by configuring a pseudo resonance control circuit, and a driver configured by controlling a drive by configuring a pseudo resonance control circuit. The turn-on / off-control circuit section for turning the element on or off is constituted.

Therefore, by reducing the switching losses, it is possible to increase the switching frequency to reduce the size and weight of the transformer and condenser, and to reduce the loss, thereby minimizing the parallel panel and reducing the volume of the power supply.

Hereinafter, exemplary embodiments of the present invention will be described in detail with reference to the accompanying drawings.

2 is a circuit diagram showing a pseudo resonance power supply device according to the present invention. In the block diagram, a driving unit 230 and a turn-on / off control circuit unit 231 constituting a pseudo resonance control circuit are used to transform a voltage. A transformer 220 which is a transformer, a first switching device 222 and a second switching device 224 connected to the transformer to perform a switching function, and a voltage detector 229 connected to a primary side winding of the transformer to measure a voltage. ) And the like.

In describing the functions and roles performed by these components, the transformer 220 has two inductances wound at n: 1. That is, it is composed of L1 210 which is a winding for the transformer before input and L2 211 which is a winding for the output after transformation. Of course, in the present invention, a single stage output is configured for understanding, but multi-output is also applicable. That is, for example, the second and third windings may be configured on the opposite side of the L1 210 in addition to the L2 211. In addition, the primary side winding L1 210 includes an intermediate tab 232 connected to the first switching element 222. The intermediate tab 232 is formed at an appropriate point of the primary side winding L1 210.

 In order to supply power to the input winding L1 210 of the transformer 220, the rectification smoothing circuit 11 shown in FIG. In other words, the power is supplied to transform the voltage. Usually, the rectification smoothing circuit receives an alternating current (AC) voltage, and consists of a diode bridge and a capacitor for smoothing it. Since the rectification smoothing circuit is already well known, detailed description thereof will be omitted. Therefore, the present invention describes that the rectified voltage or the rectified current is connected to the input winding L1 210 of the transformer 220 using the rectified voltage. That is, the input power supply voltage Vcc is connected.

As described above, the winding L1 210 is paired with the winding L2 211 to form a transformer 220. Therefore, the magnetic energy stored in the winding L1 210 is transferred to the L2 211 through resonance to generate a secondary rectified power source.

To this end, a first switching element 222 which is capable of executing turn on and turn off is connected in series to the winding L1 210. As the first switching element 222, it is preferable that an n-channel MOSFET is usually used. As is well known, the MOSFET is composed of a drain, a source, and a gate, and is widely used in a switching circuit because the drain and the source are connected by the applied voltage of the gate. Therefore, details thereof will be omitted.

When the MOSFET is applied, the drain is connected to the middle tap 232 of the primary side winding L1 210, and the source is grounded to the ground 223. In addition, the gate is connected to the driver 230, and the turn-on and turn-off are repeated by the driver 230. That is, the current flows from the drain to the source during the turn-on period of the first switching element 222. In other words, the current I1 due to the magnetic energy accumulated in the winding L1 210 flows through the drain to the ground 223 connected to the source. In contrast, the current I1 is turned off between the drain and the source of the first switching element 222 during the turn-off period, so that the current I1 does not flow.

Of course, in order for the first switching device 222 to perform such turn-on and turn-off, the driving unit 230 is configured, and the turn-on / off control circuit unit 231 for controlling this is provided. In addition, the C1 233 capacitor is configured in parallel between the source and the drain of the first switching element 222.

The turn on / off control circuit part is connected to a voltage sensing unit 229 for sensing a voltage at any point.

Before describing the turn-on / off control circuit unit 231, the voltage sensing unit 229 is described. The voltage sensing unit 229 has a drain side and a winding of the first switching element 222 as shown in FIG. 2. Using the midpoint of the L1 (210) as the reference point 221 to detect whether the voltage (V1) falls below a certain voltage. The sensed voltage V1 is transmitted to the turn on / off control circuit 231 to determine whether to turn on or turn off the first switching element 222.

That is, when the voltage falls below a predetermined voltage, the turn on / off control circuit 231 controls the driver 230 to turn on the first switching element 222 for a predetermined time. Of course, the turn on / off control circuit 231 performs a function of turning on and off the first switching element 222 for a predetermined time. Therefore, the voltage V1 sensed by the voltage sensing unit 229 which turns on the first switching element 222 for a predetermined time and then turns off to sense the voltage at the reference point 221 is almost close to the “0” voltage. When turned off, the first switching device 222 is turned on again. Of course, the turn on and turn off time is set to proceed in a certain cycle.

The driving unit 230 and the turn on / off control circuit unit 231 are configured to form the resonance control circuit unit 228. In addition, the winding L1 210 is connected to the second switching element 224 and the clamp capacitor 225 for performing the active clamp function.

The active clamp performs an active filter function such that the first switching device 222 performs resonance with the winding L1 210 during the off period. Of course, for this purpose, the second switching element 224 is configured between the clamp capacitor 225 and the winding L1 210. Similar to the first switching element 222, a MOSFET circuit element may be used for the second switching element 224. In particular, it is preferable to use a power MOSFET (also known as a power MOSFET) for implementing the present invention, but is not limited thereto.

As is well known, the MOSFET is composed of a drain, a source, and a gate, and a diode D2 234 is configured in a reverse direction between the drain and the source, and the voltage between the gate and the source has a negative voltage. Therefore, when a voltage greater than the set voltage is applied between the gate and the source, the drain and the source become conductive. Of course, the diode D1 234 is also applicable to using a built-in diode. Since the contents thereof are well known techniques, detailed descriptions of the operations will be omitted.

Of course, the second switching device 224 also has a drain side connected to the clamp capacitor 225 and a source side connected to the drain side of the first switching element 222 similar to the first switching element 222. It is connected to the primary winding L1 (210). Therefore, the first switching element 224 and the on-off is performed to be offset. That is, the second switching device 224 maintains the off state while the first switching element 222 is on, and the first switching element 222 maintains the turn on state during the off period.

That is, as described above, since the diode is built in the reverse direction between the drain and the source in the second switching device 224, the OFF state is maintained when the first switching device 222 is turned on. This is because the switching element is damaged when both the first switching element 222 and the second switching element 224 are turned on.

Due to this switching configuration, when the first switching device 222 is turned off, the first resonant capacitor 233 is gradually charged with voltage, thereby performing zero voltage switching. When the counter electromotive force is generated in the winding L1 210, the voltage between the source and the drain of the second switching element 224 is reversed, and a forward current flows through the internal diode, thereby resonating with the leakage inductance of the clamp capacitor 225 and L1 to lower the voltage. The resonance voltage becomes very small because it is resonated at a frequency. Therefore, it is possible to use the breakdown voltage of the first switching element 222 as a small one.

At this time, the source and the drain are turned on while the turn-on voltage is applied between the source and the gate of the first switching device 222 from the middle tap 234 of the L1 210, and the first switching device 222 is turned off. Only when counter electromotive force occurs in the transformer 220, the second switching element 224 is turned on so that current flows from the source side to the drain side. Accordingly, the winding L1 210 and the clamp capacitor 225 described above are to perform resonance. This resonance causes the leakage inductance of the primary winding L1 210 of the transformer 220 and the clamp capacitor 225 to resonate.

As the first switching element 222 is turned on, current flows in the primary winding L1 of the transformer 220 so that the magnetic energy accumulated in the transformer 220 is induced in the secondary winding L2 211 to rectify the diode D 226. Rectified and output by the output capacitor C (227) as a DC voltage. That is, when L1 * Cr = L2 * C (227) and a vibration current flows in an input circuit, the same vibration current will generate | occur | produce in the thrust side secondary winding L2211. Therefore, a circuit element for rectifying the induced magnetic energy is formed on the secondary winding L2 211 side for such electric resonance. That is, the forward diode 226 is configured at one end of the secondary winding L2 211, and half-wave rectification is performed by this diode D1.

The condenser 227 is configured to smooth this half-wave rectification. The smoothed voltage is output from the load 240 to generate a desired output voltage (Vout).

The configuration of the circuit diagram shown in FIG. 2 has been described. This will be described the implementation process of the present invention through this.

The pseudo resonance control unit 228 turns off the first switching element 222 after turning on the first switching element 222 for a predetermined period of time (this is called a turn-on period, which is the first time), and the voltage of the reference point 221 is changed by the transformer 220. When the lowest, the first switching device 222 is turned on. That is, since the voltage V1 between the source and the drain of the first switching device 222 is minimized, the loss when the first switching device 222 is turned off and then turned on is minimized.

The voltage at the reference point 221 immediately before the first switching element 222 is turned on is caused by resonance between the leakage inductance and the distribution capacitance of the transformer 220 and the capacitance between the source and the drain of the first switching element 222. From the turn off, resonance occurs and the maximum counter electromotive force is generated just before turning on. A diagram showing this is shown in FIG. 4A. That is, in FIG. 4A, the turn-on period and the turn-off period of the first switching element 222 are repeatedly executed to show the counter electromotive force at this time. The counter electromotive force causes the magnetic flux to change when a current flows in the winding L1 210, thereby causing induced electromotive force to interfere with the change, which becomes the counter electromotive force.

Therefore, in the turn-on period 400, since the first switching device 222 is turned on to be conductive between the drain and the source, energy accumulation does not occur without generating counter electromotive force. The accumulated magnetic energy is expressed as follows.

W = 1/2 (L1 x I12)

That is, it is proportional to the square of the winding L1 220 and the current I1. A diagram showing a graph of current I1 is shown in FIG. 4B. That is, the current I1 is a current due to energy accumulated in the transformer 220 when the drain-source of the first switching element 222 is turned on. Therefore, the first switching element 222 flows only when the turn-on period 400 and does not flow when the turn-off period 410. Of course, the conventional current also has the same shape, which is shown in Figure 3b.

However, conventionally, the resonance frequency is the same, and due to the energy accumulated in the winding L1 210, it is not sufficient to make the voltage V1 of the reference point 221 into "0" volts. That is, attenuation vibration is generated so that the voltage of the first switching element 222 is turned on at a voltage close to “0” volts without the voltage V1 dropping completely to “0” volts. A diagram showing this is shown in FIG. 3A. That is, FIG. 3A shows that the first switching device 222 is turned on at the point where the reference point 300 is not “0” volts. That is, since the use of the hatched portion 310 of the magnetic energy can be seen that the efficiency is reduced.

In the present invention, in order to eliminate the conventional inefficiency, the second switching element 224 and the clamp capacitor Cr 225 which are active clamp circuits are configured. The second switching device 224 and the clamp capacitor 225, which are the active clamp circuit, minimize the leakage inductance and resonate with a large energy. This is because the main cause of the attenuation vibration is that the leakage inductance of the transformer 220 and the parasitic capacitors of the respective parts are generated under high voltage when the first switching 222 is turned off.

Therefore, the clamp capacitor 225 preferably has a much larger capacity than the parasitic capacitor.

Therefore, the flat voltage waveform can be maintained even at a high voltage due to the resonance of the winding L1 210 and the clamp capacitor 225. Figure 4a shows this. That is, the attenuation vibration does not occur at the turn-on point 400 of the second switching element 224, and thus the electromotive force can be maintained for a certain time.

The energy accumulated in the winding L1 210 is guided to the output winding L2 211 so that energy is emitted through the diode D 226. Therefore, before the energy is exhausted through the diode 226, the second switching device 224 is turned off, so that the energy accumulated in the inductance of the winding L1 210 and the leakage inductance of the winding L1 210 is first switched. The high frequency is generated by resonating with the drain-source capacitor of the device 222 to increase the resonance Q value. In other words, as the resonance Q value increases, the voltage V1 of the reference point 221 is lowered to “0” as much as possible, thereby achieving “0” voltage switching by reliable resonance.

In other words, this means that the turn-on point 410 of the first switching element 222 reaches the "0" voltage.

Accordingly, the voltage between the drain and the source of the first switching device 222 due to the leakage inductance of the winding L1 210 may be reduced by the second switching device 224 and the clamp capacitor 225 and spikes may be required even when a multi output is desired. Since the negative voltage is eliminated, it is possible to minimize the interference between the output voltages.

FIG. 5 is a view illustrating waveforms of the output voltage Vout according to FIG. 4A. That is, since the flyback circuit is used, only the turn-off section 410 of the first switching device 222 that generates the counter electromotive force is generated, and the turn-on section 400 changes to the flat voltage. In other words, the energy stored in the transformer 220 is discharged to the output by the counter electromotive force.

The conventional output voltage waveform is similar, but unlike the output waveform according to the present invention, in the conventional output waveform, spike noise, such as ringing, is inserted into the irregular waveform.

In addition, by using a small breakdown voltage of the diode 226, the heat sink is reduced by reducing the forward voltage to reduce the loss. In addition, the switching loss is reduced, thereby increasing the resonance frequency, thereby reducing the size of the transformer 220. Since there is no spike noise such as ringing, when the multi-output power is used, each output voltage deviation due to the load variation of the output voltage can be greatly reduced. In FIG. 2, only one output side secondary winding L2 211 is configured for understanding of the present invention, but multiple outputs are possible by configuring several secondary windings in parallel.

While the present invention has been described in detail with reference to the accompanying embodiments and drawings, the present invention is not limited to the specific embodiments, and those skilled in the art or those skilled in the art are familiar with the contents of the present invention. It should be understood that numerous variations and modifications can be made without departing. Therefore, the protection scope of the present invention will be preferably determined based on the appended claims.

According to the present invention, there is an effect that it is possible to implement an effective resonance while the reverse electromotive voltage is arbitrarily added using a pseudo resonance type.

In addition, the switching loss is reduced, so that the switching frequency can be increased to reduce the size and weight of the transformer and capacitor, and the loss can be reduced, thereby minimizing the parallel panel, thereby reducing the volume of the power supply.

In addition, in the case of designing a multi-output, a difference occurs depending on the amount of current between the outputs because the coupling degree is not high at a high frequency, but it has the effect of stabilizing the output voltage between the outputs.

Claims (4)

In a switching power supply comprising a diode for releasing the accumulated energy of the input side winding induced by the output secondary winding as a current and a voltage smoothing capacitor for smoothing the voltage at the output side, A transformer consisting of an input winding formed by magnetic energy accumulation before transformation and a secondary winding for output after transformation by receiving the energy; A first switching device connected to the transformer to perform a switching function; A second switching element connected to the transformer to perform a switching function and an active clamp function; A clamp capacitor connected to the second switching element to perform an active clamp function; A voltage detector connected to the input winding of the transformer and measuring a voltage for turning on a first switching device; A driving unit configured to drive a first switching element by configuring a pseudo resonance control circuit; A turn-on / off control circuit unit configured to turn on or turn off the first switching element by forming a pseudo resonance control circuit to control the driving unit. Pseudo resonance type power supply comprising a. The method of claim 1, And the first switching element is turned on when the voltage detection unit becomes a "0" voltage. The method of claim 1, The second switching device, And the first switching element is connected to the input side winding only during a turn-off period and is turned on when a counter electromotive force is generated. The method of claim 1, The clamp capacitor has a pseudo resonance power supply, characterized in that the first switching device is turned off and the second switching device performs a resonance when turned on to generate a flat voltage.

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