JP2002044945A - Switching power circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To improve power-conversion efficiency of a compound-resonance converter that is provided with a secondary-side active-clamp circuit. SOLUTION: Stabilization of a secondary-side DC-output voltage E0 is obtained, by means of a compound-control method that variably and simultaneously controls the switching frequency and the continuity angle of a switching converter of a primary-side voltage-resonance type, according to its level. Thereby, the continuity angle of a secondary-side auxiliary switching element can be reduced so that increase of power loss that occurs along with the controlling of a constant-voltage is substantially prevented.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a switching power supply circuit provided as a power supply for various electronic devices.

[0002]

2. Description of the Related Art As a switching power supply circuit, a circuit employing a switching converter of a type such as a flyback converter or a forward converter is widely known. Since these switching converters have a rectangular switching operation waveform, there is a limit in suppressing switching noise. Also, due to its operating characteristics,
It has been found that there is a limit in improving the power conversion efficiency. Therefore, the present applicant has previously proposed various switching power supply circuits using various resonance type converters.
The resonance type converter can easily obtain high power conversion efficiency and realize low noise because the switching operation waveform is sinusoidal. It also has the advantage that it can be configured with a relatively small number of parts.

FIG. 9 is a circuit diagram showing an example of a prior art switching power supply circuit which can be constructed based on the invention proposed by the present applicant. As a basic configuration of the power supply circuit shown in this figure, a voltage resonance type converter is provided as a primary side switching converter.

In the power supply circuit shown in FIG. 1, a rectified and smoothed voltage Ei corresponding to a level that is one-time the AC input voltage VAC is generated from a commercial AC power supply (AC input voltage VAC) by a bridge rectifier circuit Di and a smoothing capacitor Ci. .

[0005] A single-ended single-end system is adopted as a voltage resonance type converter that receives and inputs the rectified smoothed voltage Ei (DC input voltage) and is intermittent. The drive system employs a self-excited configuration. In this case, the switching element Q1 forming the voltage resonance type converter includes:
A high breakdown voltage bipolar transistor (BJT; junction transistor) is selected. A primary side parallel resonance capacitor Cr is connected in parallel between the collector and the emitter of the switching element Q1. A clamp diode DD is connected between the base and the emitter. Here, the parallel resonance capacitor Cr forms a primary side parallel resonance circuit together with the leakage inductance L1 obtained in the primary winding N1 of the insulating converter transformer PIT, so that an operation as a voltage resonance type converter can be obtained. It has become.

A self-excited oscillation drive circuit including a base current limiting resistor RB, a drive winding NB, an inductor LB, and a resonance capacitor CB is connected to the base of the switching element Q1. The switching element Q1 is switched by being supplied with a base current based on an oscillation signal generated by the self-excited oscillation drive circuit. Here, the drive winding NB is wound independently of the primary winding N1 on the primary side of the insulating converter transformer PIT, and is excited by, for example, an alternating voltage obtained on the primary winding N1. Then, the self-excited oscillation drive circuit performs an oscillation operation by this excitation operation.

In this configuration, the switching frequency of the switching element Q1 is set to
It is fixedly determined by the resonance frequency of the resonance circuit formed by the series connection of the drive winding NB, the inductor LB and the resonance capacitor CB. In addition, at the time of startup, it is started by a startup current flowing from the line of the rectified smoothed voltage Ei to the base via the startup resistor Rs.

[0008] The insulating converter transformer PIT is provided for transmitting the switching output of the switching converter obtained on the primary side to the secondary side. This insulated converter transformer PIT has a primary winding N with respect to an EE type core.
1 and the secondary winding N2 are divided and wound, and a gap G is formed with respect to the center magnetic leg so that a loose coupling state with a required coupling coefficient can be obtained, and a saturated state can be obtained. I try to be hard to be.

The primary winding N1 of the insulating converter transformer PIT is connected between the line of the DC input voltage (rectified smoothed voltage Ei) and the collector of the switching element Q1. The switching element Q1 performs switching with respect to the DC input voltage. As a result, the switching output of the switching element Q1 is supplied to the primary winding N1, and an alternating voltage having a cycle corresponding to the switching frequency is generated. I do.

On the secondary side of the insulating converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. In this case, the secondary side parallel resonance capacitor C2 is connected in parallel to the secondary winding N2, so that the leakage inductance L2 of the secondary side winding N2 and the capacitance of the secondary side parallel resonance capacitor C2 are determined. A parallel resonance circuit is formed. With this parallel resonance circuit,
The alternating voltage induced in the secondary winding N2 becomes a resonance voltage.
That is, a voltage resonance operation is obtained on the secondary side.

That is, in this power supply circuit, the primary side is provided with a parallel resonance circuit for performing a voltage resonance type switching operation, and the secondary side is provided with a parallel resonance circuit for obtaining a voltage resonance operation. In the present specification, such a switching converter configured to operate with a resonance circuit provided on the primary side and the secondary side is also referred to as a “composite resonance type switching converter”.

On the secondary side of the power supply circuit formed as described above, a secondary side parallel resonance circuit composed of a secondary winding N2 and a secondary side parallel resonance capacitor C2 is connected according to the connection configuration shown in FIG. By connecting the rectifier diode DO1 and the smoothing capacitor CO1, a half-wave rectifier circuit is formed. Then, the half-wave rectifier circuit (DO1, CO1) generates the secondary-side DC output voltage EO1 which is the main. The secondary side DC output voltage EO1 is, for example, about 135V. This secondary side DC output voltage EO1 is a PWM
By being branched and input to the control circuit 42, it is used as a detection voltage.

In this case, a tap output is provided at the end of the secondary winding N2 as shown in the figure, so that the tertiary winding N3 is provided between the tap output line and the secondary ground. It is formed. A half-wave rectifier circuit composed of a rectifier diode DO2 and a smoothing capacitor CO2 is connected to the tertiary winding N3 as shown in FIG.
Is generated and output. This low voltage secondary side DC output voltage EO2
Is branched and input to the PWM control circuit 42, so that it is also used as an operation power supply.

On the secondary side of the circuit shown in FIG. 9, a secondary side active clamp circuit 40 is provided.
As shown, the secondary side active clamp circuit 40 is provided by connecting a series connection circuit of an auxiliary switching element Q3 of a MOS-FET and a clamp capacitor CCL2 in parallel with a secondary side parallel resonance circuit. . A clamp diode DD3 is connected in parallel between the drain and source of the auxiliary switching element Q3, and the clamp diode DD3 used is a MOS-FET provided as a body diode.
In this case, the auxiliary switching element Q3 of the secondary side active clamp circuit 40 is composed of the driving winding Ng2 and the resistance Rg.
It is driven by a self-excited drive circuit of 2-Cg2. As shown in the figure, the drive winding Ng2 is provided so as to be independently wound on the secondary side of the insulating converter transformer PIT, so that the alternating voltage is generated by the drive winding NB wound on the primary side. It is to be excited. The number of turns of the drive winding Ng2 is, for example, 1T, and the alternating voltage obtained in the drive winding Ng2 is:
The drive winding NB has a polarity opposite to that of the drive winding NB. As a result, the alternating voltage obtained at the drive winding Ng2 has the opposite polarity to the alternating voltage obtained at the secondary winding N2, and therefore, the auxiliary switching element Q3 is obtained at the secondary winding N2. The switching operation is performed so that the secondary side rectifier diode DO1, which is turned on / off by the alternating voltage, is turned on during the off period.

The PWM control circuit 42 is provided for stabilizing the secondary DC output voltage by performing conduction angle control (PWM control) of the auxiliary switching element Q3 of the secondary active clamp circuit 40. I have. In this case, PW
The M control circuit 42 is configured as an error amplifying circuit including a shunt regulator including a voltage dividing resistor R1-R2 and a control element Q5, and an inverting amplifier including a drive transistor Q4 and resistors R3, R4, R5, and R6. . Such a P
In the configuration of the WM control circuit 42, the secondary DC output voltage E is increased by, for example, increasing the AC input voltage or reducing the load power.
Assuming that O1 has risen, the switching frequency of the auxiliary switching element Q3 is fixed, the conduction angle is enlarged, and the ON period is extended. As a result, the period during which the rectifier diode DO1 is turned on and turned on becomes shorter, and the amount of rectified current charged in the smoothing capacitor CO1 becomes smaller, so that the level of the secondary-side DC output voltage EO1 decreases. In this way, the secondary DC output voltage is stabilized.

FIG. 10 is a waveform diagram showing the operation of the main part of the power supply circuit shown in FIG.
= 100V and the load power Po = 200W are shown. When the switching element Q1 performs the switching operation, the period TO during which the switching element Q1 is turned off
In FF1, the resonance operation of the primary side parallel resonance circuit is obtained. As a result, as shown in FIG. 10A, as the parallel resonance voltage V1 obtained at both ends of the parallel resonance capacitor Cr, the period TON1 during which the switching element Q1 is turned on is at the 0 level, and the sine wave shape during the period TOFF1. The waveform has a resonance pulse. The level of the parallel resonance voltage V1 changes according to the load power fluctuation, and tends to increase as the load power increases.

The switching output current I1 flowing to the drain or the collector of the switching element Q1 is shown in FIG.
As shown by 0 (b), it flows at the 0 level in the period TOFF1 and flows according to the waveform shown in the period TON1.
The level of the switching output current I1 also tends to increase as the load power Po increases.

Further, as shown in FIG. 10C, the secondary parallel resonance voltage V2 obtained in the secondary parallel resonance circuit (N2 // C2) is, during the period DON during which the rectifier diode DO1 is on, as shown in FIG. During a period DOFF, which is clamped at the level of the secondary side DC output voltage EO1 and turned off, a sine-wave pulse having 300 Vp in the negative polarity direction is obtained.

The secondary side active clamp circuit 40
As an operation, the clamp current ICL2 flowing through the clamp capacitor CCL2 flows during the period TON3 in which the auxiliary switching element Q3 is in the ON state according to the waveform shown in FIG. The clamp current ICL2 in this period TON3 has a negative waveform by flowing through the path of the clamp diode DD3 → the clamp capacitor CCL2 → the secondary winding N2 in the first half period, and in the second half period, the secondary winding N2 → clamping It flows on the path of the capacitor CCL2 → the auxiliary switching element Q3 (drain → source). Thus, in the period TON3, most of the secondary current flows as the clamp current ICL2,
The current I2 flowing through the secondary side parallel resonance capacitor C2 is shown in FIG.
0 (e), the auxiliary switching element Q3
Is turned off, and flows as a rectangular wave pulse in a short period at the start and end of the period TOFF3 in which is turned off.

Here, for example, the driving winding Ng2-the resistance Rg2
When the self-excited drive circuit based on -Cg2 is not provided, the waveform at the time of turning off the auxiliary switching element Q3 becomes blunt as shown by the broken line a in the clamp current ICL2 of FIG. No operation is obtained. However, the provision of the self-excited drive circuit using the drive winding Ng2-the resistors Rg2-Cg2 allows the gate current Ig of the auxiliary switching element Q3 to be as shown in FIG.
As shown in (1), since the current for turning on and turning off flows reliably in the short period at the start and end of the period TON3, the auxiliary switching element Q3 turns off sharply. As a result, the PWM control circuit 42 stabilizes the auxiliary switching element Q3 to P
WM control can be performed.

FIG. 11 shows, as constant voltage characteristics of the circuit shown in FIG. 9, fluctuation characteristics of the ON period TON1 of the switching element Q1 and the ON period TON3 of the auxiliary switching element Q3 with respect to the fluctuation of the AC input voltage VAC. Show. This figure shows that the on-period TON1 of the switching element Q1 is substantially constant while the AC input voltage VAC increases in the range of 80V to 140V. This means that the primary-side voltage resonance type converter performs a switching operation with a fixed switching frequency and a fixed on / off period within one switching cycle. On the other hand, the ON period TON3 of the auxiliary switching element Q3 changes so as to increase as the AC input voltage VAC increases. This means that
Secondary side DC output voltage EO1 with AC input voltage VAC rise
Indicates that the PWM control is performed so as to increase the conduction angle of the auxiliary switching element Q3 for the constant voltage control in accordance with the rise of. In the circuit configuration shown in FIG. 9, the switching frequency of the auxiliary switching element Q3 is fixed. Also, the ON period TON3 of the auxiliary switching element Q3
FIG. 2 shows the case where the load power Po = 200 W and the case where Po = 50 W, as shown in the figure, the Po = 50 W is longer than the load power Po = 200 W during the period T.
ON3 is longer. This is due to the fact that stabilization by PWM control is performed on the tendency that the secondary side DC output voltage EO1 increases as the load becomes lighter.

In FIG. 11, the AC input voltage V
The relationship between AC and power conversion efficiency ηAC → DC is also shown. As the power conversion efficiency, the characteristics when the load power Po = 200 W and the characteristics when Po = 50 W are shown, and the load power Po =
At 200 W, it tends to decrease near 90% as the AC input voltage VAC increases. Po = 50W
At times, the voltage tends to decrease as the AC input voltage VAC increases near 85%.

FIG. 12 shows, as a constant voltage characteristic, the ON period TON of the switching element Q1 with respect to the fluctuation of the load power.
1 and fluctuation characteristics of the ON period TON3 of the auxiliary switching element Q3. Further, in this figure, the fluctuation characteristics of power conversion efficiency ηAC → DC with respect to load power fluctuation are shown. The on-period TON1 of the switching element Q1 is almost constant irrespective of the fluctuation of the load power Po50W to 200W, and also here, the switching operation is performed with the on / off period of the primary-side voltage resonance type converter fixed. Is shown. On the other hand, the ON period TON3 of the auxiliary switching element Q3 is controlled so as to become shorter as the load power Po becomes heavier. Here, the conduction angle of the auxiliary switching element Q3 is controlled as a constant voltage control. Is performed. Also, as the power conversion efficiency ηAC → DC, the load power P
As shown in the drawing, a result is obtained in which the value f varies in the range of about 85% to 90% according to the variation of o = 50 W to 200 W.

[0024]

In the power supply circuit shown in FIG. 9, after the switching frequency and the on / off period of the primary side switching converter are fixed as described above, the secondary side active clamp circuit is provided. 40
PWM of ON period of auxiliary switching element Q3
Control is performed to perform constant voltage control. For this reason, as the AC input voltage VAC rises, the primary-side parallel resonance voltage V generated during the period in which the switching element Q1 is turned off.
A phenomenon occurs in which one pulse also rises. Then, for example, as an AC 100 V system, an AC input voltage VAC = 140 V
It is known that when the voltage rises to a certain point, the voltage rises to about 900 Vp. Also, when the AC input voltage VAC rises to 280 V as an AC 200 V system, the pulse of the primary side parallel resonance voltage V 1 rises to about 1800 V. Also, depending on the rise of the AC input voltage VAC,
The conduction angle of the auxiliary switching element Q3 is also controlled so as to increase. Therefore, when the AC input voltage VAC rises,
The power loss in the switching element Q1 and the auxiliary switching element Q3 increases. For example, in practice, when the AC input voltage VAC = 140 V, the power conversion efficiency ηAC → DC is reduced by about 2.5% compared to when VAC = 100.

Similarly, when the load power Po is 200 W
Even when the power is reduced to 50 W, the operation is controlled to increase the conduction angle of the auxiliary switching element Q3 to achieve stabilization. Therefore, the power loss in the auxiliary switching element Q3 is significantly reduced, The conversion efficiency ηAC → DC decreases by about 5%. In this way, the circuit shown in FIG. 9 has a problem that the power conversion efficiency is reduced as the AC input voltage VAC increases or the load becomes lighter.

Further, as described above, since the peak level of the primary side parallel resonance voltage pulse becomes considerably high, at least the primary side switching element Q1 has a high withstand voltage that can support 900 Vp or 1800 Vp. Must be selected. If the switching element is made of a high withstand voltage product, the switching characteristics will be reduced, so that the switching loss will increase, and the above-described increase in power loss will be unavoidable. In this case, since the heat generated by the switching element becomes considerable, for example, a heat radiating plate is required, which hinders a reduction in size, weight, and cost of the power supply circuit.
Furthermore, a switching element of a high withstand voltage product is also very expensive at present, and this is also disadvantageous in cost.

[0027]

In view of the above-mentioned problems, the present invention is configured as a switching power supply circuit as follows. That is, a switching unit formed with a main switching element for switching and outputting an input DC input voltage and a primary-side parallel resonance circuit that makes the operation of the switching unit a voltage resonance type are formed. A gap is formed so as to obtain a required coupling coefficient that is loosely coupled between the primary side and the secondary side, and the output of the switching means obtained on the primary side. And an insulating converter transformer for transmitting to the next side. In addition, a secondary parallel resonance circuit formed by connecting a secondary parallel resonance capacitor in parallel to a secondary winding wound around an insulation converter transformer, and a secondary winding wound around an insulation converter transformer DC output voltage generating means configured to obtain a secondary DC output voltage by inputting an alternating voltage obtained to a line and performing a rectification operation. In addition, by providing a series connection circuit of a clamp capacitor and a secondary-side auxiliary switching element, it is provided so as to clamp a voltage generated in the secondary winding, and is wound around the secondary side of the insulating converter transformer. Secondary active clamp means driven by a self-excited drive circuit provided with a drive winding having the following configuration. Further, according to the level of the secondary-side DC output voltage, by performing the switching frequency and conduction angle control of the main switching element, a voltage control means that performs constant voltage control of the secondary-side DC output voltage. Provide.

According to the above configuration, the primary side is provided with the primary side parallel resonance circuit for forming the voltage resonance type converter, and the secondary side is formed by the secondary winding and the secondary side parallel resonance capacitor. , A so-called composite resonance type switching converter provided with a secondary parallel resonance circuit. Based on this configuration, the secondary side is provided with secondary side active clamp means for suppressing the voltage level obtained in the secondary side parallel resonance circuit. The constant voltage control of the secondary DC output voltage is performed by simultaneously controlling the switching frequency and the conduction angle of the main switching element that forms a primary voltage resonance type switching converter according to the level of the secondary DC output voltage. It will be performed in a controlled manner. The secondary side auxiliary switching element forming the secondary side active clamp means performs switching by the self-excited drive circuit according to the switching frequency and the on / off period transmitted to the secondary side via the insulating converter transformer. However, this makes it possible to reduce the change in the conduction angle of the secondary-side auxiliary switching element according to the AC input voltage fluctuation or the load power fluctuation.

[0029]

FIG. 1 shows the configuration of a switching power supply circuit according to a first embodiment of the present invention.
In the power supply circuit shown in this figure, first, as a rectifying and smoothing circuit for inputting a commercial AC power supply (AC input voltage VAC) to obtain a DC input voltage, a full-wave rectifying circuit including a bridge rectifying circuit Di and a smoothing capacitor Ci. To generate a rectified and smoothed voltage Ei corresponding to a level that is one time the AC input voltage VAC.

As the switching converter that receives the rectified smoothed voltage Ei (DC input voltage) and is intermittent,
A voltage resonance type converter including a stone switching element Q1 and performing a so-called single-ended switching operation is provided. The voltage resonance type converter here has a separately-excited configuration, and for example, a MOS-FET is used as the switching element Q1. The drain of the switching element Q1 is connected to the positive electrode of the smoothing capacitor Ci via the primary winding N1 of the insulating converter transformer PIT, and the source is connected to the primary side ground.

The drain of the switching element Q1
A parallel resonance capacitor Cr is connected in parallel between the sources. The capacitance of the parallel resonance capacitor Cr and the primary winding N1 of the insulation converter transformer PIT
The primary side parallel resonance circuit is formed by the leakage inductance obtained as described above. The resonance operation of the parallel resonance circuit is obtained in accordance with the switching operation of the switching element Q1, so that the switching operation of the switching element Q1 is a voltage resonance type.

The drain of the switching element Q1
Between the sources, a clamp diode DD constituted by a so-called body diode provided in the MOS-FET
Are connected in parallel to form a path for a clamp current flowing during a period when the switching element is turned off.
Further, in this case, the drain of the switching element Q1 is connected to the oscillation circuit 11 in the switching drive unit 10 described below. The output of the drain input to the oscillation circuit 11 is used to variably control the ON period of switching at the time of switching frequency control as described later.

The switching element Q1 is connected to the oscillation circuit 1
1 and a drive circuit 12, which are integrally driven by a switching drive unit 10,
The switching frequency is variably controlled for constant voltage control. Note that the switching drive unit 10 in this case is provided as, for example, one integrated circuit (IC). The switching drive unit 10 is connected to the line of the rectified and smoothed voltage Ei via the starting resistor Rs. For example, at the time of starting the power, the switching driving unit 10 is started by applying the power supply voltage via the starting resistor Rs. It has been like that.

Oscillation circuit 11 in switching drive unit 10
Then, an oscillating operation is performed to generate and output an oscillating signal.
Then, the drive circuit 12 converts this oscillation signal into a drive voltage and outputs it to the gate of the switching element Q1. Thereby, the switching element Q1
Performs a switching operation based on an oscillation signal generated by the oscillation circuit 11. Therefore, the switching frequency of the switching element Q1 and the duty of the on / off period within one switching cycle are determined by the oscillation circuit 4
Is determined depending on the oscillating signal generated at.

Here, in the present embodiment, a secondary DC output voltage EO described later is input to the oscillation circuit 11 via the photocoupler 30.
Therefore, the oscillation circuit 11 performs an operation of varying the oscillation signal frequency (switching frequency fs) based on the level of the secondary DC output voltage EO. Also,
At the same time as changing the switching frequency fs, the oscillation signal waveform is controlled so that the period TON (conduction angle) during which the switching element Q1 is turned on is changed. The variable control of the period TON is performed by switching the switching resonance pulse voltage V obtained at both ends of the parallel resonance capacitor Cr.
Be made based on one level. By the operation of the oscillation circuit 12, the secondary DC output voltage EO is stabilized as described later. In particular, in the present embodiment, as described later, the auxiliary switching element Q
2 is driven by the self-excited drive circuit including the drive winding Ng1 wound around the insulating converter transformer PIT, so that not only the ON period but also the OFF period within one switching cycle is variably controlled. That is, in the present embodiment, as the composite control method, constant voltage control is performed by changing three parameters of the switching frequency of the main switching element Q1 and the ON period and the OFF period within one switching cycle.

Further, on the primary side of the power supply circuit of the present embodiment, a primary side active clamp circuit 20 for clamping the parallel resonance voltage V1 obtained at both ends of the parallel resonance capacitor Cr is provided.

In this case, the primary side active clamp circuit 20 is connected to the auxiliary switching element Q2,
A clamp capacitor CCL and a clamp diode DD2 are provided. For example, a MOS-
The body diode of the FET is used. Also,
As a drive circuit system for driving the auxiliary switching element Q2, a drive winding Ng1, a capacitor Cg1, a resistor R1,
R2.

A clamp diode DD2 is connected in parallel between the drain and source of the auxiliary switching element Q2. Here, the anode of the clamp diode DD2 is connected to the source, and the cathode is connected to the drain. The drain of the auxiliary switching element Q2 is connected via a clamp capacitor CCL to a connection point between the line of the rectified smoothed voltage Ei and the winding start end of the primary winding N1. Further, the source of the auxiliary switching element Q2 is connected to the winding end of the primary winding N1. That is, the active clamp circuit 2 of the present embodiment
A value of 0 is obtained by connecting a clamp capacitor CCL in series to the parallel connection circuit of the auxiliary switching element Q2 // clamp diode DD2. The circuit thus formed is connected in parallel with the primary winding N1 of the insulating converter transformer PIT.

As a drive circuit system for the auxiliary switching element Q2, a series connection circuit of a resistor R1, a capacitor Cg1, and a driving winding Ng1 is connected to the gate of the auxiliary switching element Q2, as shown in the figure. This series connection circuit forms a self-excited oscillation drive circuit for the auxiliary switching element Q2. Here, the drive winding Ng1 is formed so as to wind up the winding end end of the primary winding N1 in the insulating converter transformer PIT, and the number of turns in this case is, for example, 1T (turn).
As a result, a voltage excited by the alternating voltage obtained in the primary winding N1 is generated in the driving winding Ng1. In this case, a voltage having a polarity opposite to that of the primary winding N1 and the driving winding Ng1 is obtained from the relationship in the winding direction. In practice, the operation is guaranteed if the number of turns of the drive winding Ng1 is 1T, but the present invention is not limited to this.
Further, the resistor R2 is inserted between a connection point between the primary winding N1 of the insulating converter transformer PIT and the driving winding Ng1.

The insulated converter transformer PIT transmits the switching output of the switching element Q1 to the secondary side. As shown in FIG.
For example, an EE-type core in which E-type cores CR1 and CR2 made of a ferrite material are combined so that their magnetic legs face each other is provided. The wire N1 and the secondary winding N2 are wound separately. A gap G is formed for the center magnetic leg as shown in the figure.
As a result, loose coupling with a required coupling coefficient can be obtained. Gap G is E type core CR1, CR
The two center magnetic legs can be formed by making them shorter than the two outer magnetic legs. The coupling coefficient k is set to a loosely coupled state of, for example, k ≒ 0.85, so that a saturated state is hardly obtained.

The winding end of the primary winding N1 of the insulating converter transformer PIT is connected to the drain of the switching element Q1 as shown in FIG. 1, and the winding start is connected to the positive electrode of the smoothing capacitor Ci (rectified smoothing voltage Ei). ) And connected. Therefore, by supplying the switching output of the switching element Q1 to the primary winding N1, an alternating voltage having a cycle corresponding to the switching frequency is generated.

On the secondary side of the insulating converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. In this case, the secondary side parallel resonance capacitor C2 is connected in parallel to the secondary winding N2, so that the leakage inductance L2 of the secondary side winding N2 and the capacitance of the secondary side parallel resonance capacitor C2 are determined. A parallel resonance circuit is formed. With this parallel resonance circuit,
The alternating voltage induced in the secondary winding N2 becomes a resonance voltage,
Therefore, a voltage resonance operation is obtained on the secondary side. That is, this power supply circuit also includes a "parallel resonance circuit for providing a voltage resonance operation on the primary side and a parallel resonance circuit for obtaining a voltage resonance operation on the secondary side. It adopts the configuration of "switching converter".

For the secondary side of the power supply circuit formed as described above, a half of a secondary rectifier diode DO and a smoothing capacitor CO connected to the winding start end of the secondary winding N2. A wave rectifier circuit, whereby the secondary winding N
The secondary-side DC output voltage EO on the surface corresponding to a level substantially equal to the alternating voltage induced at 2 is obtained. The secondary-side DC output voltage EO is input to the oscillation circuit 11 of the primary-side switching drive unit 10 via the photocoupler 30.

By the way, the insulation converter transformer PIT
On the secondary side includes the connection relationship between the polarity (winding direction) of the primary winding N1 and the secondary winding N2 and the connection of the rectifier diode DO (DO1), and the polarity of the alternating voltage excited by the secondary winding N2. Due to the change, the inductance L1 of the primary winding N1 and the secondary winding N
The mutual inductance M with the second inductance L2 may be + M or -M. For example, when the circuit becomes equivalent to the circuit shown in FIG. 8A, the mutual inductance becomes + M, and when it becomes equivalent to the circuit shown in FIG. 8B, the mutual inductance becomes -M. If this is made to correspond to the operation on the secondary side shown in FIG. 1, in the secondary half-wave rectifier circuit, when the alternating voltage obtained in the secondary winding N2 has a positive polarity, the secondary side rectifier diode Since the rectified current flows through (DO1), this operation can be regarded as an operation mode (forward operation) of + M.

In the power supply circuit of the present embodiment, a secondary side active clamp circuit 40 is provided.
The secondary side active clamp circuit 40 is also a MOS-F
An ET auxiliary switching element Q3, a clamp capacitor CCL2, and a body diode clamp diode DD3 are provided. Further, as a drive circuit system for driving the auxiliary switching element Q3, a drive winding Ng2, a capacitor C
g2 and a resistor Rg2.

A clamp diode DD3 is connected in parallel between the drain and source of the auxiliary switching element Q3. The connection form is a clamp diode D
The anode of D3 is connected to the source, and the cathode is connected to the drain. The drain of the auxiliary switching element Q3 is connected via a clamp capacitor CCL2 to a connection point between the winding start end of the secondary winding N2 and the anode of the rectifier diode DO1.
The source of the auxiliary switching element Q2 is connected to the secondary side ground. Therefore, as the secondary-side active clamp circuit 40, the auxiliary switching element Q3
// For the parallel connection circuit of clamp diode DD3,
It is constituted by connecting a clamp capacitor CCL2 in series. The circuit thus formed is further connected in parallel to the secondary parallel resonance circuit.

As shown in the figure, the driving circuit system of the auxiliary switching element Q3 has a resistor Rg2-a capacitor Cg2-
A series connection circuit of the drive winding Ng2 is connected. Also in this case, the series connection circuit forms a self-excited drive circuit for the auxiliary switching element Q3. However, here, the resistance Rg2
And a resistor Rg2A connected to the secondary side ground, and a voltage dividing point (connection point) of the resistor Rg2 and the resistor Rg2A is connected to the gate of the auxiliary switching element Q3. The drive winding Ng2 in this case is formed so as to wind up the winding end end side of the secondary winding N2, and the number of turns in this case is, for example, 1T (turn). As a result, a voltage excited by the alternating voltage obtained in the primary winding N1 is generated in the driving winding Ng2. In this case, a voltage having a polarity opposite to that of the primary winding N2 and the driving winding Ng2 is obtained from the relationship in the winding direction. The operation of the drive winding Ng2 is guaranteed if the number of turns is 1T, but is not limited to this.

Next, the stabilizing operation in the circuit shown in FIG. 1 will be described. As described above, the secondary-side DC output voltage EO is input to the oscillation circuit 11 in the primary-side switching drive unit 10 via the photocoupler 30. The photocoupler 30 is provided to insulate the primary side and the secondary side in a DC manner. And the oscillation circuit 1
1, the input secondary side DC output voltage EO
, The frequency and the on / off duty ratio of the oscillation signal are variably outputted in accordance with the level change of the oscillation signal. This is a composite control operation in which the switching frequency of the switching element Q1 is varied and the ON / OFF period of the switching element Q1 is variably controlled. Changes, the insulation converter transformer P
The energy transmitted to the secondary side of the IT will also change. As a result, the constant-pressure secondary-side DC output voltage EO is controlled to be constant at a required level. That is, the secondary-side DC output voltage EO is controlled by controlling the switching operation on the primary side by the complex control method.
It is intended to be stable.

The waveform diagram of FIG. 2 shows the operation of the circuit shown in FIG. That is, the operation as a voltage resonance type converter provided with the primary side active clamp circuit 20 and the secondary side active clamp circuit 40 is shown. The operation shown in FIG. 2 is assumed to be obtained when the circuit shown in FIG. 1 has a configuration corresponding to the AC 100 V system, and FIGS. The operation of each section under the condition that the load power Po = 200 W is shown, and FIGS.
Has an AC input voltage VAC = 100 V and a load power Po = 5
The operation of the same part as in FIGS. 2A to 2G under the condition of 0 W is shown.

First, the operation when the load power Po = 200 W shown in FIGS. 2A to 2G will be described. In this figure, the operation mode within one switching cycle of the primary-side switching element Q1
Up to five operating modes are shown. It is during the period TON that the main switching element Q1 is controlled to be turned on. In this period TON, a mode operation is obtained. The auxiliary switching element Q2 is controlled so as to be in the off state during the period TON.

In the mode (period TON), FIG.
According to the waveform shown in FIG.
The switching output current I1 flows through the drain of the main switching element Q1 via the leakage inductance L1 obtained in the primary winding N1 of the isolated converter transformer PIT.
It flows to. At this time, the switching output current I1 has a waveform that initially reverses from a negative direction to a positive direction, as shown in a period TON of FIG. 2B. Here, during the period in which the switching output current I1 flows in the negative direction, the discharge of the parallel resonance capacitor Cr ends at the end of the immediately preceding period td2, whereby the clamp diode DD1 conducts, and the clamp diode DD1 → the primary winding N1 is connected. When the switching output current IQ1 flows through the power supply mode, a mode in which power is regenerated on the power supply side is set. Then, at the timing when the switching output current I1 (FIG. 2B) reverses from the negative direction to the positive direction, the main switching element Q1 is turned on by ZVS (Zero Volt Switching) and ZCS (Zero Current Switching). .

Then, in the next period td1, the operation is performed as a mode. During this period, the main switching element Q1 is turned off, so that the current flowing through the primary winding N1 flows through the parallel resonance capacitor Cr. At this time, the current Icr flowing through the parallel resonance capacitor Cr has a waveform that appears as a pulse due to positive polarity, as shown in FIG. This is an operation as a partial resonance mode. At this time, since the parallel resonance capacitor Cr is connected in parallel with the main switching element Q1, the main switching element Q1 is turned off by ZVS.

Subsequently, a period in which the auxiliary switching element Q2 is controlled to be in the on state and the main switching element Q1 is controlled to be in the off state is a period, which is shown in FIG. This corresponds to a period TON2 in which the voltage V2 across the auxiliary switching element Q2 is at the 0 level. This period TON2 is an operation period of the active clamp circuit 20. First, the operation as a mode is performed, and then the operation as a mode is performed.

In the operation in the previous mode, the parallel resonant capacitor Cr is charged by the current flowing from the primary winding N1, and as a result, in the mode operation, the voltage level obtained in the primary winding N1 is reduced. At the initial stage (at the start of the period TON2), the potential is equal to or higher than the voltage level across the clamp capacitor CCL1. As a result, when the conduction condition of the clamp diode DD2 connected in parallel to the auxiliary switching element Q2 is satisfied and the conduction is performed, a current flows through the path from the clamp diode DD2 to the clamp capacitor CCL1, and the clamp current ICL is After the start of the period TON2 in FIG. 2D, a saw-tooth waveform approaching the 0 level as time elapses from the negative direction is obtained. Here, for example, if the capacitance of the clamp capacitor CCL1 is selected to be 50 times or more the capacitance of the parallel resonance capacitor Cr, depending on the operation in this mode,
Most of the current flows as the clamp current ICL to the clamp capacitor CCL1, and hardly flows to the parallel resonance capacitor Cr. As a result, the slope of the parallel resonance voltage V1 applied to the main switching element Q1 during the period TON2 is made gentle, and as a result, as shown in FIG. The corners will widen. That is, a clamping operation for the parallel resonance voltage V1 is obtained. On the other hand, the parallel resonance voltage V1 obtained in the prior art circuit (the circuits in FIGS. 9 and 10) is 550 Vp
Is a pulse waveform having the following level.

Then, when the above mode is completed in the period TON2, the operation subsequently proceeds to the mode. At the start of this mode, the timing is such that the clamp current ICL shown in FIG. 2D is reversed from the negative direction to the positive direction. At this timing, the auxiliary switching element Q
2 is turned on by ZVS and ZCS at the timing when the clamp current ICL reverses from the negative direction to the positive direction. In the state where the auxiliary switching element Q2 is turned on in this manner, the resonance action of the primary side parallel resonance circuit obtained at this time causes the primary winding N1 → the drain → source of the auxiliary switching element Q2 via the clamp capacitor CCL. 2D, a waveform increasing in the positive direction is obtained as shown in FIG.

Here, although not shown, the voltage applied to the gate of the auxiliary switching element Q2 is
g, which is a pulse voltage having a rectangular waveform. Further, the gate inflow current flowing to the gate of the auxiliary switching element Q2 becomes a differential waveform by a differentiating circuit formed by the capacitor Cg and the resistor R2.
The current flows immediately after the end of the period td1 and in the period td2. The period td1 and the period td2 are threshold periods during which both the main switching element Q1 and the auxiliary switching element Q2 are turned off, and the threshold period is maintained by the flow of the gate inflow current.

The operation in the above-described mode is as follows.
The operation is terminated at a timing when the voltage V2 across the auxiliary switching element Q2, which has been set to the 0 level in the FF, starts rising, and then the operation shifts to a mode operation in the period td2. In mode,
An operation is obtained in which the parallel resonance capacitor Cr supplies a discharge current to the primary winding N1. That is, a partial resonance operation is obtained. At this time, the parallel resonance voltage V1 applied to the main switching element Q1 has a large slope due to the small capacitance of the parallel resonance capacitor Cr as described above, and as shown in FIG. hand,
It falls so as to rapidly descend to the 0 level. Then, the auxiliary switching element Q2 starts to turn off at the timing when the mode ends and the mode is started. At this time, the parallel resonance voltage V1 falls with a certain slope as described above. Then, a turn-off operation by ZVS is performed. Further, the voltage generated when the auxiliary switching element Q2 is turned off is prevented from rising sharply by discharging the parallel resonance capacitor Cr as described above. This operation is performed, for example, as shown as a voltage V2 across the auxiliary switching element Q2 in FIG.
During the period td2 (in the mode), 0 with a certain inclination
It is shown as a waveform that transitions from level to peak level. The voltage V2 across the auxiliary switching element Q2 has, for example, 300 Vp in a period TOFF2 in which the auxiliary switching element Q2 is turned off, and has a zero level during a period td1 (in a mode) which is a start period of the period TOFF2. And, during the period td2 (in the mode), which is the end period, the waveform changes from the 0 level to 300 Vp as described above. And after that,
The operations of modes 1 to 4 are repeated every switching cycle.

The operation on the secondary side is shown in FIG.
FIG. 2G shows a secondary alternating voltage V3 obtained at both ends of a secondary winding N2 (secondary parallel resonance capacitor C2). FIG. Is shown.
The secondary side alternating voltage V3 is the secondary DC voltage EO during the period DON during which the secondary side rectifier diode DO conducts and turns on.
, And has a waveform in which a peak is obtained in the negative polarity direction during the OFF period DOFF.

The auxiliary switching element Q3 of the secondary-side active clamp circuit 40 is turned off in a period DON in which the secondary-side rectifier diode DO conducts and is turned on, that is, in a period TOFF3, and the secondary-side rectifier diode DO is turned off. During a certain period DOFF, that is, during a period TON3, the clamp current IQ3 is caused to flow as shown in FIG. However, the first half period in which the clamp current IQ3 flows in the negative polarity direction is a period in which the clamp current IQ3 flows through the clamp diode DD3 → the clamp capacitor CCL2, and the second half period in which the clamp current IQ3 flows in the positive polarity direction is the clamp capacitor CCL2 → the auxiliary switching element Q3 (drain → Source).

For example, the secondary side active clamp circuit 40
Is not provided, as shown by the broken line in FIG. 2 (f), the secondary alternating voltage V3 in the period DOFF has a sinusoidal waveform having a higher peak level. When the active clamp circuit 40 is provided, the peak current in the negative direction of the secondary alternating voltage V3 is reduced by allowing the clamp current IQ3 to flow in the period DOFF (= period TON3) as described above. It will be suppressed.

The operation waveforms of the respective parts shown in FIGS. 2A to 2G are shown in FIGS. 2H to 2H under the condition that the load becomes light, for example, to the load power Po = 50 W.
It changes as shown in (n).

Here, for example, as can be seen by comparing the primary side parallel resonance voltage V1 in FIG. 2A and FIG.
The waveform shown in (h) is the main switching element Q
The period TON during which 1 is turned on is short, so that the switching frequency is higher than at the time of heavy load shown in FIG. Actually, a slight change is also obtained in the period TOFF during which the main switching element Q1 is turned on. This is because the primary-side main switching element Q1 is controlled so that the switching frequency becomes higher as the load power becomes smaller and the secondary-side output voltage EO rises. This shows that the ON / OFF period is variably controlled. That is,
The above three parameters (switching frequency fs,
This shows that the constant voltage operation by the complex control system in which the periods TON and TOFF are variably controlled is obtained.

On the other hand, the auxiliary switching element Q2 is driven at a timing according to the voltage waveform obtained in the drive winding Ng1, and the voltage obtained in the drive winding Ng1 is the alternating voltage generated in the primary winding N1. Is excited by Therefore, the auxiliary switching element Q2 is turned on during the ON period TO so that the switching operation of the main switching element Q1 is controlled as described above.
The switching frequency is also variably controlled by varying the N2 and the OFF period TOFF2. That is, in the present embodiment, regardless of whether the auxiliary switching element Q2 is driven in a self-excited manner or not, the auxiliary switching element Q2 is adapted to respond to changes in the on / off period of the main switching element Q1. Are also controlled at the same time. This is because the drive voltage level of the auxiliary switching element Q2 changes because the induced voltage of the drive winding Ng1 changes according to the load change and the change of the rectified smoothed voltage Ei and the like.

Even under such a light load condition, the operations in the modes 1 to 3 are performed at the timings shown in FIGS. 2H to 2N, so that the primary side parallel resonance voltage V1
Is suppressed to about 150 Vp, and the peak level of the voltage V2 across the auxiliary switching element Q2 is also suppressed to, for example, about 150 Vp.

In addition, as described above, as the composite control is executed in response to the load fluctuation (or the fluctuation of the AC input voltage) on the primary side, the following operation is performed on the secondary side. That is, as can be seen by comparing the secondary-side alternating voltage V3 in FIGS. 2F and 2M with the clamp current IQ3 in FIGS. 2G and 2N, the period DON increases as the load decreases.
(= Period TOFF3) is controlled to be short. On the other hand, the period DOFF (== the period TON3) changes so as to be slightly longer, but is extremely small with respect to the change in the period DON.

For reference, the values selected for the main elements in the power supply circuit shown in FIG. 1 when the experimental results shown in FIG. 2 are obtained are shown. First, MOS-FET
400 for the main switching element Q1
V / 10A low on-resistance product and MOS-F
For each of the auxiliary switching elements Q2 and Q3, which are regarded as ET, a low on-resistance product of 400V / 10A was selected.
The following values were selected for the remaining elements. Parallel resonance capacitor Cr = 1000 pF Clamp capacitor CCL = 0.033 μF Secondary parallel resonance capacitor C2 = 4700 pF Primary winding N1 = Secondary winding N2 = 43T Driving winding Ng1 = Ng2 = 1T Clamp capacitor CCL2 = 0.1 μF Capacitor Cg1 = 0.39μF Resistance R1 = 22Ω Resistance R2 = 100Ω

FIG. 3 shows a constant voltage characteristic of the power supply circuit according to the present embodiment shown in FIG. 1 as an on-period TON of the main switching element Q1 with respect to a change in the AC input voltage VAC.
1 and fluctuation characteristics of the ON period TON3 of the auxiliary switching element Q3. Also, in this figure, the relationship of the power conversion efficiency ηAC → DC with respect to the AC input voltage VAC is shown.

As shown in this figure, the AC input voltage V
According to the rise of AC, the ON period TON of the main switching element Q1 is controlled to be shortened. This is because the OFF period TOFF of the main switching element Q1 is almost constant, and the ON period TON is controlled by PWM. This shows that an operation as a complex control method in which the switching frequency is varied is obtained. The on-period TON3 of the secondary-side auxiliary switching element Q3 is changed to be slightly longer in accordance with the rise of the AC input voltage VAC, but the change is very small. You can see that.

FIG. 3 shows the power conversion efficiency ηAC → DC with respect to the fluctuation of the AC input voltage VAC.
In the circuit shown in FIG. 9, as shown in FIG. 11, the power conversion efficiency ηAC → DC has a tendency to decrease as the AC input voltage VAC increases. As shown in FIG. 3, the circuit of FIG. 1 is substantially constant irrespective of the fluctuation of the AC input voltage VAC, and tends to decrease as the AC input voltage VAC increases. It is not seen.

FIG. 4 shows, as constant voltage characteristics, fluctuation characteristics of the ON period TON of the main switching element Q1 and the ON period TON3 of the auxiliary switching element Q3 with respect to the fluctuation of the load power Po. Further, in this figure, the fluctuation characteristics of the power conversion efficiency ηAC → DC with respect to the load power Po are shown. As shown, the load power Po is 50
As the weight increases from W to 200 W, the on-period TON of the main switching element Q1 tends to increase as a constant voltage is applied by the complex control method. On the other hand, the on-period TON3 of the auxiliary switching element Q3 has a tendency to decrease, but the inclination is gentler than the change of the on-period TON of the main switching element Q1. .

As for the power conversion efficiency η AC → DC, about 90% is stably maintained in the fluctuation range of the load power Po = 50 W to 200 W. That is, also in this case, there is no tendency that the power conversion efficiency ηAC → DC is reduced in accordance with the light load condition.

As shown in FIGS. 3 and 4, the power conversion efficiency ηAC → DC does not decrease as the AC input voltage VAC increases or the load becomes lighter. The reason why the voltage is kept substantially constant is because the fluctuation of the ON period TON3 of the auxiliary switching element Q3 is reduced as described above. That is, the on-period TON3 (conduction angle) of the auxiliary switching element Q3 does not significantly increase even by the point voltage control operation corresponding to the tendency of the increase of the AC input voltage VAC or the light load. The problem of an increase in the power loss of the switching element Q3 is almost eliminated. On the other hand, in the circuit shown in FIG. 9, for example, the stabilizing operation depends on controlling the conduction angle (ON period TON3) of the auxiliary switching element Q3. If the load tends to occur, the auxiliary switching element Q3
Is increased, and an increase in power loss in the auxiliary switching element Q3 cannot be avoided.

As a result, the power conversion efficiency of the power supply circuit of the present embodiment is significantly improved. Further, since the heat generation of the auxiliary switching element Q3 is suppressed with the improvement of the power conversion efficiency, it is not necessary to provide a heat sink.

As described with reference to FIG. 2, the provision of the primary-side active clamp circuit 20 allows the peak level of the primary-side parallel resonance capacitor Cr to be about half that of the circuit shown in FIG. It can be suppressed up to. Therefore, for elements such as the main switching element Q1 and the primary side parallel resonance capacitor Cr,
A device with a lower breakdown voltage than before can be selected. For example, in the case of an AC 100 V system, the main switching element Q1 provided on the primary side and
It is sufficient to select a withstand voltage product of 400 V, and in the case of an AC 200 V system, a double withstand voltage product of 800 V may be selected.

The secondary side active clamp circuit 40
, The level of the secondary side alternating voltage (secondary parallel resonance voltage) V3 can be suppressed to about 1/2, so that the auxiliary switching element Q
3. Secondary parallel resonance capacitor C2 and rectifier diode D
As for O and the like, those having a lower breakdown voltage than before can be selected. For example, if the level required as the secondary side DC output voltage EO is up to EO = 150 V, the auxiliary switching element
What is necessary is to select a withstand voltage product of 00V. And
In this manner, the characteristics of each element are improved and the power loss can be reduced by making each element a low withstand voltage product in this manner, so that the reliability as a power supply circuit can be improved. In this respect, the power conversion efficiency can be improved. Further, since the low withstand voltage product is inexpensive and small, the cost can be reduced and the circuit can be downsized.

In the present embodiment, in the active clamp circuits 20 and 40, the auxiliary switching elements are driven by a self-excited drive circuit system. For example, as another configuration for switchingly driving the auxiliary switching element, a switching drive circuit such as an IC is provided, and a separately-excited driving circuit system for driving the auxiliary switching element by combined control is additionally provided. Can be considered. That is, the auxiliary switching element is driven by a circuit such as a separately-excited IC together with the main switching element. But,
In this configuration, a separately-excited circuit system for simultaneously performing switching frequency control and PWM control for the main switching element and a separately-excited circuit system for simultaneously performing switching frequency control and PWM control for the auxiliary switching element are provided. Must be provided. Therefore, the circuit configuration becomes more complicated and the number of parts increases.
This hinders downsizing of the circuit. On the other hand, if the configuration as described above is adopted in the present embodiment, the auxiliary switching element drive circuit system includes only a 1T winding wound around the insulating converter transformer PIT, several resistors,
And a single capacitor, which is a very simple circuit configuration, which can also realize the same operation as that of the separately excited type.

FIG. 5 shows a configuration example of a switching power supply circuit according to the second embodiment. Note that FIG.
In the figure, the same parts as those in FIG. 1 are denoted by the same reference numerals, and description thereof will be omitted.

The primary side of the power supply circuit shown in FIG.
A self-excited configuration is shown as a voltage resonance type converter circuit that performs single-ended operation by a stone switching element Q1. In this case, a high withstand voltage bipolar transistor (BJT; junction transistor) is employed as the switching element Q1.

The base of the switching element Q1 is connected to the positive electrode of the smoothing capacitor Ci (rectified smoothing voltage Ei) via the base current limiting resistor RB-starting resistor RS to obtain a base current at the time of starting from the rectifying smoothing line. Like that. In addition, a series resonance circuit for driving self-oscillation composed of a series connection circuit of a drive winding NB, a resonance capacitor CB and a base current limiting resistor RB is connected between the base of the switching element Q1 and the primary side ground. A clamp diode DD inserted between the base of the switching element Q1 and the negative electrode (primary ground) of the smoothing capacitor Ci forms a path for a clamp current flowing when the switching element Q1 is turned off. , The collector of switching element Q1 is connected to one end of primary winding N1 of insulating converter transformer PIT, and the emitter is grounded.

A parallel resonance capacitor Cr is connected in parallel between the collector and the emitter of the switching element Q1. Also in this case, the primary parallel resonance circuit of the voltage resonance type converter is formed by the capacitance of the parallel resonance capacitor Cr itself and the leakage inductance L1 on the primary winding N1 side of the insulating converter transformer PIT.

The orthogonal control transformer PRT shown in FIG.
Is a saturable reactor on which a resonance current detection winding ND, a drive winding NB, and a control winding NC are wound. This orthogonal control transformer PRT drives the switching element Q1 and is provided for constant voltage control. As the structure of the orthogonal control transformer PRT, although not shown, a three-dimensional core is formed by joining the ends of the two magnetic legs of two double U-shaped cores having four magnetic legs. . Then, a resonance current detection winding ND and a drive winding NB are wound around the predetermined two magnetic legs of the three-dimensional core in the same winding direction, and the control winding NC is connected to the resonance current detection winding. It is constructed by winding in a direction orthogonal to the winding ND and the driving winding NB.

In this case, the resonance current detecting winding ND of the orthogonal control transformer PRT is inserted in series between the positive electrode of the smoothing capacitor Ci and the primary winding N1 of the insulating converter transformer PIT, so that the switching element Q1 Is transmitted to the resonance current detection winding ND via the primary winding N1. In the orthogonal control transformer PRT, the switching output obtained in the resonance current detection winding ND is induced in the drive winding NB via the transformer coupling, so that an alternating voltage as a drive voltage is applied to the drive winding NB. appear. This drive voltage is output from the series resonance circuit (NB, CB) forming the self-excited oscillation drive circuit to the base of the switching element Q1 as a drive current via the base current limiting resistor RB. As a result, the switching element Q1 performs a switching operation at a switching frequency determined by the resonance frequency of the series resonance circuit. The switching output obtained at the collector is transmitted to the primary winding N1 of the isolated converter transformer PIT.

The insulating converter transformer PIT provided in the circuit shown in FIG. 5 has the same structure as that described with reference to FIG. 7, and the primary side and the secondary side are loosely coupled. The state is obtained.
Also on the secondary side of the power supply circuit shown in this figure, a secondary parallel resonance circuit is formed by connecting a secondary parallel resonance capacitor C2 in parallel with the secondary winding N2. Therefore, the power supply circuit also has a configuration as a complex resonance type switching converter.

On the secondary side, the rectifier circuit for obtaining the secondary side DC output voltage EO also has the same configuration as that shown in FIG. In the present embodiment, the secondary-side DC output voltage EO is input to the control circuit 1 as a detection voltage.

In the control circuit 1, the level of the control current (DC current) flowing through the control winding NC is varied according to the change in the secondary-side DC output voltage level EO, thereby being wound around the orthogonal control transformer PRT. The variable inductance LB of the driving winding NB is controlled. Thereby, the resonance condition of the series resonance circuit in the self-excited oscillation drive circuit for the switching element Q1 formed including the inductance LB of the drive winding NB changes. This is an operation of changing the switching frequency of the switching element Q1. Therefore, in the present embodiment, the low-voltage secondary-side DC output voltage level EO2 is stabilized by the switching frequency control on the primary side. In addition, even in the configuration of the constant voltage control including the orthogonal control transformer PRT, since the switching converter on the primary side is a voltage resonance type, the switching in the switching cycle can be performed simultaneously with the variable control of the switching frequency. An operation is performed as a complex control system that also performs element conduction angle control (PWM control).

The power supply circuit shown in FIG. 5 also includes the secondary-side active clamp circuit 40. Here, BJT is employed as the auxiliary switching element. When the BJT is adopted in this manner, for example, as shown in FIG.
Switching elements Q3A and Q3B, which are N transistors, are provided, and are connected in Darlington. Here, the base of the switching element Q3B is connected to the drive winding Ng.
It is connected to a self-excited drive circuit consisting of 2-capacitor Cg2-resistance Rg2, and the collectors of switching elements Q3A and Q3B are connected. The connection point of the collector is connected to the winding start end of the secondary winding N2 via the clamp capacitor CCL2. Also, the switching element Q
The emitter of 3B is connected to the base of switching element Q3A. The emitter of the switching element Q3A is connected to the secondary side ground via the cathode and anode of the clamp diode DD3. In this connection configuration, the clamp current flowing through the clamp diode DD3 is the switching element Q3A
Through a base-collector PN junction to the clamp capacitor CCL2.

Then, in the second embodiment,
Regardless of the condition of rising AC input voltage VAC and light load,
ON period TON3 of auxiliary switching element [Q3A, Q3B]
Can be small, and the power loss is reduced. Also in the circuit shown in FIG. 5, since the secondary side active clamp circuit 40 is provided, the secondary side is similar to the circuit of FIG.
The level of the secondary side alternating voltage (secondary parallel resonance voltage) V3 is suppressed to about 1/2. Therefore, low withstand voltage products can be selected for the auxiliary switching element Q3, the secondary parallel resonance capacitor C2, the rectifier diode DO, and the like.

FIG. 6 is a circuit diagram showing a configuration example of a power supply circuit according to the third embodiment. In this figure, the same parts as those in FIGS. 1 and 5 are denoted by the same reference numerals, and description thereof will be omitted. The configuration of the circuit shown in FIG. 6 is the same as that of the circuit shown in FIG. However, an IGBT (insulated gate bipolar transistor) is selected for each switching element of the main switching element Q1, the auxiliary switching element Q2, and the auxiliary switching element Q3. Therefore, in this case, as the clamp diodes DD1, DD2, and DD3 connected to the respective switching elements, not the body diodes but diode elements for these are additionally connected. For example, an IGBT with a high withstand voltage is very expensive like a MOS-FET, but in this embodiment, as described above, the switching element has a low withstand voltage by being provided with an active clamp circuit. Since a product can be selected, it is very advantageous in terms of cost reduction.

In the second embodiment, an orthogonal control transformer is used as a control transformer for performing a constant voltage control under a configuration having a self-excited resonance converter for the primary side. However, instead of the orthogonal control transformer, an oblique control transformer previously proposed by the present applicant can be adopted. Although the illustration of the structure of the oblique control transformer is omitted here,
For example, as in the case of the orthogonal control transformer, a three-dimensional core is formed by combining two sets of double U-shaped cores having four magnetic legs. Then, the control winding NC and the driving winding NB are wound around the three-dimensional core.
At this time, the relationship between the winding directions of the control winding and the drive winding is set to obliquely intersect. Specifically, any one of the control winding NC and the drive winding NB is wound around two magnetic legs adjacent to each other among the four magnetic legs, The other winding is wound around two magnetic legs which are considered to be in a diagonal positional relationship. When such an oblique control transformer is provided, even when the AC current flowing through the drive winding changes from a negative current level to a positive current level, the operation tendency that the inductance of the drive winding increases. Is obtained. As a result, the current level in the negative direction for turning off the switching element increases, and the accumulation time of the switching element is shortened.Accordingly, the fall time at the time of turning off the switching element is also shortened, The power loss of the switching element can be further reduced.

The present invention is not limited to the configurations shown in the above embodiments. For example, in the above embodiments, the main switching element and the auxiliary switching element are MOS-FET,
Although JT, IGBT and the like are adopted, other elements such as SIT (static induction thyristor) may be adopted. Further, the configuration of the switching drive unit for driving the main switching element Q1 by the separately-excited system need not be limited to that shown in each of the drawings, and may be appropriately changed to an appropriate circuit configuration. Also, the secondary-side rectifier circuit formed including the secondary-side resonance circuit is not limited to the configurations shown in the drawings as the embodiments, and other circuit configurations may be adopted. Not something.

[0091]

As described above, the present invention relates to a composite resonant converter having a secondary-side active clamp circuit, wherein the switching frequency and the conduction angle of the switching elements forming the primary-side switching converter are simultaneously controlled. The secondary side DC output voltage is made to be a constant voltage by the composite control method.

For example, when a constant voltage is to be achieved by controlling the conduction angle of the auxiliary switching element forming the secondary side active clamp circuit, the conduction angle of the auxiliary switching element is increased in accordance with the rise of the AC input voltage and light load. However, this has a problem that power loss increases.
On the other hand, in the present invention, by adopting the configuration of the constant voltage control described above, the change in the conduction angle of the secondary-side auxiliary switching element according to the change in the AC input voltage or the load is reduced. Can be. Therefore, the power loss accompanying the constant voltage control is hardly increased. As a result, the power conversion efficiency of the power supply circuit is made substantially constant without being reduced by the fluctuation of the AC input voltage or the load, and the power conversion efficiency is largely improved. In addition, since heat generation in the switching element is also suppressed, it is not necessary to provide a heat radiating plate, so that the power supply circuit can be reduced in size, weight, and cost.

The secondary side active clamp circuit is
Since the peak level of the alternating voltage obtained in the secondary winding is suppressed, for example, an auxiliary switching element forming a secondary-side active clamp circuit, a secondary-side rectifier diode,
A low withstand voltage product can be selected for the secondary side parallel resonance capacitor and the like, whereby the switching loss can be reduced, and the cost and size can be reduced. As the present invention, if an active clamp circuit is provided on the primary side with respect to the above configuration, the peak level of the parallel resonance pulse voltage generated in the primary side voltage resonance type converter is suppressed, so that the primary side voltage resonance type A low breakdown voltage product can also be selected for the main switching element, the primary side parallel resonance capacitor, and the like forming the converter, which can also contribute to a reduction in cost and size of the power supply circuit. Then, also on the primary side, the switching loss can be reduced, and the power conversion efficiency of the entire power supply circuit can be further improved.

[Brief description of the drawings]

FIG. 1 is a circuit diagram showing a configuration of a switching power supply circuit according to a first embodiment of the present invention.

FIG. 2 is a waveform chart showing an operation of a main part in the power supply circuit of the present embodiment.

FIG. 3 is an explanatory diagram showing a constant voltage characteristic with respect to an AC input voltage VAC and a fluctuation characteristic of power conversion efficiency in the power supply circuit of the present embodiment.

FIG. 4 is an explanatory diagram showing constant voltage characteristics with respect to load power and fluctuation characteristics of power conversion efficiency in the power supply circuit of the present embodiment.

FIG. 5 is a circuit diagram showing a configuration of a switching power supply circuit according to a second embodiment.

FIG. 6 is a circuit diagram showing a configuration of a switching power supply circuit according to a third embodiment.

FIG. 7 is a cross-sectional view illustrating a configuration of an insulating converter transformer.

FIG. 8 is an equivalent circuit diagram showing each operation when the mutual inductance is + M / −M.

FIG. 9 is a circuit diagram showing a configuration example of a switching power supply circuit as a prior art.

10 is a waveform chart showing an operation of a main part of the power supply circuit shown in FIG.

11 is an explanatory diagram showing constant voltage characteristics with respect to an AC input voltage VAC and fluctuation characteristics of power conversion efficiency in the power supply circuit shown in FIG. 9;

12 is an explanatory diagram showing constant voltage characteristics with respect to load power and fluctuation characteristics of power conversion efficiency in the power supply circuit shown in FIG. 9;

[Explanation of symbols]

1 control circuit, 20 primary side active clamp circuit,
30 Photocoupler, 40 Secondary active clamp circuit, 10 Switching drive unit, 11 Oscillation circuit, 1
2 drive circuit, Q1 (main) switching element, Q2, Q3, Q3A, Q3B auxiliary switching element, P
IT isolation converter transformer, N1 primary winding, N2
Secondary winding, N3 tertiary winding, Cr Primary parallel resonance capacitor, C2 Secondary parallel resonance capacitor, DO Secondary rectifier diode

Claims (2)

[Claims] 1. A switching means comprising a main switching element for switching and outputting an inputted DC input voltage, and a primary side parallel resonance circuit which makes the operation of the switching means a voltage resonance type. And a primary side parallel resonant capacitor provided in such a manner that a gap is formed so as to obtain a required coupling coefficient that is loosely coupled between the primary side and the secondary side. An insulating converter transformer for transmitting an output to the secondary side, and a secondary-side parallel resonance circuit formed by connecting a secondary-side parallel resonance capacitor in parallel to the secondary winding wound on the insulating converter transformer The rectifying operation is performed by inputting the alternating voltage obtained in the secondary winding wound around the above-mentioned isolated converter transformer, and thereby the secondary-side DC output voltage is obtained. DC output voltage generating means configured to obtain a secondary winding, comprising a series connection circuit including a clamp capacitor and a secondary auxiliary switching element connected in parallel to the secondary winding. Secondary-side active clamping means provided to clamp a voltage generated in the line, and by controlling the switching frequency and conduction angle of the main switching element according to the level of the secondary-side DC output voltage, And a voltage control means for performing constant voltage control on the secondary-side DC output voltage. 2. A primary auxiliary switching element that performs switching so as to have a predetermined on / off timing synchronized with an on / off timing of the main switching element, so that both ends of the primary parallel resonance capacitor are provided. 2. The switching power supply circuit according to claim 1, further comprising: primary-side active clamp means provided to clamp the generated primary-side parallel resonance voltage. 3.